EP1825218B1 - Verfahren zur steuerung/regelung einer physikalischen grösse eines dynamischen systems - Google Patents

Verfahren zur steuerung/regelung einer physikalischen grösse eines dynamischen systems Download PDF

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EP1825218B1
EP1825218B1 EP05811363.0A EP05811363A EP1825218B1 EP 1825218 B1 EP1825218 B1 EP 1825218B1 EP 05811363 A EP05811363 A EP 05811363A EP 1825218 B1 EP1825218 B1 EP 1825218B1
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Prior art keywords
modulation signal
resonator
deviation
vibration
phase
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French (fr)
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EP1825218A1 (de
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Günter SPAHLINGER
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Northrop Grumman Litef GmbH
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Northrop Grumman Litef GmbH
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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01C—MEASURING DISTANCES, LEVELS OR BEARINGS; SURVEYING; NAVIGATION; GYROSCOPIC INSTRUMENTS; PHOTOGRAMMETRY OR VIDEOGRAMMETRY
    • G01C19/00—Gyroscopes; Turn-sensitive devices using vibrating masses; Turn-sensitive devices without moving masses; Measuring angular rate using gyroscopic effects
    • G01C19/56—Turn-sensitive devices using vibrating masses, e.g. vibratory angular rate sensors based on Coriolis forces
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B81—MICROSTRUCTURAL TECHNOLOGY
    • B81B—MICROSTRUCTURAL DEVICES OR SYSTEMS, e.g. MICROMECHANICAL DEVICES
    • B81B7/00—Microstructural systems; Auxiliary parts of microstructural devices or systems
    • B81B7/02—Microstructural systems; Auxiliary parts of microstructural devices or systems containing distinct electrical or optical devices of particular relevance for their function, e.g. microelectro-mechanical systems [MEMS]

Definitions

  • the invention relates to a method for controlling or regulating a physical variable of a dynamic system, for example a micromechanical sensor, to a specific desired value or setpoint course using a pulse modulator, by which a series of discrete modulation signals, the control or regulation of the physical quantity cause to be generated.
  • the invention further relates to a method for the simultaneous control or regulation of at least two physical variables of a system, for example a micromechanical sensor, to specific setpoints or setpoint curves using a pulse modulator, by a sequence of discrete modulation signals, the control of the physical Sizes causes are generated.
  • Coriolis gyros also called vibration gyros
  • Coriolis gyros have a mass system that is vibrated. The vibration is usually a superposition of a variety of Einzelschwinguiigen.
  • the individual vibrations of the mass system are initially independent of each other and can each be understood abstractly as "resonators".
  • resonators For operating a Coriolis gyro at least two resonators are required: one of these resonators (first resonator) is artificially excited to oscillations, which is referred to below as "excitation oscillation".
  • the other resonator (second resonator) is excited to vibrate only when the Coriolis gyro is moved / rotated.
  • Coriolis forces occur which couple the first resonator to the second resonator, remove energy from the excitation oscillation of the first resonator and transmit this energy to the read oscillation of the second resonator.
  • the oscillation of the second resonator will be referred to as "read oscillation" hereinafter.
  • the read oscillation is tapped and a corresponding read-out signal (eg the read oscillation tapping signal) is examined to see whether changes in the amplitude of the read oscillation, which are a measure of the rotation of the Coriolis gyro, have leaked.
  • Coriolis gyros can be used both as an open loop system and as a closed loop system will be realized.
  • the amplitude of the read oscillation is continuously reset to a fixed value, preferably zero, via respective control circuits.
  • the term "resonator” is understood to mean an oscillatable mass system, which may include mechanical springs.
  • the term “oscillator” is used as a synonym in the description.
  • Fig. 1 1 shows an evaluation / control electronics 1, which includes a charge amplifier 2, an analog-to-digital converter 3, a signal separation 4, a first demodulator 5, a second demodulator 6, a control system 7, a two-dimensional pulse modulator 8, a first and second force pulse Conversion unit 9, 10, and a first to fourth force transducer electrode 11 1 to 11 4 .
  • the entirety of the components identified by reference numerals 2 to 11 forms two control circuits: a control circuit for adjusting the amplitudes, frequencies and phases of the excitation oscillation and a further control circuit for adjusting the amplitudes, frequencies and phases of the Read oscillation.
  • Fig. 1 shows, the circuit according to the invention only an analog-to-digital converter 3 and no digital-to-analog converter.
  • the digital-to-analog converters are here replaced by the two-dimensional pulse modulator 8 and the two power pulse conversion units 9, 10.
  • the two-dimensional pulse modulator 8 In order to set the amplitudes or frequencies or phases of the excitation oscillation / read oscillation of the resolver R, the two-dimensional pulse modulator 8 generates a first and a second ternary quantized output signal S 1 , S 2 , wherein the first ternary quantized output signal S 1 in the first force pulse Conversion unit 9 in power pulse signals (voltage signals) S 3 , S 4 is converted. Accordingly, the second ternary quantized output signal S 2 is converted into power pulse signals (voltage signals) S 5 , S 6 by the second power pulse conversion unit 10.
  • the ternary quantized output signals S 1 , S 2 can each assume the values 1, 0 and -1.
  • the resulting force is always positive due to the squaring of the potential.
  • the second ternary quantized output signal S 2 which is converted by the second force pulse conversion unit 10 into a fifth and sixth force pulse signal S 5 , S 6 , which are applied to the first and third force-sensor electrode 11 1 , 11 3 .
  • the parameters of the excitation oscillation are set via the force-transmitting electrodes 11 2 , 11 4
  • the parameters of the read oscillation are set / regulated via the force-transmitting electrodes 11 1 , 11 3 .
  • the charge which has flowed onto the center electrode is dependent on the capacitances of those force-transmitting electrodes 11 1 to 11 4 at which an electric field is presently applied, the charge that has flowed is a measure of the amplitudes or frequencies or other parameters of the excitation oscillation / read oscillation of the resonator R Therefore, as a function of momentary and / or temporally older output signal values of the ternary quantized output signals S 1 , S 2, the instantaneous movement / change of the movement of the resonator R can be reconstructed by the signal separation 4. If positive and negative potentials +/- U 0 occur, the signal separation 4 must continue to take account of the sign of the potential U 0 (the voltage applied to the force-transmitting electrodes 11 1 to 11 4 ) during the reconstruction.
  • the two-dimensional pulse modulator 8 is designed such that that the ternary quantized output signals S 1 and S 2 never change at the same time, because in general the charge flowing to the center electrode charge is measured summarily, ie that charge shifts resulting from a superposition of two electric fields, can only be measured as a whole , so can not be assigned to parts of the charge shift of individual electric fields.
  • the additional condition between the ternary quantized output signals S 1 and S 2 then makes it possible to obtain an unambiguous assignment of the discharged charge to a specific electric field so that a precise distinction can be made between excitation oscillation and read oscillation.
  • Another possible condition in this context is to stipulate that only one of the two signals S 1 and S 2 may assume values other than zero at a particular time.
  • the first digital readout signal S 9 is demodulated by the first demodulator 5 into a real part S 11 and an imaginary part S 12 .
  • the second digital readout signal S 10 is demodulated by the second demodulator 6 into a real part S 13 and an imaginary part S 14 .
  • the first digital readout signal S 9 contains information about the excitation oscillation
  • the second digital readout signal S 10 contains information about the read oscillation.
  • the real and imaginary parts S 11 to S 14 of the first and second digital readout signals S 9 , S 10 act on the control system 7, which generates excitation / compensation signals S 15 to S 18 as a function of these signals.
  • the signal S 15 represents the real part and the signal S 16 represents the imaginary part of the digital excitation / compensation signal for the excitation oscillation
  • the signal S 16 represents the real part
  • the signal S 18 represents the imaginary part of a digital excitation / compensation signal for the read oscillation
  • the digital excitation / compensation signals S 15 to S 18 are supplied to the two-dimensional pulse modulator 8, which generates therefrom the ternary quantized output signals S 1 , S 2 .
  • the control principle described above is not based on the regulation of excitation oscillation / read oscillation but can be used in many ways:
  • MEMS micromechanical sensors
  • electrostatic excitation or provision of a resonator in particular in the above-described Coriolis gyroscope, for example, often the resonant frequency of the resonator on be set to a predetermined value.
  • electrostatic return springs whose (positive or negative) spring constants can be adjusted by electrical voltages.
  • the resonator is usually formed of (preferably parallel connected) mechanical springs on which the oscillatory mass system is suspended, and the oscillatory mass element itself.
  • the resonant frequency of a deratigen resonator can by means of associated with Fig. 1 be set explained control technology. That is, instead of an analog, adjustable voltage, a digital pulse train is generated which "trims" the resonant frequency to a resonance frequency corresponding to the mean of the pulses. For example, can be adjusted by a corresponding switching sequence of the electrostatic return springs on natural resonances of 9000 Hz or 9200 Hz (ie by a corresponding pulse train of two pulse values) a self-resonance of 9100 Hz.
  • This has - as already mentioned - the advantage that the costly and with a relatively large power consumption afflicted digital-to-analog converter can be saved.
  • the object underlying the invention is to provide a method for controlling / regulating a physical variable of a micromechanical sensor (or, more generally, a dynamic system) using a pulse modulator which allows a suppression of possibly occurring parametric effects.
  • a method for the digital control of the spring constant of a resonator to a predetermined resonant frequency while suppressing parametric effects is to be provided.
  • the invention provides a method according to claim 1 and claim 9 ready. Furthermore, the invention provides a device according to claim 8 and claim 15. Advantageous embodiments and developments of the inventive concept can be found in the subclaims.
  • step (a) By repeatedly performing steps (a) to (c), the control of the physical quantity is effected to the target value.
  • An important principle underlying the method according to the invention is that in each iteration step, i. H. each time after performing step (a), the effects of all the modulation signals that can be generated on the current actual value of the physical quantity or on the estimate of the actual value of the physical quantity are determined. That is, the effects of the individual modulation signals are simulated before the pulse modulator actually generates a corresponding modulation signal and thus influences the instantaneous value of the physical quantity.
  • the modulation signal is selected which has the "best" effect on the physical quantity in the simulation, ie. h., leads to the best approximation of the current setpoint.
  • the advantage of such a control method is that it can be easily combined with appropriate control schemes of other physical quantities while preventing suppression of parametric effects.
  • the method according to the invention can be applied particularly advantageously to micromechanical sensors with a resonator.
  • the physical variable to be controlled or controlled in this case could be, for example, the resonant frequency of the resonator.
  • the amplitude or phase of oscillation of the resonator could also be controlled.
  • the inventive method can also be applied to dynamic systems such as pendulum systems in accelerometers, oscillators (electrical, electromechanical, mechanical) with adjustable frequency (eg for a clock frequency generation). Further suitable systems are adjustable band filters, quartz filters, etc. All physical variables relevant in connection with these systems can be regulated by the method according to the invention.
  • the invention is not limited to the dynamic systems explicitly mentioned above.
  • step (a) either the exact value of the deviation or an approximation for the deviation between the current set value and the instantaneous actual value of the physical quantity can be determined.
  • This problem can be counteracted by simulating the resonator's oscillation response, which would result from applying the resonator (for a given initial amplitude and initial phase) with a modulation signal sequence, to control the resonant frequency of the resonator, and thus selecting the modulation signal sequence in that in the simulation the most accurate possible approximation of a vibration setpoint course of the resonator results.
  • the frequency of the vibration setpoint curve is the resonant frequency of the resonator to be controlled.
  • the "real" resonator is supplied with the modulation signal sequence thus obtained.
  • vibration response herein is meant the response of the resonator to the modulation signal sequence, i. understood the resulting from the modulation signal sequence natural vibration (decay) of the resonator.
  • the decay process carried out in the simulation is to be understood such that in the simulation the resonator experiences an initial excursion (initial amplitude, initial phase) and then is left to itself, and the modulation signal sequence whose effect on the decay process is tested in the simulation.
  • the phase and amplitude of the decay (damped) decay settles (to the ideal set point), it has “nothing to do” with the phase and amplitude of the initial excursion (this is "constraint” and independent of the modulation burst).
  • the attenuated, decaying inherent oscillation process of the real resonator is simulated as a function of a modulation signal sequence, and in each iteration step or time step (discrete-time digital modulation method) the oscillation response resulting from the previous modulation signals is compared with an ideal natural oscillation which the system controls at the resonant frequency to be controlled would have.
  • the "effective total deviation” thus represents a “global” error representing the sum of phase error and amplitude error for a producible modulation signal. Keeping the "global” error as small as possible (step e)) always causes the physical quantity that contributes the most to the total deviation to be given priority, ie. h., which has the largest "regulation needs".
  • the method according to the invention can be applied particularly advantageously to micromechanical sensors with a resonator.
  • the physical variables to be controlled could be, for example, a resonance frequency of the resonator as well as amplitudes or phases of the excitation oscillation and / or read oscillation of the resonator.
  • a damped, decaying inherent oscillation process of the real resonator is simulated as a function of a modulation signal sequence, and in each iteration step or clock cycle the natural oscillation resulting from the previous modulation signals is compared with an "ideal" natural oscillation which has an ideal system at the resonant frequency to be controlled would.
  • the effects of all the generatable modulation signals on the instantaneous vibration state of the simulated real resonator are determined, and in the next clock cycle the resonator in the simulation is loaded with the modulation signal selected in step b), i. H. for which the total effective deviation (for all physical quantities to be controlled simultaneously) is lowest.
  • the comparison of the hypothetical resulting natural vibration profiles with the natural vibration desired value curve can be carried out, for example, by comparing corresponding amplitudes and phases of the profiles with one another.
  • a total deviation which is given is calculated for each producible modulation signal the sum of the deviations between the instantaneous setpoints and corresponding amplitude and phase simulation values that would result from maintaining this modulation signal or changing over to that modulation signal.
  • the total deviations here represent the resonance frequency deviation approximations to be determined.
  • total deviation thus represents a "global” error, which in this embodiment represents the sum of phase errors and amplitude errors with respect to a producible modulation signal.
  • total effective error represents a global error, part of this global error being the total deviations determined in this embodiment, and another part of the global error resulting from the deviations of at least one other physical quantity to be controlled. Keeping the global error as small as possible (step b)) always causes the physical quantity which contributes the most to the overall effective deviation to be given priority, ie. h., which has the largest "regulation needs".
  • the desired resonant frequency of the resonator is automatically set to the desired value.
  • the invention further provides a device for the simultaneous control or regulation of at least two physical variables of a dynamic system to specific setpoint values / desired value curves.
  • the device has a pulse modulator, by which a series of discrete modulation signals, which causes the control / regulation of the physical quantities, can be generated.
  • the device further comprises a calculation unit which calculates for each producible modulation signal an effective total deviation which is given by the sum of the exact values or the approximation for the deviations between the instantaneous desired values and corresponding actual values of the physical quantities resulting from the maintenance of this Modulation signal or the transition to this modulation signal would result.
  • a decision unit connected to the calculation unit is provided which, depending on the total effective deviations calculated by the calculation unit, decides for which modulation signal the calculated total effective deviation would be lowest and drives the pulse modulator to generate a corresponding modulation signal.
  • the method according to the invention makes it possible to simultaneously set the resonant frequency of a resonator as well as to stimulate or reset oscillation amplitudes of the resonator.
  • the following is with reference to the Fig. 2 to 19 a preferred embodiment for controlling the resonant frequency of the resonator explained.
  • T an (initially) arbitrary, constant time.
  • s ⁇ 1 t 1 T ⁇ - ⁇ t s ⁇ 0 ⁇ d ⁇
  • s ⁇ 2 t 1 T ⁇ - ⁇ t s ⁇ 1 t d ⁇
  • the components identified by reference numeral 20 represent operators that multiply corresponding input signals (or states) by the specified factors (or matrices).
  • the components identified by reference numeral 21 represent integrators that integrate corresponding input signals in consideration of the specified factors.
  • the components identified by reference numeral 22 are delay elements.
  • the components identified by reference numeral 23 are summing and subtracting nodes, respectively.
  • the transition matrix A * is expressed as equations (40) and (41)
  • M - 1 + 1 H 0 T d 2 m H 1
  • the complex variable s always has a well-defined amplitude
  • and Pase ⁇ arc ( S ).
  • the matrix M ( n ) can be routed via the summing node 23 and the delay 22. However, M ( n ) goes into M ( n -1) ( Fig. 9 ).
  • the chain circuit K ( n ) M ( n -1) M -1 ( n ) no longer yields (ia) the unit matrix, so that the in Fig. 10 structure shown results.
  • K n 4 k n - 1 m - d 2 4 k n m - d 2 0 0 1
  • a first branch 25 and a second branch 26 can be seen.
  • Each branch generates an initial state s a (first branch) or s b (second branch) from an input state s .
  • one of the output states s a or s b is supplied to the restrainer 22 and thus supplied to the two branches 25, 26 again in the next clock cycle as a new input state s .
  • Each branch simulates the effect of a modulation signal (represented by A a * . A b * ) to the current vibration state (represented by the input state s ).
  • the switch selects either s a or s b as the valid state s .
  • the matrices A a ' and A b ' are the transition matrices of the symmetric system, the upstream and downstream matrices M a . b - 1 . M a . b are the correction matrices that A a . b ' into the transition matrices A a . b * of the original (unbalanced) system convict.
  • the internal signals S'a, b (which are the states which the symmetric system would have) are available in addition to the true states S a, b .
  • the reason for providing the states of the symmetric system is that they are particularly well suited as an indicator of instantaneous frequency and moment-amplitude.
  • the replica of the real vibrator can also be used directly with the transition matrices A a . b * realize the unbalanced system, without having to do without the symmetric state variables, such as Fig. 12 shows.
  • the modulation of the spring constant should be such that the decay process of the replica of the real oscillator in amplitude and phase as closely as possible follows a predetermined desired function.
  • the Fig. 13 shows a device for determining the magnitude of the deviation in the approximation of the transient process.
  • the input signal s' is the vector of the (symmetrized) state variables of the replica. From this, by interpreting the state variable vector as a complex pointer, the phase of the given setpoint function is subtracted by taking e - jn ⁇ 0 t multiplied. From the result ⁇ e 1 , e 2 ⁇ the phase ⁇ e is determined, which is the phase deviation from the specification.
  • the amplitude of s ' is determined by magnitude. After subtracting the amplitude a ( n ) of the specification, the amplitude deviation a e results. From phase and amplitude deviation can finally derive a total error e by z. B. the sum of squares of the two deviations is formed. On the right in the figure, a simplified symbolic representation ⁇ E> of the arrangement is shown.
  • the module ⁇ E> can therefore be regarded as a comparison unit, by which an approximation for the deviation between the instantaneous desired value and the instantaneous actual value of the resonant frequency of the real oscillator to be regulated can be determined.
  • the module ⁇ E> can also be regarded as a comparison unit, by which an exact value (actually a summation of two exact values) for the deviation between the current setpoint and the instantaneous actual value of the resonant frequency to be controlled in the simulated simulated vibrator can be determined.
  • or e
  • the two possible future symmetrized states will be S a ' and S b ' evaluated with two blocks ⁇ E> with respect to the error compared to the default signal.
  • a downstream decision maker 27 compares the two errors e a and e b and selects as the next switch position that which picks up the state with the smaller error.
  • a phase and amplitude reference generator 28 whose frequency ⁇ 0 can be set produces the reference phase n ⁇ 0 T and the reference amplitude a ( n ) corresponding to the values of the ideal oscillator.
  • the arrangement shown now controls the switch so that the decay of the replica follows the amount and phase of the default in the middle.
  • the spring constant of the true oscillator can be controlled with the modulator signal generated by the decoder, which results in this real switched oscillator emulating an ideal non-switched oscillator with resonant frequency ⁇ 0 .
  • the presented system has two shortcomings. One is more practical, the other of a fundamental nature.
  • the practical shortcoming is related to the fact that the amplitude of the signal of the replica, which follows the amplitude of the default signal, is an exponentially decaying function. This means that the signals are getting smaller and smaller until no correct function is possible because of occurring numerical problems. This problem can be corrected by adding the reciprocal of the signals concerned multiplied by decaying exponential function.
  • the generated modulation signal remains the same. This is achieved by using the transition matrices A a * and A b * multiplied by the factor ⁇ defined above.
  • the replica is attenuated.
  • the real physical oscillator controlled thereby is not attenuated thereby, since the modulation signal remains unaffected by the measure.
  • the measure is to ensure only the numerically stable kontiuierlichen operation of the modulation signal generator.
  • a b * such as M a - 1 .
  • M b - 1 result from A * and M -1 by replacing ⁇ by ⁇ a , ⁇ b .
  • the result of a simulation with the presented system will be discussed.
  • the ratios are selected so that the real physical oscillator can be switched to a self-resonance of 9000 Hz or 9200 by means of the switchable spring constant.
  • a frequency specification of 9100 Hz was chosen.
  • the initial conditions were chosen so that the phases of real Schwinger and replica were the same.
  • the real vibrator was then left to itself; the associated decay process 30 is in Fig. 16 shown.
  • the result is a nice, exponential decay, and applying this function to a Fourier transform results in a sharp resonance peak 31 at 9100 Hz ( Fig. 17 ).
  • the two simulations are each assigned an effective total deviation (in the first replica: ea1 for the first modulation signal, and eb1 for the second modulation signal, in the second replica: ea2 for the first modulation signal, and eb2) given from the sum of the deviations (sum from ea1 and ea2 or from eb1 and eb2 for the second modulation signal) between the instantaneous setpoint values of the simulation and corresponding values regulated in the simulation, which would result from the maintenance of the modulation signal or the changeover to the modulation signal , Then, the total effective deviations with respect to the same modulation signal from both simulations are added (via summing nodes 33 and 34), thus giving the sums e a and e b .
  • the modulation signal is selected for which the sum calculated in the previous step ( e a or e b ) is the lowest.
  • the free parameter of this approximation is the pulse sequence sought, which is generated by the approximation process. It has been shown that it is necessary to make the approximation of the decay process simultaneously by two replicas with two ⁇ / 2 offset phase positions. It It has also been shown that, without altering the result, it is possible to proceed to evanescent systems for the replicas, permitting continuous operation of the pulse modulator.
  • the pulses generated in this way control in the manner indicated, the electrostatic spring (s) of the real oscillator whose resonant frequency can then be set in the desired manner. Parametric effects are effectively suppressed.
  • Fig. 20 shows a complex representation of a possible embodiment 100 of a pulse modulator.
  • the complex input signal x ( t ) comprises a real part and an imaginary part, both of which are represented as digital values.
  • the complex feedback signal 102 is subtracted from the complex input signal x ( t ), the difference of these two complex signals representing the control deviation.
  • the (also complex) content of the delay element 103 is added to this difference in the adder node 101.
  • the content of the delay element 103 is fed via the signal line 104 to the adder node 101.
  • the delay element 103 together with the signal line 104 forms a complex integrator stage, which integrates the complex control deviation, that is to say the difference between the input signal and the feedback signal.
  • the integrated signal 105 is amplified in the amplifier stage 106 in accordance with the factor "a", and the amplified signal 107 is supplied to the first multiplier stage 108. There, the amplified signal 107 with the complex mixed signal e -j ⁇ 0 t multiplied so as to obtain the signal 109 up-converted to the frequency ⁇ 0 .
  • Block 110 determines the real part of the complex up-mixed signal 109, and the thus obtained real part 111 of the up-mixed signal is provided to the quantizer 112.
  • the quantizer 112 is implemented as a ternary quantizer, which converts the respective input signal into the three possible values -1, 0, +1 of a pulse signal with the aid of comparators.
  • the quantized pulse signal y (t) generated in this way can be tapped at the output of the quantizer 112.
  • the complex feedback signal 102 obtained in this way by multiplying a real and a complex number is supplied to the adder node 101 at the input of the circuit.
  • a corresponding modulation signal sequence is generated from the complex compensation signal x ( t ) (pulse signal y ( t ), corresponds to the signals S1 and S2 in FIG Fig. 1 ), which restores the read oscillation of the resonator R or causes excitation of the excitation oscillation of the resonator R.
  • the pulse modulator 100 has the disadvantage that the quantization method used by the pulse modulator 100 is not suitable to be combined with methods for controlling other physical quantities (eg, a method for controlling the resonant frequency of a resonator). These disadvantages can be avoided if instead of in Fig. 20 shown pulse modulator 100 of in Fig. 22 shown pulse modulator 200 is used. This will be discussed in the following description.
  • Modulation state / modulation signal E 1 E 2 force frequency a 0 0 0 ⁇ a b 0 U 0 F 0 ⁇ b c U 0 0 - F 0 ⁇ c d U 0 U 0 0 ⁇ d
  • FIG. 22 A preferred embodiment of a corresponding control unit 200 is shown, which could serve to control the amplitude and phase of the excitation oscillation / read oscillation of the resonator instead of the pulse modulator 100.
  • the control unit 200 has a first summing node 201, a second summing node 202, a third summing node 203, a delay element 204, a switching element 205, a first to fourth error block 206 to 209 and a decision maker 210.
  • Pulse modulator 100 shown A major difference from the in Fig. 20 Pulse modulator 100 shown that instead of the quantizer 112 used there, the decision 210 is used.
  • the input signal x ( t ) is first applied to signal lines 211 to 214, wherein the signal x ( t ) in the summing node 202 the signal e - j ⁇ 0 t is added up and the signal e -j ⁇ from the signal x ( t ) in the summing node 203 0 t is deducted.
  • Correspondingly modified / unchanged signals are fed to the error blocks 206 to 209, which determine the deviation of the signals supplied to them from the instantaneous desired value of the input signal x ( t ) or convert the signals supplied to them so that corresponding deviations can be detected by the decision maker 210.
  • Corresponding error signals output signals of error blocks 206 to 209 are fed to the decider 210 which, by evaluating the error signals, decides which input signal of the error blocks 206 to 209 has the smallest deviation from the current nominal value, and controls the switching element 205 such that the input signal of the corresponding input signal Error block for which the detected deviation is the least, is applied to the input of the delay element 204 (one of the signals applied to the taps 215 to 218).
  • the signal stored in this way in the current clock cycle in the delay element 204 is supplied to the node 201 in the next clock cycle, which adds the signal to the input signal x ( t ).
  • Each of the taps 215 to 218 corresponds to a modulation state / modulation signal a), b), c) and d) given in the above table.
  • the resonator R upon connection of the tap 216 to the input of the delay element 204, the resonator R is supplied with the modulation signal b) (ie, the force F 0 is applied to the control electrodes), when the tap 217 is connected to the input of the delay element 204, the resonator R is supplied with the modulation signal c) (ie, the force -F 0 is applied to the control electrodes), etc.
  • control units 100, 200 are similar to each other.
  • an integrated error between the complex input signal (input signal) x ( t ) and the modulation signal down-converted by - ⁇ 0 is minimized.
  • the error is again mixed up and quantized ternary, wherein the quantizer 112 determines the modulation state. This creates a closed loop that minimizes the integrated error.
  • all possible modulation states a) to d) are tried at a certain point in time and analyzed with regard to the respectively occurring integrated error.
  • the modulation signal (more precisely: the force) Zero, therefore, the integrated error remains unchanged (except for x ( t )), while in states b) and c) the downsampled modulation signal is subtracted / added.
  • the complex integrated error is evaluated by error blocks 206 to 214.
  • the decider 210 thus finds the optimal modulation state a), b), c) or d) and sets the switching element 205 in the appropriate position, so that the integration of the error depending on the selected state of modulation with one of the taps 215 to 218 applied signals can be done.
  • the corresponding modulating signal according to the table is generated by a pulse generating unit (not shown here) driven by the decider 210 and applied to the resonator.
  • the signals can be two-dimensional or complex signals.
  • the error blocks 206 through 214 form the square of the absolute value of their input signal and pass corresponding signals to the decider 210.
  • a method has already been described which provides a control signal for a switchable spring constant, so that a resonator composed of a mass and a switchable spring approximates as closely as possible a resonator having a predetermined resonant frequency.
  • Fig. 19 a control system is shown which can switch between two resonance frequencies.
  • the control unit 300 has a first branch 301 and a second branch 301 '.
  • the first branch 301 includes a delay element 302, first to fourth operators 303 to 306, fifth to eighth operators 307 to 310, first to fourth subtraction nodes 311 to 314, first to fourth error blocks 315 to 318, and a switching element 319 .
  • the second branch 301 has a delay element 302 ', first to fourth operators 303' to 306 ', fifth to eighth operators 307' to 310 ', first to fourth subtraction nodes 311' to 314 ', first to fourth error blocks 315 'to 318', and a switching element 319 '.
  • the output signal of the delay element 302 represents the instantaneous state of the simulated oscillation process of the resonator.
  • the output signal is fed to the inputs of the operators 303 to 306, each operator simulating the influence of one of the four modulating signals that can be generated on the current state of the vibration simulation.
  • the output signals of the operators 303 to 306 are transformed by means of the operators 307 to 310 into a form suitable for error evaluation, the signal e -j ⁇ from each of the transformed signals 0 t subtracted (at the summing nodes 311 to 314) and the signals thus obtained the error blocks 315 to 318 supplied.
  • the error blocks 315 to 318 determine the deviation or a measure of the deviation of the output signals generated by the operators 303 to 306 from the current setpoint value ( e -j ⁇ 0 t ) and pass corresponding deviation signals to adder nodes 321-324.
  • the letters a), b), c) and d) denote the respective modulation state given in the above table, the influence of which on the resonator is to be tested.
  • the mode of operation of the second branch 301 ' corresponds to the mode of operation of the first branch 301.
  • the deviation signals determined in the second branch 301' by the error blocks 315 'to 318' are likewise forwarded to the adder nodes 321 to 324.
  • a deviation signal generated by the first branch 301 is added to a deviation signal generated by the second branch 301 ', wherein in each adder node deviation signals are added of the same modulation signal (modulation state) are added.
  • the added deviation signals are supplied to the decider 320.
  • the decider 320 controls the switching elements 319 and 319 'simultaneously and in such a manner that the one output signal of the operators 303 to 306 or 303' to 306 'is applied to the input of the delay element 302 or 302', whose associated deviation signal is summed with the corresponding error signal of the respective other branch, with respect to amplitude and phase of the least deviation from the current set value (e j ⁇ 0 t ).
  • ⁇ 0 represents the resonance frequency to which the resonator is to be regulated.
  • the resonator R When the tap 331/331 ') is connected to the input of the delay element 302/302', the resonator R is supplied with the modulation signal b) (ie, the force F 0 is applied to the control electrodes), when the tap 332) / 332 is connected ') is applied to the input of the delay element 302/302', the resonator R with the modulation signal c) (ie, the force -F 0 is applied to the control electrodes), etc ..
  • a damped system is preferably simulated.
  • a reference carrier ej ⁇ t is provided, which is the default for the transient in the upper loop.
  • the reference carrier is multiplied by j , resulting in a phase shift of ⁇ / 2.
  • x a , b , c , d , are of the form:
  • a x * cos ⁇ x T sin ⁇ x t / ⁇ x t - ⁇ x t sin ⁇ x t cos ⁇ x t ;
  • M x - 1 - 1 / ⁇ x t 0 0 1
  • a controller 400 is shown in which the control units 200 and 300 are off Fig. 22 and 23 are combined together to form a unit by means of which the resonance frequency ⁇ as well as the amplitude / phase of the excitation oscillation / read oscillation of the resonator can be regulated to specific values at the same time.
  • the deviation signals generated by the error blocks 206 to 209 are added to the summing nodes 401 to 404 to the deviation signals generated by the error blocks 315 to 318.
  • the outputs of summing nodes 401-404 are in turn added to the deviation signals generated by error blocks 315-318. Only deviation signals with respect to the same modulation signal are added, thus obtaining "global" deviation signals for each producible modulation signal.
  • a summary error criterion is applied by adding the "single errors" determined by the control units 200, 300 for the given states a), b), c) and d).
  • the state with the smallest sum error is selected by the decider 420 and determines the current modulation state and the switch position (the switches 205, 319 and 319 'are switched simultaneously and in an identical manner). Keeping the sum error as small as possible causes the physical quantity which contributes the most to the sum error, that is, which has the greatest "control need", to be always given priority.
  • the error blocks 206 through 209, 315 through 318 and 315 'through 318' may be configured differently to weight different deviation signals differently.
  • the individual errors for the regulation of the resonance frequency and for the regulation of the excitation oscillation / read oscillation could be weighted differently, or form the magnitude and phase deviation with respect to the predefined function, and from this an error criterion can be derived.
  • the excitation takes place via the (complex) baseband signal x ( t ) separated according to in-phase and quadrature components by high mixing to the resonance frequency ⁇ .
  • the tuning to this frequency ⁇ can only take place exactly if the parameters of the real oscillator (such as ⁇ a , ⁇ b , ⁇ c , ⁇ d ) are known with sufficient accuracy. If this is not the case, the elements of the matrices can be activated via additional auxiliary control loops A x * and M x - 1 .
  • x a, b, c, d are controlled to the correct values (ie, ⁇ a , ⁇ b , ⁇ c , ⁇ d are determined, and depending thereon, the matrices A x * and M x - 1 .
  • x a , b , c , d formed.
  • a control unit 500 is shown, which is composed of a control unit 200 'and a control unit 300'.
  • the effect of the modulation signal c) on the simulated self-oscillation process is not investigated since the effect is identical to the effect that the modulation signal b) would have on the simulated natural oscillation process.
  • the effect of the modulation signal d) on the oscillation process of the resonator is not investigated, since the effect is identical to the effect that the modulation signal a) would have on the oscillation process of the resonator.
  • the configurations of the adder nodes also fall correspondingly differently.
  • Appendix 1 Conversion of the general to a symmetric state variable form
  • transition matrix A a 11 a 12 a 21 a 22
  • Appendix 2 Exponentiation of a square matrix of 2nd order

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  • Engineering & Computer Science (AREA)
  • Physics & Mathematics (AREA)
  • General Physics & Mathematics (AREA)
  • Radar, Positioning & Navigation (AREA)
  • Remote Sensing (AREA)
  • Microelectronics & Electronic Packaging (AREA)
  • Computer Hardware Design (AREA)
  • Gyroscopes (AREA)
  • Apparatuses For Generation Of Mechanical Vibrations (AREA)
  • Feedback Control In General (AREA)
  • Oscillators With Electromechanical Resonators (AREA)
  • Vibration Prevention Devices (AREA)
  • Measuring Fluid Pressure (AREA)
  • Micromachines (AREA)
EP05811363.0A 2004-11-24 2005-11-21 Verfahren zur steuerung/regelung einer physikalischen grösse eines dynamischen systems Active EP1825218B1 (de)

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DE102004056699A DE102004056699A1 (de) 2004-11-24 2004-11-24 Verfahren zur Steuerung/Regelung einer physikalischen Größe eines dynamischen Systems, insbesondere eines mikromechanischen Sensors
PCT/EP2005/012449 WO2006056389A1 (de) 2004-11-24 2005-11-21 Verfahren zur steuerung/regelung einer physikalischen grösse eines dynamischen systems

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DE102006043412A1 (de) 2006-09-15 2008-03-27 Litef Gmbh Mikroelektromechanischer Sensor sowie Betriebsverfahren für einen mikroelektromechanischen Sensor
JP4576441B2 (ja) * 2008-03-21 2010-11-10 日立オートモティブシステムズ株式会社 角速度センサ
DE102008057281A1 (de) * 2008-11-14 2010-05-20 Northrop Grumman Litef Gmbh Simulationsverfahren für das Betriebsverhalten eines Corioliskreisels
DE102014010671A1 (de) * 2014-05-23 2015-12-17 Gerd Reime Verfahren zur Ermittlung wenigstens eines physikalischen Parameters mittels einer Sensoreinheit
JP6880600B2 (ja) * 2016-08-18 2021-06-02 セイコーエプソン株式会社 回路装置、物理量検出装置、電子機器及び移動体
DE102018202093A1 (de) * 2018-02-12 2019-08-14 Robert Bosch Gmbh Verfahren und Vorrichtung zur Berechnung von Datenmodellen in sicherheitskritischen Systemen
DE102020206003A1 (de) * 2020-05-13 2021-11-18 Robert Bosch Gesellschaft mit beschränkter Haftung Verfahren zum Betreiben eines mikroelektromechanischen Gyroskops, Gyroskop

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US5992233A (en) * 1996-05-31 1999-11-30 The Regents Of The University Of California Micromachined Z-axis vibratory rate gyroscope
DE19635923C1 (de) * 1996-09-04 1998-02-26 Litef Gmbh Verfahren zur Antriebsanregung von Schwingern zur kapazitiven Messung von Kraft, Beschleunigung und/oder Drehraten
FI112298B (fi) * 1996-12-19 2003-11-14 Kone Corp Menetelmä ja laitteisto moottorin värähtelyjen vaimentamiseksi
JPH10262395A (ja) * 1997-03-18 1998-09-29 Toa Medical Electronics Co Ltd ステッピングモータの駆動装置
DE19739903A1 (de) * 1997-09-11 1999-04-01 Bosch Gmbh Robert Sensorvorrichtung
US6360602B1 (en) * 1999-07-29 2002-03-26 Litton Systems, Inc. Method and apparatus reducing output noise in a digitally rebalanced accelerometer
GB0008365D0 (en) * 2000-04-06 2000-05-24 British Aerospace Control syste for a vibrating structure gyroscope
US6718823B2 (en) * 2002-04-30 2004-04-13 Honeywell International Inc. Pulse width modulation drive signal for a MEMS gyroscope
DE10362031B4 (de) 2003-05-08 2008-05-29 Litef Gmbh Betriebsverfahren für einen Corioliskreisel und dafür geeignete Auswerte-/Regelelektronik
US6995622B2 (en) * 2004-01-09 2006-02-07 Robert Bosh Gmbh Frequency and/or phase compensated microelectromechanical oscillator

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AU2005309009A1 (en) 2006-06-01
WO2006056389A9 (de) 2006-07-13
KR20070065449A (ko) 2007-06-22
CN101065641A (zh) 2007-10-31
EP1825218A1 (de) 2007-08-29
CA2599603C (en) 2013-01-08
KR20100020042A (ko) 2010-02-19
CA2599603A1 (en) 2006-06-01
NO20071679L (no) 2007-06-11
RU2007111703A (ru) 2008-12-27
WO2006056389A1 (de) 2006-06-01
NO339405B1 (no) 2016-12-12
AU2005309009B2 (en) 2009-03-19
RU2363929C2 (ru) 2009-08-10
JP4802195B2 (ja) 2011-10-26
JP2008520983A (ja) 2008-06-19
KR101031858B1 (ko) 2011-05-02
US7490015B2 (en) 2009-02-10
US20070286294A1 (en) 2007-12-13
CN101065641B (zh) 2010-10-20

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