EP1249113A1 - Procede et appareil permettant d'effectuer une correction de voie dans une liaison de donnees numeriques - Google Patents

Procede et appareil permettant d'effectuer une correction de voie dans une liaison de donnees numeriques

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Publication number
EP1249113A1
EP1249113A1 EP00969598A EP00969598A EP1249113A1 EP 1249113 A1 EP1249113 A1 EP 1249113A1 EP 00969598 A EP00969598 A EP 00969598A EP 00969598 A EP00969598 A EP 00969598A EP 1249113 A1 EP1249113 A1 EP 1249113A1
Authority
EP
European Patent Office
Prior art keywords
equalizer
value
precoder
ffe
dfe
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP00969598A
Other languages
German (de)
English (en)
Inventor
Heikki Laamanen
Janne Väänänen
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Infinera Oy
Original Assignee
Tellabs Oy
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Tellabs Oy filed Critical Tellabs Oy
Publication of EP1249113A1 publication Critical patent/EP1249113A1/fr
Withdrawn legal-status Critical Current

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03343—Arrangements at the transmitter end
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/03777—Arrangements for removing intersymbol interference characterised by the signalling
    • H04L2025/03802—Signalling on the reverse channel

Definitions

  • the invention relates to a method according to the preamble of claim 1 for implementing channel equalization on a digital communications path.
  • the invention also relates to an apparatus according to claim 17 suited for perform- ing channel equalization on a digital communications path, as well as a transmitter according to claim 22 and a receiver according to claim 23.
  • the bit stream is converted in a transmitter (TX) into an analog signal that is capable of passing through the communications channel.
  • the communications channel may be a radio path, copper wireline or fiber-optic cable.
  • the receiver (RX) performs a recovery of the sent bit stream as error- free as possible.
  • the bit stream reconstruction performed in the receiver is complicated by signal distortion and noise summed with the signal on the communica- tions channel. Due to these side-effects, a portion of the reconstructed bits are erroneous (e.g., on an average, 1 bit per 10 bits may be erroneous).
  • the signal distortion originating from the transmission path is generally compensated for by means of equalizers that are located in the receiver, the transmitter or partially in both of these.
  • the equalizers may be of a fixed or adaptive type. Respectively, the effect of noise is compensated for by means of different coding techniques such as Reed-Solomon coding, convolution coding, trellis coding, turbo coding and others.
  • a generally used correction method of channel distortion is the use of a linear adap- tive equalizer (FFE).
  • FFE linear adap- tive equalizer
  • a linear equalizer alone may give an insufficient correction on certain channels. This kind of situation maybe encountered when the transfer function of the signal band includes zero points, whereby certain frequency components cannot be passed over the communications channel 2.
  • a feedback equalizer is used to compensate for the distortion caused by the spectral nulls of the signal band.
  • the use of a feedback equalizer is often advantageous inasmuch it improves the noise tolerance of the system.
  • the feedback equalizer is located in the receiver, it is called a decision- feedback equalizer (DFE), while an equalizer located in the transmitter is called a Tomlinson-Harashima precoder.
  • DFE decision- feedback equalizer
  • Tomlinson-Harashima precoder an equalizer located in the transmitter
  • a system may also have both a DFE and a TML.
  • the linear equalizer may be situated in the receiver, the transmitter or a portion of the equalizer may be in the transmitter while the other portion is in the receiver.
  • FFE Feedforward equalizer also known as a linear equalizer
  • a digital communications channel is examined in terms of the training phase of its adaptive equalizers.
  • the line code used on the channel may be implemented using either pulse-amplitude modulation (PAM), quadrature-amplitude modulation (QAM) or carrierless amplitude and phase modulation (CAP).
  • PAM pulse-amplitude modulation
  • QAM quadrature-amplitude modulation
  • CAP carrierless amplitude and phase modulation
  • FIG. 1 is shown a model for a system implemented using conventional techniques, wherein the receiver is provided with an adaptive linear equalizer (FFE) and an adaptive decision-feedback equalizer (DFE) (cf. Lee & Messerschmitt).
  • FFE adaptive linear equalizer
  • DFE adaptive decision-feedback equalizer
  • CHN channel noise model
  • the outgoing bit stream is coded into symbols (S) that are sent through the channel 2.
  • the output signal of the channel 2 is processed by equalizers (FFE and DFE), and the decisions on symbols (S') are made from the equalized signal.
  • the decision resulting in the resolved symbol (S') is also called the estimated received symbol.
  • Both adaptive equalizers are adapted to the characteristics of the channel 2 during the training period carried out when a connection is being established.
  • the equalizers are also continually adjusted during the period of data transmission in order to compensate for possible changes in the channel 2.
  • the equalizers are adapted and controlled on the basis of the detection error (e) of the receive signal.
  • FIG. 2 is shown another system according to the prior art (cf. Lee & Messer- schmitt).
  • the receiver has an adaptive linear equalizer (FFE), while the transmitter has a feedback equalizer (of the TML type).
  • FFE adaptive linear equalizer
  • DFE decision-feedback equalizer
  • the tap-weight values of the decision-feedback equalizer (DFE) are transmitted over an upstream auxiliary channel to the transmitter, wherein they are utilized in the configuration of a Tomlinson-Harashima precoder (TML).
  • TTL Tomlinson-Harashima precoder
  • the linear equalizer (FFE) of the receiver is adjusted during the data transmission state, but due to the fixed configuration of the decision-feedback equalizer (TML) of the receiver, the latter equalizer will not be adjusted.
  • a benefit of Tomlinson- Harashima precoding over a DFE is that precoding does not cause feedback of a detection error as is the case in a DFE.
  • a really complex problem evokes from the feedback of erroneous decision-making in the detector.
  • a single erroneous decision may cause loss of connection when in a system using a DFE.
  • the communications channel 2 may include analog bandstop filters serving to eliminate radio-frequency interference, for instance.
  • the positions of the spectral nulls caused by the analog bandstop filters in the frequency spectrum may vary as the component values of the filters change with temperature. This kind of variation in the characteristics of the communications channel 2 cannot be compensated for simply by adjust- ing the linear equalizer.
  • Another complication arises from the incapacity of the system to cope in an optimal manner with varying noise conditions if the decision- feedback equalizer is not adjusted during the data transmission state.
  • the system comprises a linear equalizer (FFE), a Tomlinson-Harashima precoder (TML) and decision-feedback equalizer (DFE).
  • FFE linear equalizer
  • TTL Tomlinson-Harashima precoder
  • DFE decision-feedback equalizer
  • the system comprises the FFE and the DFE alone.
  • TTL precoder
  • the tap coefficient values of the DFE are sent to the precoder (TML) included in the transmitter and the tap coefficient values of the DFE are reset to zero.
  • TTL precoder
  • both the FFE and the DFE are adjusted, but not the precoder (TML).
  • a benefit of this arrangement is that the problems associated with such changes in the communications channel characteristics and noise conditions that cannot be coped with merely by adjusting the linear equalizer are overcome, because also the DFE of the receiver can be adjusted during the data transmission state.
  • a disadvantage still remains from the risk of erroneous decision feedback due to the
  • the tap coefficients of the DFE in the receiver maybe assumed to have smaller values than in the situation illustrated in FIG. 1 inasmuch a portion of the feedback equalization is performed already in the transmitter. Consequently, also the effect of erroneous decision feedback is less severe than in the configuration shown in FIG. 1.
  • the system performance remains substantially dependent on how large changes may occur in the characteristics of the communications channel 2 and system noise condition in regard to the preceding situation prevailed during the training period.
  • a straightforward approach to improve the system shown in FIG. 2 or 3 would be to compute the incremental values of tap coefficient adjustments in the receiver from the detector error and symbol decisions in the same manner as when adjusting a DFE, but then transmitting the computed incremental values of adjustment over an auxiliary channel of the reverse transmit direction to the transmitter. These incremental adjustment values are then used for updating the tap coefficient values of the precoder in the transmitter. Accordingly, the precoder could be adjusted also during the data transmission state, whereby the receiver DFE would become redundant or the high values of its tap coefficients can be limited. However, it can be shown that this kind of equalizer adjustment method is not practicable in a general case.
  • the goal of the invention is achieved by way of adjusting a conventional Tomlinson- Harashima precoder during the data transmission state.
  • the adjustment is implement- ed in a system according to the invention so that the detector input signal of the receiver is transmitted back to the transmitter over an auxiliary channel of the reverse transmit direction and the tap coefficients of the precoder are then adjusted on the basis of the detector error and with the help of the precoder delay line using, e.g., an LMS algorithm.
  • the transmitter according to the invention is characterized by what is stated in the characterizing part of claim 22.
  • the receiver according to the invention is characterized by what is stated in the characterizing part of claim 23.
  • the invention offers significant benefits.
  • the invention allows the precoder to adapt in a continuous manner also during the data transmission state to changes occurring in the noise conditions on the commu- nications channel 2 and also to changes in the properties of analog filters due to temperature variations and to drift caused by other factors.
  • FIG. 1 shows a block diagram of a system of the prior art for implementing channel equalization
  • FIG. 2 shows a block diagram of a second system of the prior art for implementing channel equalization
  • FIG. 3 shows a block diagram of a third system of the prior art for implementing channel equalization
  • FIG. 4 shows a simplified block diagram of a system according to the invention
  • FIG. 5 shows a more detailed block diagram of a system according to the invention and its mathematical model
  • FIG. 6 shows a block diagram of a system according to the invention not having equalizers at its receive end
  • FIG. 7 shows a block diagram of a system according to the invention having merely a linear equalizers at its receive end
  • FIG. 8 shows a block diagram of a system according to the invention having only a decision-feedback equalizer at its receive end
  • FIG. 9 shows a block diagram of a system according to the invention having both a linear equalizer and a decision-feedback equalizer
  • FIG. 10 shows a block diagram of a system according to the invention that has the linear equalizer divided into two separate equalizers in a cascaded configuration and further includes a decision-feedback equalizer;
  • FIG. 11 shows a block diagram of a system according to the invention that has the linear equalizer paralleled with an adaptive filter (that is, a second linear equalizer) and further includes a decision-feedback equalizer.
  • an adaptive filter that is, a second linear equalizer
  • Each one of the equalizers in FIGS. 1-11 maybe adjustable or fixed except for the precoder TML that according to the invention in all cases is adjustable.
  • the invention concerns a method and apparatus suitable for implementing a communications system, wherein a Tomlinson-Harashima precoder is adjusted during the data transmission state, see FIG. 4.
  • a Tomlinson-Harashima precoder is adjusted during the data transmission state, see FIG. 4.
  • FIG. 5 therein is shown a discrete-time model of a system equipped with Tomlinson-Harashima precoding.
  • the result of the modulo operation which is an integral part of the precoding step, is included in the transmitted symbol.
  • Basics on Tomlinson-Harashima precoding can be found, e.g., in cited reference (Lee & Messerschmitt).
  • C(z _1 ) transfer function of communications channel 2 (includes fixed filters, modulation systems, etc.)
  • the precoder output is:
  • the detector input is:
  • index i 1 ...n (number of taps).
  • superindex notation " * " refers to a complex conjugate.
  • the tap coefficients of a Tomlinson- Harashima precoder can be adjusted by the least mean squares algorithm (LMS) using the error difference between the detector input and the transmitted symbol (dk - Sk), and the values contained by the precoder delay line.
  • LMS least mean squares algorithm
  • the formula of the estimated values is replaced by the following control algorithm of tap coefficients:
  • a practical problem in the method according to the invention arises from the requirement of a correct mutual phase between the receiver input signal values (d k ) and the values of the elements (b k-1 , b k - 2 , bk -3 , ...) of the precoder delay line.
  • this detail can be handled by synchronizing the information on the error variable to the precoder delay line content elements with the help of the line frame synchronization information.
  • a functional apparatus needs memory elements for storage of the symbol and delay line content elements until the moment when the information related to their respective error variable has been submitted to the transmitter.
  • the method according to the invention for adjusting the tap coefficients of a Tomlinson-Harashima precoder in the data transmission state is accomplished as follows:
  • the error variable (u k or ⁇ k ) to be used in the adjustment of the precoder is defined as the difference (d k - S k or d k - S' k ) between the signal (d k ) detected at receiver and the sent symbol (S) or, respectively, the estimated symbol (S').
  • the information (d k or ⁇ k ) used in the determination of the error variable value is transmitted from the receiver to the transmitter over an auxiliary channel of the reverse transmit direction.
  • FIGS. 6, 7, 8, 9, 10 and 11 Systems according to the invention using the precoder adjustment based on the control scheme of the invention applied during the data transmission state is illustrated in FIGS. 6, 7, 8, 9, 10 and 11.
  • the systems shown therein comprise a Tomlinson- Harashima precoder (TML) that is adjusted by means of the method according to the invention.
  • TTL Tomlinson- Harashima precoder
  • the computation of the error variable value (subtraction operation) needed for the adjustment of the precoder can be made in either the transmitter or the receiver.
  • the system illustrated in FIG. 6 is functional only on such channels that do not cause precursor intersymbol interference (precursor ISI), that is, an interaction between successive symbols prior to the instant of decision-making.
  • precoder is adjusted based on the error (ei) that represents the situation preceding detection.
  • the system illustrated in FIG. 6 is functional only on such channels that do not cause precursor intersymbol interference (precursor ISI), that is, an interaction between successive symbols prior to the instant of decision-making.
  • precursor ISI precursor intersymbol interference
  • the system shown in FIG. 7 has both a precoder and a linear equalizer in the receiver.
  • the precoder and the linear equalizer are adjusted based on the error (ei) corresponding to the situation preceding detection.
  • the system shown in FIG. 8 has a precoder and a decision-feedback equalizer (DFE) in the receiver.
  • the precoder is adjusted based on the error (ei) corresponding to the situation before the receive signal is corrected by the effect of the DFE.
  • the DFE is adjusted in a conventional manner based on the value of the error e 2 .
  • the system shown in FIG. 8 is functional only on such channels that do not cause precursor ISI.
  • the stop band generated by the linear equalizer manages to eliminate the interference, it also causes in the data signal a distortion that must be compensated for by adapting the decision-feedback equalizer to the new situation.
  • the adjustment rate of the decision-feedback equalizer must be in the same order with the adjustment rate of the linear equalizer.
  • FIGS. 9, 10 and 11 Embodiments according to the invention serving to solve the above-described problems are illustrated in FIGS. 9, 10 and 11.
  • the system shown in FIG. 9 has a precoder and a linear equalizer (FFE) in the receiver, complemented with a decision-feedback equalizer (DFE).
  • the precoder is adjusted based on the value of the error (ej) in the signal immediately after the linear equalizer .
  • the linear equalizer and the DFE are adjusted in a conventional manner based on the value of the error e 2 .
  • a situation is discussed wherein fast adaptation of equalization is needed and adaptation of the linear equalizer alone is insufficient.
  • the linear equalizer and the DFE have a fast response and hence are quickly adapted to the changed situation.
  • the precoder adapts at its inherent adaptation rate so as to make the error ei smaller.
  • the linear equalizer and the DFE detect the adaptation of the precoder in the same manner as any slow change in the communications channel 2 and adjust themselves accordingly.
  • the effect of post cursor ISI is eliminated from the output signal of the linear equalizer as soon as the precoder has reached a new steady state.
  • the tap coefficients of the DFE should be (almost) zero and, in a practical situation, the tap coefficients have been adjusted to zero or a low value during the adaptation period of the precoder.
  • the method according to the invention makes it possible to configure a system wherein the DFE of the receiver can respond to needs of fast adaptation, but still the compensation settings of the DFE for channel distortion can be reflected to the transmitter at the speed determined by the adjustment rate of the precoder.
  • This arrangement limits the increase of tap coefficients in the DFE and thus reduces the feedback of receive error.
  • the DFE is involved with the adjustment process of the precoder, because the DFE affects the error e 2 on the basis of which the linear equalizer is adjusted that in turn affects the error ei . Resultingly, the system stability is deteriorated at certain mutual adjustment rates of the precoder, the linear equalizer and the DFE. Hence, the system illustrated in FIG. 9 requires careful design in respect to these adjustment rates.
  • a situation, wherein the DFE does not affect the adjustment of the precoder, can be accomplished by using systems illustrated in FIGS. 10 and 11.
  • the linear equalizer is divided in two cascaded separate linear equalizers (FFE1 and FFE2).
  • the precoder and the first linear equalizer (FFE1) are adjusted based on the error (ej) measurable at the output of the first linear equalizer.
  • the second linear equalizer (FFE2) and the DFE are adjusted in a conventional manner based on error e 2 .
  • the precoder seeks determined by its inherent adjustment rate toward a situation wherein no feedback equalization is needed in the receiver. This arrangement avoids the use of high tap coefficient values in the DFE.
  • sampling rates of the cascaded adaptive filters FFE1 and FFE2 can be equal or different. Furthermore, the sampling rates can be equal to the symbol rate, multiples thereof or rational number multiples thereof. According to a preferred embodiment, the sampling rate of FFE 1 is a multiple of the symbol rate (fractional spaced equal- izer) and the sampling rate of FFE2 is equal to the symbol rate. According to another preferred embodiment, the sampling rates of both filters FFE1 and FFE2 can be equal to or multiples of the symbol rate.
  • the linear equalizer (FFE) is paralleled by another adaptive FIR filter (AFIR) that also is a linear equalizer.
  • AFIR adaptive FIR filter
  • the precoder and the adaptive filter (AFIR) are adjusted based on the error (ei) measurable at the output of the adaptive filter.
  • the linear equalizer (FFE) and the DFE are adjusted in a conventional manner based on error e 2 .
  • the precoder seeks determined by its inherent adjustment rate toward a situation wherein no feedback equalization is need- ed in the receiver, thus avoiding high tap coefficient values in the DFE.
  • An advantage of this arrangement is that the adaptive filter AFIR needs adjustment and coefficient computation only when the precoder is being adjusted. Hence, the AFIR filter can be implemented computationally without the need for complicated ASIC design.
  • the linear equalizer and/or the decision-feedback equalizer (DFE) of the receiver may in certain cases be such that needs no adjustment during the data transmission state.
  • the linear equalizer and/or the DFE may comprise an entirely fixed filter configuration, whereby this filter is not adjusted even during the training period.
  • the replacement of an adaptive equalizer by an entirely fixed configuration or a configuration which is not adjustable during the training period compromises the system capability of adapting to changes in the communications channel parameters.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)

Abstract

L"invention concerne un procédé et un appareil permettant d"effectuer une égalisation de voie dans une voie de communication numérique. Selon ce procédé, un train de bits sortant est codé en symboles ; la distorsion de voie est compensée à l"aide d"un précodage de symboles (TML), les symboles précodés sont envoyés via une voie de communication (2, CHN) ; des moyens permettent de récupérer les symboles passés par la voie de communication (2, CHN) et le moyen de traitement des signaux du récepteur les transforme en un train de bits. Pendant l"état de transmission de données, le précodage (TML) est réglé sur la base des valeurs de contenu de la ligne de retard de précodeur et de la valeur d"une telle/de telles variable(s) d"erreur dépendant de la différence entre le signal mesurable au niveau du récepteur et la valeur du symbole envoyée (S) par l"émetteur ou évaluée (S") par le récepteur. Dans ce récepteur, le signal est mesuré en un point où cette différence atteint la valeur minimale de sa valeur absolue lorsque le réglage est un état stable.
EP00969598A 2000-01-18 2000-10-16 Procede et appareil permettant d'effectuer une correction de voie dans une liaison de donnees numeriques Withdrawn EP1249113A1 (fr)

Applications Claiming Priority (3)

Application Number Priority Date Filing Date Title
FI20000100A FI107307B (fi) 2000-01-18 2000-01-18 Menetelmä ja laitteisto kanavakorjauksen tekemiseksi digitaalisella tietoliikenneyhteydellä
FI20000100 2000-01-18
PCT/FI2000/000897 WO2001054368A1 (fr) 2000-01-18 2000-10-16 Procede et appareil permettant d"effectuer une correction de voie dans une liaison de donnees numeriques

Publications (1)

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EP1249113A1 true EP1249113A1 (fr) 2002-10-16

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EP00969598A Withdrawn EP1249113A1 (fr) 2000-01-18 2000-10-16 Procede et appareil permettant d'effectuer une correction de voie dans une liaison de donnees numeriques

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US (1) US7512191B2 (fr)
EP (1) EP1249113A1 (fr)
CN (1) CN1193560C (fr)
AU (1) AU2000279270A1 (fr)
FI (1) FI107307B (fr)
MX (1) MXPA02006996A (fr)
WO (1) WO2001054368A1 (fr)

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US20030035495A1 (en) 2003-02-20
FI20000100A0 (fi) 2000-01-18
MXPA02006996A (es) 2002-12-13
CN1425239A (zh) 2003-06-18
FI107307B (fi) 2001-06-29
US7512191B2 (en) 2009-03-31
CN1193560C (zh) 2005-03-16
WO2001054368A1 (fr) 2001-07-26
AU2000279270A1 (en) 2001-07-31

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