EP0542595A1 - Microstrip antenna device especially for satellite telephone transmissions - Google Patents

Abstract

An antenna comprises a first dielectric layer (D1) including on one side an earth-plane (PM1), and on the other a first conductive slab (P1) of chosen shape. A second dielectric layer (D2) surmounts the first on the first slab side, and supports, on the other side, facing the first slab, a second conductive slab (P2) of chosen shape. A third dielectric layer (DR) surmounts the second. The second slab (P2) is of smaller size than that of the first slab (P1), and this first slab (P1) is fed from the bottom, at at least one chosen point (FR1) situated between its centre and its periphery. Advantageously, the first slab (P1) is connected to a run-through (TR1) of the first-plane joining up with a feed circuit (DL1, DL2) implanted in the dielectric substrate of three-plate structure (SDH, SDB). <IMAGE>

Description

The invention relates to microstrip or "microstrip" antenna devices.

Many antenna structures have already been described in this area. The simplest microstrip radiating structure comprises a dielectric layer, carrying on one side a conductive pad of selected shape, and on the other a conductive plane which is called ground plane. To obtain an antenna, it is necessary to define the mode of supply of this structure with microwave energy.

The idea of providing a stack of overlapping tiles was described in the article by LONG & WALTON, "A dual-frequency stacked circular disk antenna", IEEE Transactions on Antenna and Propagation, Vol. AF 27, No. 2, March 1979. Other proposals have since been made.

With regard to the supply of antennas with two superimposed blocks, two very different cases must be distinguished from the functional point of view, depending on whether the supply takes place at the level of the upper block or the lower block (the closest to ground plan).

In the case where the supply is made at the level of the lower pad, it is most often a connection at the periphery of this pad. In addition, an upper block larger than the lower block is systematically provided (see in particular the article by TULINTSEFF, ALI, and KONG, "Input impedance of a probe-fed stacked circular microstrip antenna", IEEE Transactions on Antennas and Propagation, Vol. 39, No. 3, March 1991).

Those skilled in the art know that the development of antennas with overlapping pavers is particularly delicate. Attempts have been made to model its properties. We will quote as for example the article by COCK and CHRISTODOULOU, "Design of a two-layer, capacitively coupled, microstrip patch antenna element for broad band applications", IEEE Symposium on antenna propagation, 1987. Despite these attempts, it remains extremely difficult to predict by modeling and understanding the behavior of microstrip structures comprising two or more overlapping pavers.

The Applicant has in particular posed the problem of producing an antenna conforming to electronic scanning, intended for the communication system with mobiles such as aircraft (so-called SATCOM system).

This system is intended to work with the group of geostationary satellites managed by the INMARSAT organization. With regard to at least aircraft applications, the telecommunications service offered is governed by an international standard called ARINC 741.

Technically, it is a question of developing an antenna capable of operating on the one hand in transmission, on the other hand in reception, in two very neighboring bands, namely a little more than 1.5 gigahertz for reception, and just over 1.6 gigahertz for the broadcast.

The electronic scanning function is necessary for this antenna, due to the movement of the carrier mobile, which is assumed here to be an aircraft. It is also necessary to choose between a roof antenna, or two side antennas. In the case of two lateral antennas, the aforementioned ARINC standard has defined two acceptable official footprints, delimiting the volume in which the planned antenna must be inscribed.

The antenna must also be conformal, that is to say capable of adapting to the exact wall shape of the carrier mobile. It must also be not very thick, in order to minimize aerodynamic drag, and of course designed to respect the mechanical characteristics required for the structure of the aircraft.

During the research which it carried out, the Applicant found that it was possible to design a microstrip antenna going practically opposite to the solutions accepted until now by those skilled in the art.

The present invention therefore provides an antenna element fundamentally different from those known up to now.

This antenna element is of the type comprising a first dielectric layer comprising on one side a ground plane, and on the other a first conductive pad of selected shape, a second dielectric layer, which surmounts the first layer, on the side of the first block, and supports on the other side, opposite the first block, a second conductive block of selected shape, a third dielectric layer surmounting the second, as well as microwave power supply means of one of the conductive blocks.

According to the invention, the second block is smaller in size than that of the first block, and this first block is fed from below, at at least one chosen point, located between its center and its periphery.

With this structure, it appeared possible to build an operational antenna, subject to choosing the position of the point in question, according to the respective sizes of the first and second blocks, the dielectric characteristics of the first and second dielectric layers, as well as those of the third dielectric layer, which preferably has dielectric constants significantly greater than those of the other two.

According to another aspect of the invention, the first block is connected to a crossing of the ground plane joining a supply circuit implanted in a dielectric substrate of triplate type structure. More particularly, the triplate structure comprises a substrate layer implanted between the ground plane already mentioned and a low ground plane; between the two ground planes are provided conductive crossings defining a peripheral shielding of the feed part of the antenna element. Preferably, a Wilkinson divider is provided capable of supplying the lower block at two points forming with its center a substantially isosceles right triangle, while the respective signals brought to these two points are in quadrature. The Wilkinson divider is located at an intermediate level of the substrate layer, in accordance with the three-ply structure. This intermediate level serves in practice as a level of distribution of the power between a central connector for the whole of the antenna, and the different antenna elements which will constitute it, in the application as an array antenna.

In an advantageous embodiment, the two blocks are of generally circular shape, and these two blocks are substantially coaxial, that is to say that they are located on the same perpendicular to the planes of the dielectric layers.

• la figure 1 est un schéma de principe général d'un élément d'antenne, en perspective éclatée ;
• la figure 2 est une vue en coupe partielle fragmentée d'un élément d'antenne ;
• la figure 3 est une vue partielle détaillée (superposée) du branchement du pavé inférieur à son alimentation par diviseur de Wilkinson ;
• la figure 4 est une vue de dessous des 24 diviseurs de Wilkinson, pour une antenne à 24 éléments, interconnectés à un connecteur central ;
• la figure 5 est une vue de dessus de 24 pavés inférieurs, correspondant précisément à la figure 4 ; et
• la figure 6 est un diagramme représentant le coefficient de réflexion de l'antenne en fonction de la fréquence.

Other characteristics and advantages of the invention will appear on examining the detailed description below, and the appended drawings, in which:

• Figure 1 is a general block diagram of an antenna element, in exploded perspective;
• Figure 2 is a fragmentary partial sectional view of an antenna element;
• Figure 3 is a detailed partial view (superimposed) of the connection of the lower block to its supply by Wilkinson divider;
• Figure 4 is a bottom view of the 24 Wilkinson dividers, for a 24-element antenna, interconnected to a central connector;
• FIG. 5 is a top view of 24 lower blocks, corresponding precisely to FIG. 4; and
• FIG. 6 is a diagram representing the reflection coefficient of the antenna as a function of the frequency.

Those skilled in the art know that shape is important in microstrip devices. Furthermore, the drawings are, for the most part, certain. They can therefore be incorporated into the description, not only to make it easier to understand, but also to contribute to the definition of the invention, if necessary.

In FIGS. 1 and 2, the reference PM0 designates a lower ground plane, which can be assembled using an insulating adhesive on a sheet of metal to be incorporated into the wall of the aircraft. This lower ground plane is surmounted by two dielectric layers SDB and SDH (respectively low and high). The SDH layer is in turn surmounted by another PM1 ground plane. The whole forms a three-ply structure, with appropriate metallizations etched between the SDB and SDH layers, or more exactly on one of these layers.

Basically, these metallizations include a supply line L, which is then subdivided in the manner of a Wilkinson divider, shown diagrammatically in FIG. 1, but better visible in FIGS. 3 and 4. This divider comprises two branches DL1 and DL2 which first move away from each other, to meet at a level where they are connected to a resistor RLL implanted in the thickness of the SDB layer, but without joining the lower ground plane PM0. Then, the two branches DL1 and DL2 move apart again to join respective supply points EL1 and EL2.

These points EL1 and EL2 are connected by crossings TR1 and TR2 (not connected to the ground plane PM1) to supply points FR1 and FR2 provided on the lower block P1, or pilot block, engraved on the upper face of a dielectric layer D1 placed above the ground plane PM1.

As can be seen in FIGS. 3 and 4, the end portions of the etchings DL1 and DL2 are of different lengths, so that, electromagnetically, the signals available at the points FR1 and FR2 are substantially in quadrature with each other. Correlatively, the feed points FR1 and FR2 of the block P1 are substantially located at right angles to each other.

These two points are located at distances d1 and d2 from the center of block P1 which are in principle equal. We will come back later on the choice of these distances. However, it is possible to indicate immediately that these distances d1 and d2 are in principle between 50% and 100% of the radius of the block P1 (denoted DP1 / 2 in FIG. 3).

Above block P1 is provided a second dielectric layer D2, with the same dielectric constant as layer D1, but of greater thickness, as visible in FIG. 2. In the upper part, layer D2 receives by etching a second block conductor P2 (coupled block), which is in principle circular and coaxial with block P1, but has a smaller diameter than that of block P1.

The antenna element is completed with an additional dielectric layer DR, forming a radome, and in principle having a dielectric constant significantly greater than that of the layers D1 and D2.

In FIGS. 2 and 4, it further appears that the line L continues until a passage through a metallized hole, towards a general microwave connector CCH, of the coaxial type, located behind the metal sheet underlying the lower ground plane PM0.

Furthermore, comparing Figures 2 and 4, it appears that this connector is provided for each pad with a peripheral shield in the form of a horseshoe, passing through the entire dielectric layer SDB. This shielding could be defined by a layer conductive continues. The Applicant has found that it is sufficient to provide a certain number of through pads, surrounding the location of the CCH crossing, with a spacing between these pads which remains sufficiently less than the wavelength of the processed microwave signals.

Similarly, the peripheral pads such as BP11, BP12 and BP13 define a shielding of the supply of the antenna element considered, with respect to the neighboring antenna elements, and with respect to the outside.

On the other hand, it will be noted that above the ground plane PM1, there is no provision for isolation of the antenna element from its neighbors.

FIG. 5 shows how 24 antenna elements can be arranged to form a conforming electronic scanning antenna, satisfying the conditions of the problem posed. As already indicated, these antenna elements are connected to a general connector, with 24 pins (at least). Upstream of this connector, an individual reciprocal phase shift treatment is provided for each antenna element, using DPH phase shifters, as shown diagrammatically in FIG. 2.

• la hauteur et la constante diélectrique des trois couches DR, D2 et D1;
• les diamètres des pavés P1 et P2; et
• le rayon d = d1 = d2 des deux points d'alimentation du pavé inférieur P1.

The main parameters involved in such an antenna are:

• the height and the dielectric constant of the three layers DR, D2 and D1;
• the diameters of the blocks P1 and P2; and
• the radius d = d1 = d2 of the two supply points of the lower block P1.

• a) un comportement bi-fréquences comportant une très bonne adaptation (meilleure que -20 décibels), sur deux fréquences F1 et F2;
• b) un comportement large bande, assurant au moins une adaptation de -10 décibels entre des fréquences F3 et F4 contenant l'intervalle de fréquence F1 et F2.

The problem posed, in the particular application targeted, is to obtain a dual behavior from the unitary antenna element (Figure 6):

• a) dual-frequency behavior with very good adaptation (better than -20 decibels), on two frequencies F1 and F2;
• b) broadband behavior, ensuring at least an adaptation of -10 decibels between frequencies F3 and F4 containing the frequency interval F1 and F2.

The Applicant has observed that, provided that the frequencies F1 and F2 are not too far apart from each other, and as soon as the parameters of heights and dielectric constants of the three aforementioned layers are fixed, there is practically only one solution. , in terms of the radii of the two blocks and the supply radius of the block P1, which makes it possible to satisfy the conditions which have just been recalled.

Any modification of one of the parameters makes it very difficult to find a situation likely to satisfy said conditions.

Although the phenomena in question are not yet fully understood, it seems that, in the general case, everything happens as if only one of the two blocks P1 and P2 resonates at the working frequency. On the other hand, there is a very small area, in the antenna definition parameters, for which the two blocks interact, while exhibiting typically dual-frequency behavior, as desired. And it is still necessary to seek the optimal point of this dual-frequency behavior, to meet the operating conditions desired for the antenna, as recalled above.

In particular, it has been found that it is practically very difficult to operate the antenna element, without adding an upper layer of DR radome to it.

• épaisseur de la couche DR : 1,5 à 2,5 mm;
• constante diélectrique relative de la couche DR : de 4 à 5 ; - épaisseur de la couche D2 : environ 4,8 mm;
• épaisseur de la couche D1 : environ 1,6 mm;
• constantes diélectriques relatives des couches D1 et D2 ainsi que SDB et SDH : environ 2 ;
• diamètre du pavé P1 : environ 70 mm;
• diamètre du pavé P2 : environ 60 mm;
• rayon des points d'alimentation FR1 et FR2 : entre 0,5 et 0,7 fois le rayon du pavé P1.

The Applicant has thus been able to produce antennas meeting the following parameters:

• thickness of the DR layer: 1.5 to 2.5 mm;
• relative dielectric constant of the DR layer: from 4 to 5; - thickness of the layer D2: approximately 4.8 mm;
• thickness of layer D1: approximately 1.6 mm;
• relative dielectric constants of layers D1 and D2 as well as SDB and SDH: approximately 2;
• diameter of paver P1: approximately 70 mm;
• diameter of the block P2: approximately 60 mm;
• radius of feed points FR1 and FR2: between 0.5 and 0.7 times the radius of block P1.

• coefficient de réflexion meilleur que -20 dB à la fréquence centrale de réception (1,545 GHz) ;
• coefficient de réflexion meilleur que -20 dB à la fréquence centrale d'émission (1,645 GHz) ;
• comportement passe-bande à mieux que -10 dB entre 1,53 et 1,66 GHz.

Such antennas can satisfy the conditions set for the SATCOM working band, namely:

• reflection coefficient better than -20 dB at the central reception frequency (1.545 GHz);
• reflection coefficient better than -20 dB at the central emission frequency (1.645 GHz);
• bandpass behavior at better than -10 dB between 1.53 and 1.66 GHz.

We will now focus on the constitution of the array of antenna elements as illustrated in Figures 4 and 5.

First of all, it was indicated above that each lower block is supplied at two points located on two rays substantially perpendicular to each other.

It appeared advantageous to distribute the two feed points properly, and this in a different way for the 24 antenna elements illustrated. The Applicant has found that this makes it possible to reduce the rate of ellipticity of the antenna, taking into account the fact that the latter operates in circular polarization, and with electronic scanning. To this end, it is possible either to distribute the feeding points substantially at random, or to experimentally search for an optimal configuration from the point of view of this ellipticity rate (for example as in FIG. 5).

The electronic scanning array antenna thus obtained has proved capable of operating for depointing angles up to 60 °, with sufficiently low side lobe levels, and a gain of at least 12 decibels compared to an antenna. isotropic.

A good compromise between the loss of gain and the level of the secondary lobes has been obtained by applying a law of lighting which is weakly weighted in amplitude. It can be a Taylor law, of circular type 20 decibels, these indications being understandable for the person skilled in the art.

The phase shifters associated with each of the antenna elements can be integrated into the beam orientation unit (or BSU for "Beam Steering Unit"), housed inside the aircraft.

Advantageously, phase shifters with lines switched by PIN diodes are used, controlled by 4-bit binary words, which gives a resolution of 22.5 °.

The distributor, integrated into the phase shift block, provides amplitude weighting according to the aforementioned law.

In the particular application targeted, the antenna must operate simultaneously in transmission and in reception, at relatively similar frequencies. With regard to the calibration of the electronic scanning phase shifters, the network should be phased out or "phased" over a band of approximately 8%.

Rather than calculating the phase law at the mid-frequency of the band, the Applicant has found that it is preferable to take into account the use of two distinct frequency bands, as well as the quantization and the nature of the phase shifters. (switched lines). To this end, it uses the calibration method described below.

Let be an element Ai of a conforming antenna, therefore not planar, with coordinates (in the center) Xi, Yi, Zi. When we want to spot the main beam in the direction U, V at the frequency f, it is necessary to apply to this antenna element Ai a theoretical phase shift DPi, which is a function of f, U and V which the man of the art: $$\text{DPi (f, U, V)}$$

In practice, we use a calibration table TC (n, F) where n is an integer (or other discrete variable) representing the required state of the phase shifter, with 0 <= n <= N, while we also limit ourselves at discrete values of the frequency F. This is written: $$\text{DQi (F, n)}$$

In the example, we take 101 frequency points in the band 1.53 - 1.66 GHz; and N = 15, with n defined on 4 bits. This process only "phases" the network correctly for one frequency. However, the antenna has essentially dual-frequency behavior.

où ¦ désigne la valeur absolue.The Applicant then established a "distance" between the theoretical phase and the tabulated phase, for the two frequencies f1 and f2, in particular of the form:

where ¦ denotes the absolute value.

The calibration then consists in seeking a priori, for each direction of sight and each antenna element, the value of n which minimizes this function DDi.
The phase shifters are controlled accordingly. Of course, this calibration can be stored.

The present invention is not necessarily limited to the embodiment described, nor to the intended application. The antenna element can itself be used for other applications, provided that the new structure is preserved. Consider also the combination of a microstrip element and a tri-plate power supply, in the same dielectric stack.

The polarization may be other than the circular polarization of the embodiment described.

Another feature of the invention is that it can avoid, for the layers D1 and D2, the use of dielectrics of low constant, or porous, or even consisting of a gas.

Claims (12)

Antenna device, of the type comprising a first dielectric layer (D1) comprising on one side a ground plane (PM1), and on the other a first conductive pad (P1) of selected shape, a second dielectric layer (D2 ), which overcomes the first layer, on the side of the first block, and supports on the other side, opposite the first block, a second conductive block (P2) of chosen shape, a third dielectric layer (DR) overcoming the second, as well as microwave power supply means for one of the conducting blocks, characterized, in combination, in that the second block (P2) is smaller in size than that of the first block (P1), and in that the first block (P1) is fed from below, at least one chosen point (FR1), located between its center and its periphery. Device according to claim 1, characterized in that the first block (P1) is connected to a crossing (TR1) of the ground plane joining a supply circuit (DL1, DL2) implanted in a dielectric substrate of tri-plate structure ( SDH, SDB). Device according to claim 2, characterized in that the two blocks (P1, P2) are of generally circular shape. Device according to claim 3, characterized in that the two blocks (P1, P2) are substantially coaxial. Device according to one of Claims 2 to 4, characterized in that the dielectric materials of the first and second layers (D1, D2) and of the substrate (SDH, SDB) have a dielectric constant of the order of 2, and in that that the thickness ratio of the second (D2) and first (D1) layers is of the order of 3. Device according to claim 5, characterized in that the dielectric material of the third layer (DR) has a dielectric constant of the order of 4. Device according to one of claims 2 to 6, characterized in that the triplate structure comprises a substrate layer (SDH, SDB) implanted between the ground plane (PM1) and a low ground plane (PMB), with crossings conductive (BP11-BP13) defining a peripheral shielding, and a Wilkinson divider (DL1, DL2, RLL), located at an intermediate level of the substrate layer, and suitable for supplying the lower block at two points (FR1, FR2) forming with its center a substantially isosceles right triangle. Device according to claim 7, characterized in that the dielectric material of the substrate layer (SDH, SDB) has substantially the same dielectric constant as that of the first and second layers (D1, D2). Device according to one of the preceding claims, characterized in that it comprises a network of first (P1) and second (P2) pavers respectively located on the same first and second dielectric layers. Device according to claim 9, characterized in that it is associated with controlled phase shifters (DPH) giving it an electronic scanning function. Device according to claim 10, taken in combination with one of claims 7 and 8, characterized in that the pairs of feed points (FR1, FR2) of the lower blocks are distributed to improve the ellipticity rate of the antenna, with strong depointings. Device according to one of claims 10 and 11, characterized in that the phase shifters (DPH) are calibrated on the basis of a distance function between theoretical and actual values, for both of the two central frequencies of the antenna.

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