WO2024066511A1 - 故障穿越方法及变换器 - Google Patents

故障穿越方法及变换器 Download PDF

Info

Publication number
WO2024066511A1
WO2024066511A1 PCT/CN2023/101717 CN2023101717W WO2024066511A1 WO 2024066511 A1 WO2024066511 A1 WO 2024066511A1 CN 2023101717 W CN2023101717 W CN 2023101717W WO 2024066511 A1 WO2024066511 A1 WO 2024066511A1
Authority
WO
WIPO (PCT)
Prior art keywords
vector
current
filter inductor
value
converter
Prior art date
Application number
PCT/CN2023/101717
Other languages
English (en)
French (fr)
Inventor
张美清
董明轩
周玖洋
辛凯
Original Assignee
华为数字能源技术有限公司
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 华为数字能源技术有限公司 filed Critical 华为数字能源技术有限公司
Publication of WO2024066511A1 publication Critical patent/WO2024066511A1/zh

Links

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H02M1/325—Means for protecting converters other than automatic disconnection with means for allowing continuous operation despite a fault, i.e. fault tolerant converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/493—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode the static converters being arranged for operation in parallel

Definitions

  • the present application relates to the field of power electronics technology, and in particular to a fault ride-through method and a converter using the fault ride-through method.
  • Power sources such as wind power, photovoltaic power, and energy storage are gradually becoming the main power sources for the power grid.
  • the above power sources are connected to the power grid through converters (or converters).
  • the traditional grid-following control method of the grid-connected converter makes the new energy present the characteristics of a current source. It cannot provide frequency and voltage support for the power grid like traditional synchronous generators, nor can it operate under off-grid conditions, which makes the stability and power supply reliability of the power system with a high proportion of new energy sources extremely challenging.
  • Grid-forming control is a control method that can make the power electronic converter have the external characteristics of a synchronous generator, which can provide the necessary voltage and frequency support for the power grid, can operate stably under grid-connected or off-grid conditions, and has broad application prospects in the power system.
  • the converter uses a grid-forming control method with grid-forming capability to establish a stable AC voltage and frequency for the grid.
  • the power electronic devices have poor overcurrent capability and are not capable of providing a large short-circuit current for a short period of time during a grid short-circuit fault.
  • the converter will automatically generate a large short-circuit current according to the grid fault situation, causing damage to the power electronic devices.
  • one fault ride-through method applied to converters is to add hardware equipment to the power grid to provide current to the power grid in the event of a fault. This method is not conducive to cost control due to the addition of hardware equipment.
  • Another fault ride-through method is to switch to a grid-following current source control strategy during the fault. During the fault, the converter loses the ability to form a network, which is not conducive to grid stability.
  • Another fault ride-through method is to use a combination of virtual impedance and current limiting, but this method is difficult to accurately limit the current amplitude, mainly including that the virtual impedance is too large, the current amplitude is small, and the current capacity is not fully utilized to support the power grid; the virtual impedance is too small, the voltage loop is saturated, and the voltage recovery speed is slow after the fault is removed. And the virtual impedance is different according to different fault conditions and grid operating conditions, and it is difficult to determine the parameters. In addition, this method is prone to reduce system stability.
  • the present application provides a fault ride-through method, which is applied to a converter, wherein the converter includes a conversion circuit and a filter inductor, wherein an input end of the conversion circuit is used to connect to a DC source, a first end of the filter inductor is used to connect to an output end of the conversion circuit, and a second end of the filter inductor is used to connect to a load;
  • the fault ride-through method is executed by the converter, and the fault ride-through method includes: detecting the internal potential vector of the first end of the filter inductor and the terminal voltage vector of the second end; when the amplitude of the terminal voltage vector at the second end of the filter inductor decreases to less than or equal to a preset terminal voltage extreme value, reducing the internal potential vector of the first end of the filter inductor to reduce the amplitude of the filter inductor current vector on the filter inductor to less than or equal to a preset current limit value, wherein the internal potential vector, the terminal voltage vector and the filter induct
  • the above fault ride-through method determines that the voltage outputted from the output terminal (i.e., the second terminal) of the conversion device is abnormal (low voltage) when the amplitude of the terminal voltage vector is detected to be less than the preset terminal voltage extreme value.
  • the amplitude of the internal potential vector is reduced, and the amplitude of the filter inductor voltage drop vector can be constrained, so that the amplitude of the filter inductor current vector can be constrained within the preset current limit value (less than or equal to the current limit value).
  • the current limiting value is less than or equal to an upper limit value that can be tolerated by the hardware capability of the converter or allowed for normal operation.
  • the amplitude of the terminal voltage vector at the second end of the filter inductor is less than the preset terminal voltage extreme value and the filter inductor current vector is greater than the upper limit value
  • the internal potential vector of the first end of the filter inductor is controlled by the converter to decrease so that the filter inductor current vector is reduced to less than or equal to the upper limit value.
  • the amplitude of the filter inductor voltage drop vector can be effectively suppressed by reducing the internal potential vector, thereby controlling the amplitude of the filter inductor current vector to be within the current limit value.
  • the upper limit value is greater than a rated current value of the converter.
  • a current vector feedback value of the second end of the filter inductor or a current vector reference value of the converter is used as the amplitude of the filter inductor current vector.
  • the vector value with a larger absolute value between the current vector feedback value and the current vector reference value is used as the amplitude of the filter inductor current vector.
  • the current vector reference value is the current vector reference value of the second end under the voltage loop control
  • the current vector feedback value is the current value of the second end output actually detected.
  • the current vector reference value and the current vector feedback value are basically consistent. If the converter is working abnormally, there may be a difference between the current vector reference value and the current vector feedback value. When either the current vector reference value or the current vector feedback value is greater than the current limit value, it indicates that the converter is working abnormally. Therefore, the current vector reference value and the current vector feedback value with a larger absolute value are compared with the current limit value. When the current vector reference value and the current vector feedback value with a larger absolute value are greater than the current limit value, a fault is determined.
  • the fault ride-through method when the filter inductor current vector is greater than the upper limit value, further includes: starting a duration timing; when the duration is greater than a preset duration, switching the current limit value from a first limit value to a second limit value, wherein the first limit value is greater than the second limit value.
  • switching the current limit value from the first limit value to the second limit value includes: switching the current limit value from the first limit value to the second limit value in a step-wise, ramp-wise, or exponential manner.
  • the fault ride-through method further includes: when the duration is less than or equal to the preset duration, maintaining the current limit value unchanged.
  • the duration is the duration that the amplitude of the filter inductor current vector is between the maximum current limit boundary value and the rated current boundary value, and is also equal to the duration that the current limit value is set to the first limit value.
  • a preset duration is also set.
  • the preset duration is set to the limit duration that the amplitude of the filter inductor current vector is between the maximum current limit boundary value and the rated current boundary value. That is, when the duration that the amplitude of the filter inductor current vector at the second end is between the maximum current limit boundary value and the rated current boundary value is greater than the preset duration, the converter is at risk of damage.
  • the fault ride-through method further includes: when the amplitude of the filter inductor current vector is less than or equal to the current amplitude limit value, clearing the duration to zero.
  • reducing the internal potential vector of the first end of the filter inductor includes: generating an internal potential reference correction value based on the difference between the amplitude of the filter inductor current vector and the current limit value; generating an internal potential vector instruction based on the internal potential reference correction value and the internal potential vector reference value of the converter; and reducing the internal potential vector according to the internal potential vector instruction.
  • the converter can maintain its grid-building capability during output voltage drops, ensuring the voltage stability of the output to the load.
  • the fault ride-through method further includes: correcting a current vector reference value according to the internal potential vector instruction and the voltage vector feedback value; setting the smaller absolute value of the corrected current vector reference value and the current limit value as the current vector instruction; and generating a modulation wave according to the current vector instruction, the current vector feedback value and the phase reference value in the converter, wherein the modulation wave is used to modulate the power signal provided by the DC source to provide it to the load.
  • the current vector command is used to determine how much the filter inductor current needs to be corrected.
  • the corrected current vector reference value calculated in the previous step may not be equal to the current limit value.
  • a converter comprising: a conversion circuit, wherein an input end of the conversion circuit is used to connect to a DC source; a filter inductor, wherein a first end of the filter inductor is used to connect to an output end of the conversion circuit, and a second end of the filter inductor is used to connect to a load; a detection circuit, wherein the detection circuit is connected to the first end and the second end, and is used to detect an internal potential vector of the first end of the filter inductor and a terminal voltage vector of the second end; and a controller, wherein the controller is connected to the detection circuit, and is used to reduce the internal potential vector of the first end of the filter inductor when the amplitude of the terminal voltage vector at the second end of the filter inductor is reduced to less than or equal to a preset terminal voltage extreme value, so as to reduce the amplitude of the current vector on the filter inductor to less than or equal to a preset current limit value, wherein the
  • the above converter includes a controller and a detection circuit.
  • the amplitude of the terminal voltage vector is detected to be less than the preset terminal voltage extreme value, it is judged that the voltage outputted from the output terminal (i.e., the second terminal) of the conversion device is abnormal (low voltage).
  • the amplitude of the internal potential vector is reduced, and the amplitude of the filter inductor voltage drop vector can be constrained, so that the amplitude of the filter inductor current vector can be constrained within the preset current limit value (less than or equal to the current limit value).
  • the above converter achieves the above beneficial effects through the above control process without adding or modifying the hardware structure, and is also conducive to cost saving.
  • the current limiting value is less than or equal to an upper limit value that can be tolerated by the hardware capability of the converter or allowed for normal operation.
  • the amplitude of the terminal voltage vector at the second end of the filter inductor is less than the preset terminal voltage extreme value and the filter inductor current vector is greater than the upper limit value
  • the internal potential vector of the first end of the filter inductor is controlled by the converter to decrease so that the filter inductor current vector is reduced to less than or equal to the upper limit value.
  • the internal potential vector can effectively suppress the amplitude of the filter inductor voltage drop vector, thereby controlling the amplitude of the filter inductor current vector to be within the current limit value.
  • the controller is used to: generate an internal potential reference correction value based on the difference between the amplitude of the filter inductor current vector and the current limit value; generate an internal potential vector instruction based on the internal potential reference correction value and the internal potential vector reference value of the converter; and reduce the internal potential vector based on the internal potential vector instruction.
  • the converter can maintain its grid-building capability during output voltage drops, ensuring the voltage stability of the output to the load.
  • the controller is used to: correct the current vector reference value according to the internal potential vector instruction and the voltage vector feedback value; set the smaller absolute value of the corrected current vector reference value and the current limit value as the current vector instruction; and generate a modulation wave according to the current vector instruction, the current vector feedback value and the phase reference value in the converter, and the modulation wave is used to modulate the power signal provided by the DC source to provide it to the load.
  • the current vector command is used to determine how much the filter inductor current needs to be corrected.
  • the corrected current vector reference value calculated in the previous step may not be equal to the current limit value.
  • FIG1 is a schematic diagram of a converter, a DC source, and a load according to a first embodiment of the present application.
  • FIG. 2 is a schematic diagram of the control link of the controller in the converter in FIG. 1 .
  • FIG3 is a flow chart of a fault ride-through method according to the first embodiment of the present application.
  • FIG. 4 is a schematic diagram showing the constraint relationship among the internal potential vector, the terminal voltage vector and the filter inductor voltage drop vector during the normal output voltage period of the converter in FIG. 2 .
  • FIG. 5 is a schematic diagram showing the constraint relationship among the internal potential vector, the terminal voltage vector and the filter inductor voltage drop vector of the converter in FIG. 2 during the output voltage drop period.
  • FIG6 is another flow chart of the fault ride-through method according to the first embodiment of the present application.
  • FIG. 7 is a schematic diagram of an application scenario of the converter and the DC source according to the first embodiment of the present application.
  • FIG8 is a flow chart of a fault ride-through method according to the second embodiment of the present application.
  • FIG. 9 is a schematic diagram showing the constraint relationship among the internal potential vector, the terminal voltage vector and the filter inductor voltage drop vector during the normal output voltage period of the second embodiment of the present application.
  • FIG. 10 is a schematic diagram showing the constraint relationship between the internal potential vector, the terminal voltage vector and the filter inductor voltage drop vector of the converter according to the second embodiment of the present application when the output voltage drops.
  • FIG. 11 is a waveform diagram showing the voltage and current waveforms varying with time during the low voltage ride-through process of the fault ride-through method of the second embodiment of the present application.
  • the input end of the converter 1 of this embodiment is connected to the DC source 2, and the output end is connected to the load 3. That is, the DC source 2, the converter 1 and the load 3 are electrically connected in sequence, and the converter 1 is connected between the DC source 2 and the load 3.
  • the DC source 2 can be a DC power supply module such as photovoltaic, wind power, and battery, which is used to provide a power signal.
  • the converter 1 is used to modulate the power signal output by the DC source 2 to a target voltage and a target frequency and then provide it to an external load 3.
  • the load 3 is, for example, a household electrical appliance.
  • the converter 1 includes a conversion circuit 11 and a filter inductor 12.
  • the input end of the conversion circuit 11 is the input end of the converter 1, which is connected to the DC source 2.
  • the filter inductor 12 includes a first end M and a second end N.
  • the first end M is connected to the output end of the conversion circuit 11, and the second end N is the output end of the converter 1, which is connected to the load 3.
  • the conversion circuit 11 is used to convert the DC power signal output by the DC source 2 into an AC signal.
  • the filter inductor 12 is used to filter out the high-frequency band in the power signal output by the conversion circuit 11.
  • the power signal is used to power the load 3.
  • the above-mentioned target voltage and target frequency match the load 3.
  • the power signal output by the converter 1 can be single-phase, three-phase or multi-phase to match the load 3.
  • the power signal output by the converter 1 is single-phase.
  • the output voltage of the output end of the converter 1 may drop (for example, three-phase short circuit, single-phase short circuit, phase-to-phase short circuit, etc.).
  • the amplitude of the terminal voltage vector Vt of the second end N drops to less than the terminal voltage extreme value (the system can preset a terminal voltage extreme value as a trigger standard for judging low voltage ride-through), and the amplitude of the filter inductor current vector of the filter inductor 12 increases to exceed the preset current limit value.
  • the converter 1 will be disconnected from the grid, and the converter 1 cannot normally supply power to the load 3.
  • the present embodiment provides a fault ride-through method and a converter 1 using the fault ride-through method, which is used to limit the amplitude of the filter inductor current vector (that is, the output current of the converter 1, that is, the current output by the converter to the load) within the current limit value during the period when the output voltage at the output end of the converter 1 drops, and quickly increase the output voltage at the output end of the converter 1 to ensure stable operation of the load 3.
  • the filter inductor current vector that is, the output current of the converter 1, that is, the current output by the converter to the load
  • the converter 1 of this embodiment further includes a detection circuit 13 and a controller 14.
  • the detection circuit 13 is respectively connected to the first terminal M and the second terminal N of the filter inductor 12.
  • the controller 14 is respectively connected to the conversion circuit 11 and the detection circuit 13.
  • the detection circuit 13 is used to detect the internal potential vector E of the first terminal M and the terminal voltage vector V t of the second terminal N.
  • the controller 14 is used to control the conversion circuit 11 according to the signal obtained by the detection circuit 13.
  • the conversion circuit 11 is controlled by the controller 14 and modulates the power signal provided by the DC source 2 so that the converter 1 finally outputs a power signal of corresponding voltage and frequency to the load 3.
  • the converter 1 may also include other circuit function modules, which are not described in detail in this application.
  • control link of the controller 14 in the converter 1 is shown in FIG2 .
  • the control process of the controller 14 includes voltage correction 211 , voltage loop 212 , current selection 213 , current loop 214 , modulation wave generation 215 and current amplitude limitation 216 .
  • the voltage loop 212 generates a current vector reference value according to the obtained internal potential vector command and the voltage vector feedback value.
  • the current selection link 213 is used to generate a current vector reference value according to the obtained current vector reference value and the preset current limit.
  • the current loop 214 generates a modulation vector reference value according to the obtained current vector command and the current vector feedback value.
  • the modulation wave generation 215 link generates a modulation wave according to the obtained modulation vector reference value and the phase reference value. Among them, the current vector feedback value and the current vector reference value are also used for the current limiting 216 link to obtain the internal potential vector correction value.
  • the internal potential vector command is generated by the voltage correction 211 link according to the set internal potential vector reference value and the obtained internal potential vector correction value.
  • the converter 1 In one pair of ratios, the converter 1 generates a modulation wave by controlling the converter through a voltage loop and a current loop.
  • the controller 14 in the converter 1 adds a voltage correction 211, a current selection 213 and a current limit 216 control link, so that the converter 1 can achieve fault ride-through when the output voltage drops.
  • FIG. 2 shows only part of the control links of the controller 14 .
  • the controller 14 may also work in other necessary control links such as phase control, which will not be elaborated in this application.
  • the fault ride-through method of this embodiment includes:
  • Step S11 detecting an internal potential vector of the first end and a terminal voltage vector of the second end of the filter inductor
  • Step S12 determining whether the amplitude of the terminal voltage vector at the second end of the filter inductor is reduced to less than or equal to a preset terminal voltage extreme value
  • step S13 is executed to reduce the internal potential vector of the first end of the filter inductor to reduce the amplitude of the filter inductor current vector on the filter inductor to be less than or equal to a preset current limit value.
  • step S14 is executed to maintain the internal potential vector of the first end.
  • the filter inductor current vector on the filter inductor 12 will exceed the preset current limit value Imax.
  • the filter inductor current vector is a three-phase abc alternating current vector; or a two-phase static ⁇ current vector, or a dq axis rotating current vector.
  • the vector value of the current vector feedback value detected at the second terminal N and the current vector reference value in the converter 1, whichever has a larger absolute value, is used as the amplitude of the filter inductor current vector.
  • the current vector reference value is the current vector reference value of the second terminal N under the voltage loop control
  • the current vector feedback value is the current value actually detected output from the second terminal N.
  • the current vector reference value or the current vector feedback value is compared with the current limit value Imax.
  • the current vector reference value or the current vector feedback value is greater than the current limit value Imax, it is determined that the output voltage drops.
  • the current vector reference value or the current vector feedback value is less than or equal to the current limit value Imax, it is determined that the converter 1 is working normally.
  • the current vector reference value and the current vector feedback value may not be compared.
  • the absolute value of the current vector is used instead of the absolute value of the current vector feedback value or the current vector reference value, and the current vector feedback value or the current vector reference value is directly used as the amplitude of the filter inductor current vector.
  • the vector value of the current vector reference value and the current vector feedback value with a larger absolute value is used as the amplitude of the filter inductor current vector, which is conducive to improving the accuracy of judging whether the converter is working abnormally.
  • the current limit value Imax is a preset current value. In this embodiment, the current limit value Imax is less than or equal to the upper limit value that the hardware capability of the converter 1 can bear or that can be allowed for normal operation. In this embodiment, the upper limit value is greater than the rated current value of the converter 1. When the converter 1 is working, the amplitude of the filter inductor current vector of the filter inductor 12 must be controlled within the current limit value Imax (that is, less than or equal to the current limit value Imax).
  • the amplitude of the filter inductor current vector can be allowed to be greater than the rated current value. By making the amplitude of the filter inductor current vector greater than the rated current value, it is beneficial to quickly increase the output voltage when the output voltage of the converter 1 drops.
  • the filter inductance value of the filter inductor 12 changes substantially little, and it can be considered that the change of the filter inductor voltage drop vector Vf of the filter inductor 12 mainly depends on the change of the filter inductor current vector.
  • the vector boundary V1 of the filter inductor voltage drop vector Vf corresponding to the current limit value of the converter 1 can be obtained.
  • Limiting the filter inductor current vector output by the converter 1 can be achieved by limiting the amplitude of the filter inductor voltage drop vector Vf.
  • the amplitude of the filter inductor voltage drop vector Vf falls within the vector boundary V1 (less than or equal to). If the output voltage of the converter 1 drops, resulting in the amplitude of the terminal voltage vector Vt dropping, the internal potential vector E, the terminal voltage vector Vt and the filter inductor voltage drop vector Vf at this time still maintain the aforementioned constraint relationship.
  • step S13 the value of the internal potential vector is reduced to E.
  • the filter inductor voltage drop vector Vf can be effectively constrained not to exceed the vector boundary V1. Accordingly, this also realizes that the amplitude of the filter inductor current vector is limited to within the current amplitude limit value Imax.
  • the fault ride-through method and converter 1 of this embodiment are beneficial to quickly increase the current of the second terminal N when the output voltage of the converter 1 drops, while maintaining the amplitude of the filter inductor current vector within the current limit value Imax, and support the voltage of the second terminal N, so that the output end of the converter 1 can stably supply power to the load 3.
  • step S13 specifically includes:
  • Step S131 generating an internal potential reference correction value according to the difference between the amplitude of the filter inductor current vector and the current amplitude limit value;
  • Step S132 generating an internal potential vector instruction according to the internal potential reference correction value and the internal potential vector reference value of the converter, and reducing the internal potential vector according to the internal potential vector instruction.
  • step S131 the current vector feedback value and the current vector reference value, whichever has a larger absolute value, are subtracted from the current limit value Imax, and the obtained current error is used to calculate the internal potential reference correction value based on a preset algorithm.
  • the preset algorithm includes a combination of one or more links in a proportional link, an integral link, a cross-coupling link, and a nonlinear control link.
  • step S132 the internal potential reference correction value and the preset internal potential vector reference value are weightedly added to calculate the internal potential vector instruction.
  • the internal potential vector instruction can be used to determine the specific amplitude of the internal potential vector of the first terminal M that needs to be reduced.
  • step S14 is executed: the internal potential vector value of the first terminal M is maintained. That is, the internal potential reference correction value is 0, and the internal potential vector value is maintained unchanged.
  • the converter fault ride-through method of this embodiment further includes, after step S132:
  • Step S15 correcting the current vector reference value according to the internal potential vector command and the voltage vector feedback value
  • Step S16 setting the smaller absolute value between the corrected current vector reference value and the current limit value as the current vector instruction
  • Step S17 generating a modulation wave according to the current vector instruction, the current vector feedback value and the phase reference value in the converter, wherein the modulation wave is used to modulate the power signal provided by the DC source to provide it to the load.
  • the preset current vector reference value can be corrected according to the internal potential vector command and the voltage vector feedback value to obtain a new current vector reference value.
  • the current vector command is used to determine how much the amplitude of the filter inductor current vector needs to be corrected.
  • the corrected current vector reference value calculated in the previous step may not be equal to the current limit value.
  • by setting the current vector command to the smaller absolute value of the corrected current vector reference value and the current limit value it is beneficial to ensure that the amplitude of the filter inductor current vector is constrained within the current limit value.
  • the control module 21 is also pre-set with a phase reference value.
  • step S17 of this embodiment a corresponding modulation wave is generated according to the phase reference value, the current vector command set in the previous step, and the current vector feedback value obtained.
  • the modulation wave is used to control the operation of the conversion circuit 11 in the converter 1.
  • the conversion circuit 11 includes a plurality of switching elements, and the modulation wave is used to control each switching element to be turned on or off to modulate the power signal output by the second terminal N. In this embodiment, when the fault of the converter is removed, the output voltage is restored.
  • the controller 14 of this embodiment includes a terminal that can automatically perform numerical calculations and/or information processing according to pre-set or stored instructions, and its hardware includes but is not limited to a microprocessor, a dedicated integrated circuit, a programmable gate array, a digital processor, and an embedded device.
  • the controller 14 also includes a memory, which is used to store program codes and various data.
  • the storage The device may include read-only memory (ROM), random access memory (RAM), programmable read-only memory (PROM), erasable programmable read-only memory (EPROM), one-time programmable read-only memory (OTPROM), electronically erasable programmable read-only memory (EEPROM), compact disc read-only memory (CD-ROM) or other optical disc storage, magnetic disk storage, magnetic tape storage, or any other computer-readable medium that can be used to carry or store data.
  • ROM read-only memory
  • RAM random access memory
  • PROM programmable read-only memory
  • EPROM erasable programmable read-only memory
  • OTPROM one-time programmable read-only memory
  • EEPROM electronically erasable programmable read-only memory
  • CD-ROM compact disc read-only memory
  • CD-ROM compact disc read-only memory
  • the controller 14 may include an integrated circuit, for example, a single packaged integrated circuit, or a plurality of packaged integrated circuits with the same or different functions, including a microprocessor, a digital processing chip, a graphics processor, and a combination of various control chips, etc.
  • the controller 14 executes various functions and processes data by running or executing programs or modules stored in the memory, and calling data stored in the memory.
  • the above-mentioned integrated unit implemented in the form of a software function module can be stored in a computer-readable storage medium.
  • the above-mentioned software function module is stored in a storage medium, including a number of instructions for enabling a computer device (which can be a personal computer, terminal, or network device, etc.) or a processor to execute part of the method described in each embodiment of the present application.
  • the memory stores program codes, and the controller 14 can call the program codes stored in the memory to execute related functions.
  • the memory stores multiple instructions, and the multiple instructions are executed by the controller 14 to perform a fault crossing method.
  • the specific implementation method of the controller 14 for the above instructions can refer to the description of the relevant steps in the corresponding embodiments of Figures 3 and 6, which will not be repeated here.
  • the above-mentioned fault ride-through method and converter 1 of this embodiment are applied in off-grid scenarios, including large off-grid microgrids, small and medium-sized industrial and commercial microgrids and other application scenarios.
  • the fault ride-through method and converter 1 can also be applied to a grid-connected scenario.
  • the fault ride-through method and converter 1 in this embodiment are applied to a grid-connected scenario, the above-mentioned current limit value needs to be adjusted according to the grid fault setting.
  • the converter 1 may be combined with a new energy power supply system such as a wind power supply system 4 and a photovoltaic power supply system 5 to be applied in an off-grid scenario.
  • a new energy power supply system such as a wind power supply system 4 and a photovoltaic power supply system 5 to be applied in an off-grid scenario.
  • the fault ride-through method and converter of the present embodiment when detecting that the amplitude of the terminal voltage vector is lower than the preset terminal voltage extreme value (that is, the amplitude of the filter inductor current vector is greater than the preset current limit value), determines that the output voltage of the output terminal of the converter 1 is abnormal (low voltage).
  • the internal potential vector is correspondingly adjusted to constrain the amplitude of the filter inductor voltage drop vector, and there is a corresponding relationship between the amplitude of the filter inductor voltage drop vector and the amplitude of the filter inductor current vector.
  • the filter inductor current vector output by the converter can be constrained. This is conducive to achieving rapid current response on the basis of avoiding the amplitude of the filter inductor current vector (that is, the amplitude of the converter output current) from exceeding the current limit value, and at the same time quickly restoring the voltage at the output terminal of the converter after the fault is removed. Furthermore, the present application The fault ride-through method and the converter using the method achieve the above beneficial effects through the above control process without adding or modifying the hardware structure, which is also conducive to cost saving.
  • the main difference between the fault ride-through method of this embodiment and the fault ride-through method of the first embodiment lies in the setting method of the current limit value.
  • the fault ride-through method specifically includes:
  • Step S21 detecting an internal potential vector of the first end and a terminal voltage vector of the second end of the filter inductor
  • Step S22 determining whether the amplitude of the terminal voltage vector at the second end of the filter inductor is reduced to less than or equal to a preset terminal voltage extreme value
  • step S22 If the judgment in step S22 is yes, then executing step S23, reducing the internal potential vector of the first end of the filter inductor to reduce the amplitude of the filter inductor current vector on the filter inductor to be less than or equal to the preset current limit value, and starting the timing of the duration;
  • Step S24 determining whether the duration is greater than a preset duration
  • step S25 is executed to switch the current amplitude limit value from the first amplitude limit value to the second amplitude limit value, the first amplitude limit value is greater than the second amplitude limit value, and step S21 is executed again.
  • step S21 is substantially the same as step S11 in the first embodiment and will not be described in detail.
  • the converter 1 when the converter 1 is operating normally, the amplitude of the filter inductor current vector outputted by the second terminal N is within the rated current boundary value. Furthermore, the converter 1 also has a maximum current limit boundary value. The maximum current limit boundary value is greater than the rated current boundary value. The converter 1 can operate within the rated current boundary value, or between the maximum current limit boundary value and the rated current boundary value. If the operating current exceeds the maximum current limit boundary value, there is a risk of damaging the converter 1.
  • boundary V1 corresponds to the filter inductor voltage drop vector value when the operating current of converter 1 is the rated current boundary value
  • boundary V corresponds to the filter inductor voltage drop vector value when the operating current of converter 1 is the maximum current limit boundary value
  • the constraint relationship among the internal potential vector E of the first terminal M, the terminal voltage vector Vt of the second terminal N and the filter inductor voltage drop vector Vf of the filter inductor 12 is shown in FIG9.
  • the filter inductor voltage drop vector Vf falls within the vector boundary V1 (less than or equal to). If the output voltage of the converter 1 drops, resulting in the amplitude drop of the terminal voltage vector Vt, the internal potential vector E, the terminal voltage vector Vt and the filter inductor voltage drop vector Vf still maintain the aforementioned constraint relationship.
  • step S23 the value of the internal potential vector is reduced to E.
  • the filter inductor voltage drop vector Vf can be effectively constrained not to exceed the vector boundary V2. That is, the amplitude of the filter inductor current vector is limited within the current amplitude limit value Imax.
  • a first amplitude limit value ILmt1 and a second amplitude limit value ILmt2 are preset, wherein the first amplitude limit value ILmt1 is greater than the second amplitude limit value ILmt1.
  • the first amplitude limit value ILmt1 is not greater than the maximum current amplitude limit boundary value, and when the amplitude of the filter inductor current vector is equal to the first amplitude limit value ILmt1, the filter inductor voltage drop vector Vf is located at the vector boundary V2.
  • the second amplitude limit value ILmt2 is not greater than the rated current boundary value, and when the amplitude of the filter inductor current vector is equal to the second amplitude limit value ILmt2, the filter inductor voltage drop vector Vf is located at the vector boundary V1.
  • the current limit value is first set to the first limit value ILmt1.
  • the timing process of the duration T is started in step S23.
  • the duration T is the duration during which the amplitude of the filter inductor current vector is between the maximum current limit boundary value and the rated current boundary value, and is also equal to the duration during which the current limit value Imax is set to the first limit value ILmt1.
  • the amplitude of the filter inductor current vector can exceed the rated current boundary value, thereby facilitating the improvement of the current response speed and quickly supporting the voltage on the load 3.
  • step of reducing the internal potential vector in step S23 is similar to step S31 and step S32 in the first embodiment.
  • a preset time length Tset is also set.
  • the preset time length Tset is set to the limit time length that the amplitude of the filter inductor current vector is between the maximum current limit boundary value and the rated current boundary value. That is, when the duration of the amplitude of the filter inductor current vector being between the maximum current limit boundary value and the rated current boundary value is greater than the preset time length Tset, the converter 1 is at risk of being damaged.
  • step S25 is executed to reduce the current limit value Imax, switching Imax from the first limit value ILmt1 to ILmt2 to constrain the amplitude of the filter inductor current vector to be within the rated current boundary value.
  • the current limit value is switched from the first limit value ILmt1 to the second limit value ILmt2 in a step manner. That is, the current limit value is directly switched from the first limit value ILmt1 to the second limit value ILmt2.
  • the current limit value is switched from the first limit value ILmt1 to the second limit value ILmt2 in a ramp manner. That is, the current limit value is switched from the first limit value ILmt1 for multiple times, and finally switches to the second limit value ILmt2, and the process of value switching changes linearly.
  • the current limit value is switched from the first limit value ILmt1 to the second limit value ILmt2 in an exponential manner. That is, the current limit value is switched from the first limit value ILmt1 for multiple times, and finally switches to the second limit value ILmt2, and the process of value switching changes exponentially.
  • This embodiment controls the specific value of the reduction of the internal potential vector E so that the filter inductor voltage drop vector Vf is just located at the boundary V1 or the boundary V2. At this time, the amplitude of the filter inductor current vector is equal to the maximum current limit value or the rated current limit value. This is beneficial to quickly respond to the current demand on the load 3 and support the voltage on the load 3 while keeping the amplitude of the filter inductor current vector within the maximum current limit value.
  • step S26 is executed to maintain the internal potential vector value of the first terminal M and reset the duration T to zero.
  • the voltage vector correction value is 0 to keep the internal potential vector value unchanged.
  • step S24 is judged as no, indicating that the time duration of converter 1 operating at the maximum current limit value and the rated current limit value has not reached the preset time duration Tset, then step S27 is executed, and the current limit value Imax is still maintained at the first limit value ILmt1, so that the current at the output end of converter 1 is quickly increased to quickly support the output end voltage.
  • the fault ride-through method of this embodiment further includes, after step S23:
  • the smaller absolute value of the corrected current vector reference value and the current limit value is set as the current vector instruction.
  • a modulation wave is generated according to the current vector instruction, the current vector feedback value and the phase reference value in the converter, and the modulation wave is used to modulate the power signal provided by the DC source to provide it to the load.
  • FIG. 11 shows the variation of the per-unit value Udp of the d-axis component of the terminal voltage (voltage output to the load) of the second terminal over time
  • (b) shows the variation of the per-unit value Uqp of the q-axis component of the terminal voltage of the second terminal over time
  • (c) shows the variation of the per-unit value Ip of the d-axis component of the current vector command and the actual value of the converter (the waveforms of the two values are basically overlapped, with a slight difference at 2.5 seconds and 3.1 seconds)
  • (d) shows the variation of the per-unit value Iq of the q-axis component of the output current vector command and the actual value of the converter (the waveforms of the two values are basically overlapped, with a slight difference at 2.5 seconds and 3.1 seconds).
  • the horizontal axis of the above (a), (b), (c), and (d) is in seconds, and the vertical axis is in pu.
  • the fault ride-through method of this embodiment is also applied to the controller 14 of the converter 1, and the controller 14 is specifically as described in the first embodiment.
  • the converter 1 of this embodiment can be applied to any power grid scenario as described in the first embodiment.
  • the fault ride-through method and converter of this embodiment can achieve all the beneficial effects described in Example 1. On this basis, by increasing the current limit value (first setting the current limit value to a larger first limit value ILmt1, and then switching to a smaller second limit value ILmt2), it is also beneficial to further improve the current response speed and voltage recovery speed of converter 1, so as to better support the function of converter 1, thereby improving the stability of the output voltage of converter 1.

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Inverter Devices (AREA)

Abstract

本申请提供一种故障穿越方法，应用于变换器，变换器包括变换电路及滤波电感，变换电路的输入端用于连接直流源，滤波电感的第一端用于连接变换电路的输出端，滤波电感的第二端用于连接负载；故障穿越方法由变换器执行，故障穿越方法包括：检测滤波电感的第一端的内电势矢量和第二端的端电压矢量；在滤波电感的第二端的端电压矢量的幅值降低到小于或等于预设的端电压极值时，减小滤波电感的第一端的内电势矢量，以减小滤波电感上的滤波电感电流矢量的幅值至小于或等于预设的电流限幅值，其中内电势矢量、端电压矢量和滤波电感电流矢量构成三角函数关系。本申请还提供一种变换器。

Description

故障穿越方法及变换器
相关申请的交叉引用
本申请要求在2022年9月26日提交中国专利局、申请号为202211177155.3、申请名称为“故障穿越方法及变换器”的中国专利的优先权，其全部内容通过引用结合在本申请中。
技术领域
本申请涉及电力电子技术领域，尤其涉及一种故障穿越方法及应用该故障穿越方法的变换器。
背景技术
风电、光伏、储能等供电电源逐渐成为电网的主力电源。上述供电电源通过变换器(或称变流器)接入电网。并网变换器的传统电网跟随型(Grid-following)控制方法使得新能源呈现出电流源特性，无法像传统同步发电机一样为电网提供频率和电压支撑，也不能在离网条件下运行，使得高比例新能源的电力系统稳定性及供电可靠性受到极大挑战。构网型(Grid-forming)控制是一种可以使电力电子变流器具有同步发电机的外特性的控制方法，可以为电网提供必要的电压和频率支撑，能够在并网或离网条件稳定运行，在电力系统中具有广泛的应用前景。
变换器采用具有构网能力的构网型(Grid-forming)控制方式，以为电网建立起稳定的交流电压和频率。但是电力电子器件过流能力较差，不具备在电网短路故障期间短时提供较大短路电流的能力。在短路故障期间，如果不对构网型控制的变换器进行任何控制，则变换器根据电网故障情况会自动产生较大短路电流，造成电力电子器件损坏。
目前，一种应用于变换器的故障穿越方式是在电网中增加硬件设备以在故障时为电网提供电流。此方式由于增加硬件设备，不利于成本控制。另一种故障穿越方式是在故障期间切换为电网跟随型的电流源控制策略，故障期间变换器失去了构网能力，不利于电网稳定。还有一种故障穿越方式为采用虚拟阻抗和电流限幅结合，但该方式难以精确限制电流幅值，主要包括虚拟阻抗太大，电流幅值小，没有充分利用电流能力对电网提供支撑；虚拟阻抗太小，电压环存在饱和，故障切除后电压恢复速度慢。且虚拟阻抗根据不同故障情况、电网运行工况不同，参数确定困难。另外，该方式易导致系统稳定性降低。
发明内容
本申请第一方面提供一种故障穿越方法，应用于变换器，所述变换器包括变换电路及滤波电感，所述变换电路的输入端用于连接直流源，所述滤波电感的第一端用于连接所述变换电路的输出端，所述滤波电感的第二端用于连接负载；所 述故障穿越方法由所述变换器执行，所述故障穿越方法包括：检测所述滤波电感的所述第一端的内电势矢量和所述第二端的端电压矢量；在所述滤波电感的所述第二端的所述端电压矢量的幅值降低到小于或等于预设的端电压极值时，减小所述滤波电感的所述第一端的所述内电势矢量，以减小所述滤波电感上的滤波电感电流矢量的幅值至小于或等于预设的电流限幅值，其中所述内电势矢量、所述端电压矢量和所述滤波电感电流矢量构成三角函数关系。
上述故障穿越方法，在检测到端电压矢量的幅值小于预设的端电压极值时判断变换装置输出端(也即第二端)输出的电压异常(低电压)，此时基于第一端的内电势矢量、第二端的端电压矢量及滤波电感的滤波电感电流矢量三者之间的约束关系，以及滤波电感压降矢量与滤波电感电流矢量两者的对应关系，调小内电势矢量的幅值，可以约束滤波电感压降矢量的幅值，从而可以约束滤波电感电流矢量的幅值在预设的电流限幅值之内(小于等于电流限幅值)。这有利于在避免滤波电感电流矢量的幅值超出电流限幅值的基础上，实现在输出的电压异常时变换器进行快速电流响应；在输出的电压异常期间维持变换装置的构网能力，快速提供电压支撑；在故障切除后，迅速恢复变换器输出的电压。进一步的，上述故障穿越方法及应用该方法的变换器，通过上述控制流程实现上述有益效果，无需新增或修改硬件结构，还有利于节约成本。
于一些实施例中，所述电流限幅值小于或等于所述变换器的硬件能力所能够承受或者正常工作所能允许的上限值，在所述滤波电感的所述第二端的所述端电压矢量的幅值小于所述预设的端电压极值且所述滤波电感电流矢量大于所述上限值时，通过所述变换器控制所述滤波电感的所述第一端的所述内电势矢量减少，以使所述滤波电感电流矢量降低到小于或等于所述上限值。
如此，当变换器的输出电压异常，导致端电压矢量的幅值减小时，通过减小内电势矢量，可以有效抑制滤波电感压降矢量的幅值，从而控制滤波电感电流矢量的幅值在电流限幅值之内。
于一些实施例中，所述上限值大于所述变换器的额定电流值。
在变换装置输出至负载的电压异常(低电压)时，通过使得上限值大于额定电流边界值，有利于快速提供电压支撑。
于一些实施例中，将所述滤波电感的所述第二端的电流矢量反馈值或所述变换器的电流矢量参考值作为所述滤波电感电流矢量的幅值。
电流矢量反馈值和电流矢量参考值中任一者大于预设的电流限幅值都可认为是变换装置的输出端输出电压异常。
于一些实施例中，将所述电流矢量反馈值和所述电流矢量参考值中绝对值较大的矢量值作为所述滤波电感电流矢量的幅值。
电流矢量参考值为电压环控制下第二端的电流矢量参考值，该电流矢量反馈值为实际检测到的第二端输出的电流值。在变换器正常工作时，电流矢量参考值与电流矢量反馈值是基本一致的。若变换器工作异常，电流矢量参考值与电流矢量反馈值可能存在差异。电流矢量参考值与电流矢量反馈值中任一者大于电流限幅值时，都表示变换器工作异常。因此以电流矢量参考值与电流矢量反馈值两者中绝对值较大者与电流限幅值进行比较。当电流矢量参考值与电流矢量反馈值两者中绝对值较大者大于电流限幅值时，判断故障。
于一些实施例中，所述滤波电感电流矢量大于所述上限值时，所述故障穿越方法还包括：启动持续时长的计时；当所述持续时长大于预设时长时，将所述电流限幅值由第一限幅值切换为第二限幅值，其中，所述第一限幅值大于所述第二限幅值。
于一些实施例中，所述将所述电流限幅值由第一限幅值切换为第二限幅值，包括：通过阶跃式、斜坡式或指数式方式将所述电流限幅值由所述第一限幅值切换为所述第二限幅值。
通过增大电流限幅值，还有利于进一步提升变换器的电流响应速度和电压恢复水平和速度。
于一些实施例中，所述故障穿越方法还包括：当所述持续时长小于或等于所述预设时长时，保持所述电流限幅值不变。
持续时长为滤波电感电流矢量的幅值位于最大电流限幅边界值与额定电流边界值之间的持续时长，同时也等于电流限幅值被设定为第一限幅值的持续时长。为了避免损坏变换器，还设置预设时长。预设时长设定为滤波电感电流矢量的幅值位于该最大电流限幅边界值与该额定电流边界值之间的极限时长。也即，当第二端的滤波电感电流矢量的幅值位于该最大电流限幅边界值与该额定电流边界值之间的持续时长大于预设时长，变换器就有损坏风险。
于一些实施例中，所述故障穿越方法还包括：当所述滤波电感电流矢量的幅值小于或等于所述电流限幅值时，将所述持续时长清零。
当确认所述滤波电感电流矢量的幅值小于或等于所述电流限幅值时，判断未发生故障，无需进行低电压穿越，无需计时。
于一些实施例中，所述减小所述滤波电感的所述第一端的所述内电势矢量，包括：根据所述滤波电感电流矢量的幅值与所述电流限幅值之间的差值，生成内电势参考修正值；根据所述内电势参考修正值和所述变换器的内电势矢量参考值生成内电势矢量指令；根据所述内电势矢量指令减小所述内电势矢量。
如此，使得变换器在输出电压跌落期间维持构网能力，保障输出至负载的电压稳定。
于一些实施例中，所述故障穿越方法还包括：根据所述内电势矢量指令和电压矢量反馈值，修正电流矢量参考值；将修正后的所述电流矢量参考值与所述电流限幅值二者中绝对值较小者，设定为电流矢量指令；以及根据所述电流矢量指令、电流矢量反馈值和所述变换器内的相位参考值生成调制波，所述调制波用于调制所述直流源提供的电力信号以提供至所述负载。
电流矢量指令用于确定滤波电感电流具体需要被修正多少数值。在前序步骤中计算出来的修正后的电流矢量参考值与电流限幅值可能不相等，通过将电流矢量指令设定为修正后的电流矢量参考值与电流限幅值中绝对值较小者，有利于确保滤波电感电流被约束于电流限幅值之内。
本申请第二方面提供一种变换器，所述变换器包括：变换电路，所述变换电路的输入端用于连接直流源；滤波电感，所述滤波电感的第一端用于连接所述变换电路的输出端，所述滤波电感的第二端用于连接负载；检测电路，所述检测电路连接所述第一端和所述第二端，用于检测所述滤波电感的所述第一端的内电势矢量和所述第二端的端电压矢量；控制器，所述控制器连接所述检测电路，用于在所述滤波电感的第二端的端电压矢量的幅值降低到小于或等于预设的端电压极值时，减小所述滤波电感第一端的所述内电势矢量，以减小所述滤波电感上电流矢量的幅值至小于或等于预设的电流限幅值，其中所述内电势矢量、所述端电压矢量和所述滤波电感电流矢量构成三角函数关系。
上述变换器，包括控制器和检测电路，在检测到端电压矢量的幅值小于预设的端电压极值时判断变换装置输出端(也即第二端)输出的电压异常(低电压)，此时基于第一端的内电势矢量、第二端的端电压矢量及滤波电感的滤波电感电流矢量三者之间的约束关系，以及滤波电感压降矢量与滤波电感电流矢量两者的对应关系，调小内电势矢量的幅值，可以约束滤波电感压降矢量的幅值，从而可以约束滤波电感电流矢量的幅值在预设的电流限幅值之内(小于等于电流限幅值)。这有利于在避免滤波电感电流矢量的幅值超出电流限幅值的基础上，实现在输出的电压异常时变换器进行快速电流响应；在输出的电压异常期间维持变换装置的构网能力，快速提供电压支撑；在故障切除后，迅速恢复变换器输出的电压。进一步的，上述变换器，通过上述控制流程实现上述有益效果，无需新增或修改硬件结构，还有利于节约成本。
于一些实施例中，所述电流限幅值小于或等于所述变换器的硬件能力所能够承受或者正常工作所能允许的上限值，在所述滤波电感的所述第二端的所述端电压矢量的幅值小于所述预设的端电压极值且所述滤波电感电流矢量大于所述上限值时，通过所述变换器控制所述滤波电感的所述第一端的所述内电势矢量减少，以使所述滤波电感电流矢量降低到小于或等于所述上限值。
如此，当变换器的输出电压异常，导致端电压矢量的幅值减小时，通过减小 内电势矢量，可以有效抑制滤波电感压降矢量的幅值，从而控制滤波电感电流矢量的幅值在电流限幅值之内。
于一些实施例中，所述控制器用于：根据所述滤波电感电流矢量的幅值与所述电流限幅值之间的差值，生成内电势参考修正值；根据所述内电势参考修正值和所述变换器的内电势矢量参考值生成内电势矢量指令；根据所述内电势矢量指令减小所述内电势矢量。
如此，使得变换器在输出电压跌落期间维持构网能力，保障输出至负载的电压稳定。
于一些实施例中，所述控制器用于：根据所述内电势矢量指令和电压矢量反馈值，修正电流矢量参考值；将修正后的所述电流矢量参考值与所述电流限幅值二者中绝对值较小者，设定为电流矢量指令；以及根据所述电流矢量指令、电流矢量反馈值和所述变换器内的相位参考值生成调制波，所述调制波用于调制所述直流源提供的电力信号以提供至所述负载。
电流矢量指令用于确定滤波电感电流具体需要被修正多少数值。在前序步骤中计算出来的修正后的电流矢量参考值与电流限幅值可能不相等，通过将电流矢量指令设定为修正后的电流矢量参考值与电流限幅值中绝对值较小者，有利于确保滤波电感电流被约束于电流限幅值之内。
附图说明
图1为本申请实施例一的变换器与直流源和负载的架构图。
图2为图1中变换器内控制器的控制环节示意图。
图3为本申请实施例一的故障穿越方法的一流程图。
图4为图2中变换器输出电压正常期间的内电势矢量、端电压矢量及滤波电感压降矢量三者之间的约束关系示意图。
图5为图2中变换器在输出电压跌落期间的内电势矢量、端电压矢量及滤波电感压降矢量三者之间的约束关系示意图。
图6为本申请实施例一的故障穿越方法的另一流程图。
图7为本申请实施例一的变换器与直流源的一应用场景示意图。
图8为本申请实施例二的故障穿越方法的流程图。
图9为本申请实施例二的输出电压正常期间的内电势矢量、端电压矢量及滤波电感压降矢量三者之间的约束关系示意图。
图10为本申请实施例二的变换器在输出电压跌落的内电势矢量、端电压矢量及滤波电感压降矢量三者之间的约束关系示意图。
图11为模拟本申请实施例二的故障穿越方法在低电压穿越过程中电压电流随时间变化的波形示意图。
具体实施方式
下面结合本申请实施例中的附图对本申请实施例进行描述。
实施例一
请参阅图1，本实施例的变换器1的输入端连接直流源2，输出端连接负载3。也即直流源2、变换器1及负载3依次电连接，变换器1连接于直流源2与负载3之间。直流源2可为光伏、风电、蓄电池等直流供电模块，用于提供电力信号。变换器1用于调制直流源2输出的电力信号至目标电压和目标频率后提供至外部的负载3。负载3例如为家庭用电设备。
本实施例中，变换器1包括变换电路11及滤波电感12。变换电路11的输入端为变换器1的输入端，连接直流源2。滤波电感12包括第一端M和第二端N。第一端M连接变换电路11的输出端，第二端N为变换器1的输出端，连接负载3。变换电路11用于将直流源2输出的直流的电力信号转换为交流信号。滤波电感12用于滤除变换电路11输出的电力信号中的高频波段。电力信号用于为负载3供电。本实施例中，上述目标电压和目标频率与负载3匹配。本实施例中，变换器1输出的电力信号可以为单相、三相或多相，以匹配负载3。例如，对于负载3为家庭用电设备的场景，变换器1输出的电力信号为单相。
本实施例中，在一些场景中，变换器1的输出端(也即滤波电感12的第二端N)的输出电压可能跌落(例如三相短路、单相短路、相间短路等)。变换器1的输出端输出电压跌落时，第二端N的端电压矢量Vt的幅值跌落至小于端电压极值(系统可预设一端电压极值作为判断低电压穿越的触发标准)，滤波电感12的滤波电感电流矢量的幅值增大至超过预设的电流限幅值，此时变换器1将脱网，变换器1无法正常为负载3供电。
本实施例提供一种故障穿越方法及应用该故障穿越方法的变换器1，用于在变换器1的输出端输出电压跌落期间，限制滤波电感电流矢量的幅值(也即变换器1的输出电流，也即变换器输出至负载的电流)在电流限幅值内，迅速提升变换器1的输出端的输出电压，保证负载3工作稳定。
本实施例的变换器1，还包括检测电路13及控制器14。检测电路13分别连接滤波电感12的第一端M和第二端N。控制器14分别连接变换电路11和检测电路13。检测电路13用于检测第一端M的内电势矢量E和第二端N的端电压矢量Vt。控制器14用于根据检测电路13获取的信号控制变换电路11。变换电路11受控于控制器14，对直流源2提供的电力信号进行调制，以使得变换器1最终输出相应电压、频率的电力信号至负载3。本实施例中，变换器1还可包括其他电路功能模块，本申请不作过多描述。
本实施例中，变换器1中控制器14的控制环节如图2所示。控制器14的控制流程包括电压修正211、电压环212、电流选择213、电流环214、调制波生成215以及电流限幅216。
电压环212根据获取的内电势矢量指令和电压矢量反馈值生成电流矢量参考值。电流选择213环节用于根据获取的电流矢量参考值和预先设定的电流限幅 值生成电流矢量指令。电流环214根据获取的电流矢量指令和电流矢量反馈值生成调制矢量参考值。调制波生成215环节根据获取的调制矢量参考值和相位参考值生成调制波。其中，该电流矢量反馈值和电流矢量参考值还用于供电流限幅216环节获取内电势矢量修正值。该内电势矢量指令由电压修正211环节根据设定的内电势矢量参考值和获取的内电势矢量修正值生成。
在一对比例中，变换器1通过电压环和电流环控制变换器生成调制波。本实施例中，变换器1内控制器14通过新增电压修正211、电流选择213及电流限幅216控制环节，使得变换器1可在输出电压跌落时实现故障穿越。
图2所示仅为控制器14的部分控制环节，控制器14还可工作于例如相位控制的其他必要控制环节，本申请不对此赘述。
因此请参阅图3，本实施例的故障穿越方法包括：
步骤S11，检测所述滤波电感的所述第一端的内电势矢量和所述第二端的端电压矢量；
步骤S12，判断所述滤波电感的第二端的端电压矢量的幅值是否降低到小于或等于预设的端电压极值；
若判断为是，则执行步骤S13，减小所述滤波电感的第一端的所述内电势矢量，以减小所述滤波电感上的滤波电感电流矢量的幅值至小于或等于预设的电流限幅值。
若判断为否，则执行步骤S14，保持所述第一端的内电势矢量。
本实施例中，当变换器1的输出端的端电压矢量的幅值降低到小于或等于预设的端电压极值时，滤波电感12上的滤波电感电流矢量会超过预设的电流限幅值Imax。本实施例步骤S12中通过检测滤波电感电流矢量是否超过电流限幅值Imax来判断端电压矢量的幅值是否降低到小于或等于预设的端电压极值。本实施例中，滤波电感电流矢量为三相abc交流电流矢量；或称两相静αβ电流矢量，亦或称dq轴旋转电流矢量。
本实施例中，将在第二端N检测到的电流矢量反馈值与变换器1内的电流矢量参考值两者中绝对值较大者的矢量值作为上述滤波电感电流矢量的幅值。
电流矢量参考值为电压环控制下第二端N的电流矢量参考值，该电流矢量反馈值为实际检测到的第二端N输出的电流值。在变换器1正常工作时，电流矢量参考值与电流矢量反馈值是基本一致的。若变换器1工作异常，电流矢量参考值与电流矢量反馈值可能存在差异。
电流矢量参考值与电流矢量反馈值中任一者大于电流限幅值Imax时，都表示变换器1工作异常，变换器1的输出电压跌落，因此以电流矢量参考值与电流矢量反馈值两者中绝对值较大者与电流限幅值Imax进行比较。当电流矢量参考值与电流矢量反馈值两者中绝对值较大者大于电流限幅值Imax时，判断输出电压跌落。当电流矢量参考值与电流矢量反馈值两者中绝对值较大者小于等于电流限幅值Imax时，判断变换器1工作正常。
于本申请其他实施例中，也可不比较电流矢量参考值和电流矢量反馈值两者 的绝对值，而直接将电流矢量反馈值或电流矢量参考值作为滤波电感电流矢量的幅值。本实施例中，将所述电流矢量参考值和所述电流矢量反馈值中绝对值较大者的矢量值作为滤波电感电流矢量的幅值，有利于提升判断变换器是否工作异常的精准度。
电流限幅值Imax为预先设定的电流值。本实施例中，电流限幅值Imax小于或等于变换器1的硬件能力所能够承受或者正常工作所能允许的上限值。本实施例中，该上限值大于变换器1的额定电流值。变换器1工作时，滤波电感12的滤波电感电流矢量的幅值要控制在电流限幅值Imax以内(也即小于等于电流限幅值Imax)。
在变换器1的输出电压跌落时，可以允许滤波电感电流矢量的幅值大于额定电流值。通过使得滤波电感电流矢量的幅值大于额定电流值，有利于在变换器1的输出电压跌落时快速提升输出电压。
本实施例中，滤波电感12的滤波电感值变化基本较小，可认为该滤波电感12的滤波电感压降矢量Vf变化主要取决于滤波电感电流矢量的变化。根据变换器1的电流限幅值可以得到与其对应的滤波电感压降矢量Vf的矢量边界V1。限制变换器1输出的滤波电感电流矢量，可通过限制滤波电感压降矢量Vf的幅值实现。
变换器1正常工作时，第一端M的内电势矢量E、第二端N的端电压矢量Vt和滤波电感12的滤波电感电流矢量三者之间存在三角函数关系，而滤波电感压降矢量Vf随滤波电感电流矢量变化，因此变换器1正常工作时，第一端M的内电势矢量E、第二端N的端电压矢量Vt和滤波电感12的滤波电感压降矢量Vf三者之间也存在三角函数关系。内电势矢量E、端电压矢量Vt、滤波电感压降矢量Vf之间的约束关系如附图4所示，也即：Vt+Vf＝E。其中，滤波电感压降矢量Vf的幅值落在矢量边界V1以内(小于等于)。如果考虑变换器1的输出电压跌落，导致端电压矢量Vt的幅值跌落，此时的内电势矢量E、端电压矢量Vt及滤波电感压降矢量Vf三者之间仍然保持前述的约束关系。
但，请参阅图5，如果第一端M的内电势矢量维持恒定电压源模式，保持为E’(如虚线所示)，由于端电压矢量Vt的幅值跌落，此时滤波电感压降矢量Vf’的幅值将超出矢量边界V1，这也就意味着此时滤波电感12上的滤波电感电流矢量的幅值也超出了变换器1的允许最大运行电流边界值。
因此本实施例中，在步骤S13时，将内电势矢量的值减小为E，如此，根据内电势矢量E、第二端N的端电压矢量Vt和滤波电感压降矢量Vf三者之间的约束关系，可有效约束滤波电感压降矢量Vf不超出矢量边界V1。相应地，这也即实现了滤波电感电流矢量的幅值被限制在该电流限幅值Imax以内。
因此本实施例的故障穿越方法和变换器1，有利于在保持滤波电感电流矢量的幅值不超出电流限幅值Imax的基础上，在变换器1的输出电压跌落时迅速提升第二端N的电流，支撑第二端N的电压，从而使得变换器1的输出端稳定为负载3供电。
本实施例中，步骤S13具体包括：
步骤S131，根据所述滤波电感电流矢量的幅值与所述电流限幅值之间的差值，生成内电势参考修正值；
步骤S132，根据所述内电势参考修正值和所述变换器的内电势矢量参考值生成内电势矢量指令，根据所述内电势矢量指令减小所述内电势矢量。
本实施例中，步骤S131时将电流矢量反馈值和电流矢量参考值两者中绝对值较大者与电流限幅值Imax作差，获得的电流误差用于基于预设算法计算获得内电势参考修正值。本实施例中，上述预设算法包括比例环节、积分环节、交叉耦合环节以及非线性控制环节中的一个或多个环节的组合。
本实施例中，步骤S132时，对所述内电势参考修正值和预先设定好的内电势矢量参考值进行加权相加，以计算得到内电势矢量指令。所述内电势矢量指令即可用于确定具体需要对第一端M的内电势矢量进行减小的幅值。
本实施例中，若步骤S12中判断为否，则表征此时变换器1工作正常，输出电压正常，则执行步骤S14：保持第一端M的内电势矢量值。也即，内电势参考修正值为0，保持内电势矢量值不变。
请参阅图6，本实施例的变换器故障穿越方法，在步骤S132之后还包括：
步骤S15，根据所述内电势矢量指令和电压矢量反馈值，修正电流矢量参考值；
步骤S16，将修正后的所述电流矢量参考值与所述电流限幅值二者中绝对值较小者，设定为电流矢量指令；以及
步骤S17，根据所述电流矢量指令、电流矢量反馈值和所述变换器内的相位参考值生成调制波，所述调制波用于调制所述直流源提供的电力信号以提供至所述负载。
本实施例中，在步骤S15时，根据内电势矢量指令和电压矢量反馈值可以修正预先设定好的电流矢量参考值，得到新的电流矢量参考值。步骤S16中，电流矢量指令用于确定滤波电感电流矢量的幅值具体需要被修正多少数值。在前序步骤中计算出来的修正后的电流矢量参考值与电流限幅值可能不相等，本实施例中通过将电流矢量指令设定为修正后的电流矢量参考值与电流限幅值中绝对值较小者，有利于确保滤波电感电流矢量的幅值被约束于电流限幅值之内。
控制模块21还预先设定有相位参考值，本实施例步骤S17中，根据相位参考值、前述步骤设定的电流矢量指令以及获取的电流矢量反馈值，生成相应的调制波，该调制波用于控制变换器1中变换电路11工作。本实施例中，变换电路11包括多个开关元件，该调制波用于控制各个开关元件导通或关断，以调制第二端N输出的电力信号。本实施例中，在变换器的故障切除时，恢复输出电压。
本实施例的控制器14，在一些实施例中，包括一种能够按照事先设定或存储的指令，自动进行数值计算和/或信息处理的终端，其硬件包括但不限于微处理器、专用集成电路、可编程门阵列、数字处理器及嵌入式设备等。在一些实施例中，控制器14还包括存储器，存储器用于存储程序代码和各种数据。所述存储 器可以包括只读存储器(Read-Only Memory，ROM)、随机存储器(Random Access Memory，RAM)、可编程只读存储器(Programmable Read-Only Memory，PROM)、可擦除可编程只读存储器(Erasable Programmable Read-Only Memory，EPROM)、一次可编程只读存储器(One-time Programmable Read-Only Memory，OTPROM)、电子擦除式可复写只读存储器(Electrically-Erasable Programmable Read-Only Memory，EEPROM)、只读光盘(Compact Disc Read-Only Memory，CD-ROM)或其他光盘存储器、磁盘存储器、磁带存储器、或者能够用于携带或存储数据的计算机可读的任何其他介质。
在一些实施例中，所述控制器14可以包括集成电路，例如可以包括单个封装的集成电路，也可以包括多个相同功能或不同功能封装的集成电路，包括微处理器、数字处理芯片、图形处理器及各种控制芯片的组合等。所述控制器14通过运行或执行存储在所述存储器内的程序或者模块，以及调用存储在所述存储器内的数据，以执行各种功能和处理数据。
上述以软件功能模块的形式实现的集成的单元，可以存储在一个计算机可读取存储介质中。上述软件功能模块存储在一个存储介质中，包括若干指令用以使得一台计算机设备(可以是个人计算机，终端，或者网络设备等)或处理器(processor)执行本申请各个实施例所述方法的部分。
所述存储器中存储有程序代码，且所述控制器14可调用所述存储器中存储的程序代码以执行相关的功能。在本申请的一个实施例中，所述存储器存储多个指令，所述多个指令被控制器14所执行以故障穿越方法。具体地，控制器14对上述指令的具体实现方法可参考图3和图6对应实施例中相关步骤的描述，在此不赘述。
本实施例的上述故障穿越方法和变换器1，应用于离网场景中，包括大型离网型微电网，中小型工商业微电网等应用场景。
于本申请其他实施例中，故障穿越方法和变换器1也可应用于并网场景中。当本实施例中故障穿越方法和变换器1应用于并网场景时，需要根据电网故障设置调整上述的电流限幅值取值。
请参阅图7，于本申请其他实施例中，变换器1可与风电供电系统4、光伏供电系统5等新能源供电系统结合，以应用于离网场景中。
综上，本实施例的故障穿越方法和变换器，在检测到端电压矢量的幅值低于预设的端电压极值(也即滤波电感电流矢量的幅值大于预设的电流限幅值)时判断变换器1的输出端输出电压存在异常(低电压)，此时基于第一端的内电势矢量、第二端的端电压矢量及滤波电感的滤波电感电流矢量三者之间的矢量约束关系，相应调小内电势矢量，可以约束滤波电感压降矢量的幅值，而滤波电感压降矢量的幅值与滤波电感电流矢量的幅值存在对应关系，通过约束滤波电感压降矢量可以约束变换器输出的滤波电感电流矢量。这有利于在避免滤波电感电流矢量的幅值(也即变换器输出电流的幅值)超出电流限幅值的基础上，实现快速电流响应，同时在故障切除后迅速恢复变换器的输出端的电压。进一步的，本申请的 故障穿越方法及应用该方法的变换器，通过上述控制流程实现上述有益效果，无需新增或修改硬件结构，还有利于节约成本。
实施例二
本实施例的故障穿越方法与实施例一中的故障穿越方法，主要区别在于电流限幅值的设定方式。
请参阅图8，本实施例中，故障穿越方法具体包括：
步骤S21，检测所述滤波电感的所述第一端的内电势矢量和所述第二端的端电压矢量；
步骤S22，判断所述滤波电感的第二端的端电压矢量的幅值是否降低到小于或等于预设的端电压极值；
若步骤S22判断为是，则执行步骤S23，减小所述滤波电感的第一端的所述内电势矢量，以减小所述滤波电感上的滤波电感电流矢量的幅值至小于或等于预设的电流限幅值，并启动持续时长的计时；
步骤S24，判断所述持续时长是否大于预设时长；
若步骤S24判断为是，则执行步骤S25，将所述电流限幅值由第一限幅值切换为第二限幅值，所述第一限幅值大于所述第二限幅值，并再次执行步骤S21。
本实施例中，步骤S21与实施例一中步骤S11基本相同，不再赘述。
如实施例一中所述的，变换器1正常工作时，第二端N输出的滤波电感电流矢量的幅值在额定电流边界值以内。进一步的，变换器1还具有最大电流限幅边界值。该最大电流限幅边界值大于该额定电流边界值。变换器1可工作于额定电流边界值以内，也可工作于该最大电流限幅边界值与该额定电流边界值之间，工作电流超过该最大电流限幅边界值即有损坏变换器1的风险。
请参阅图9，根据上述的最大电流限幅边界值和额定电流边界值，可以确定对应的滤波电感压降矢量值的边界V1和V2，其中边界V1对应变换器1的工作电流为额定电流边界值时的滤波电感压降矢量值，边界V对应变换器1的工作电流为最大电流限幅边界值时的滤波电感压降矢量值。
变换器1正常工作时，第一端M的内电势矢量E、第二端N的端电压矢量Vt和滤波电感12的滤波电感压降矢量Vf三者之间的约束关系如附图9所示。其中，滤波电感压降矢量Vf落在矢量边界V1以内(小于等于)。如果考虑变换器1的输出电压跌落，导致端电压矢量Vt的幅值跌落，此时的内电势矢量E、端电压矢量Vt及滤波电感压降矢量Vf三者之间仍然保持前述的约束关系。
但，请参阅图10，如果第一端M的内电势矢量维持恒定电压源模式，保持为E’(如虚线所示)，由于端电压矢量Vt的幅值跌落，此时滤波电感压降矢量Vf’将直接超出矢量边界V2，这也就意味着此时滤波电感电流矢量的幅值也超出了变换器1的最大电流限幅边界值。
因此本实施例中，在步骤S23时，将内电势矢量的值减小为E，如此，根据内电势矢量E、第二端N的端电压矢量Vt和滤波电感压降矢量Vf三者之间的约束关系，可有效约束滤波电感压降矢量Vf不超出矢量边界V2。相应地，这也 即实现了滤波电感电流矢量的幅值被限制在该电流限幅值Imax以内。
本实施例中，预设第一限幅值ILmt1和第二限幅值ILmt2，其中第一限幅值ILmt1大于第二限幅值ILmt1。第一限幅值ILmt1不大于最大电流限幅边界值，滤波电感电流矢量的幅值等于第一限幅值ILmt1时，滤波电感压降矢量Vf位于矢量边界V2。第二限幅值ILmt2不大于额定电流边界值，滤波电感电流矢量的幅值等于第二限幅值ILmt2时，滤波电感压降矢量Vf位于矢量边界V1。
本实施例中，先将电流限幅值设定为第一限幅值ILmt1。当步骤S22中判断滤波电感电流矢量的幅值大于电流限幅值(也即第一限幅值ILmt1)时，在步骤S23时开始启动持续时长T的计时流程。持续时长T为滤波电感电流矢量的幅值位于该最大电流限幅边界值与该额定电流边界值之间的持续时长，同时也等于电流限幅值Imax被设定为第一限幅值ILmt1的持续时长。
本实施例中，通过设定电流限幅值为大于额定电流边界值的第一限幅值ILmt1，使得滤波电感电流矢量的幅值可超出额定电流边界值，从而有利于提升电流响应速度，快速支撑负载3上的电压。
本实施例中，步骤S23减小内电势矢量的步骤如实施例一中步骤S31和步骤S32。
本实施例中，为了避免损坏变换器1，还设置预设时长Tset。预设时长Tset设定为滤波电感电流矢量的幅值位于该最大电流限幅边界值与该额定电流边界值之间的极限时长。也即，当滤波电感电流矢量的幅值位于该最大电流限幅边界值与该额定电流边界值之间的持续时长大于预设时长Tset，变换器1就有损坏风险。
若步骤S24中判断持续时长T大于预设时长Tset，则执行步骤S25，减小电流限幅值Imax，将Imax由第一限幅值ILmt1切换为ILmt2，以约束滤波电感电流矢量的幅值于该额定电流边界值之内。
步骤S24中，于一些实施例中，电流限幅值从第一限幅值ILmt1切换至第二限幅值ILmt2的方式采用阶跃式。也即，电流限幅值从第一限幅值ILmt1直接切换至第二限幅值ILmt2。于另一些实施例中，电流限幅值从第一限幅值ILmt1切换至第二限幅值ILmt2的方式采用斜坡式。也即，电流限幅值从第一限幅值ILmt1开始进行多次数值切换，最终切换至第二限幅值ILmt2，并且数值切换的过程呈线性变化。于又一些实施例中，电流限幅值从第一限幅值ILmt1切换至第二限幅值ILmt2的方式采用指数式。也即，电流限幅值从第一限幅值ILmt1开始进行多次数值切换，最终切换至第二限幅值ILmt2，并且数值切换的过程呈指数式变化。
本实施例通过控制内电势矢量E减小的具体值，使得滤波电感压降矢量Vf刚好位于边界V1或边界V2，此时滤波电感电流矢量的幅值等于最大电流限幅值或额定电流限幅值，这有利于在保持滤波电感电流矢量的幅值不超出最大限流限幅的基础上，迅速响应负载3上的电流需求，支撑负载3上的电压。
本实施例中，若步骤S22判断为否，则执行步骤S26，保持第一端M的内电势矢量值，并将持续时长T清零。也即，当步骤S22中判断滤波电感电流矢量 的幅值未超出电流限幅值时，电压矢量修正值为0，以保持内电势矢量值不变。
本实施例中，若步骤S24判断为否，表示变换器1工作于最大电流限幅值与额定电流限幅值的时长还未达到预设时长Tset，则执行步骤S27，还是保持电流限幅值Imax为第一限幅值ILmt1，使得变换器1的输出端的电流快速拉升，快速支撑输出端电压。
本实施例的故障穿越方法，在步骤S23之后还包括：
根据所述内电势矢量指令和电压矢量反馈值，修正电流矢量参考值；
将修正后的所述电流矢量参考值与所述电流限幅值二者中绝对值较小者，设定为电流矢量指令；以及
根据所述电流矢量指令、电流矢量反馈值和所述变换器内的相位参考值生成调制波，所述调制波用于调制所述直流源提供的电力信号以提供至所述负载。
上述步骤如实施例一中步骤S15-S17所述。
请参阅图11，图11中(a)图表示第二端的端电压(输出至负载的电压)d轴分量标幺值Udp随时间的变化，(b)图表示第二端的端电压q轴分量标幺值Uqp随时间的变化，(c)图表示变换器的电流矢量指令和实际值的d轴分量标幺值Ip(两个值的波形基本重叠，在2.5秒和3.1秒处稍有差别)随时间的变化，(d)图表示变换器输出电流矢量指令和实际值的q轴分量标幺值Iq(两个值的波形基本重叠，在2.5秒和3.1秒处稍有差别)随时间的变化。上述(a)、(b)、(c)、(d)图横坐标为单位为秒，纵坐标单位为pu。
以图11中(c)图为例，可以看出，在2.5秒左右第二端输出的电压异常，此时变换器的电流矢量指令快速响应，且在2.6秒时快速切换限幅值，在3.4秒异常切除后，如图11中(a)图所示，迅速恢复第二端的输出电压正常水平。
本实施例的故障穿越方法，其同样应用于变换器1的控制器14中，控制器14具体如实施例一中所述。且本实施例的变换器1可应用于如实施例一中所述的任意电网场景。
本实施例的故障穿越方法和变换器，可以实现如实施例一中所述的所有有益效果，在此基础上，通过增大电流限幅值(先设定电流限幅值为较大的第一限幅值ILmt1，再切换为较小的第二限幅值ILmt2)，还有利于进一步提升变换器1的电流响应速度和电压恢复速度，以更好地支撑变换器1的功能，从而提升变换器1的输出电压的稳定性。
本技术领域的普通技术人员应当认识到，以上的实施方式仅是用来说明本发明，而并非用作为对本发明的限定，只要在本发明的实质精神范围之内，对以上实施例所作的适当改变和变化都落在本发明要求保护的范围之内。

Claims (15)

  1. 一种故障穿越方法，应用于变换器，所述变换器包括变换电路及滤波电感，所述变换电路的输入端用于连接直流源，所述滤波电感的第一端用于连接所述变换电路的输出端，所述滤波电感的第二端用于连接负载；
    其特征在于，所述故障穿越方法由所述变换器执行，所述故障穿越方法包括：
    检测所述滤波电感的所述第一端的内电势矢量和所述第二端的端电压矢量；
    在所述端电压矢量的幅值降低到小于或等于预设的端电压极值时，减小所述滤波电感的所述第一端的所述内电势矢量，以减小所述滤波电感上的滤波电感电流矢量的幅值至小于或等于预设的电流限幅值，其中所述内电势矢量、所述端电压矢量和所述滤波电感电流矢量构成三角函数关系。
  2. 如权利要求1所述的故障穿越方法，其特征在于，所述电流限幅值小于或等于所述变换器的硬件能力所能够承受或者正常工作所能允许的上限值，在所述滤波电感的所述第二端的所述端电压矢量的幅值小于所述预设的端电压极值且所述滤波电感电流矢量大于所述上限值时，通过所述变换器控制所述滤波电感的所述第一端的所述内电势矢量减少，以使所述滤波电感电流矢量降低到小于或等于所述上限值。
  3. 如权利要求2所述的故障穿越方法，其特征在于，所述上限值大于所述变换器的额定电流值。
  4. 如权利要求1-3中任一项所述的故障穿越方法，其特征在于，将所述滤波电感的所述第二端的电流矢量反馈值或所述变换器的电流矢量参考值作为所述滤波电感电流矢量的幅值。
  5. 如权利要求4所述的故障穿越方法，其特征在于，将所述电流矢量反馈值和所述电流矢量参考值中绝对值较大的矢量值作为所述滤波电感电流矢量的幅值。
  6. 如权利要求2-5中任一项所述的故障穿越方法，其特征在于，所述滤波电感电流矢量大于所述上限值时，所述故障穿越方法还包括：
    启动持续时长的计时；
    当所述持续时长大于预设时长时，将所述电流限幅值由第一限幅值切换为第二限幅值，其中，所述第一限幅值大于所述第二限幅值。
  7. 如权利要求6所述的故障穿越方法，其特征在于，所述将所述电流限幅值由第一限幅值切换为第二限幅值，包括：
    通过阶跃式、斜坡式或指数式方式将所述电流限幅值由所述第一限幅值切换为所述第二限幅值。
  8. 如权利要求6或7所述的故障穿越方法，其特征在于，所述故障穿越方法还包括：
    当所述持续时长小于或等于所述预设时长时，保持所述电流限幅值不变。
  9. 如权利要求6-8中任一项所述的故障穿越方法，其特征在于，所述故障穿越方法还包括：
    当所述滤波电感电流矢量的幅值小于或等于所述电流限幅值时，将所述持续时长清零。
  10. 如权利要求1-9中任一项所述的故障穿越方法，其特征在于，所述减小所述滤波电感的所述第一端的所述内电势矢量，包括：
    根据所述滤波电感电流矢量的幅值与所述电流限幅值之间的差值，生成内电势参考修正值；
    根据所述内电势参考修正值和所述变换器的内电势矢量参考值生成内电势矢量指令；
    根据所述内电势矢量指令减小所述内电势矢量。
  11. 如权利要求10所述的故障穿越方法，其特征在于，所述故障穿越方法还包括：
    根据所述内电势矢量指令和电压矢量反馈值，修正电流矢量参考值；
    将修正后的所述电流矢量参考值与所述电流限幅值二者中绝对值较小者，设定为电流矢量指令；以及
    根据所述电流矢量指令、电流矢量反馈值和所述变换器内的相位参考值生成调制波，所述调制波用于调制所述直流源提供的电力信号以提供至所述负载。
  12. 一种变换器，其特征在于，所述变换器包括：
    变换电路，所述变换电路的输入端用于连接直流源；
    滤波电感，所述滤波电感的第一端用于连接所述变换电路的输出端，所述滤波电感的第二端用于连接负载；
    检测电路，所述检测电路连接所述第一端和所述第二端，用于检测所述滤波电感的所述第一端的内电势矢量和所述第二端的端电压矢量；
    控制器，所述控制器连接所述检测电路，用于在所述滤波电感的第二端的端电压矢量的幅值降低到小于或等于预设的端电压极值时，减小所述滤波电感第一端的所述内电势矢量，以减小所述滤波电感上电流矢量的幅值至小于或等于预设的电流限幅值，其中所述内电势矢量、所述端电压矢量和所述滤波电感电流矢量 构成三角函数关系。
  13. 如权利要求12所述的变换器，其特征在于，所述电流限幅值小于或等于所述变换器的硬件能力所能够承受或者正常工作所能允许的上限值，在所述滤波电感的所述第二端的所述端电压矢量的幅值小于所述预设的端电压极值且所述滤波电感电流矢量大于所述上限值时，通过所述变换器控制所述滤波电感的所述第一端的所述内电势矢量减少，以使所述滤波电感电流矢量降低到小于或等于所述上限值。
  14. 如权利要求12或13所述的变换器，其特征在于，所述控制器用于：
    根据所述滤波电感电流矢量的幅值与所述电流限幅值之间的差值，生成内电势参考修正值；
    根据所述内电势参考修正值和所述变换器的内电势矢量参考值生成内电势矢量指令；
    根据所述内电势矢量指令减小所述内电势矢量。
  15. 如权利要求14所述的变换器，其特征在于，所述控制器用于：
    根据所述内电势矢量指令和电压矢量反馈值，修正电流矢量参考值；
    将修正后的所述电流矢量参考值与所述电流限幅值二者中绝对值较小者，设定为电流矢量指令；以及
    根据所述电流矢量指令、电流矢量反馈值和所述变换器内的相位参考值生成调制波，所述调制波用于调制所述直流源提供的电力信号以提供至所述负载。
PCT/CN2023/101717 2022-09-26 2023-06-21 故障穿越方法及变换器 WO2024066511A1 (zh)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
CN202211177155.3A CN115622379A (zh) 2022-09-26 2022-09-26 故障穿越方法及变换器
CN202211177155.3 2022-09-26

Publications (1)

Publication Number Publication Date
WO2024066511A1 true WO2024066511A1 (zh) 2024-04-04

Family

ID=84860423

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/CN2023/101717 WO2024066511A1 (zh) 2022-09-26 2023-06-21 故障穿越方法及变换器

Country Status (2)

Country Link
CN (1) CN115622379A (zh)
WO (1) WO2024066511A1 (zh)

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN115622379A (zh) * 2022-09-26 2023-01-17 华为数字能源技术有限公司 故障穿越方法及变换器

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN101860038A (zh) * 2009-01-23 2010-10-13 通用电气公司 用于控制并网发电系统的系统和方法
US20180115255A1 (en) * 2016-10-20 2018-04-26 Ge Energy Power Conversion Technology Ltd Method for controlling a photovoltaic inverter for network fault ride-through and a photovoltaic inverter system
CN110048455A (zh) * 2019-04-24 2019-07-23 湖南大学 具有弱电网故障穿越能力的下垂控制逆变器及其控制方法
CN115622379A (zh) * 2022-09-26 2023-01-17 华为数字能源技术有限公司 故障穿越方法及变换器

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN101860038A (zh) * 2009-01-23 2010-10-13 通用电气公司 用于控制并网发电系统的系统和方法
US20180115255A1 (en) * 2016-10-20 2018-04-26 Ge Energy Power Conversion Technology Ltd Method for controlling a photovoltaic inverter for network fault ride-through and a photovoltaic inverter system
CN110048455A (zh) * 2019-04-24 2019-07-23 湖南大学 具有弱电网故障穿越能力的下垂控制逆变器及其控制方法
CN115622379A (zh) * 2022-09-26 2023-01-17 华为数字能源技术有限公司 故障穿越方法及变换器

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
"Doctoral Dissertation", 1 May 2017, HUAZHONG UNIVERSITY OF SCIENCE AND TECHNOLOGY, CN, article WANG, SHUO: "Basic Theory and Key Technology of Virtual Synchronous Controlled DFIG-based Wind Turbine", pages: 1 - 200, XP009554319 *

Also Published As

Publication number Publication date
CN115622379A (zh) 2023-01-17

Similar Documents

Publication Publication Date Title
EP2930840B1 (en) Power conversion device
Newman et al. A dynamic voltage restorer (DVR) with selective harmonic compensation at medium voltage level
US10084315B2 (en) Power conversion device with an autonomous operation function
JP6159271B2 (ja) 電力変換装置および電力変換装置の制御方法
WO2024066511A1 (zh) 故障穿越方法及变换器
US20180212460A1 (en) Uninterruptible power supply
JP6273874B2 (ja) 系統連系用電力変換装置の制御装置、及び系統連系用電力変換装置
US12074535B2 (en) Control device and power conversion device
CN113315401B (zh) 一种微型逆变器及其并网输出控制方法和装置
JP5580095B2 (ja) 系統連系インバータ装置
JP2009247162A (ja) 単相電圧型交直変換装置
US20240305096A1 (en) Microgrid System, Power Converter, and Control Method for Power Converter
JP6842953B2 (ja) 電力変換装置
CN112042072A (zh) 电力系统稳定化体系
WO2013123433A2 (en) Negative sequence current compensation controller and method for power conversion system
JP6361398B2 (ja) 系統連系用電力変換装置の制御装置、及び系統連系用電力変換装置
US11527966B2 (en) Power conversion apparatus that judges system power failure based on system frequency and voltage
JP2016032325A (ja) 系統連系用電力変換装置
US12068610B2 (en) Control device
EP4060883A1 (en) Power converter testing device and testing method
CN111130119B (zh) 一种空载线路合闸过电压抑制系统及其控制方法
JP7402775B2 (ja) 電力変換装置
JP5616412B2 (ja) 単相電圧型交直変換装置
JP6110093B2 (ja) インバータ装置、および制御方法
JPH11178350A (ja) 電源装置

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 23869759

Country of ref document: EP

Kind code of ref document: A1