WO2024051181A1 - 一种双向谐振型直流变换器的控制方法及控制电路 - Google Patents

一种双向谐振型直流变换器的控制方法及控制电路 Download PDF

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Publication number
WO2024051181A1
WO2024051181A1 PCT/CN2023/091097 CN2023091097W WO2024051181A1 WO 2024051181 A1 WO2024051181 A1 WO 2024051181A1 CN 2023091097 W CN2023091097 W CN 2023091097W WO 2024051181 A1 WO2024051181 A1 WO 2024051181A1
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Prior art keywords
circuit
resonant
converter
primary
bidirectional
Prior art date
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PCT/CN2023/091097
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English (en)
French (fr)
Inventor
王腾飞
庄加才
徐江涛
刘威
潘岱栋
徐君
Original Assignee
阳光电源股份有限公司
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Publication of WO2024051181A1 publication Critical patent/WO2024051181A1/zh

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present application relates to the field of power electronics technology, and in particular to a control method and control circuit of a bidirectional resonant DC converter.
  • bidirectional DC power conversion is gradually required by users and the market.
  • isolation systems are used between different energy systems, such as two-way vehicle power supplies, two-way charging piles, etc.
  • Commonly used bidirectional DC converters are mostly DAB (dual active bridge) topology and CLLC topology.
  • the DAB topology controls gain and power through phase shifting, but its soft switching range is limited.
  • the switching tube loss Large the CLLC topology is a resonant topology, which achieves gain and power control through frequency modulation and has high efficiency.
  • the existing CLLC resonant converter has its secondary side switch operating in a synchronous rectification state. If a wide range of voltage gain needs to be achieved, the switching frequency adjustment range will be large; in addition, with the development of devices, SiC ( The application of silicon carbide (SiC) devices and GaN (gallium nitride) devices is gradually increasing. When the device itself flows current in the reverse direction, if the drive is not turned on, it will generate a conduction voltage of 3V or even higher. At the same time, as the switching frequency increases, the reverse current will The proportion of current flow time in the entire switching cycle increases, which will produce larger conduction losses in synchronous rectification situations, increase the difficulty of device heat dissipation, and lead to a decrease in system power.
  • SiC silicon carbide
  • GaN gallium nitride
  • This application provides a control method and control circuit for a bidirectional resonant DC converter to reduce the adjustment range of switching frequency and reduce conduction loss when achieving a wide range of voltage gain.
  • the first aspect of this application provides a control method for a bidirectional resonant DC converter.
  • the bidirectional resonant DC converter includes: a transformer, a primary circuit, a secondary circuit, and a device provided between the transformer and the primary circuit and/or Or the resonant cavity between the secondary circuit; the primary circuit and the secondary circuit Both are single-phase full-bridge circuits, and each switch tube in the single-phase full-bridge circuit is equipped with an anti-parallel diode or body diode; the control method includes:
  • the switching frequency of the primary circuit and the secondary circuit is set to be greater than the resonant frequency of the resonant cavity
  • the delay times are equal and less than a preset value, so that each switch tube in the corresponding bridge arm of the secondary circuit remains on at zero voltage; or,
  • the method further includes:
  • a driving control signal of the primary circuit is generated and output according to the switching frequency.
  • the switching tubes of the two half-bridge arms in the same bridge arm are complementary and conductive;
  • the switching tubes of the half-bridge arms at different positions in different bridge arms have the same action.
  • the input electrical parameters and/or output electrical parameters include: at least one of input current, input voltage, output current and output voltage.
  • a second aspect of the present application provides a control circuit for a bidirectional resonant DC converter.
  • the bidirectional resonant DC converter includes: a transformer, a primary circuit, a secondary circuit, and a device provided between the transformer and the primary circuit and/or Or the resonant cavity between the secondary circuits; both the primary circuit and the secondary circuit are single-phase full-bridge circuits, and each switch tube in the single-phase full-bridge circuit has an anti-parallel diode. Or body diode;
  • the control circuit includes: primary side drive circuit, secondary side drive circuit, zero-crossing detection circuit, control module, and input sampling circuit and/or output sampling circuit; wherein,
  • the input sampling circuit is used to sample the input electrical parameters of the bidirectional resonant DC converter, and the output sampling circuit is used to sample the output electrical parameters of the bidirectional resonant DC converter;
  • the zero-crossing detection circuit is used to detect whether the current in the resonant cavity crosses zero and generate a zero-crossing signal
  • the control module is configured to receive the zero-crossing signal and the input electrical parameters and/or the output electrical parameters, execute the control method of the bidirectional resonant DC converter as described in any one of the above first aspects, and
  • the primary-side drive circuit controls the operation of each switch tube in the primary-side circuit, and the secondary-side drive circuit controls the operation of each switch tube in the secondary-side circuit.
  • the zero-crossing detection circuit is used to detect whether the current in the resonant cavity crosses zero, specifically for:
  • the input sampling circuit is used to sample the input current and input voltage on the DC side of the primary circuit
  • the output sampling circuit is used to sample the output current and output voltage on the DC side of the secondary circuit.
  • the resonant cavity includes: at least one resonant inductor module, and at least one resonant capacitor module;
  • the number of the resonant inductor module and the resonant capacitor module is both 1, they are respectively arranged on the primary side and the secondary side of the transformer, or both are arranged on the same side of the transformer;
  • each of the resonant inductor modules is installed on the primary side and the secondary side of the transformer respectively; when the number of the resonant capacitor modules is greater than 1, each of the resonant capacitor modules is installed on the primary side and the secondary side of the transformer respectively. Set on the primary side and secondary side of the transformer.
  • the control method of the bidirectional resonant DC converter controls the corresponding switch tube in each bridge arm of the secondary circuit to delay for a corresponding period of time before turning off after detecting the commutation of the secondary resonant current. After turning off, the current is commutated to the anti-parallel diode or body diode of the complementary switch tube in the same bridge arm. Therefore, after the corresponding dead time, the complementary switch tube in each bridge arm of the secondary circuit is controlled to conduct, so that the secondary circuit Each switch tube in the switch only has a working condition where the reverse current is not turned on during the dead time, and the conduction loss caused by this is greatly reduced.
  • the delay time of each bridge arm of the secondary circuit is determined based on the input electrical parameters and/or output electrical parameters of the bidirectional resonant DC converter and the required gain, so
  • the switching frequency of the primary circuit and the secondary circuit is set to be greater than the resonant frequency of the resonant cavity, the gain required by the bidirectional resonant DC converter can also be achieved; thereby reducing the frequency adjustment range in wide voltage output situations, It has a significant effect on increasing the switching frequency.
  • Figure 1 is a schematic structural diagram of a bidirectional resonant DC converter and its control circuit provided by an embodiment of the present application;
  • Figure 2a is a schematic structural diagram of a resonant cavity provided by an embodiment of the present application.
  • Figure 2b is another structural schematic diagram of the resonant cavity provided by the embodiment of the present application.
  • Figure 2c is another structural schematic diagram of the resonant cavity provided by the embodiment of the present application.
  • Figure 2d is another structural schematic diagram of the resonant cavity provided by the embodiment of the present application.
  • Figure 2e is another structural schematic diagram of the resonant cavity provided by the embodiment of the present application.
  • Figure 3 is a schematic structural diagram of a specific circuit of the bidirectional resonant DC converter and the control circuit of the bidirectional resonant DC converter provided by the embodiment of the present application;
  • Figure 4 is a flow chart of a control method for a bidirectional resonant DC converter provided by an embodiment of the present application
  • FIG. 5 is a signal waveform diagram of the control method of the bidirectional resonant DC converter provided by the embodiment of the present application.
  • This application provides a control method for a bidirectional resonant DC converter to reduce the adjustment range of the switching frequency and reduce the conduction loss when achieving a wide range of voltage gain.
  • the bidirectional resonant DC converter includes: a transformer T, a primary circuit 101, a secondary circuit 102, and a resonant cavity disposed between the transformer T and the primary circuit 101 and/or the secondary circuit 102.
  • Figure 1 takes the resonant cavity 103 disposed between the transformer T and the primary circuit 101 as an example.
  • the resonant cavity 103 can also be disposed between the transformer T and the secondary circuit 102, or the resonant cavity 103 can be disposed between the transformer T and the secondary circuit 102.
  • Each resonant element in the cavity 103 can also be disposed on both sides of the transformer T respectively.
  • the resonant cavity 103 includes: at least one resonant inductor module and at least one resonant capacitor module; and when the number of the resonant inductor module Lr and the resonant capacitor module Cr is both 1, they are respectively disposed at The primary side and secondary side of the transformer T (as shown in Figure 2a), or they are both arranged on the same side of the transformer T (as shown in Figure 2b); when the number of resonant inductor modules is greater than 1, each resonant inductor module is Set on the primary and secondary sides of the transformer T (Lrp and Lrs shown in Figure 2c, Figure 2d and Figure 2e); when the number of resonant capacitor modules is greater than 1, each resonant capacitor module is set on the transformer T respectively.
  • the primary side and the secondary side (Crp and Crs as shown in Figure 2e and Figure 3); the case where the resonant cavity 103 includes two resonant capacitor modules Crp and Crs and a resonant inductor module Lrs arranged on the secondary side is not carried out. shown in the figure, but they are all within the protection scope of this application.
  • the primary circuit 101 and the secondary circuit 102 are both single-phase full-bridge circuits, as shown in FIG. 3 , and each switch tube in the two single-phase full-bridge circuits has an anti-parallel diode or a body diode.
  • the bidirectional resonant DC converter can be a single-phase resonant converter of the CLLC type, LLC type and SRC type in the prior art.
  • CLLC type, LLC type and SRC type in the prior art.
  • the specific structure and connection relationship can be found in the prior art, and will not be discussed one by one here. Repeat.
  • the input electrical parameters and/or output electrical parameters may specifically include: at least one of input current, input voltage, output current and output voltage of the bidirectional resonant DC converter.
  • the input of the DC side of the primary circuit 101 can be sampled through the input sampling circuit 203 shown in Figure 1 current and input voltage, that is, the input current and input voltage of the bidirectional resonant DC converter; and the output current and output voltage of the DC side of the secondary circuit 102 are sampled through the output sampling circuit 204, that is, the bidirectional resonant DC converter of output current and output voltage.
  • the input sampling circuit 203 and the output sampling circuit 204 respectively transmit the respective sampled signals to the control module 201, and the control module 201 can obtain the input current, input voltage, output current and output voltage of the bidirectional resonant DC converter based on them. .
  • control module 201 can determine the respective delay times dt1 and dt2 of the two rectifier bridge arms in the secondary circuit based on one or more signals of the input electrical parameters and/or the output electrical parameters; and, both The delay times dt1 and dt2 may be equal or unequal, depending on the specific application environment, and both are within the protection scope of this application.
  • the gain of the bidirectional resonant DC converter is greater than 1; when the switching frequency is greater than the resonant frequency, the gain of the bidirectional resonant DC converter is The gain of the converter is less than 1; if the wide-range voltage gain requirement is to be met, the switching frequency adjustment range in the existing technology is larger.
  • step S103 the control module 201 in Figure 1 limits the adjustment range of the switching frequency to above the resonant frequency based on the input electrical parameters and/or output electrical parameters and the internal preset reference signal, and through step S102 Corresponding delay times dt1 and dt2 are respectively set for the two bridge arms in the secondary circuit 102 so that the bidirectional resonant DC converter can achieve its required gain, thereby achieving a wide range of voltage gain with a smaller switching frequency adjustment range.
  • steps S102 and S103 are not limited, and they can be executed one after another or at the same time, depending on the specific application environment, and both are within the protection scope of this application.
  • the control module 201 can generate and output the driving control signal of the primary side circuit 101 according to the switching frequency, and realize driving of the primary side circuit 101 through the primary side driving circuit 205 in FIG. 1 .
  • the control module 201 can also generate and output the drive control signal of the secondary side circuit 102 according to the switching frequency and the two delay times dt1 and dt2, and pass the secondary side driver in Figure 1
  • the dynamic circuit 206 realizes the driving of the secondary circuit 102; the specific control process of the secondary circuit 102 by the control module 201 is step S104.
  • the primary side resonance current Ip and/or the secondary side resonance current Is of the bidirectional resonant DC converter can be detected through the zero-crossing detection circuit 202 shown in FIG. 1, and the control module 201 can detect the The detection result of 202 is used to judge whether the secondary resonance current Is crosses zero; in practical applications, if the excitation inductance of the transformer T is large, the zero-crossing position difference between the original secondary resonance current Ip and Is is very small. At this time, it can A corresponding detection circuit is only provided on one side of the transformer T for zero-crossing detection to save costs. It is preferable to only perform zero-crossing detection on the secondary side resonant current Is.
  • the turn-off time of the turned-off switch tube is set to: the time after the corresponding delay time starting from the zero-crossing moment, and then after the dead time after the turn-off time, the corresponding complementary switches in the secondary circuit are controlled.
  • the switch tube is conducting.
  • the primary circuit 101 and the secondary circuit 102 are both single-phase full-bridge circuits.
  • the two in the same bridge arm The switching tubes of the half-bridge arms are complementary to each other and conductive; moreover, in different bridge arms of the primary circuit 101, the switching tubes of the half-bridge arms at different positions have the same action.
  • the primary circuit 101 includes a first bridge arm and a second bridge arm connected in parallel; wherein, the first bridge arm is composed of a first switch tube S1 and a second switch tube S2 connected in series, and their drives are complementary and are 50 %; the second bridge arm is composed of the third switch tube S3 and the fourth switch tube S4 connected in series, and their driving signals are complementary and equal to 50%; at the same time, the driving signals of the first switch tube S1 and the third switch tube S3 are the same.
  • the secondary circuit 102 includes a third bridge arm and a fourth bridge arm connected in parallel; wherein, the third bridge arm is composed of a fifth switch transistor S5 and a sixth switch transistor S6 connected in series, and their drive is complementary and is 50%; The four bridge arms are composed of a seventh switching tube S7 and an eighth switching tube S8 connected in series, and their driving is complementary and is 50%.
  • the resonant element (Crs) on the secondary side and the secondary winding of the transformer T are connected between the two nodes.
  • the zero-crossing detection circuit 202 detects the zero-crossing of the secondary resonant current
  • a zero-crossing signal is generated and transmitted to the control module 201.
  • the control module 201 uses the zero-crossing moment as the starting point of the delay, and delays two corresponding delays respectively.
  • the driving signal is sent to the third bridge arm and the fourth bridge arm of the secondary side circuit 102 via the secondary side driving circuit 206; the delay times dt1 and dt2 may be equal or different.
  • the two bridge arm switches in the primary side circuit 101 operate at time t0, and the secondary side resonant current Is is detected to cross zero at time t1, and a delay time dt1 to time t2 is delayed to make the fourth bridge arm seventh
  • the switch S7 and the eighth switch S8 operate, and a delay time dt2 to t3 is delayed to cause the fifth switch S5 and the sixth switch S6 of the third bridge arm to operate.
  • t2-t1 dt1
  • t3-t1 dt2.
  • the gain is less than 1; if the output gain required by the bidirectional resonant DC converter is less than 1, the two delay times dt1 and dt2 can be made equal; and the two delay times dt1 and dt2 are all smaller than the first preset value, so that each switch tube in the corresponding bridge arm of the secondary circuit 102 remains on at zero voltage; in addition, in order to ensure that each switch tube in the corresponding bridge arm of the primary circuit 101 also remains on at zero voltage, at the same time
  • the first preset value is generally less than 1 us, and its specific value is not limited and depends on its specific application environment.
  • the two delay times dt1 and dt2 can be made different.
  • the delay time dt2 can be set to be less than 1/4 of the switching period; and the delay of the fourth bridge arm
  • the time dt1 is less than a second preset value, that is, the delay time dt1 is as small as possible, in order to achieve soft switching of the fourth bridge arm as the primary condition; in addition, in order to ensure that each switch tube in the corresponding bridge arm of the primary side circuit 101 maintains zero voltage Turn on, and at the same time reduce
  • the delay time dt1 of the fourth bridge arm can also be set to change in the same direction as the gain required by the bidirectional resonant DC converter; while the delay time dt2 of the third bridge arm can be as small as possible.
  • the soft switching of the third bridge arm can be achieved. That is, when the two delay times dt1 and dt2 are not equal, As long as there is one delay time to meet the gain requirement, the other delay time only needs to implement the soft switching of the corresponding bridge arm; how to set the two delay times specifically depends on the application environment, both of which are described in this application. within the scope of protection.
  • the soft switching implementation of other switch tubes in the primary circuit 101 is the same as that of the first switch tube S1.
  • the secondary circuit 102 when the fourth bridge arm is operating (time t2), the secondary resonant current Is has completed commutation at time t1, and the secondary resonant current Is flows from the other end of the eighth switch S8 to the second switch S8.
  • the drive of the eighth switch S8 is turned off, the secondary resonant current Is is reversed to the seventh switch S7, and the current flow direction becomes from the other end of the seventh switch S7 to the fourth node D.
  • the seventh switch S7 is driven to open, realizing zero-voltage turn-on of the seventh switch S7, that is, soft switching is realized.
  • the soft switching of the eighth switch S8 is the same as that of the seventh switch S7.
  • the original secondary switch tube only has a reverse current flow process during the dead time between the complementary switch tube off and its own drive is not turned on. During other times when the current flows, the drive is in the open state, so the drive is not turned on.
  • the reverse current flow process is limited to the dead time. In practical applications, by reasonably controlling the dead time, the reverse current flow time when the drive is not turned on can be controlled, thereby reducing the reverse conduction voltage drop and conduction loss.
  • the control method of the bidirectional resonant DC converter provided in this embodiment is to control the corresponding switch tubes in each bridge arm of the secondary circuit to delay for a corresponding period of time before turning off after detecting the commutation of the secondary resonant current. Since the current after shutdown is commutated to the anti-parallel diode or body diode of the complementary switch tube in the same bridge arm, after the corresponding dead time, the complementary switch tube in each bridge arm of the secondary circuit is controlled to conduct, so that the secondary side Each switch tube in the circuit only has a working condition where the reverse current is not turned on during the dead time, and the conduction loss caused by this is greatly reduced.
  • the delay time of each bridge arm of the secondary circuit is determined based on the input electrical parameters and/or output electrical parameters of the bidirectional resonant DC converter and the required gain, so
  • the gain required by the bidirectional resonant DC converter can also be achieved; thereby reducing the frequency adjustment range in wide voltage output situations. , which has a significant effect on increasing the switching frequency.
  • resonance can achieve soft switching, switching losses can be effectively reduced, further improving frequency and system efficiency.
  • the above control method is when the bidirectional resonant DC converter transmits power from the primary side to the secondary side.
  • the implementation method is the same as that of the primary side. It is the same when transmitting from side to secondary side. It can be seen as exchanging the names of the primary side and the secondary side. That is, the power input side is named the primary side and the power output side is named the secondary side. Then the above control method is used. There is no Any further descriptions are within the protection scope of this application.
  • the bidirectional resonant DC converter includes: a transformer T, a primary circuit 101 , the secondary circuit 102 and the resonant cavity 103 disposed between the transformer T and the primary circuit 101 and/or the secondary circuit 102;
  • Figure 1 shows the resonant cavity 103 being disposed between the transformer T and the primary circuit 101 as an example.
  • the resonant cavity 103 can also be disposed between the transformer T and the secondary circuit 102, or each resonant element in the resonant cavity 103 can be disposed on both sides of the transformer T respectively.
  • the resonant cavity 103 includes: at least one resonant inductor module and at least one resonant capacitor module; and when the number of the resonant inductor module Lr and the resonant capacitor module Cr is both 1, they are respectively disposed at The primary side and secondary side of the transformer T (as shown in Figure 2a), or they are both arranged on the same side of the transformer T (as shown in Figure 2b); when the number of resonant inductor modules is greater than 1, each resonant inductor module is Set on the primary and secondary sides of the transformer T (Lrp and Lrs shown in Figure 2c, Figure 2d and Figure 2e); when the number of resonant capacitor modules is greater than 1, each resonant capacitor module is set on the transformer T respectively.
  • the primary side and the secondary side (Crp and Crs as shown in Figure 2e and Figure 3); the case where the resonant cavity 103 includes two resonant capacitor modules Crp and Crs and a resonant inductor module Lrs arranged on the secondary side is not carried out. shown in the figure, but they are all within the protection scope of this application.
  • the number and position of the resonant inductor module and the resonant capacitor module can be modified according to the needs of the circuit; moreover, the resonant inductor module is generally implemented with one inductor, and the realization form of multiple inductors in series and parallel is not excluded; the resonance
  • the capacitor module can be implemented by one capacitor or multiple capacitors connected in series and parallel; It depends on the specific application environment, and they are all within the protection scope of this application. Taking the situation shown in Figure 2b and Figure 2e as an example to illustrate, the relationship between parameters under the two structures is:
  • Np is the number of turns of the primary winding of transformer T
  • Ns is the number of turns of the secondary winding of transformer T.
  • the primary circuit 101 and the secondary circuit 102 are both single-phase full-bridge circuits, as shown in FIG. 3 , and each switch tube in the two single-phase full-bridge circuits has an anti-parallel diode or a body diode.
  • the bidirectional resonant DC converter can be a single-phase resonant converter of the CLLC type, LLC type and SRC type in the prior art.
  • CLLC type, LLC type and SRC type in the prior art.
  • the specific structure and connection relationship can be found in the prior art, and will not be discussed one by one here. Repeat.
  • control circuit 205 includes: primary side drive circuit 205, secondary side drive circuit 206, zero-crossing detection circuit 202, control module 201, and , input sampling circuit 203 and/or output sampling circuit 204 (both shown in Figures 1 and 3 as examples of including both at the same time); where:
  • the input sampling circuit 203 is used to sample the input electrical parameters of the bidirectional resonant DC converter; the output sampling circuit 204 is used to sample the output electrical parameters of the bidirectional resonant DC converter.
  • the input sampling circuit 203 is disposed on the DC side of the primary circuit 101, and the output end is connected to an input end of the control module 201.
  • the input sampling circuit 203 is specifically used to sample the input current of the bidirectional resonant DC converter. and the input voltage;
  • the output sampling circuit 204 is arranged on the DC side of the secondary circuit 102, and the output terminal is connected to the other input terminal of the control module 201.
  • the output sampling circuit 204 is specifically used to sample the bidirectional resonant DC converter. output current and output voltage.
  • the zero-crossing detection circuit 202 is used to detect whether the current in the resonant cavity 103 crosses zero, and when the zero-crossing information indicates that the secondary resonance current Is of the bidirectional resonant DC converter crosses zero, it generates a zero-crossing signal and outputs it to the control module 201 the corresponding input terminal. Since both the primary side resonant current Ip and the secondary side resonant current Is cross zero, it can indicate that the secondary side resonant current Is crosses zero.
  • the zero-crossing detection circuit 202 detects the resonant cavity 103 When the current in the DC converter crosses zero, the specific method may be: detecting whether the secondary resonance current Is and/or the primary resonance current Ip of the bidirectional resonant DC converter crosses zero. That is, in practical applications, the zero-crossing detection circuit 202 can determine the delay starting point of each bridge arm in the secondary circuit 102 by detecting the zero-crossing condition of the primary resonant current Ip and/or the secondary resonant current Is.
  • a corresponding detection circuit can be set up on only one side of the transformer T for zero-crossing detection, thereby saving costs. It is preferable to detect the secondary resonant current.
  • the current Is performs zero-crossing detection.
  • the control module 201 adopts the control method provided by the above embodiment, since under various output gain requirements, the switching frequency is greater than the resonant frequency, that is, the bidirectional resonant DC converter operates in an over-resonant state, and its converter
  • the excitation inductance of T does not participate in the work, so the excitation inductance can be very large, which facilitates the zero-crossing detection circuit 202 to realize the above-mentioned zero-crossing detection at low cost.
  • the control module 201 is used to receive the zero-crossing signal and the input electrical parameters and/or output electrical parameters, execute the control method of the bidirectional resonant DC converter as described in any of the above embodiments, and pass the primary side drive circuit 205
  • the operation of each switching tube in the primary side circuit 101 is controlled, and the operation of each switching tube in the secondary side circuit 102 is controlled through the secondary side driving circuit 206 .
  • the specific process and principle of the control method can be referred to the above embodiments, and will not be described again here.
  • the tasks of each module in the control circuit are:
  • the input sampling circuit 203 and the output sampling circuit 204 sample the input and output side voltages and currents of the bidirectional resonant DC converter, and transmit the sampled sampling signals to the control module 201, so that the control module 201 obtains the above-mentioned Input electrical parameters and/or output electrical parameters.
  • the control module 201 determines the respective delay times dt1 and dt2 of the two bridge arms on the secondary side based on one or more signals among the above-mentioned input electrical parameters and/or output electrical parameters.
  • the control module 201 generates the switching frequency of the primary and secondary circuits based on the above-mentioned input electrical parameters and/or output electrical parameters and the internal preset reference signal. Each switching tube of the primary circuit 101 will directly generate the switching frequency based on the control module 201 operating switching frequency.
  • the zero-crossing detection circuit 202 detects the primary side resonant current Ip and/or the secondary side resonant current Is, preferably the secondary side rectified current (that is, the secondary side resonant current Is). When its zero-crossing is detected, the zero-crossing The detection circuit 202 outputs a zero-crossing signal to the control module 201, so that the control module 201 can determine the zero-crossing of the secondary resonant current Is and obtain its zero-crossing moment.
  • control module 201 After receiving the zero-crossing signal, the control module 201 delays two delay times dt1 and dt2 respectively, and outputs a drive control signal that controls the actions of the fourth bridge arm and the third bridge arm.
  • the control module 201 generates drive control signals for the primary circuit 101 and the secondary circuit 102 respectively.
  • the drive control signal of the primary circuit 101 generates the drive signal of the primary circuit 101 through the primary drive circuit 205 for control.
  • Each switch tube of the primary circuit 101 operates; the drive control signal of the secondary circuit 102 is generated by the secondary drive circuit 206 to generate a drive signal of the secondary circuit 102, which is used to control the operation of each switch tube of the secondary circuit 102.
  • the control circuit provided in this embodiment realizes soft switching of the secondary side switch tube by performing resonance zero-crossing detection on the bidirectional resonant DC converter and combining it with delay time control. Compared with traditional synchronous rectification control, it is more effective. It reduces the duration of reverse current flowing when the driver is not turned on, and has obvious efficiency advantages in synchronous rectification applications of SiC and GaN devices; at the same time, combined with the control of the delay time, the output voltage gain is changed, thereby reducing the need for wide voltage output applications.
  • the frequency adjustment range has a significant effect on increasing the switching frequency.

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Abstract

本申请提供一种双向谐振型直流变换器的控制方法及控制电路，该控制方法在检测到副边谐振电流换向后，控制副边电路各桥臂中相应开关管分别延时一段相应的时间后再关断，由于关断后电流换流至同桥臂内互补开关管的反并二极管或体二极管，所以经过相应死区时间后，再控制副边电路各桥臂中互补的开关管导通，使得副边电路中各开关管仅在死区时间内存在驱动未开通反向走电流的工况，因此导通损耗降低。而且，各延时时间是根据双向谐振型直流变换器的输入电参数和/或输出电参数和所需的增益确定得到的，所以在设置开关频率大于谐振频率的情况下，也可以实现所需的增益；进而可以减小宽电压输出场合的频率调节范围，对于提高开关频率具有明显作用。

Description

一种双向谐振型直流变换器的控制方法及控制电路
本申请要求于2022年09月07日提交中国专利局、申请号为202211089489.5、发明名称为“一种双向谐振型直流变换器的控制方法及控制电路”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本申请涉及电力电子技术领域，特别涉及一种双向谐振型直流变换器的控制方法及控制电路。
背景技术
在能源互连的发展趋势下，不同储能系统间的能量交互变得普遍，其中，双向直流功率变换需求逐渐被用户和市场所需要。通常为保证安全，不同能量系统间会采用隔离系统，如双向车载电源，双向充电桩等。常用的双向直流变换器多为DAB(双有源桥)拓扑和CLLC拓扑，其中DAB拓扑通过移相控制增益和功率，但其软开关范围受限，在电池电压范围较宽时，开关管损耗大；CLLC拓扑则是谐振型拓扑，通过调频实现增益和功率控制，效率较高。
但是，现有的CLLC谐振变换器，其副边开关管工作在同步整流状态，若需要实现宽范围电压增益，则会导致其开关频率的调节范围较大；另外，随着器件发展，SiC(碳化硅)器件、GaN(氮化镓)器件应用逐渐增加，其器件本身反向走电流时，若未开通驱动将会产生3V甚至更高的导通电压，同时随着开关频率提高，反向走电流时间占比整个开关周期中的占比增大，在同步整流场合下将产生较大的导通损耗，增加了器件散热的困难，并导致系统功率下降。
因此，如何发展一种改善上述技术问题的控制方案，成为一项迫切需求。
发明内容
本申请提供一种双向谐振型直流变换器的控制方法及控制电路，以在实现宽范围电压增益时降低开关频率的调节范围，并降低导通损耗。
为实现上述目的，本申请提供如下技术方案：
本申请第一方面提供了一种双向谐振型直流变换器的控制方法，双向谐振型直流变换器包括：变压器、原边电路、副边电路以及设置于所述变压器与所述原边电路和/或所述副边电路之间的谐振腔；所述原边电路和所述副边电路 均为单相全桥电路，且所述单相全桥电路中的各开关管均带有反并联二极管或体二极管；所述控制方法包括：
获取所述双向谐振型直流变换器的输入电参数和/或输出电参数；
根据所述输入电参数和/或输出电参数和所述双向谐振型直流变换器所需的增益，确定所述副边电路中各桥臂的延时时间；
根据所述输入电参数和/或输出电参数以及预设参考信号，设置所述原边电路和所述副边电路的开关频率大于所述谐振腔的谐振频率；
在所述双向谐振型直流变换器的副边谐振电流过零时，以过零时刻为起点，在各所述延时时间之后分别进行对所述副边电路中相应开关管的关断控制；再在死区时间后，控制所述副边电路中互补的开关管导通。
可选的，各所述延时时间相等，且小于预设值，使所述副边电路对应桥臂中的各开关管保持零电压开通；或者，
所述双向谐振型直流变换器所需的增益大于1时，各所述延时时间不等。
可选的，各所述延时时间不等时，存在一个所述延时时间，随所述双向谐振型直流变换器所需的增益同向变化。
可选的，在设置所述原边电路和所述副边电路的开关频率大于所述谐振腔的谐振频率之后，还包括：
根据所述开关频率生成并输出所述原边电路的驱动控制信号。
可选的，所述单相全桥电路中，同一桥臂中两半桥臂的开关管互补导通；
所述原边电路中，不同桥臂中不同位置半桥臂的开关管动作相同。
可选的，所述输入电参数和/或输出电参数，包括：输入电流、输入电压、输出电流及输出电压中的至少一种。
本申请第二方面提供了一种双向谐振型直流变换器的控制电路，双向谐振型直流变换器包括：变压器、原边电路、副边电路以及设置于所述变压器与所述原边电路和/或所述副边电路之间的谐振腔；所述原边电路和所述副边电路均为单相全桥电路，且所述单相全桥电路中的各开关管均带有反并联二极管或体二极管；所述控制电路包括：原边驱动电路、副边驱动电路、过零检测电路、控制模块以及输入采样电路和/或输出采样电路；其中，
所述输入采样电路用于采样所述双向谐振型直流变换器的输入电参数，所述输出采样电路用于采样所述双向谐振型直流变换器的输出电参数；
所述过零检测电路用于检测所述谐振腔中的电流是否过零，并生成过零信号；
所述控制模块用于接收所述过零信号及所述输入电参数和/或所述输出电参数，执行如上述第一方面任一种所述的双向谐振型直流变换器的控制方法，并通过所述原边驱动电路控制所述原边电路中各开关管动作，通过所述副边驱动电路控制所述副边电路中各开关管动作。
可选的，所述过零检测电路用于检测所述谐振腔中的电流是否过零时，具体用于：
检测所述双向谐振型直流变换器的副边谐振电流和/或原边谐振电流是否过零。
可选的，所述输入采样电路用于采样所述原边电路直流侧的输入电流和输入电压；
所述输出采样电路用于采样所述副边电路直流侧的输出电流和输出电压。
可选的，所述谐振腔，包括：至少一个谐振电感模块，以及，至少一个谐振电容模块；
所述谐振电感模块与所述谐振电容模块的个数均为1时，两者分别设置于所述变压器的原边和副边，或者，均设置于所述变压器的同一边；
所述谐振电感模块的个数大于1时，各所述谐振电感模块分别设置于所述变压器的原边和副边；所述谐振电容模块的个数大于1时，各所述谐振电容模块分别设置于所述变压器的原边和副边。
本申请提供的双向谐振型直流变换器的控制方法，其在检测到副边谐振电流换向后，控制副边电路各桥臂中相应开关管分别延时一段相应的时间后再关断，由于关断后电流换流至同桥臂内互补开关管的反并二极管或体二极管，所以经过相应死区时间后，再控制副边电路各桥臂中互补的开关管导通，使得副边电路中各开关管仅在死区时间内存在驱动未开通反向走电流的工况，由此带来的导通损耗大大降低。而且，其副边电路各桥臂的延时时间是根据双向谐振型直流变换器的输入电参数和/或输出电参数和所需的增益确定得到的，所以 在设置原边电路和副边电路的开关频率大于谐振腔的谐振频率的情况下，也可以实现该双向谐振型直流变换器所需的增益；进而可以减小宽电压输出场合的频率调节范围，对于提高开关频率具有明显作用。
附图说明
为了更清楚地说明本发明实施例或现有技术中的技术方案，下面将对实施例或现有技术描述中要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本发明的实施例，对于本领域普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据提供的附图获得其他的附图。
图1为本申请实施例提供的双向谐振型直流变换器及其控制电路的结构示意图；
图2a为本申请实施例提供的谐振腔的一种结构示意图；
图2b为本申请实施例提供的谐振腔的另一种结构示意图；
图2c为本申请实施例提供的谐振腔的另一种结构示意图；
图2d为本申请实施例提供的谐振腔的另一种结构示意图；
图2e为本申请实施例提供的谐振腔的另一种结构示意图；
图3为本申请实施例提供的双向谐振型直流变换器的一种具体电路及双向谐振型直流变换器的控制电路的结构示意图；
图4为本申请实施例提供的双向谐振型直流变换器的控制方法的流程图；
图5为本申请实施例提供的双向谐振型直流变换器的控制方法的信号波形图。
具体实施方式
下面将结合本发明实施例中的附图，对本发明实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例仅仅是本发明一部分实施例，而不是全部的实施例。基于本发明中的实施例，本领域普通技术人员在没有做出创造性劳动前提下所获得的所有其他实施例，都属于本发明保护的范围。
在本申请中，术语“包括”、“包含”或者其任何其他变体意在涵盖非排他性的包含，从而使得包括一系列要素的过程、方法、物品或者设备不仅包括那些要素，而且还包括没有明确列出的其他要素，或者是还包括为这种过程、方法、物品或者设备所固有的要素。在没有更多限制的情况下，由语句“包括 一个……”限定的要素，并不排除在包括要素的过程、方法、物品或者设备中还存在另外的相同要素。
本申请提供一种双向谐振型直流变换器的控制方法，以在实现宽范围电压增益时降低开关频率的调节范围，并降低导通损耗。
该双向谐振型直流变换器，如图1所示，包括：变压器T、原边电路101、副边电路102以及设置于变压器T与原边电路101和/或副边电路102之间的谐振腔103；图1以谐振腔103设置于变压器T与原边电路101之间为例进行展示，实际应用中，该谐振腔103也可以设置于变压器T与副边电路102之间，或者，该谐振腔103中的各谐振元件还可以分别设置于变压器T的两边。具体的，该谐振腔103包括：至少一个谐振电感模块，以及，至少一个谐振电容模块；而且，当该谐振电感模块Lr与该谐振电容模块Cr的个数均为1时，两者分别设置于变压器T的原边和副边(如图2a所示)，或者，均设置于变压器T的同一边(如图2b所示)；当谐振电感模块的个数大于1时，各谐振电感模块分别设置于变压器T的原边和副边(如图2c、图2d及图2e中所示的Lrp和Lrs)；当谐振电容模块的个数大于1时，各谐振电容模块分别设置于变压器T的原边和副边(如图2e和图3中所示的Crp和Crs)；该谐振腔103中包括两个谐振电容模块Crp和Crs以及一个设置于副边的谐振电感模块Lrs的情况未进行图示，但均在本申请的保护范围内。而且，该原边电路101和副边电路102均为单相全桥电路，如图3中所示，且两个单相全桥电路中的各开关管均带有反并联二极管或体二极管。也即，该双向谐振型直流变换器可以为现有技术中的CLLC型、LLC型和SRC型的单相谐振变换器，其具体结构及连接关系可以参见现有技术，此处不再一一赘述。
不论该双向谐振型直流变换器采用上述何种拓扑，其控制方法，均可参见图4，具体包括：
S101、获取双向谐振型直流变换器的输入电参数和/或输出电参数。
该输入电参数和/或输出电参数，具体可以包括：该双向谐振型直流变换器的输入电流、输入电压、输出电流及输出电压中的至少一种。实际应用中，具体可以通过图1中所示的输入采样电路203采样原边电路101直流侧的输入 电流和输入电压，也即该双向谐振型直流变换器的输入电流和输入电压；并通过输出采样电路204采样副边电路102直流侧的输出电流和输出电压，也即该双向谐振型直流变换器的输出电流和输出电压。
输入采样电路203和输出采样电路204分别将各自采样得到的采样信号传输给控制模块201，控制模块201可以根据其获取到该双向谐振型直流变换器的输入电流、输入电压、输出电流及输出电压。
S102、根据输入电参数和/或输出电参数以及双向谐振型直流变换器所需的增益，确定副边电路中各桥臂的延时时间。
实际应用中，该控制模块201具体可以依据该输入电参数和/或输出电参数中的一种或几种信号确定副边电路中两整流桥臂各自的延时时间dt1与dt2；而且，两个延时时间dt1与dt2可以相等，也可以不相等，视其具体应用环境而定即可，均在本申请的保护范围内。
S103、根据输入电参数和/或输出电参数以及预设参考信号，设置原边电路和副边电路的开关频率大于谐振腔的谐振频率。
当原边电路101和副边电路102的开关频率小于谐振腔103的谐振频率时，该双向谐振型直流变换器的增益大于1；当该开关频率大于该谐振频率时，该双向谐振型直流变换器的增益小于1；若要满足宽范围电压增益要求，则现有技术中开关频率的调节范围较大。
本实施例通过步骤S103，由图1中的控制模块201依据该输入电参数和/或输出电参数以及内部的预设参考信号，将开关频率的调节范围限定为谐振频率以上，并通过步骤S102为副边电路102中两桥臂分别设置相应的延时时间dt1与dt2，使双向谐振型直流变换器实现其所需的增益，进而以较小的开关频率调节范围实现了宽范围电压增益。
实际应用中，步骤S102和S103的顺序不限，可以先后执行，也可以同时执行，视其具体应用环境而定即可，均在本申请的保护范围内。
在得到该开关频率之后，控制模块201即可根据该开关频率生成并输出原边电路101的驱动控制信号，并通过图1中的原边驱动电路205实现对于原边电路101的驱动。同时，控制模块201还可以根据该开关频率和两个延时时间dt1与dt2，生成并输出副边电路102的驱动控制信号，并通过图1中的副边驱 动电路206实现对于副边电路102的驱动；控制模块201对于副边电路102的具体控制过程也即步骤S104。
S104、在双向谐振型直流变换器的副边谐振电流过零时，以过零时刻为起点，在各延时时间之后分别进行对副边电路中相应开关管的关断控制；再在死区时间后，控制副边电路中互补的开关管导通。
具体的，可以通过图1中所示的过零检测电路202来检测该双向谐振型直流变换器的原边谐振电流Ip和/或副边谐振电流Is，并由控制模块201根据过零检测电路202的检测结果来实现对于副边谐振电流Is是否过零的判断；实际应用中，若变压器T励磁电感很大，则原副边谐振电流Ip和Is的过零位置相差很小，此时可以仅在变压器T的一边设置相应的检测电路进行过零检测，节约成本，优选仅对该副边谐振电流Is进行过零检测。
对于上述各种拓扑形式的谐振型直流变换器，只要其副边谐振电流过零，则区别于现有技术中对于副边电路的同步整流控制，本实施例均将副边电路中即将被控关断的开关管的关断时刻设置为：以过零时刻为起点的对应延时时间之后的时刻，然后再在该关断时刻之后的死区时间后，控制副边电路中各对应互补的开关管导通。
以图3所示CLLC型的单相谐振变换器为例进行说明，原边电路101和副边电路102均为单相全桥电路，这两个单相全桥电路中，同一桥臂中两半桥臂的开关管互补导通；而且，原边电路101的不同桥臂中，不同位置半桥臂的开关管动作相同。具体的，该原边电路101包含并联连接的第一桥臂与第二桥臂；其中，第一桥臂由第一开关管S1与第二开关管S2串联连接组成，其驱动互补且为50％；第二桥臂由第三开关管S3与第四开关管S4串联连接组成，其驱动互补且为50％；同时，第一开关管S1与第三开关管S3的驱动信号相同。第一开关管S1与第二开关管S2之间具有第一节点A，第三开关管S3与第四开关管S4之间具有第二节点B，两节点之间连接原边侧的谐振元件(Lrp和Crp)以及变压器T的原边绕组。该副边电路102包含并联连接的第三桥臂与第四桥臂；其中，第三桥臂由第五开关管S5与第六开关管S6串联连接组成，其驱动互补且为50％；第四桥臂由第七开关管S7与第八开关管S8串联连接组成，其驱动互补且为50％。第五开关管S5与第六开关管S6之间具有第三 节点C，第七开关管S7与第八开关管S8之间具有第四节点D，两节点之间连接副边侧的谐振元件(Crs)以及变压器T的副边绕组。
在过零检测电路202检测到副边谐振电流过零时，生成过零信号并传输到控制模块201，然后由控制模块201将过零时刻作为延时起点，分别延时两个对应的延时时间dt1与dt2，经副边驱动电路206将驱动信号送至副边电路102的第三桥臂与第四桥臂；延时时间dt1与dt2可以相等也可以不等。
如图5所示，原边电路101中两桥臂开关管在t0时刻动作，t1时刻检测到副边谐振电流Is过零，延时一个延时时间dt1到t2时刻使第四桥臂第七开关管S7与第八开关管S8动作，延时一个延时时间dt2到t3时刻使第三桥臂第五开关管S5与第六开关管S6动作。其中，t2-t1＝dt1，t3-t1＝dt2。
谐振腔103在开关频率大于谐振频率时，增益小于1；若双向谐振型直流变换器需要的输出增益小于1，则可使两个延时时间dt1与dt2相等；且两个延时时间dt1与dt2均小于第一预设值，以使副边电路102对应桥臂中的各开关管保持零电压开通；另外，为保证原边电路101对应桥臂中各开关管也保持零电压开通，同时降低副边电路102对应桥臂中的各开关管关断损耗，该第一预设值一般小于1us，其具体取值不做限定，视其具体应用环境而定即可。
若双向谐振型直流变换器需要的输出增益大于1，则可使两个延时时间dt1与dt2不等。而且，两个延时时间dt1与dt2不等时，可选的，可以设置dt1<dt2；其中，第三桥臂的延时时间dt2随双向谐振型直流变换器所需的增益同向变化，实际应用中，为保证该双向谐振型直流变换器的增益能够随该延时时间dt2增加而单调上升，可设置该延时时间dt2小于开关周期的1/4；而第四桥臂的延时时间dt1小于一个第二预设值，也即延时时间dt1尽可能小，以实现第四桥臂软开关为首要条件；另外，为保证原边电路101对应桥臂中各开关管保持零电压开通，同时降低副边电路102对应桥臂(也即第四桥臂)中的各开关管关断损耗，此第二预设值一般小于1us，其具体取值不做限定，视其具体应用环境而定即可。
需要说明的是，实际应用中，也可以设置第四桥臂的延时时间dt1随双向谐振型直流变换器所需的增益同向变化；而第三桥臂的延时时间dt2尽可能小，只要能实现第三桥臂的软开关即可。也即，两个延时时间dt1与dt2不等时， 只要存在一个延时时间用来满足增益需求即可，另一个延时时间只需实现相应桥臂的软开关；具体如何设置两个延时时间，视其应用环境而定，均在本申请的保护范围内。
通过图5可以看出，由于开关频率大于谐振频率，谐振电流相位滞后于电压相位，因此在原边电路101中第一桥臂与第二桥臂动作时，原边谐振电流Ip还未过零(t0时刻)，此时在第二开关管S2驱动关断后，原边谐振电流Ip由第二开关管S2换向至第一开关管S1，电流方向为从第一节点A向第一开关管S1的另一端，此时将第一开关管S1驱动打开，实现第一开关管S1的零电压开通，即实现了软开关。原边电路101中其他开关管的软开关实现同第一开关管S1相同。对于副边电路102而言，第四桥臂动作时(t2时刻)，副边谐振电流Is在t1时刻已经换向完成，副边谐振电流Is流向为从第八开关管S8的另一端向第四节点D，在t2时刻，第八开关管S8的驱动关断，副边谐振电流Is换向至第七开关管S7，电流流向变为从第七开关管S7的另一端向第四节点D，此时第七开关管S7驱动打开，实现第七开关管S7的零电压开通，即实现了软开关，第八开关管S8的软开关实现与第七开关管S7相同。第三桥臂动作时(t3时刻)，副边谐振电流Is相对于t2时刻更大，其他情况与第四桥臂完全相同，因此更容易实现软开关。
由上述分析可见，原副边开关管仅在互补开关管关断与自身驱动未打开的死区时间内存在电流反向流动过程，其他流过电流时间内驱动都为打开状态，因此驱动未打开下电流反向流动过程限制在死区时间内，实际应用中通过合理控制死区时间，即可控制驱动未打开下电流反向流动时间，进而降低反向导通压降，减小导通损耗。
本实施例提供的该双向谐振型直流变换器的控制方法，在检测到副边谐振电流换向后，控制副边电路各桥臂中相应开关管分别延时一段相应的时间后再关断，由于关断后电流换流至同桥臂内互补开关管的反并二极管或体二极管，所以经过相应死区时间后，再控制副边电路各桥臂中互补的开关管导通，使得副边电路中各开关管仅在死区时间内存在驱动未开通反向走电流的工况，由此带来的导通损耗大大降低。而且，其副边电路各桥臂的延时时间是根据双向谐振型直流变换器的输入电参数和/或输出电参数和所需的增益确定得到的，所 以在设置原边电路和副边电路的开关频率大于谐振腔的谐振频率的情况下，也可以实现该双向谐振型直流变换器所需的增益；进而可以减小宽电压输出场合的频率调节范围，对于提高开关频率具有明显作用。另外，由于谐振均能实现软开关，可以有效降低开关损耗，进一步提升频率和系统效率。
值得说明的是，上述控制方法为该双向谐振型直流变换器由原边侧向副边侧进行功率传输时的情况，当功率由副边侧向原边侧进行传输时，其实现方式与原边侧向副边侧传输时相同，可以看作将原边与副边互换名称，也即令功率输入侧命名为原边而功率输出侧命名为副边，然后再采用上述控制方法，此处不再赘述，均在本申请的保护范围内。
本申请另一实施例还提供了一种双向谐振型直流变换器的控制电路，如图1和图3中所示，此时，该双向谐振型直流变换器包括：变压器T、原边电路101、副边电路102以及设置于变压器T与原边电路101和/或副边电路102之间的谐振腔103；图1以谐振腔103设置于变压器T与原边电路101之间为例进行展示，实际应用中，该谐振腔103也可以设置于变压器T与副边电路102之间，或者，该谐振腔103中的各谐振元件还可以分别设置于变压器T的两边。
具体的，该谐振腔103包括：至少一个谐振电感模块，以及，至少一个谐振电容模块；而且，当该谐振电感模块Lr与该谐振电容模块Cr的个数均为1时，两者分别设置于变压器T的原边和副边(如图2a所示)，或者，均设置于变压器T的同一边(如图2b所示)；当谐振电感模块的个数大于1时，各谐振电感模块分别设置于变压器T的原边和副边(如图2c、图2d及图2e中所示的Lrp和Lrs)；当谐振电容模块的个数大于1时，各谐振电容模块分别设置于变压器T的原边和副边(如图2e和图3中所示的Crp和Crs)；该谐振腔103中包括两个谐振电容模块Crp和Crs以及一个设置于副边的谐振电感模块Lrs的情况未进行图示，但均在本申请的保护范围内。实际应用中，可以根据电路需要对谐振电感模块与谐振电容模块的数量和位置进行衍变；而且，该谐振电感模块一般采用一个电感来实现，也不排除多个电感串并联的实现形式；该谐振电容模块可以由一个电容来实现，也可以由多个电容串并联来实现； 视其具体应用环境而定即可，均在本申请的保护范围内。以图2b和图2e所示情况为例进行说明，两种结构下的各参数关系为：
其中，Np为变压器T原边绕组的匝数，Ns为变压器T副边绕组的匝数。
谐振频率fr的计算式为：
而且，该原边电路101和副边电路102均为单相全桥电路，如图3中所示，且两个单相全桥电路中的各开关管均带有反并联二极管或体二极管。
也即，该双向谐振型直流变换器可以为现有技术中的CLLC型、LLC型和SRC型的单相谐振变换器，其具体结构及连接关系可以参见现有技术，此处不再一一赘述。
不论该双向谐振型直流变换器采用上述何种拓扑，参见图1和图3，其控制电路均包括：原边驱动电路205，副边驱动电路206，过零检测电路202，控制模块201，以及，输入采样电路203和/或输出采样电路204(图1和图3中均以同时包括两者为例进行展示)；其中：
输入采样电路203用于采样该双向谐振型直流变换器的输入电参数；输出采样电路204用于采样该双向谐振型直流变换器的输出电参数。具体的，该输入采样电路203设置于原边电路101的直流侧，且输出端与控制模块201的一个输入端相连，该输入采样电路203具体用于采样该双向谐振型直流变换器的输入电流和输入电压；该输出采样电路204设置于副边电路102的直流侧，且输出端与控制模块201的另一个输入端相连，该输出采样电路204具体用于采样该双向谐振型直流变换器的输出电流和输出电压。
过零检测电路202用于检测谐振腔103中的电流是否过零，并在过零信息表征双向谐振型直流变换器的副边谐振电流Is过零时，生成过零信号输出至该控制模块201的相应输入端。由于原边谐振电流Ip和副边谐振电流Is过零时都可以表征副边谐振电流Is过零，所以，过零检测电路202检测谐振腔103 中的电流是否过零时，具体可以是：检测副边谐振电流Is和/或双向谐振型直流变换器的原边谐振电流Ip是否过零。也即，实际应用中，该过零检测电路202可通过检测原边谐振电流Ip和/或副边谐振电流Is的过零情况来确定副边电路102中各桥臂的延时起点。而且，若变压器T的励磁电感很大，原副边谐振电流过零位置相差很小，则可以仅在变压器T的一边设置相应的检测电路进行过零检测，节约成本，其中优选对副边谐振电流Is进行过零检测。当控制模块201采用上述实施例所提供的控制方法时，由于各种输出增益要求下，开关频率均大于谐振频率，也即该双向谐振型直流变换器都工作在过谐振状态，其变流器T的励磁电感不参与工作，所以励磁电感可以很大，方便过零检测电路202以低成本实现上述过零检测。
该控制模块201用于接收该过零信号及该输入电参数和/或输出电参数，执行如上述任一实施例所述的双向谐振型直流变换器的控制方法，并通过原边驱动电路205控制原边电路101中各开关管动作，通过副边驱动电路206控制副边电路102中各开关管动作。该控制方法的具体过程及原理参见上述实施例即可，此处不再一一赘述。
为了实现该控制方法，该控制电路中各模块的工作分别是：
(1)输入采样电路203与输出采样电路204采样该双向谐振型直流变换器的输入与输出侧电压与电流，并将采样得到的采样信号传输至该控制模块201，使该控制模块201获取上述输入电参数和/或输出电参数。
(2)控制模块201依据上述输入电参数和/或输出电参数中一种或多种信号，确定副边侧两桥臂各自的延时时间dt1与dt2。
(3)控制模块201依据上述输入电参数和/或输出电参数以及内部的预设参考信号，产生原副边电路的开关频率，其中原边电路101的各开关管将直接依据控制模块201产生的开关频率工作。
(4)过零检测电路202检测原边谐振电流Ip和/或副边谐振电流Is，优选检测副边整流电流(也即该副边谐振电流Is)，在检测到其过零时，过零检测电路202通过输出过零信号到控制模块201，使控制模块201可以确定副边谐振电流Is过零并获得其过零时刻。
(5)控制模块201接收到过零信号后，分别延时两个延时时间dt1与dt2，输出控制第四桥臂与第三桥臂动作的驱动控制信号。
(6)控制模块201分别产生原边电路101和副边电路102的驱动控制信号，其中，原边电路101的驱动控制信号经原边驱动电路205生成原边电路101的驱动信号，用于控制原边电路101各开关管工作；副边电路102的驱动控制信号经副边驱动电路206生成副边电路102的驱动信号，用于控制副边电路102各开关管工作。
本实施例提供的该控制电路通过对该双向谐振型直流变换器进行谐振过零检测，结合延时时间控制，实现了副边侧开关管的软开关，相对于传统同步整流控制而言，有效减小了驱动未开反向流过电流的持续时间，在SiC以及GaN器件同步整流应用场合具有明显的效率优势；同时结合控制延时时间大小，改变输出电压增益，从而减小宽电压输出场合的频率调节范围，对于提高开关频率具有明显作用。
本说明书中的各个实施例之间相同相似的部分互相参见即可，每个实施例重点说明的都是与其他实施例的不同之处。尤其，对于系统或系统实施例而言，由于其基本相似于方法实施例，所以描述得比较简单，相关之处参见方法实施例的部分说明即可。以上所描述的系统及系统实施例仅仅是示意性的，其中所述作为分离部件说明的单元可以是或者也可以不是物理上分开的，作为单元显示的部件可以是或者也可以不是物理单元，即可以位于一个地方，或者也可以分布到多个网络单元上。可以根据实际的需要选择其中的部分或者全部模块来实现本实施例方案的目的。本领域普通技术人员在不付出创造性劳动的情况下，即可以理解并实施。
专业人员还可以进一步意识到，结合本文中所公开的实施例描述的各示例的单元及算法步骤，能够以电子硬件、计算机软件或者二者的结合来实现，为了清楚地说明硬件和软件的可互换性，在上述说明中已经按照功能一般性地描述了各示例的组成及步骤。这些功能究竟以硬件还是软件方式来执行，取决于技术方案的特定应用和设计约束条件。专业技术人员可以对每个特定的应用来使用不同方法来实现所描述的功能，但是这种实现不应认为超出本发明的范围。
对所公开的实施例的上述说明，本说明书中各实施例中记载的特征可以相互替换或者组合，使本领域专业技术人员能够实现或使用本发明。对这些实施例的多种修改对本领域的专业技术人员来说将是显而易见的，本文中所定义的一般原理可以在不脱离本发明的精神或范围的情况下，在其它实施例中实现。因此，本发明将不会被限制于本文所示的这些实施例，而是要符合与本文所公开的原理和新颖特点相一致的最宽的范围。

Claims (10)

  1. 一种双向谐振型直流变换器的控制方法，其特征在于，双向谐振型直流变换器包括：变压器、原边电路、副边电路以及设置于所述变压器与所述原边电路和/或所述副边电路之间的谐振腔；所述原边电路和所述副边电路均为单相全桥电路，且所述单相全桥电路中的各开关管均带有反并联二极管或体二极管；所述控制方法包括：
    获取所述双向谐振型直流变换器的输入电参数和/或输出电参数；
    根据所述输入电参数和/或输出电参数以及所述双向谐振型直流变换器所需的增益，确定所述副边电路中各桥臂的延时时间；
    根据所述输入电参数和/或输出电参数以及预设参考信号，设置所述原边电路和所述副边电路的开关频率大于所述谐振腔的谐振频率；
    在所述双向谐振型直流变换器的副边谐振电流过零时，以过零时刻为起点，在各所述延时时间之后分别进行对所述副边电路中相应开关管的关断控制；再在死区时间后，控制所述副边电路中互补的开关管导通。
  2. 根据权利要求1所述的双向谐振型直流变换器的控制方法，其特征在于，各所述延时时间相等，且小于预设值，使所述副边电路对应桥臂中的各开关管保持零电压开通；或者，
    所述双向谐振型直流变换器所需的增益大于1时，各所述延时时间不等。
  3. 根据权利要求2所述的双向谐振型直流变换器的控制方法，其特征在于，各所述延时时间不等时，存在一个所述延时时间，随所述双向谐振型直流变换器所需的增益同向变化。
  4. 根据权利要求1至3任一项所述的双向谐振型直流变换器的控制方法，其特征在于，在设置所述原边电路和所述副边电路的开关频率大于所述谐振腔的谐振频率之后，还包括：
    根据所述开关频率生成并输出所述原边电路的驱动控制信号。
  5. 根据权利要求1至3任一项所述的双向谐振型直流变换器的控制方法，其特征在于，所述单相全桥电路中，同一桥臂中两半桥臂的开关管互补导通；
    所述原边电路中，不同桥臂中不同位置半桥臂的开关管动作相同。
  6. 根据权利要求1至3任一项所述的双向谐振型直流变换器的控制方法，其特征在于，所述输入电参数和/或输出电参数，包括：输入电流、输入电压、输出电流及输出电压中的至少一种。
  7. 一种双向谐振型直流变换器的控制电路，其特征在于，双向谐振型直流变换器包括：变压器、原边电路、副边电路以及设置于所述变压器与所述原边电路和/或所述副边电路之间的谐振腔；所述原边电路和所述副边电路均为单相全桥电路，且所述单相全桥电路中的各开关管均带有反并联二极管或体二极管；所述控制电路包括：原边驱动电路、副边驱动电路、过零检测电路、控制模块以及输入采样电路和/或输出采样电路；其中，
    所述输入采样电路用于采样所述双向谐振型直流变换器的输入电参数，所述输出采样电路用于采样所述双向谐振型直流变换器的输出电参数；
    所述过零检测电路用于检测所述谐振腔中的电流是否过零，并生成过零信号；
    所述控制模块用于接收所述过零信号及所述输入电参数和/或所述输出电参数，执行如权利要求1至6任一项所述的双向谐振型直流变换器的控制方法，并通过所述原边驱动电路控制所述原边电路中各开关管动作，通过所述副边驱动电路控制所述副边电路中各开关管动作。
  8. 根据权利要求7所述的双向谐振型直流变换器的控制电路，其特征在于，所述过零检测电路用于检测所述谐振腔中的电流是否过零时，具体用于：
    检测所述双向谐振型直流变换器的副边谐振电流和/或原边谐振电流是否过零。
  9. 根据权利要求7所述的双向谐振型直流变换器的控制电路，其特征在于，所述输入采样电路用于采样所述原边电路直流侧的输入电流和输入电压；
    所述输出采样电路用于采样所述副边电路直流侧的输出电流和输出电压。
  10. 根据权利要求7至9任一项所述的双向谐振型直流变换器的控制电路，其特征在于，所述谐振腔，包括：至少一个谐振电感模块，以及，至少一个谐振电容模块；
    所述谐振电感模块与所述谐振电容模块的个数均为1时，两者分别设置于所述变压器的原边和副边，或者，均设置于所述变压器的同一边；
    所述谐振电感模块的个数大于1时，各所述谐振电感模块分别设置于所述变压器的原边和副边；所述谐振电容模块的个数大于1时，各所述谐振电容模块分别设置于所述变压器的原边和副边。
PCT/CN2023/091097 2022-09-07 2023-04-27 一种双向谐振型直流变换器的控制方法及控制电路 WO2024051181A1 (zh)

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Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2003324956A (ja) * 2002-05-09 2003-11-14 Origin Electric Co Ltd 直列共振型ブリッジインバータ回路の制御方法及び直列共振型ブリッジインバータ回路
CN114430235A (zh) * 2022-04-01 2022-05-03 浙江富特科技股份有限公司 电源变换装置及系统
CN114465491A (zh) * 2022-02-10 2022-05-10 山东大学 一种经济型双向隔离dc/dc四象限变换器及控制方法
CN114583967A (zh) * 2022-02-17 2022-06-03 中国科学院电工研究所 两相并联升压电路的隔离型直流升压变换器及其控制方法
CN115765467A (zh) * 2022-09-07 2023-03-07 阳光电源股份有限公司 一种双向谐振型直流变换器的控制方法及控制电路

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2003324956A (ja) * 2002-05-09 2003-11-14 Origin Electric Co Ltd 直列共振型ブリッジインバータ回路の制御方法及び直列共振型ブリッジインバータ回路
CN114465491A (zh) * 2022-02-10 2022-05-10 山东大学 一种经济型双向隔离dc/dc四象限变换器及控制方法
CN114583967A (zh) * 2022-02-17 2022-06-03 中国科学院电工研究所 两相并联升压电路的隔离型直流升压变换器及其控制方法
CN114430235A (zh) * 2022-04-01 2022-05-03 浙江富特科技股份有限公司 电源变换装置及系统
CN115765467A (zh) * 2022-09-07 2023-03-07 阳光电源股份有限公司 一种双向谐振型直流变换器的控制方法及控制电路

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