WO2022170645A1 - 一种应用于大功率范围的低失真d类功放 - Google Patents

一种应用于大功率范围的低失真d类功放 Download PDF

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WO2022170645A1
WO2022170645A1 PCT/CN2021/078065 CN2021078065W WO2022170645A1 WO 2022170645 A1 WO2022170645 A1 WO 2022170645A1 CN 2021078065 W CN2021078065 W CN 2021078065W WO 2022170645 A1 WO2022170645 A1 WO 2022170645A1
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output
circuit
current
capacitor
voltage
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French (fr)
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张金路
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张金路
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    • HELECTRICITY
    • H03ELECTRONIC CIRCUITRY
    • H03FAMPLIFIERS
    • H03F1/00Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32Modifications of amplifiers to reduce non-linear distortion
    • HELECTRICITY
    • H03ELECTRONIC CIRCUITRY
    • H03FAMPLIFIERS
    • H03F3/00Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F3/217Class D power amplifiers; Switching amplifiers

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  • the invention belongs to the technical field of audio power amplifiers, and discloses a low-distortion class D power amplifier applied to a large power range.
  • Class D power amplifier simply refers to an amplification mode in which the power amplifying element is in a switching state.
  • the principle is generally to compare the audio signal with the high-frequency fixed-frequency signal to obtain the modulation signal of the audio signal on the fixed-frequency carrier, that is, convert it into a PWM signal, and amplify the PWM signal into a high-voltage, high-current signal through a switching amplifier.
  • the high-power PWM signal can finally be restored to the high-power audio signal through a low-pass filter.
  • the distortion of digital class D power amplifiers has a unique source.
  • a good digital power amplifier circuit can be designed to meet high dynamic performance, but the fluctuation of switching frequency is inevitably introduced into the control loop, and this fluctuation is sampled by the switching comparator. Reflected to the output, resulting in relatively large distortion of the digital power amplifier. Therefore, the control loop should not only adjust the dynamic control characteristics, but also adjust the phase delay characteristics of the switching frequency. Theoretically, a perfect phase delay characteristic is designed, which can make the sum of the voltage fluctuations of the positive and negative trip points of the comparator to be 0. At this time, although the fluctuation exists, it does not affect the output, that is, the nonlinearity of this high-frequency fluctuation can be eliminated. After doing this, the power amplifier distortion can achieve a level of less than 0.0002% of the full audio range.
  • the present invention discloses a low-distortion class D power amplifier applied to a large power range, and its specific technical scheme is as follows:
  • a low-distortion class D power amplifier applied to a large power range comprising: a voltage control module, a current self-excited inner-loop oscillation circuit, a MOS switch circuit, an output LC filter circuit, a nonlinear compensation module and a self-excited oscillation frequency adjustment module.
  • the current self-excited inner loop oscillation circuit is composed of a comparator and a current feedback circuit connected to the output LC capacitor current sampling feedback circuit of the comparator.
  • the input terminal of the voltage control module is connected to the input voltage VIN, and the output terminal outputs the control voltage VM to the comparator's Input terminal, the output terminal of the comparator is connected to the input terminal of the MOS switch circuit, the output terminal of the MOS switch circuit is connected to the input terminal of the output LC filter circuit, the output terminal of the output LC filter circuit is connected to the voltage control module, and the output terminal of the LC filter circuit is connected to the output terminal of the LC filter circuit.
  • the capacitor current is fed back to the comparator through the output LC capacitor current sampling feedback circuit; the input end of the nonlinear compensation module is connected to the input voltage VIN, and the output end outputs the nonlinear compensation function signal to the output end of the voltage control module; the self-excited The input end of the oscillation frequency adjustment module is connected with the input voltage VIN, and the output end is connected with the comparator.
  • the voltage control module is provided with a high-frequency phase control circuit, and the high-frequency phase control circuit is provided with a plurality of high-frequency zero-pole points to precisely adjust the phase of the switching frequency point.
  • the output LC capacitor current sampling feedback circuit performs sampling feedback on the capacitor current of the output LC filter circuit, using one of the following methods:
  • the output capacitor of the output LC filter circuit is connected in parallel with a small capacitor.
  • the small capacitor current is a proportionally reduced mirror image of the output capacitor.
  • the virtual ground generation and current detection module is used to form a virtual ground at the lower end of the small capacitor to absorb its current.
  • the ground generation and current detection module output obtains the output capacitor current signal;
  • the output capacitor of the output LC filter circuit is connected in parallel with a small capacitor.
  • the small capacitor current is a proportionally reduced mirror image of the output capacitor.
  • the virtual ground generation and current detection module is used to form a virtual ground at the lower end of the capacitor to absorb its current.
  • the stage circuit controls the series resistance of the output terminal to cancel the low frequency part of the mirror current, and the virtual ground generation and the output of the current detection module obtain the high frequency part of the output capacitor current signal;
  • the output capacitor of the output LC filter circuit is connected to a current detection sensor, and the sampling current is detected by the current detection sensor, and the current detection sensor includes a current transformer or a resistor;
  • the output capacitor of the output LC filter circuit is connected in parallel with a small capacitor, the small capacitor is connected in series with the resistor and then grounded, and the sampling is amplified several times by the resistor voltage amplifier to obtain an approximate output capacitor current signal;
  • the output capacitor of the output LC filter circuit is connected in parallel with a small capacitor, and the small capacitor is connected in series with the resistor to control the output of the previous stage circuit.
  • the sampling is amplified several times by the resistor voltage amplifier to obtain the high frequency part of the approximate output capacitor current signal.
  • the virtual ground generation and current detection module uses an OPA860 chip or a transconductance amplifier to absorb current to form a virtual ground and convert the current into a voltage.
  • the self-excited oscillation frequency adjustment module adopts an analog circuit or is calculated by MCU to obtain and adjust the hysteresis value.
  • the analog circuit includes an RC filter circuit, the RC filter circuit is connected to the input voltage VIN, and the output voltage signal VIN2 is used for Instead of outputting the voltage OUT signal, output the absolute value of the voltage signal VIN2 and the power supply voltage VCC to the function f2 (abs(VIN2), VCC) operation circuit to obtain the hysteresis value required when the frequency is stable, and according to the hysteresis value, the frequency is stabilized by the hysteresis amplitude control circuit, wherein the f2(abs(VIN2), VCC) analog circuit realizes fitting using a linear function or a broken line method.
  • the nonlinear compensation module uses an analog circuit or MCU calculation to eliminate the full signal amplitude distortion of the circuit, the analog circuit is provided with an RC filter circuit, and the input terminal of the RC filter circuit is connected to the input voltage VIN, and the output voltage VIN2, At the same time, the power supply voltage VCC is input to the nonlinear compensation function f1 (VIN2, VCC) circuit to obtain the compensation value of the circuit.
  • analog circuit of the nonlinear compensation function f1 (VIN2, VCC) implements fitting using the broken line method.
  • the output capacitor current sampling method of the present invention adopts the mirror mode, using small capacitors in parallel, which can reduce the current to be sampled many times, and then use the mirror capacitor current acquisition method, the mirror capacitor current acquisition method has the advantages of extremely low distortion rate and low cost;
  • the well-designed high-frequency phase control of the circuit can greatly eliminate the influence of high-frequency ripple in the loop; the self-excited oscillation frequency adjustment module can adjust the oscillation frequency according to the input signal amplitude, which can bring two benefits.
  • the increase of the minimum operating frequency can speed up the control loop speed; 2: After the frequency change becomes smaller, the phase delay design of the switching frequency of the control loop becomes simpler; after adding the frequency adjustment circuit, the distortion of the medium and small amplitude signals will become very small, However, other factors will produce nonlinear distortion under very large signal amplitude.
  • the nonlinear compensation circuit can eliminate the remaining distortion, so that the circuit has extremely low distortion in a very large amplitude range.
  • Fig. 1 is the circuit schematic diagram of the present invention
  • Fig. 2 is the schematic diagram of the circuit adopted by the existing basic voltage control module
  • FIG. 3, FIG. 4, FIG. 5, and FIG. 6 are circuit schematic diagrams of the voltage control module according to the embodiment of the present invention.
  • FIG. 7 and FIG. 8 are the high-frequency ripple and sampling point waveform diagrams of the existing basic voltage control module when the small signal and the large signal are respectively;
  • 9 and 10 are waveform diagrams of high-frequency ripple and sampling points of the voltage control module provided with the high-frequency phase control circuit according to the present invention when the small signal and the large signal are respectively;
  • FIG. 11 is a BODE diagram of a voltage control module and a basic voltage control module provided with a high-frequency phase control circuit according to the present invention
  • FIG. 12 is a BODE diagram of a voltage control module circuit provided with a high-frequency phase control circuit according to the present invention.
  • Fig. 13, Fig. 14, Fig. 15, Fig. 16, Fig. 17, Fig. 18 are the circuit schematic diagrams of the present invention for sampling and feeding back the capacitor current of the output LC filter circuit;
  • Fig. 19a is a circuit schematic diagram of a virtual ground generation and current detection module of the present invention.
  • 19b is a schematic circuit diagram of the virtual ground generation and current detection module of the present invention when the OPA860 chip is used;
  • 19c is a schematic circuit diagram of the virtual ground generation and current detection module of the present invention using a transconductance amplifier
  • FIG. 20 is a schematic diagram of the broken-line fitting of the nonlinear compensation function of the present invention.
  • 1-voltage control module 2-comparator, 3-MOS switch circuit, 4-output LC filter circuit, 5-nonlinear compensation module, 6-output LC capacitor current sampling feedback circuit, 7-self-oscillation frequency adjustment module.
  • a low-distortion class D power amplifier applied to a large power range includes: a voltage control module 1, a current self-excited inner ring oscillator circuit, a MOS switch circuit 3, an output LC filter circuit 4, and a nonlinear compensation module.
  • the current self-excited inner loop oscillation circuit is composed of a comparator 2 and a capacitor current sampling feedback circuit 6 connected to the output LC of the comparator 2 by current feedback, and the input of the voltage control module 1 is composed of The terminal is connected to the input voltage VIN, the output terminal outputs the control voltage VM to the input terminal of the comparator 2, the output terminal of the comparator 2 is connected to the input terminal of the MOS switch circuit 3, and the output terminal of the MOS switch circuit 3 is connected to the output terminal of the LC filter circuit 4.
  • the input end of the output LC filter circuit 4 is connected to the voltage control module 1, and the capacitance current of the output LC filter circuit 4 is fed back to the comparator 2 through the capacitance current sampling feedback circuit 6 of the output LC; the nonlinear compensation module 5.
  • the first input terminal is connected to the input voltage VIN
  • the second input terminal is connected to the power supply voltage VCC
  • the output terminal outputs the nonlinear compensation function signal to the input terminal of the comparator 2
  • the first input terminal of the self-oscillation frequency adjustment module 7 is connected to The input voltage VIN
  • the second input terminal is grounded to the power supply voltage VCC
  • the output terminal is connected to the comparator 2 .
  • the circuit of the existing basic voltage control module includes an operational amplifier U1, a resistor R1, a resistor R2, a capacitor C1, and a capacitor C2.
  • the capacitor C1 and the resistor R2 are connected in parallel and then connected to the LC filter circuit 4.
  • the other end is connected to one end of the capacitor C2 and then one end of the resistor R1, the other end of the capacitor C2 is connected to the output end of the operational amplifier U1, the inverting input end of the operational amplifier U1 is connected to one end of the resistor R1, and the other end of the resistor R1
  • the other end is connected to the input voltage VIN, and the output of this circuit is VM1;
  • this circuit can meet the dynamic characteristics of the whole circuit, because the current inner loop of the latter stage is equivalent to the first-order characteristics of the current source driving the capacitor parallel resistance load, and the voltage outer loop is actually very It is easy to stabilize. After adding the differential characteristic of feedback capacitance, it can achieve the characteristics of fast response and no overshoot.
  • the basic circuit does not deal with high-frequency ripple, but high-frequency ripple is the main source of nonlinear distortion of the circuit.
  • the voltage control module 1 of the present invention is provided with a high-frequency phase control circuit for processing high-frequency ripple, and the high-frequency phase control circuit includes the following circuit structure:
  • a resistor R3 is added on the basic voltage control module circuit. After the resistor R3 and the capacitor C1 are connected in series, the resistor end is connected to the output end of the LC filter circuit 4; the output of the circuit is VM2;
  • a resistor R3, a resistor R4 and a capacitor C3 are added. After the resistor R4 and the capacitor C3 are connected in series, the resistor terminal is connected to the output terminal of the LC filter circuit 4. The resistor R3 and capacitor C1 After the series connection, the resistance end is connected to the output end of the LC filter circuit 4; the output of this circuit is VM3;
  • a resistor R3 and a resistor R5 are added on the basic voltage control module circuit. After the resistor R3 and the capacitor C1 are connected in series, the resistor end is connected to the output end of the LC filter circuit 4, and the resistor R5 is set on the capacitor. Between C2 and the output of operational amplifier U1; the output of this circuit is VM4;
  • a first-level circuit is added to the basic voltage control module circuit.
  • the added circuit includes operational amplifier U2, resistor R3, resistor R5, resistor R6, resistor R7, resistor R8, resistor R10, capacitor C4 and capacitor C5, after the resistor R3 and the capacitor C1 are connected in series, the resistor end is connected to the output end of the LC filter circuit 4, the resistor R5 is set between the capacitor C2 and the output end of the operational amplifier U1, and one end of the capacitor C4 is connected to the output end of the resistor R7 One end, the other end of the resistor R7 is connected to one end of the resistor R6 and then connected to the reverse input end of the operational amplifier U2, the other end of the capacitor C4 is connected to the other end of the resistor R6 and then connected to the output end of the operational amplifier U1; the output of the circuit for VM6;
  • the above circuits can retain the dynamic characteristics of the audio frequency, but multiple high-frequency zero-pole points can be set according to the actual circuit needs to accurately adjust the phase of the switching frequency point.
  • the resistor R3 can make the high-frequency phase lag, and the resistor R5 can make the phase lead, which is complicated.
  • the purpose of the phase control is to set an appropriate phase slope in the switching frequency range.
  • Figure 6 shows that some circuit structures need to add a first-level circuit on the original basis, and a certain phase control can also be added to this level of circuit. When the high-frequency ripple phase is precisely adjusted, the distortion rate can be reduced by two orders of magnitude, and the full audio range can reach below 0.0002%.
  • Figures 7 and 8 show the high-frequency ripple and sampling point waveforms of the basic voltage control module, where VM1 is the output of the basic voltage control module, sample is the comparator flip point, and Figure 7 shows that the signal input level is 0 , the duty cycle of the high-frequency switch is 50%. Figure 8 shows that when a large signal is input, the duty cycle deviates from 50%. It can be seen that the sampling point of VM1 does not reflect the average voltage of VM1 when the switch is turned over, and actually has a large nonlinearity, which is the main nonlinear distortion of the circuit.
  • Figures 9 and 10 show the high-frequency ripple and sampling point waveform diagrams of the voltage control module provided with the high-frequency phase control circuit of the present invention, and the phase of the switching frequency point is shifted back by nearly 90 degrees.
  • the average value of the voltage of the positive and negative switching sampling points of VM2 is exactly the average value of VM2, and the influence of the switching frequency ripple is eliminated at this time. More complex phase control can be controlled more precisely, and after this nonlinear distortion is eliminated, the circuit distortion can reach extremely low levels.
  • VM1 is a basic circuit that meets the dynamic characteristics of the voltage outer loop.
  • the dynamic characteristics of the following circuits are similar, but the phase phase is adjusted near the switching frequency point of 500KHZ.
  • VM2 is the simplest circuit that roughly meets the requirements.
  • the amplitude of the switching frequency point is greatly attenuated, that is, the reduction of the control voltage ripple is conducive to reducing nonlinearity.
  • the phase of the switching frequency point is greatly adjusted, making the positive and negative of the comparator.
  • the sum of the voltage fluctuations of VM2 at the transition time point is 0, and the high-frequency ripple factor is canceled.
  • Both VM3 and VM4 can be modified on the basis of VM2, including the phase slope, so that the phase control is more accurate.
  • the sampling feedback of the capacitor current of the output LC filter circuit 4 adopts one of the following methods: As shown in FIG. 13 , the output capacitor of the output LC filter circuit 4 is connected in series with a current transformer.
  • the transformer detection can work stably, but the magnetic components may have a certain nonlinearity, which is not suitable when extremely low distortion is required;
  • the output capacitor of the output LC filter circuit 4 is connected in series with a resistor, and the resistor voltage is amplified to obtain the capacitor current.
  • the disadvantage is that the resistor has a certain power loss and heat generation, and has a little impact on the output voltage;
  • the output capacitor of the output LC filter circuit 4 is connected in parallel with a small capacitor, which is connected in series with a resistor and then grounded.
  • the voltage of the resistor is amplified several times by the resistor voltage amplifier as an approximate output capacitor current signal.
  • a small capacitor is connected in series with a resistor, what is actually obtained is the low-pass filtered signal of the output capacitor current.
  • Using a larger resistor without amplification can actually work, but the high-frequency part of the current sampling is attenuated more, and the output impedance parameter will become worse.
  • the output capacitor of the output LC filter circuit 4 is connected in parallel with a small capacitor, and the small capacitor is connected in series with the resistor to control the output of the previous stage circuit.
  • the pre-stage feedback circuit will maintain the virtual ground at the lower end of the small capacitor, but because the pre-stage only controls the audio part, the high-frequency part is still filtering.
  • the output capacitor of the output LC filter circuit 4 is connected in parallel with a small capacitor.
  • the current of the small capacitor is a proportionally reduced mirror image of the output capacitor.
  • the virtual ground generation and current detection module is used to form a virtual ground at the lower end of the small capacitor to absorb its The current, virtual ground generation and the output of the current detection module get the output capacitor current signal, and at the same time can realize the detection of the current on the small capacitor, this method can perfectly reflect the output capacitor current, and the distortion is small.
  • the front-stage circuit is used to control the output VM(n) series resistance to cancel the low-frequency part of the mirror current, and the remaining high-frequency part is
  • the virtual ground generation and current detection module absorbs and converts it into a high-frequency voltage signal, which is compared with the hysteresis voltage above and below the 0 level to generate a PWM wave.
  • the virtual ground generation and current detection module can use a chip like OPA860 or a transconductance amplifier to absorb current to form a virtual ground and convert the current into a voltage.
  • the feedback of the capacitor current of the output LC filter circuit 4 includes the output inductor current and the load current.
  • the inductor current feedback provides circuit dynamic characteristics and stability, and the load current feedback makes the output impedance extremely low.
  • the self-excited oscillation frequency adjustment module 7 is used to dynamically adjust the stability of the switching frequency and greatly improve the circuit performance.
  • the self-excited oscillation frequency adjustment module 7 adopts an analog circuit or is calculated by MCU to obtain and adjust the hysteresis value, thereby stabilizing the frequency.
  • the analog circuit includes an RC filter circuit, and the RC filter circuit is connected to the input used to replace the output voltage.
  • the voltage VIN, the output voltage signal VIN2, the output voltage signal VIN2 can simulate the phase lag of the output voltage OUT relative to the input voltage VIN, so the output voltage signal VIN2 can replace the output voltage OUT for calculation, output the voltage signal VIN2
  • the MCU calculation implementation can obtain any f2() function.
  • the nonlinear distortion of the circuit also comes from another aspect.
  • the output voltage OUT is close to the power supply voltage VCC, and the output inductor current is no longer a triangular wave, but a little circular arc deformation.
  • the average value of the peak-to-peak value of the current no longer represents the average value of the current. nonlinear properties.
  • the nonlinear compensation module 5 of the present invention uses an analog circuit or MCU calculation to eliminate the full signal amplitude distortion of the circuit.
  • the analog circuit can default to constant VCC and use the broken line method to fit, and the MCU calculation and implementation method can obtain any f1() function. .
  • the analog circuit is provided with an RC filter circuit, the input terminal of the RC filter circuit is connected to the input voltage VIN, the output terminal outputs a signal to the nonlinear compensation function circuit, and the power supply voltage VCC is input to the nonlinear compensation function f1 (VIN2, VCC) circuit at the same time , to obtain the compensation value of the circuit, thereby eliminating the nonlinear distortion of the circuit.
  • the power supply voltage VCC can be ignored because it is relatively stable.
  • the non-linear compensation function analog circuit implementation can default to constant VCC and use the broken line method to fit, and the circuit implementation is relatively simple.

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Abstract

本发明属于音频功放技术领域,公开一种应用于大功率范围的低失真D类功放,包括:电压控制模块、电流自激内环振荡电路、MOS开关电路、输出LC滤波电路、非线性补偿模块和自激振荡频率调节模块,所述电流自激内环振荡电路由比较器和电流反馈连接比较器的输出LC电容电流采样反馈电路组成,所述电压控制模块设有高频相位控制电路用于调节电路开关频率相位延迟,消除高频波动的引起的非线性,自激振荡频率调节模块根据输入信号幅度调节振荡频率,缩窄开关频率变化范围增加电压环路适应性并减小非线性失真,输出LC电容电流采样反馈电路采用输出电容并联小电容的电流镜的方式,所述非线性补偿模块对电路进行非线性补偿,进一步消除电路全信号幅度失真。

Description

一种应用于大功率范围的低失真D类功放 技术领域
本发明属于音频功放技术领域,公开一种应用于大功率范围的低失真D类功放。
背景技术
D类功放简单来说是指功率放大元件处于开关工作状态的一种放大模式。原理实现上一般是将音频信号与高频固定频率信号比较,得到音频信号在固定频率的载波上的调制信号,即转换成了PWM信号,通过开关放大器把PWM信号放大成高电压、大电流的大功率PWM信号,最终通过低通滤波器就可将大功率的音频信号还原。
数字D类功放的失真相对于模拟功放有独特的来源,好的数字功放电路可以设计到满足高动态性能,但是开关频率的波动不可避免的引入到控制环路中,这个波动被开关比较器取样反映到输出,造成数字功放比较大的失真。所以控制环路不但要调节动态控制特性,而且要调节开关频率的相位延迟特性。理论上设计完美的相位延迟特性,可以使得比较器的正负跳变点电压波动相加和为0,此时波动虽然存在,但是不影响输出,也就是可以消除这个高频波动的非线性。做到这点后功放失真可以实现全音频范围小于0.0002%级别。
现有数字功放性能高的都是自激振荡工作模式,得到高性能的同时,也存在随信号幅度变大后振荡频率变低的问题,大信号时频率比小信号变低50%甚至更多,这使得控制环路必须降低速度来迁就大信号幅度的工作状态。同时频率大幅度变化,使得高频相位控制变得困难,所以在大信号幅度工作时失真急剧上升。并且,大信号幅值时,输出电压OUT接近电源VCC,输出电感电流不再是三角波,而是有点圆弧变形,频率变低后更为严重,此时电流峰峰值的平均值不再代表电流的平均值,此时产生复杂的非线性特性。
发明内容
为了解决现有技术中存在的上述技术问题,本发明公开了一种应用于大功率范围的低失真D类功放,其具体技术方案如下:
一种应用于大功率范围的低失真D类功放,包括:电压控制模块、电流自激内环振荡电路、MOS开关电路、输出LC滤波电路、非线性补偿模块和自激振荡频率调节模块,所述电流自激内环振荡电路由比较器和电流反馈连接比较器的输出LC电容电流采样反馈电路组成,所述电压控制模块的输入端接输入电压VIN,输出端输出控制电压VM至比较器的输入端,所述比较器的输出端连接MOS开关电路的输入端,MOS开关电路的输出端连接输出LC滤波电路的输入端,输出LC滤波电路的输出端与电压控制模块连接,输出LC滤波电路的电容电流经输出LC电容电流采样反馈电路反馈至比较器;所述非线性补偿模块输入端接入输 入电压VIN,输出端输出非线性补偿函数信号至电压控制模块的输出端;所述自激振荡频率调节模块输入端接输入电压VIN,输出端连接比较器。
进一步的,所述的电压控制模块设有高频相位控制电路,所述高频相位控制电路设置多个高频零极点精确调整开关频率点的相位。
进一步的,所述输出LC电容电流采样反馈电路,对输出LC滤波电路的电容电流进行采样反馈,采用如下其中一种方式:
a:输出LC滤波电路的输出电容并联一个小电容,所述小电容电流是输出电容按比例减小的镜像,使用虚地生成及电流检测模块在小电容下端形成虚地吸收它的电流,虚地生成及电流检测模块输出得到输出电容电流信号;
b:输出LC滤波电路的输出电容并联一个小电容,所述小电容电流是输出电容按比例减小的镜像,使用虚地生成及电流检测模块在电容下端形成虚地吸收它的电流,在前级电路控制输出端串联电阻来抵消镜像电流的低频部分,虚地生成及电流检测模块输出得到输出电容电流信号的高频部分;
c:输出LC滤波电路的输出电容连接电流检测传感器,通过所述电流检测传感器检测得到采样电流,所述电流检测传感器包括电流互感器或电阻;
d:输出LC滤波电路的输出电容并联一个小电容,小电容串联电阻再接地,采样经电阻电压放大器放大数倍得到近似输出电容电流信号;
e:输出LC滤波电路的输出电容并联一个小电容,小电容串联电阻再接前级电路控制输出,采样经电阻电压放大器放大数倍得到近似输出电容电流信号的高频部分。
进一步的,所述虚地生成及电流检测模块输入端对外表现为低阻,吸收外部灌入电流,输出端输出从输入端灌入电流的检测值,得到输出电压信号V=K*I,其中K为常量,I为所述外部灌入电流。
进一步的,所述虚地生成及电流检测模块使用OPA860芯片或跨导放大器吸收电流形成虚地并把电流转换成电压。
进一步的,所述自激振荡频率调节模块采用模拟电路或通过MCU计算得到并调节滞回值,所述模拟电路包括RC滤波电路,所述RC滤波电路接入输入电压VIN,输出电压信号VIN2用于代替输出电压OUT信号,输出所述电压信号VIN2的绝对值和电源电压VCC至函数f2(abs(VIN2),VCC)运算电路,得到频率稳定时需要的滞回值,并根据所述滞回值,通过滞回幅度控制电路稳定频率,其中,所述f2(abs(VIN2),VCC)模拟电路实现使用线性函数或折线法拟合。
进一步的,所述非线性补偿模块,使用模拟电路或通过MCU计算来消除电路全信号 幅度失真,所述模拟电路设有RC滤波电路,RC滤波电路的输入端接输入电压VIN,输出电压VIN2,同时将电源电压VCC输入至非线性补偿函数f1(VIN2,VCC)电路,得到电路的补偿值。
进一步的,所述非线性补偿函数f1(VIN2,VCC)模拟电路实现使用折线法拟合。
本发明输出电容电流采样采用镜像方式,使用小电容并联,可以把需要采样的电流减小很多倍,然后使用采集镜像电容电流,镜像电容电流采集方式有失真率极低和低成本优势;电压环路加入精心设计的高频相位控制可以大幅消除高频纹波在环路里的影响;通过自激振荡频率调节模块根据输入信号幅度调节振荡频率,可以带来两个好处。1:最低工作频率提高可以使得加快控制环路速度,2:频率变化变小后,控制环路开关频率相位延迟设计变得更简单;加入频率调节电路后中小幅度信号失真会变得非常小,但是非常大信号幅度下还会产生其他因素非线性失真,非线性补偿电路可以消除剩下的失真,使得电路在非常大的幅度范围失真极低。
附图说明
图1为本发明的电路原理图;
图2为现有的基础电压控制模块采用电路的原理图;
图3、图4、图5、图6均为本发明实施例的电压控制模块电路原理图;
图7和图8为分别小信号和大信号时现有的基础电压控制模块的高频纹波及采样点波形图;
图9和图10为分别小信号和大信号时本发明的设有高频相位控制电路的电压控制模块高频纹波及采样点波形图;
图11为本发明的设有高频相位控制电路的电压控制模块和基础电压控制模块BODE图;
图12为本发明的设有高频相位控制电路的电压控制模块电路BODE图;
图13、图14、图15、图16、图17、图18为本发明的对输出LC滤波电路的电容电流进行采样反馈的各个电路原理图;
图19a为本发明的虚地生成及电流检测模块电路原理图;
图19b为本发明的虚地生成及电流检测模块使用OPA860芯片时的电路原理图;
图19c为本发明的虚地生成及电流检测模块使用跨导放大器时的电路原理图;
图20为本发明的非线性补偿函数的折线法拟合示意图;
图中,1-电压控制模块,2-比较器,3-MOS开关电路,4-输出LC滤波电路,5-非线性补偿模块,6-输出LC电容电流采样反馈电路,7-自激振荡频率调节模块。
具体实施方式
为了使本发明的目的、技术方案和技术效果更加清楚明白,以下结合说明书附图和 实施例,对本发明作进一步详细说明。
如图1所示,一种应用于大功率范围的低失真D类功放,包括:电压控制模块1、电流自激内环振荡电路、MOS开关电路3、输出LC滤波电路4、非线性补偿模块5和自激振荡频率调节模块7,所述电流自激内环振荡电路由比较器2和电流反馈连接比较器2的输出LC的电容电流采样反馈电路6组成,所述电压控制模块1的输入端接输入电压VIN,输出端输出控制电压VM至比较器2的输入端,所述比较器2的输出端连接MOS开关电路3的输入端,MOS开关电路3的输出端连接输出LC滤波电路4的输入端,输出LC滤波电路4的输出端与电压控制模块1连接,输出LC滤波电路4的电容电流经输出LC的电容电流采样反馈电路6反馈至比较器2;;所述非线性补偿模块5第一输入端接入输入电压VIN,第二输入端接电源电压VCC,输出端输出非线性补偿函数信号至比较器2的输入端;所述自激振荡频率调节模块7第一输入端接输入电压VIN,第二输入端接地电源电压VCC,输出端连接比较器2。
如图2所示的是现有的基础电压控制模块的电路,包括运算放大器U1,电阻R1、电阻R2、电容C1、电容C2,所述电容C1和电阻R2并联连接后一端接LC滤波电路4的输出端,另一端接电容C2的一端后接电阻R1的一端,电容C2的另一端接运算放大器U1的输出端,所述运算放大器U1的反相输入端接电阻R1的一端,电阻R1的另一端接输入电压VIN,此电路的输出为VM1;该电路能满足整个电路的动态特性,因为后级电流内环等效为电流源驱动电容并联电阻负载的一阶特性,电压外环实际很容易稳定,加入反馈电容微分特性后,可以达到快速响应及无超调的特性,但是该基础电路对高频纹波没有处理,然而高频纹波是电路非线性失真的主要来源。
本发明所述的电压控制模块1设有高频相位控制电路,用于处理高频纹波,所述高频相位控制电路包括如下电路结构:
如图3所示,在基础的电压控制模块电路上,增设电阻R3,所述电阻R3和电容C1串联连接后电阻端连接LC滤波电路4的输出端;该电路的输出为VM2;
如图4所示,在基础的电压控制模块电路上,增设电阻R3、电阻R4和电容C3,所述电阻R4和电容C3串联后电阻端连接LC滤波电路4的输出端,电阻R3和电容C1串联连接后电阻端连接LC滤波电路4的输出端;该电路的输出为VM3;
如图5所示,在基础的电压控制模块电路上,增设电阻R3和电阻R5,所述电阻R3和电容C1串联连接后电阻端连接LC滤波电路4的输出端,所述电阻R5设置在电容C2和运算放大器U1的输出端之间;该电路的输出为VM4;
如图6所示,在基础的电压控制模块电路上,增设一级电路,增设的电路包括运算放大器 U2、电阻R3、电阻R5、电阻R6、电阻R7、电阻R8、电阻R10、电容C4和电容C5,所述电阻R3和电容C1串联连接后电阻端连接LC滤波电路4的输出端,所述电阻R5设置在电容C2和运算放大器U1的输出端之间,所述电容C4一端接电阻R7的一端,电阻R7的另一端与电阻R6的一端相连后接入运算放大器U2的反向输入端,电容C4的另一端与电阻R6的另一端相连后接运算放大器U1的输出端;该电路的输出为VM6;
上述这些电路在音频的动态特性都能保留,但是可以根据实际电路需要设置多个高频零极点精确调整开关频率点的相位,其中电阻R3可以使得高频相位滞后,电阻R5使得相位超前,复杂的相位控制目的是:在开关频率范围,设置合适的相位斜率。图6是某些电路结构需要在原来的基础上加一级电路,这级电路上也可以加入一定的相位控制。当高频纹波相位精确调整好后,失真率可以降低两个数量级,全音频范围达到0.0002%以下。
如图7和图8所示为基础电压控制模块的高频纹波及采样点波形图,其中,VM1为基础电压控制模块的输出,sample为比较器翻转点,图7为信号输入电平为0时,高频开关占空比为50%,图8为大信号输入时,占空比偏离50%。可以看到VM1在开关翻转时采样点并不能反映VM1的平均电压,实际有很大的非线性,为电路的主要非线性失真。
如图9和图10所示为本发明的设有高频相位控制电路的电压控制模块高频纹波及采样点波形图,把开关频率点相位往后推移了接近90度。使得不同输入信号幅度时,VM2的正负开关切换采样点电压平均值正好为VM2的平均值,此时开关频率纹波的影响被消除。更复杂的相位控制可以控制的更精确,此非线性失真消除后,电路失真可以达到极低水平。
如图11和图12所示,VM1为满足电压外环动态特性的基本电路,后面几个电路动态特性类似,但是在开关频率点500KHZ附近相位相位做了调整。VM2为最简单的大致满足要求的电路,跟VM1比较开关频率点幅值大幅衰减,也就是控制电压纹波减小利于减小非线性,同时开关频率点相位大幅调整,使得比较器的正负跳变时间点VM2电压波动相加和为0,高频纹波因素得以抵消。VM3和VM4都可以在VM2基础上做包括相位斜率的修正,使得相位控制的更精确。
本发明的对输出LC滤波电路4的电容电流进行采样反馈,采用如下其中一种方式:如图13所示,输出LC滤波电路4的输出电容串联电流互感器,因为输出电容只有交流,所以电流互感器检测是可以稳定工作的,不过磁性元件可能有一定的非线性,需要极低失真度时不是太合适;
如图14所示,输出LC滤波电路4的输出电容串联电阻,电阻电压放大获得电容电流,缺点是所述电阻有一定的功率损耗发热,并且对输出电压有一点影响;
如图15所示,输出LC滤波电路4的输出电容并联一个小电容,小电容串联电阻再接地, 该电阻的电压放大经电阻电压放大器数倍作为近似输出电容电流信号。小电容串联电阻后,实际得到的是输出电容电流的低通滤波信号,电阻越小高频衰减越小,所以最好是用小电阻电压放大使用。用大些的电阻不做放大使用实际也可以工作,只是电流采样高频部分衰减更多,输出阻抗参数会变差。
如图16所示,输出LC滤波电路4的输出电容并联一个小电容,小电容串联电阻再接前级电路控制输出,采样经电阻电压放大器放大数倍得到近似输出电容电流信号的高频部分,前级反馈电路会维持小电容下端虚地,不过因为前级只是控制音频部分,高频部分还是滤波特性。
如图17所示,输出LC滤波电路4的输出电容并联一个小电容,小电容电流是输出电容按比例减小的镜像,使用虚地生成及电流检测模块在小电容下端形成虚地吸收它的电流,虚地生成及电流检测模块输出得到输出电容电流信号,同时可以实现检测小电容上的电流,此方式可以完美反映输出电容电流,并且失真很小。
如图18所示为实际应用电路中,为了使电流自激比较器工作在0点附近,使用前级电路控制输出VM(n)串联电阻来抵消镜像电流的低频部分,剩余的高频部分被虚地生成及电流检测模块吸收转换成高频电压信号,此信号与0电平上下的滞回电压比较产生PWM波。
具体的,如图19a所示,所述虚地生成及电流检测模块输入端外接低阻,吸收外部灌入电流,输出端输出从输入端灌入电流的检测值,得到输出电压信号V=K*I,其中K为常量,I为所述外部灌入电流。
如图19b和图19c所示,所述虚地生成及电流检测模块可以使用类似OPA860芯片或跨导放大器吸收电流形成虚地并把电流转换成电压。
所述输出LC滤波电路4的电容电流的反馈包含输出电感电流和负载电流,电感电流反馈提供了电路动态特性和稳定性,负载电流反馈使得输出阻抗达到极低程度。
由于自激振荡随着输出电压幅值变大,占空比发生变化,同时开关频率也会大幅变低,所述开关频率是输出电压、输出电感、电源电压和比较器滞回值的函数,并通过所述函数计算出频率稳定时需要的滞回值,开关频率变低使得电压控制模块1的环路必须适应低开关频率因而性能降低,同时也使得高频相位控制变复杂,所以本发明采用所述自激振荡频率调节模块7,用于动态调节开关频率的稳定性,大幅提升电路性能。
所述自激振荡频率调节模块7采用模拟电路或通过MCU计算得到并调节滞回值,从而稳定频率,所述模拟电路包括RC滤波电路,所述RC滤波电路接入用于代替输出电压的输入电压VIN,输出电压信号VIN2,所述输出电压信号VIN2可以模拟输出电压OUT相对 于输入电压VIN的相位滞后量,因而输出电压信号VIN2可以替代输出电压OUT用于计算,输出所述电压信号VIN2的绝对值和电源电压VCC至函数f2(abs(VIN2),VCC),得到频率稳定时需要的滞回值,并根据所述滞回值,通过滞回幅度控制电路稳定频率,其中,所述f2(abs(VIN2),VCC)模拟电路实现可做一定简化使用线性函数或折线法拟合。MCU计算实现方式可以得到任意的f2()函数。
电路的非线性失真除了高频纹波对控制电路的影响还来自另外一方面。大信号幅值时,输出电压OUT接近电源电压VCC,输出电感电流不再是三角波,而是有点圆弧变形,此时电流峰峰值的平均值不再代表电流的平均值,此时产生复杂的非线性特性。本发明的非线性补偿模块5,使用模拟电路或通过MCU计算来消除电路全信号幅度失真,模拟电路实现可默认VCC恒定并使用折线法拟合,MCU计算实现方式可以得到任意的f1()函数。
所述模拟电路设有RC滤波电路,RC滤波电路的输入端接输入电压VIN,输出端输出信号至非线性补偿函数电路,同时将电源电压VCC输入至非线性补偿函数f1(VIN2,VCC)电路,得到电路的补偿值,从而消除电路非线性失真。如图20所示,实际电路为了简化,电源电压VCC因为比较稳定可以忽略掉,非线性补偿函数模拟电路实现可以默认VCC恒定并使用折线法拟合,电路实现比较简单。

Claims (8)

  1. 一种应用于大功率范围的低失真D类功放,包括:电压控制模块(1)、电流自激内环振荡电路、MOS开关电路(3)、输出LC滤波电路(4),其特征在于,还包括非线性补偿模块(5)和自激振荡频率调节模块(7),所述电流自激内环振荡电路由比较器(2)和电流反馈连接比较器(2)的输出LC电容电流采样反馈电路(6)组成,所述电压控制模块(1)的输入端接输入电压VIN,输出端输出控制电压VM至比较器(2)的输入端,所述比较器(2)的输出端连接MOS开关电路(3)的输入端,MOS开关电路(3)的输出端连接输出LC滤波电路(4)的输入端,输出LC滤波电路(4)的输出端与电压控制模块(1)连接,输出LC滤波电路(4)的电容电流经输出LC电容电流采样反馈电路(6)反馈至比较器(2);所述非线性补偿模块(5)第一输入端接入输入电压VIN,第二输入端接电源电压VCC,输出端输出非线性补偿函数信号至比较器(2)的输入端;所述自激振荡频率调节模块(7)第一输入端接输入电压VIN,第二输入端接地电源电压VCC,输出端连接比较器(2)。
  2. 如权利要求1所述的一种应用于大功率范围的低失真D类功放,其特征在于,所述的电压控制模块(1)设有高频相位控制电路,所述高频相位控制电路设置多个高频零极点精确调整开关频率点的相位。
  3. 如权利要求1所述的一种应用于大功率范围的低失真D类功放,其特征在于,所述输出LC电容电流采样反馈电路(6),对输出LC滤波电路(4)的电容电流进行采样反馈,采用如下其中一种方式:
    a:输出LC滤波电路(4)的输出电容并联一个小电容,所述小电容电流是输出电容按比例减小的镜像,使用虚地生成及电流检测模块在电容下端形成虚地吸收它的电流,虚地生成及电流检测模块输出得到输出电容电流信号;
    b:输出LC滤波电路(4)的输出电容并联一个小电容,所述小电容电流是输出电容按比例减小的镜像,使用虚地生成及电流检测模块在小电容下端形成虚地吸收它的电流,在前级电路控制输出端串联电阻来抵消镜像电流的低频部分,虚地生成及电流检测模块输出得到输出电容电流信号的高频部分;
    c:输出LC滤波电路(4)的输出电容连接电流检测传感器,通过所述电流检测传感器检测得到采样电流,所述电流检测传感器包括电流互感器或电阻;
    d:输出LC滤波电路(4)的输出电容并联一个小电容,小电容串联电阻再接地,采样经电阻电压放大器放大数倍得到近似输出电容电流信号;
    e:输出LC滤波电路(4)的输出电容并联一个小电容,小电容串联电阻再接前级电路控制 输出,采样经电阻电压放大器放大数倍得到近似输出电容电流信号的高频部分。
  4. 如权利要求3所述的一种应用于大功率范围的低失真D类功放,其特征在于,所述虚地生成及电流检测模块输入端对外表现为低阻,吸收外部灌入电流,输出端输出从输入端灌入电流的检测值,得到输出电压信号V=K*I,其中K为常量,I为所述外部灌入电流。
  5. 如权利要求4所述的一种应用于大功率范围的低失真D类功放,其特征在于,所述虚地生成及电流检测模块使用OPA860芯片或跨导放大器吸收电流形成虚地并把电流转换成电压。
  6. 如权利要求1所述的一种应用于大功率范围的低失真D类功放,其特征在于,所述自激振荡频率调节模块(7)采用模拟电路或通过MCU计算得到并调节滞回值,所述模拟电路包括RC滤波电路,所述RC滤波电路接入输入电压VIN,输出电压信号VIN2用于代替输出电压OUT信号,输出所述电压信号VIN2的绝对值和电源电压VCC至函数f2(abs(VIN2),VCC)运算电路,得到频率稳定时需要的滞回值,并根据所述滞回值,通过滞回幅度控制电路稳定频率,其中,所述f2(abs(VIN2),VCC)模拟电路实现使用线性函数或折线法拟合。
  7. 如权利要求1所述的一种应用于大功率范围的低失真D类功放,其特征在于,所述非线性补偿模块(5),使用模拟电路或通过MCU计算来消除电路全信号幅度失真,所述模拟电路设有RC滤波电路,RC滤波电路的输入端接输入电压VIN,输出电压VIN2,同时将电源电压VCC输入至非线性补偿函数f1(VIN2,VCC)电路,得到电路的补偿值。
  8. 如权利要求7所述的一种应用于大功率范围的低失真D类功放,其特征在于,所述非线性补偿函数f1(VIN2,VCC)模拟电路实现使用折线法拟合。
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