WO2020238208A1 - 基于空域反馈的面向混合大规模mimo阵列的数字预失真结构 - Google Patents

基于空域反馈的面向混合大规模mimo阵列的数字预失真结构 Download PDF

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WO2020238208A1
WO2020238208A1 PCT/CN2019/130500 CN2019130500W WO2020238208A1 WO 2020238208 A1 WO2020238208 A1 WO 2020238208A1 CN 2019130500 W CN2019130500 W CN 2019130500W WO 2020238208 A1 WO2020238208 A1 WO 2020238208A1
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signal
feedback
unit
field
feedback unit
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French (fr)
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陈文华
刘昕
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清华大学
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/02—Transmitters
    • H04B1/04—Circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/06—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station
    • H04B7/0613—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission
    • H04B7/0615—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission of weighted versions of same signal
    • H04B7/0617—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission of weighted versions of same signal for beam forming
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/38—Synchronous or start-stop systems, e.g. for Baudot code
    • H04L25/40—Transmitting circuits; Receiving circuits
    • H04L25/49—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/02—Transmitters
    • H04B1/04—Circuits
    • H04B2001/0408—Circuits with power amplifiers
    • H04B2001/0425—Circuits with power amplifiers with linearisation using predistortion
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/02—Transmitters
    • H04B1/04—Circuits
    • H04B2001/0408—Circuits with power amplifiers
    • H04B2001/0441—Circuits with power amplifiers with linearisation using feed-forward

Definitions

  • the present disclosure relates to the field of mobile communication technology, and in particular to a digital predistortion structure and method for hybrid massive MIMO arrays based on spatial feedback.
  • the power amplifier is one of the most energy-consuming devices in the RF front-end, and its performance directly affects the overall transmitter efficiency.
  • high-order modulation methods such as CDMA and OFDM are generally used. While these modulation methods improve the spectrum efficiency, they also have a peak-to-average ratio (PAPR) problem.
  • PAPR peak-to-average ratio
  • a radio frequency power amplifier amplifies a non-constant envelope signal, due to the inconsistent gain characteristics of the power amplifier within the signal power dynamic range, nonlinear distortion will be superimposed in the output, deteriorating communication quality.
  • RF power amplifiers often work near the saturation zone to ensure high-efficiency output. At this time, the distortion caused by the power amplifier will be more serious, not only deteriorating its own communication quality, but even causing interference to nearby users. Therefore, effective power amplifier linearization technology must be used to ease the contradiction between efficiency and linearity.
  • DPD Digital predistortion
  • PA linearization technology is currently the most commonly used PA linearization technology in base stations, and its structure is shown in Figure 1. Based on accurate behavioral modeling, DPD can predict the nonlinear distortion of PA and eliminate them by adding appropriate correction signals.
  • a feedback loop is needed to collect the output signal of the power amplifier and process the transmission signal in the baseband digital domain. Therefore, the most basic condition for applying traditional DPD technology is that the number of digital channels and the number of power amplifiers are the same, which can ensure the one-to-one correspondence between the predistorter, power amplifier and feedback loop.
  • FIG. 2 is a schematic diagram of a massive MIMO system architecture based on a hybrid architecture.
  • one digital stream corresponds to an analog beamforming array, and the high gain of the large-scale antenna array generates strong directional beams to achieve spatial multiplexing.
  • the application of digital predistortion technology faces challenges-the configuration of the predistorter and feedback link.
  • the beam-oriented digital predistortion technology (BO-DPD) is proposed (see X. Liu et al., "Beam-Oriented Digital Predistortion for 5G Massive MIMO Hybrid Beamforming Transmitters," IEEE Trans.Microw.TheoryTechn ., vol.66, no.7, pp.3419-3432, July 2018.), linearize the composite signal in the beam direction of the sub-array.
  • the BO-DPD scheme can effectively solve the configuration problem of the predistorter in the hybrid beamforming array.
  • the scheme requires a complex feedback link and signal processing algorithm.
  • the BO-DPD scheme needs to output for each power amplifier Configure the feedback link at the end.
  • RF power amplifiers and antennas are often integrated in the same chip. Removing the bulky couplers and isolators between the power amplifier and the antenna has become a reality. foregone conclusion. How to configure the feedback link is another challenge faced by the DPD technology application in the hybrid beamforming system.
  • a digital predistortion structure for a hybrid massive MIMO array based on spatial feedback including: multiple sub-arrays; each sub-array includes:
  • the input signal is predistorted according to the feedback signal to obtain the predistorted signal
  • Analog domain part including: analog beamforming network, transmitting unit, near-field feedback unit or far-field feedback unit, feedback signal synthesizer;
  • the analog beamforming network processes the predistortion signal
  • the transmitting unit transmits the processed predistorted signal
  • the near-field feedback unit or the far-field feedback unit receives the transmitted signal from the transmitting unit and generates a received signal
  • the feedback signal synthesizer receives the received signal of the near-field feedback unit or the far-field feedback unit, obtains the feedback signal and sends it to the digital domain part.
  • a digital predistortion method for hybrid massive MIMO array based on spatial feedback including:
  • the digital domain part performs predistortion processing on the input signal according to the feedback signal to obtain a predistortion signal
  • the analog beamforming network processes the predistortion signal
  • the transmitting unit transmits the processed predistorted signal
  • the near-field feedback unit or the far-field feedback unit receives the transmitted signal from the transmitting unit and generates the received signal
  • the feedback signal synthesizer receives the received signal of the near-field feedback unit or the far-field feedback unit, obtains the feedback signal and sends it to the digital domain part.
  • Figure 1 is a typical structure diagram of digital predistortion in the prior art.
  • FIG. 2 is a structure diagram of a massive MIMO array based on a hybrid beamforming structure in the prior art.
  • Fig. 3 is a structural diagram of beam directional digital predistortion in the prior art.
  • FIG. 4 is a structural diagram of a hybrid massive MIMO array based on spatial diversity feedback according to an embodiment of the disclosure.
  • Figure 5(a) is a diagram of a single feedback configuration in the near-field feedback airspace.
  • Figure 5(b) is a single feedback configuration diagram in the far-field feedback airspace.
  • Figure 6(a) is a configuration diagram of near-field feedback spatial diversity feedback.
  • Figure 6(b) is a configuration diagram of far-field feedback spatial diversity feedback.
  • Figure 7 (a), Figure 7 (b), Figure 7 (c) are three types of feedback signal synthesizer structure diagram.
  • Fig. 8(a) is a diagram of the HFSS electromagnetic simulation antenna array
  • Fig. 8(b) is a diagram of the antenna element numbering of the HFSS electromagnetic simulation antenna array.
  • the present disclosure proposes a spatial feedback-based digital predistortion structure and method for hybrid massive MIMO arrays. Compared with the original BO-DPD scheme, it omits The feedback link configured after each power amplifier is used to couple the output information of the array with a small amount of feedback antennas, and the feedback signal is simply processed to approximate the main beam signal to realize the linearization of the sub-array beam.
  • a digital predistortion structure for hybrid massive MIMO arrays based on spatial feedback As shown in FIG. 4, sub-array 1, ..., sub-array i, ..., sub-array p, ..., etc. Sub-arrays. In a hybrid massive MIMO array, each sub-array generates a directional beam for the user through analog beamforming, and the input signals of each sub-array are x 1 (n),..., x i (n),..., x p (n), ... not relevant. Therefore, each sub-array can be equivalent to a phased array.
  • the predistorter is configured for each sub-array to linearize the transmission signal of the respective sub-array. According to the basic principle of the antenna array, the main beam signal of the array is the in-phase superposition of the signals of each channel. In this embodiment, obtaining an approximate main beam signal directly or through calculation is a very critical step.
  • the sub-array p includes: a digital domain part and an analog domain part.
  • the digital domain part includes: predistorter, predistortion trainer, feedback signal processing and main beam signal approximation processor.
  • the predistorter performs predistortion processing on the input signal x p (n) to obtain a predistortion signal z(n), and the predistortion signal z(n) is sent to the analog domain part through the transmission link.
  • the feedback signal processing and main beam signal approximation processor receives the feedback signal y F (n) via the feedback link, and approximates the feedback signal y F (n) to the main beam signal y R (n).
  • the predistortion trainer uses the input signal x p (n) and the main beam signal y R (n) to perform reverse modeling, identify and update the predistorter, and realize the linearization of the sub-array main beam signal.
  • the analog domain part includes: analog beamforming network, N-channel antenna unit, feedback signal synthesizer.
  • the analog beamforming network processes a channel of predistortion signal z(n), and outputs N channels of signals s 1 (n), ..., s N (n).
  • Each antenna unit includes: power amplifier and antenna. As shown in Figure 5(a), one of the N antenna units is used as the feedback unit, and the remaining N-1 antenna units are used as the transmitting unit. Any antenna unit can be used as a feedback unit, and a suitable unit can be selected according to the characteristics of the array and needs.
  • the N- 1 signals of the N signals s 1 (n), ..., s N (n) are respectively sent to the N-1 transmitting unit as the input signal of the power amplifier of the transmitting unit.
  • the power amplifier amplifies the input signal and transmits it through the antenna.
  • the feedback unit since the feedback unit is located in the near-field area of the array, the distance difference between each channel of the transmitting unit and the feedback unit cannot be ignored.
  • the feedback unit receives The signals of each channel may have phase cancellation, resulting in incomplete output information of each power amplifier fed back.
  • the behavioral characteristics of the power amplifier are mainly controlled by the amplitude of the input signal, and are basically independent of the phase of the input signal. Changing the phase of the input signal will not affect the nonlinear characteristics of the power amplifier. Therefore, in this embodiment, by controlling the weight of the analog beamforming network, the phase difference of each signal is adjusted so that the signal received by the feedback unit is the in-phase superposition of each signal, thereby ensuring the integrity of the feedback information to the greatest extent.
  • the received signal of the feedback unit is not considered. If the analog beamforming network is not considered, the received signal of the feedback unit is not considered.
  • y p is the output signal of the power amplifier of the p-th transmitting unit.
  • a phase shifter is configured before each power amplifier, and N phase shifters are formed to form an analog beamforming network, and the predistortion signal z(n) is processed. Adjust the phase offset of the phase shifter to change the phase of the input signal of the power amplifier of each transmitting unit, so that the input signal of the power amplifier of the p-th transmitting unit is Since the phase of the input signal basically does not affect the nonlinearity of the power amplifier, the output signal of the corresponding p-th transmitting unit power amplifier is Instead of y p , the received signal of the feedback unit of this embodiment is
  • the receiving signal of the feedback unit is formed by the in-phase superposition of the output signals of the power amplifiers of the transmitting units.
  • the feedback signal y F (n) is sent to the signal processing feedback signal to the main beam via a feedback link processor approximated .
  • the output signals of the power amplifiers of each transmitting unit reach the feedback unit in the same phase, but since the distance difference between the above-mentioned output signals to the feedback unit cannot be ignored, the feedback unit The strength of each output signal received is different. Therefore, in this scenario, the received signal of the feedback unit is not exactly the same as the main beam signal, and can be equivalent to an array sidelobe signal.
  • a hybrid massive MIMO array-oriented digital predistortion structure based on spatial feedback.
  • the same or similar features as those in the first embodiment will not be repeated. The following will only focus on the differences from the first embodiment. The characteristics of the case.
  • the N antenna units are all transmitting units, and a feedback unit is also provided.
  • the feedback unit is located in the far-field area outside the sub-array, and the feedback unit can be arranged at any point in the far-field area, and an appropriate placement position for the feedback unit can be selected according to the position of the sub-array and the characteristics of the sub-array.
  • the transmission coefficient from the p-th transmitting unit to the feedback unit is Since the feedback unit is located in the far field area, the attenuation value from each transmitting unit to the feedback unit is approximately the same, so the transmission coefficient is expressed as
  • the analog beamforming coefficient of the p-th transmitting unit is w p
  • the feedback unit receives the signal from the transmitting unit, the feed phase of each transmitting unit is controlled so that Then the signal received by the feedback unit is
  • the transmitting beam of the sub-array is directed to the feedback unit, that is, the signal received by the feedback unit is the in-phase superposition of the output signals of the power amplifiers of each transmitting unit.
  • the phase of the transmission coefficient is mainly determined by the characteristics of the array, so the phase shift coefficient w p is easy to design in advance. For example, in a uniform linear array, ⁇ is the wavelength of the carrier wave, d is the distance between the array units, and ⁇ is the angle between the position of the transmitting unit and the feedback unit.
  • the phase shift coefficient can be calculated Receiving a reception signal synthesizer receiving the feedback signal of the feedback unit, and the feedback unit as the feedback signal y F (n), the feedback signal y F (n) is sent to the signal processing feedback signal to the main beam via a feedback link processor approximated .
  • a hybrid massive MIMO array-oriented digital predistortion structure based on spatial diversity feedback.
  • the same or similar features as those in the above embodiments will not be repeated. The following will only focus on the differences from the above embodiments. Characteristics.
  • this embodiment proposes a feedback structure of the airspace diversity, as shown in FIG. 6(a).
  • the multiple antenna units among the N antenna units are used as feedback units, and the remaining antenna units are used as transmitting units.
  • the multiple feedback units are symmetrically arranged in the antenna array, as shown in the example shown in Fig. 6(a), in the first row, first column, first row, last column, last row, first column, and last row, last column of the antenna array.
  • the antenna unit as the feedback unit.
  • the signals of each feedback unit are added together to ensure as much as possible that the amplitudes of the transmitted signals of each channel in the added signal of each channel of feedback signal are similar.
  • the antenna array includes N antenna units in total, of which L antenna units are used as feedback units, and the remaining NL antenna units are transmitting units.
  • the coupling coefficient from the p-th transmitting unit to the k-th feedback unit is
  • each feedback unit receives signals in time sharing.
  • the phase of the output signal of the power amplifier of each transmission unit is controlled so that the received signal of the k-th feedback unit is still the in-phase superposition of the output signal of the power amplifier of each transmission unit. That is, when the k-th feedback unit is working, the analog beamforming network adjusts the phase of the input signal of the power amplifier of each transmitting unit so that the input signal of the p-th transmitting unit power amplifier is At this time, the received signal of the k-th feedback unit is:
  • the feedback signal synthesizer receives the received signals of each feedback unit, and adds the received signals of each feedback unit to obtain the total feedback signal as the feedback signal y F (n), and the feedback signal y F (n) is sent via the feedback link To the feedback signal processing and main beam signal approximation processor.
  • the feedback signal y F (n) is expressed as:
  • the position of the feedback unit is symmetrically designed (as in Figure 6(a), the transmitter units at the four corners of the square array are selected as the feedback unit), so that
  • the feedback signal processing and main beam signal approximation processor receives the feedback signal y F (n) via the feedback link, and approximates the feedback signal y F (n) to the main beam signal y R (n).
  • the main beam information can be directly obtained without complicated calculations.
  • the total feedback signal and the main beam The signal difference is only one coefficient, which can be eliminated by normalization.
  • the feedback signal synthesizer adopts the structure shown in FIG. 7(a).
  • the feedback signal synthesizer includes a radio frequency combiner.
  • the radio frequency combiner adds the received signals of each feedback unit, and sends the feedback signal y F (n) to the feedback signal processing and main beam signal approximation processor through a feedback link .
  • the feedback signal synthesizer may also adopt the structure shown in FIG. 7(b) and FIG. 7(c).
  • the feedback signal synthesizer includes a multiplexer, and the multiplexer sequentially sends the received signals of each feedback unit to the feedback signal processing and main beam signal approximation processor through a feedback link.
  • the feedback signal processing and main beam signal approximation processor adds the received signals of each feedback unit to obtain the feedback signal y F (n).
  • multiple feedback links are included, and the feedback signal synthesizer sends the received signals of each feedback unit to the feedback signal processing and main beam signal approximation processor through each feedback link.
  • the feedback signal processing and main beam signal approximation processor adds the received signals of each feedback unit to obtain the feedback signal y F (n).
  • the fourth embodiment of the present disclosure is a hybrid massive MIMO array based on spatial diversity feedback.
  • features that are the same as or similar to the foregoing embodiment will not be repeated, and only the features that are different from the foregoing embodiment will be described below.
  • the N antenna units are all transmitting units, and multiple feedback units are additionally provided, and the multiple feedback units are located in the far field area outside the sub-array, as shown in FIG. 6(b).
  • the far-field diversity structure of Fig. 6(b) is improved in that there is no need to adjust the analog beamforming network to direct the beam to the feedback unit during feedback reception.
  • This structure can by designing the position of the feedback unit, when the beam direction of the transmitting unit antenna changes, it can also ensure that one or more of the feedback units receive strong radiation signals, so as to ensure that a relatively complete array nonlinearity is collected. information.
  • the following is an example of a uniform linear array to illustrate the feedback unit position design method.
  • the antenna array is assumed to be a uniform linear array including N-channel transmitting units, and two feedback units are arranged, respectively located in the far field area with an angle of ⁇ 1 and ⁇ 2 with the transmitting array.
  • the array beam direction is ⁇ m
  • the simulated beamforming coefficient of the p-th transmitting unit is According to the far-field superposition, the signals received by the two feedback units are
  • the phase difference between the in-phase superimposition direction (main beam direction and sidelobe direction) and the nearest null direction signal is which is among them Indicates the phase of the main beam or sidelobe direction, Is the phase of the superimposed signal in the null direction; ⁇ null represents the null direction, ⁇ s represents the sidelobe or main beam azimuth, and ⁇ m represents the main beam direction; ⁇ is the carrier wavelength, and d is the distance between the transmitting units.
  • the angular difference between the main beam direction and the sidelobe direction has nothing to do with the feed phase difference ⁇ . Therefore, when the angles between the two feedback units and the transmitting unit meet It can be ensured that no matter where the beam direction points, at least one feedback unit can receive a strong signal.
  • the feedback signal synthesizer can adopt the structure shown in Figs. 7(a), 7(b) and 7(c).
  • the feedback signal synthesizer includes a radio frequency combiner, which sends the received signals of each feedback unit to the feedback signal processing and main beam signal approximation processor through a feedback link.
  • the feedback signal synthesizer includes a multiplexer, and the multiplexer sequentially sends the received signals of each feedback unit to the feedback signal processing and main beam signal approximation processor through a feedback link.
  • multiple feedback links are included, and the feedback signal synthesizer sends the received signals of each feedback unit to the feedback signal processing and main beam signal approximation processor through each feedback link.
  • the feedback signal processing and main beam signal approximation processor calculates the main beam signal according to the strength of the received signal of each feedback unit. Specifically,
  • the received signal of the feedback unit with the highest strength is approximated as the main beam signal, or the received signals of several larger feedback units are added together, and the result of the addition is approximately Main beam signal.
  • the received signals of each feedback unit are added, and the result of the addition is approximately the main beam signal.
  • the present disclosure uses a small amount of feedback antennas to couple the transmission information of the array, and uses the symmetry of the array and the feedback unit to restore the digital predistortion linearization target-the main beam signal, and solves the feedback configuration problem of the hybrid beamforming array DPD scheme. It is worth noting that the position of the feedback antenna in the present disclosure can be flexibly configured, that is, it can be a few units in the transmitting array, or it can be erected and configured separately in space.
  • the embodiment of the present disclosure provides a digital predistortion method for hybrid massive MIMO array based on spatial feedback, including the following steps:
  • the digital domain part performs predistortion processing on the input signal according to the feedback signal to obtain a predistortion signal
  • the analog beamforming network processes the predistortion signal
  • the transmitting unit transmits the processed predistorted signal
  • the near-field feedback unit or the far-field feedback unit receives the transmitted signal from the transmitting unit and generates a received signal
  • the feedback signal synthesizer receives the received signal of the near-field feedback unit or the far-field feedback unit, obtains the feedback signal and sends it to the digital domain part.
  • the weight of the analog beamforming network is controlled, and the receiving signal of the near-field feedback unit is used by the N-1 transmitting unit
  • the output signal of the power amplifier is composed of in-phase superposition, and the feedback signal synthesizer uses the received signal of the near-field feedback unit as the feedback signal.
  • the received signal of the far field feedback unit is in phase with the output signal of the power amplifier of the N transmitting unit Superposition composition; the feedback signal synthesizer uses the received signal of the far-field feedback unit as the feedback signal.
  • the phase of the input signal of the power amplifier of each transmitting unit is adjusted through the analog beamforming network, so that each path is close to
  • the received signal of the field feedback unit is the in-phase superposition of the output signals of the power amplifiers of each transmission unit; the feedback signal synthesizer adds the received signals of the near-field feedback units of each path to obtain the feedback signal.
  • the feedback signal synthesizer will The received signal of the far-field feedback unit is sent to the feedback signal processing and main beam signal approximation processor of the digital domain part; when the strength of the received signal of each feedback unit differs greatly, the feedback signal processing and the main beam signal approximation processor will The received signal of the largest feedback unit is approximately the main beam signal, or the received signals of several larger feedback units are added together, and the result of the addition is approximately the main beam signal; when the strength of the received signals of each feedback unit is similar , The feedback signal processing and main beam signal approximation processor adds the received signals of each feedback unit, and the result of the addition is approximate to the main beam signal.
  • Figure 8(a) shows the HFSS electromagnetic simulation antenna array
  • Figure 8(b) shows the antenna element numbers.
  • the present disclosure has conducted a joint simulation verification based on the HFSS electromagnetic simulation platform and the MATLAB platform.
  • the unit spacing is ⁇ /2 (half wavelength), and the center frequency 3.5GHz.
  • the simulation signal is a 10MHz bandwidth LTE signal
  • the power amplifier model is based on 64 different behavior models extracted from the GaN Doherty power amplifier.
  • Table 1 and Table 2 show the similarity between the total feedback signal and the main beam signal obtained when different feedback units are selected, and the predistortion performance.
  • the simulation results show the effectiveness of the near-field spatial feedback, and the performance of diversity feedback is better than single feedback.
  • test results verify the effectiveness of the far-field airspace feedback and show that the performance of the diversity feedback scheme is better than that of the single feedback scheme.

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Abstract

本公开提供了一种基于空域反馈的面向混合大规模MIMO阵列的数字预失真结构及方法，包括：多个子阵列；每个子阵列包括：数字域部分，根据反馈信号对输入信号进行预失真处理，得到预失真信号；模拟域部分，包括：模拟波束成形网络、发射单元、近场反馈单元或远场反馈单元、反馈信号综合器；模拟波束成形网络对预失真信号进行处理；发射单元发射处理后的预失真信号；近场反馈单元或远场反馈单元接收发射单元的发射信号，生成接收信号；反馈信号综合器接收近场反馈单元或远场反馈单元的接收信号，得到反馈信号并发送给数字域部分。

Description

基于空域反馈的面向混合大规模MIMO阵列的数字预失真结构 技术领域
本公开涉及移动通信技术领域，尤其涉及一种基于空域反馈的面向混合大规模MIMO阵列的数字预失真结构及方法。
背景技术
由于频谱资源的不可再生性，在现代通信系统中，频谱资源是最紧张和稀缺的资源之一。近年来，随着科技水平和生活水平的不断进步，人们对通信容量和传输速率提出了更高的需求。为了在不增加频谱资源和天线发射功率的情况下，成倍提高系统的信道容量，充分利用空间资源的多输入多输出(MIMO)技术成为了目前4G移动通信的核心技术。在下一代(5G)移动通信系统中，将采用大规模MIMO技术以进一步扩大空间自由度，从而提高系统容量、频谱效率和传输速率。
功率放大器(PA)作为射频前端中最耗能的器件之一，它的性能优劣直接影响着整个发射机效率。在现代通信系统中，一般采用CDMA、OFDM等高阶调制方式，这些调制方式在提高频谱效率的同时，也产生了高峰均比(PAPR)问题。射频功放在放大非恒包络信号时，由于在信号功率动态范围内功放增益特性不一致，将在输出中叠加产生非线性失真，恶化通信质量。通常情况下，射频功放往往工作在饱和区附近以保证高效率的输出，此时功放带来的失真将更加严重，不仅恶化自身的通信质量，甚至会对邻近用户造成干扰。因此，必须采用有效的功放线性化技术以缓和效率和线性度的矛盾。
数字预失真(DPD)技术是目前基站中最常用的PA线性化技术，其结构如图1所示。基于精确的行为建模，DPD可以预测PA的非线性失真，并通过添加适当的校正信号来消除它们。在基本的DPD模块中，需要反馈回路采集功放的输出信号，并在基带数字域对传输信号进行处理。因此，应用传统DPD技术的最基本条件是数字通道数量和功放数量相同，能够保证预失真器、功放及反馈回路的一一对应关系。
在即将到来的5G时代，大规模MIMO技术将成为5G关键技术之一。综合考虑成本和系统效率，基于混合架构的大规模MIMO系统具有十分光 明的应用前景。图2为基于混合架构的大规模MIMO系统架构示意图。在混合大规模MIMO阵列中，一路数字流对应一个模拟波束成形阵列，通过大规模天线阵列的高增益产生强定向性波束，实现空间复用。然而，在混合大规模MIMO系统中，数字预失真技术的应用面临着挑战——预失真器和反馈链路的配置问题。一方面，由于射频功放数量远大于数字信号流数，传统DPD技术中数字预失真器和功放的一一对应关系无法保证。为了解决这一问题，提出了波束定向数字预失真技术(BO-DPD)(参见X.Liu et al.，“Beam-Oriented Digital Predistortion for 5G Massive MIMO Hybrid Beamforming Transmitters，”IEEE Trans.Microw.Theory Techn.，vol.66，no.7，pp.3419-3432，July 2018.)，针对子阵列波束方向的合成信号进行线性化。BO-DPD方案能够有效解决混合波束成形阵列中预失真器的配置问题，然而，该方案需要复杂的反馈链路和信号处理算法，如图3所示，BO-DPD方案需要为每个功放输出端配置反馈链路。在5G技术绿色、小型化、高集成度的要求下，在大规模MIMO阵列的设计中，射频功放和天线往往集成在同一芯片中，去掉功放和天线之间笨重的耦合器、隔离器已成定局。如何配置反馈链路是混合波束成形系统中DPD技术应用面临的另一挑战。
在目前的已发表文献中，针对混合大规模MIMO系统的数字预失真方案成果较少，并且都采用了复杂的为每个单元配置反馈或简单的配置个别单元反馈的方式，前者在实际系统难以实现，后者则由于无法采集所有功放的输出信息，造成非线性信息缺失而影响DPD算法性能。
公开内容
根据本公开的一个方面，提供了一种基于空域反馈的面向混合大规模MIMO阵列的数字预失真结构，包括：多个子阵列；每个子阵列包括：
数字域部分，根据反馈信号对输入信号进行预失真处理，得到预失真信号；
模拟域部分，包括：模拟波束成形网络、发射单元、近场反馈单元或远场反馈单元、反馈信号综合器；
模拟波束成形网络对预失真信号进行处理；
发射单元发射处理后的预失真信号；
近场反馈单元或远场反馈单元接收发射单元的发射信号，生成接收信 号；
反馈信号综合器接收近场反馈单元或远场反馈单元的接收信号，得到反馈信号并发送给数字域部分。
根据本公开的另一个方面，提供了一种基于空域反馈的面向混合大规模MIMO阵列的数字预失真方法，包括：
数字域部分根据反馈信号对输入信号进行预失真处理，得到预失真信号；
模拟波束成形网络对预失真信号进行处理；
发射单元发射处理后的预失真信号；
近场反馈单元或远场反馈单元接收发射单元的发射信号，生成接收信号；
反馈信号综合器接收近场反馈单元或远场反馈单元的接收信号，得到反馈信号并发送给数字域部分。
为使本公开的上述目的、特征和优点能更明显易懂，下文特举优选实施例，并配合所附附图，作详细说明如下。
附图说明
为了更清楚地说明本公开实施例的技术方案，下面将对实施例中所需要使用的附图作简单地介绍。应当理解，以下附图仅示出了本公开的某些实施例，因此不应被看作是对范围的限定。对于本领域普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据这些附图获得其他相关的附图。
图1为现有技术的数字预失真典型结构图。
图2为现有技术的基于混合波束成形结构的大规模MIMO阵列结构图。
图3为现有技术的波束定向数字预失真结构图。
图4为本公开实施例的基于空域分集反馈的混合大规模MIMO阵列结构图。
图5(a)为近场反馈空域单一反馈配置图。
图5(b)为远场反馈空域单一反馈配置图。
图6(a)为近场反馈空域分集反馈配置图。
图6(b)为远场反馈空域分集反馈配置图。
图7(a)、图7(b)、图7(c)为三种反馈信号综合器的结构图。
图8(a)为HFSS电磁仿真天线阵列图，图8(b)为HFSS电磁仿真天线阵列的天线单元编号图。
具体实施方式
本公开针对混合大规模MIMO阵列中预失真反馈难配置的问题，提出了一种基于空域反馈的面向混合大规模MIMO阵列的数字预失真结构及方法，与原始的BO-DPD方案相比，省略了每路功率放大器后配置的反馈链路，利用少量的反馈天线耦合阵列输出信息，对反馈信号进行简单的处理以近似主波束信号，实现针对子阵列的波束的线性化。
为使本公开的目的、技术方案和优点更加清楚明白，以下结合具体实施例，并参照附图，对本公开进一步详细说明。其中一些但并非全部的实施例将被示出。实际上，本公开的各种实施例可以许多不同形式实现，而不应被解释为限于此数所阐述的实施例。在不冲突的情况下，本公开中的实施例及实施例中的特征可以相互组合。
本公开第一实施例的一种基于空域反馈的面向混合大规模MIMO阵列的数字预失真结构，如图4所示，子阵列1、…、子阵列i、…、子阵列p、…等多个子阵列。在混合大规模MIMO阵列中，每个子阵列通过模拟波束成形产生一个针对用户的定向性波束，各个子阵列的输入信号x 1(n)、...、x i(n)、...、x p(n)、...不相关。因此，每个子阵列可以等效成一个相控阵。预失真器针对每个子阵列配置，以线性化各自子阵列的发射信号。由天线阵列基本原理可知，阵列主波束信号为各路信号的同相叠加。在本实施例中，直接获得或通过计算得到近似的主波束信号是十分关键的一步。
在图4中，子阵列p包括：数字域部分和模拟域部分。
数字域部分包括：预失真器、预失真训练器、反馈信号处理与主波束信号近似处理器。
预失真器对输入信号x p(n)进行预失真处理，得到预失真信号z(n)，预失真信号z(n)经发射链路发送给模拟域部分。
反馈信号处理与主波束信号近似处理器经反馈链路接收反馈信号y F(n)，将反馈信号y F(n)近似为主波束信号y R(n)。
预失真训练器利用输入信号x p(n)和主波束信号y R(n)进行逆向建模，识别和更新预失真器，实现子阵列主波束信号的线性化。
模拟域部分包括：模拟波束成形网络、N路天线单元、反馈信号综合器。
模拟波束成形网络对一路预失真信号z(n)进行处理，输出N路信号s 1(n)、…、s N(n)。
N路天线单元形成天线阵列。每路天线单元包括：功率放大器和天线。如图5(a)所示，其中，N路天线单元中的一路天线单元作为反馈单元，剩余N-1路天线单元作为发射单元。任意天线单元都可作为反馈单元，根据阵列特性和需要选择合适的单元即可。
N路信号s 1(n)、…、s N(n)中的N-1路信号分别发送给N-1路发射单元，作为发射单元功率放大器的输入信号。功率放大器将输入信号放大后通过天线发射。
本实施例中，由于反馈单元位于阵列近场区，各路发射单元与反馈单元的距离差不可忽略，在子阵列中由于各路信号具有强相关性(仅存在相位差)，反馈单元接收的各路信号可能存在相位抵消的情况，导致反馈的各路功放输出信息不全。幸运的是，功放的行为特性主要受输入信号的幅度控制，与输入信号的相位基本无关，改变输入信号的相位不会对功放的非线性特性产生影响。因此，本实施例通过控制模拟波束成形网络权重，调节各路信号相位差，使得反馈单元接收到的信号是各路信号的同相叠加，从而在最大程度上保证反馈信息的完整度。
如果不考虑模拟波束成形网络，反馈单元的接收信号为
Figure PCTCN2019130500-appb-000001
其中，
Figure PCTCN2019130500-appb-000002
为第p路发射单元到反馈单元的耦合系数，可通过阵列设计时的仿真，或者出厂测试提前得到；y p为第p路发射单元的功率放大器的输出信号。
本实施例中，每路功放前配置一个移相器，组成了N路移相器组成模拟波束成形网络，对预失真信号z(n)进行处理。调节移相器的相位偏移量 以改变各路发射单元功率放大器输入信号的相位，使第p路发射单元功率放大器的输入信号为
Figure PCTCN2019130500-appb-000003
由于该输入信号相位基本不会影响功率放大器的非线性，则对应的第p路发射单元功率放大器的输出信号为
Figure PCTCN2019130500-appb-000004
而非y p，本实施例的反馈单元的接收信号为
Figure PCTCN2019130500-appb-000005
由此可见，反馈单元的接收信号由各路发射单元功率放大器的输出信号的同相叠加构成。
反馈信号综合器接收反馈单元的接收信号，并将反馈单元的接收信号作为反馈信号y F(n)，反馈信号y F(n)经反馈链路发送至反馈信号处理与主波束信号近似处理器。
在图5(a)所示的近场反馈中，可保证各路发射单元功率放大器的输出信号到达反馈单元的相位相同，但由于上述各路输出信号到达反馈单元的路程差不可忽略，反馈单元接收到各路输出信号的强度是不同的。因此，该场景下反馈单元的接收信号与主波束信号不完全相同，可以等效为阵列副瓣信号。
本公开第二实施例基于空域反馈的面向混合大规模MIMO阵列的数字预失真结构，为简要起见，与第一实施例相同或相似的特征不再赘述，以下仅重点描述其不同于第一实施例的特征。
如图5(b)所示，本实施例中，N路天线单元均为发射单元，另设一反馈单元。反馈单元位于子阵列外的远场区，反馈单元可配置在远场区任一点，可根据子阵列位置、子阵列特性为反馈单元选择合适的放置位置。设反馈单元位于发射阵列远场区某点，第p个发射单元到反馈单元的传输系数为
Figure PCTCN2019130500-appb-000006
由于反馈单元位于远场区，各发射单元到反馈单元的 衰减值近似相同，则传输系数表示为
Figure PCTCN2019130500-appb-000007
设第p个发射单元的模拟波束成形系数为w p，当反馈单元接收发射单元的信号时，控制各发射单元的馈电相位使
Figure PCTCN2019130500-appb-000008
则反馈单元接收到的信号为
Figure PCTCN2019130500-appb-000009
此时子阵列的发射波束指向反馈单元，即反馈单元接收到的信号为各路发射单元功率放大器的输出信号的同相叠加。值得注意的是，在远场反馈环境下，传输系数的相位主要由阵列特性决定，因此移相系数w p是便于提前设计的。例如，在均匀直线阵中，
Figure PCTCN2019130500-appb-000010
λ为载波波长，d为阵列单元间距，φ为发射单元与反馈单元位置的夹角。此时，可计算得到移相系数
Figure PCTCN2019130500-appb-000011
反馈信号综合器接收反馈单元的接收信号，并将反馈单元的接收信号作为反馈信号y F(n)，反馈信号y F(n)经反馈链路发送至反馈信号处理与主波束信号近似处理器。
本公开第三实施例基于空域分集反馈的面向混合大规模MIMO阵列的数字预失真结构，为简要起见，与上述实施例相同或相似的特征不再赘述，以下仅重点描述其不同于上述实施例的特征。
在第一和第二实施例的空域单一反馈结构的基础上，本实施例提出了空域分集反馈结构，如图6(a)所示。N路天线单元中的多路天线单元作为反馈单元，剩余的天线单元作为发射单元。多路反馈单元对称设置于天线阵列中，如图6(a)所示的示例，在天线阵列的第一行第一列、第一行最后一列、最后一行第一列、以及最后一行最后一列的天线单元作为反馈单元。将各反馈单元的信号相加，以尽可能保证在各路反馈信号的相加信 号中，各路发射信号的幅度相似。
天线阵列共包括N路天线单元，其中L路天线单元作为反馈单元，其余N-L路天线单元为发射单元。第p路发射单元到第k路反馈单元的耦合系数为
Figure PCTCN2019130500-appb-000012
本实施例中，各路反馈单元分时接收信号。在第k路反馈单元工作的时隙内，控制各路发射单元功率放大器的输出信号的相位，使第k路反馈单元的接收信号仍是各路发射单元功率放大器的输出信号的同相叠加。即第k路反馈单元工作时，模拟波束成形网络调节各路发射单元功率放大器输入信号的相位，使第p路发射单元功率放大器的输入信号为
Figure PCTCN2019130500-appb-000013
此时该第k路反馈单元的接收信号为：
Figure PCTCN2019130500-appb-000014
反馈信号综合器接收各路反馈单元的接收信号，并将各路反馈单元的接收信号相加，得到总反馈信号作为反馈信号y F(n)，反馈信号y F(n)经反馈链路发送至反馈信号处理与主波束信号近似处理器。其中，反馈信号y F(n)表示为：
Figure PCTCN2019130500-appb-000015
根据阵列几何结构，对称设计反馈单元位置(如在图6(a)中，选择方形阵列四个角落处的发射单元作为反馈单元)，使
Figure PCTCN2019130500-appb-000016
其中，
Figure PCTCN2019130500-appb-000017
为各路发射单元的功率放大器的输出信号的同相叠加。反馈信号处理与主波束信号近似处理器经反馈链路接收反馈信号y F(n)，将反馈信号y F(n)近似为主波束信号y R(n)。
由此可见，本实施例通过设计反馈单元位置使总反馈信号中各路发射信号的幅度近似相同，则可以在不经过复杂运算的情况下直接获取主波束信息，此时总反馈信号和主波束信号仅相差一个系数，通过归一化可以消除。
本实施例中，反馈信号综合器采用图7(a)所示的结构。反馈信号综合器包括射频合路器，射频合路器将各路反馈单元的接收信号相加，通过一个反馈链路将反馈信号y F(n)发送给反馈信号处理与主波束信号近似处理器。本实施例中，反馈信号综合器也可以采用图7(b)和图7(c)所示的结构。在图7(b)中，反馈信号综合器包括复用器，复用器依次将各路反馈单元的接收信号通过一个反馈链路发给反馈信号处理与主波束信号近似处理器。反馈信号处理与主波束信号近似处理器将各路反馈单元的接收信号相加，得到反馈信号y F(n)。在图7(c)中，包括多个反馈链路，反馈信号综合器将各路反馈单元的接收信号通过各个反馈链路发给反馈信号处理与主波束信号近似处理器。反馈信号处理与主波束信号近似处理器将各路反馈单元的接收信号相加，得到反馈信号y F(n)。
本公开第四实施例基于空域分集反馈的混合大规模MIMO阵列，为简要起见，与上述实施例相同或相似的特征不再赘述，以下仅重点描述其不同于上述实施例的特征。
本实施例中，N路天线单元均为发射单元，另设多路反馈单元，多路反馈单元位于子阵列外的远场区，如图6(b)所示。图6(b)的远场分集结构与图5(b)中的远场单一反馈方式相比，改进之处在于反馈接收时 无需调节模拟波束成形网络使波束指向反馈单元。
该结构能够通过设计反馈单元的位置，在发射单元天线的波束方向变化时，也能保证各路反馈单元中有一路及以上接收到较强的辐射信号，从而保证采集到较为完整的阵列非线性信息。下面通过均匀直线阵的例子说明反馈单元位置设计方式。
设天线阵列为包括N路发射单元的均匀直线阵，设置两路反馈单元，分别位于与发射阵列夹角为φ 1和φ 2的远场区。当阵列波束方向为φ m时，第p个发射单元的模拟波束成形系数为
Figure PCTCN2019130500-appb-000018
根据远场叠加性，则两个反馈单元接收的信号分别为
Figure PCTCN2019130500-appb-000019
Figure PCTCN2019130500-appb-000020
对于均匀直线阵的远场响应而言，同相叠加方向(主波束方向及副瓣方向)和相距最近的零陷方向信号的相位差为
Figure PCTCN2019130500-appb-000021
即
Figure PCTCN2019130500-appb-000022
其中
Figure PCTCN2019130500-appb-000023
表示主波束或者副瓣方向相位，
Figure PCTCN2019130500-appb-000024
为零陷方向叠加信号的相位；φ null表示零陷方向，φ s代表副瓣或主波束方位角，φ m表示主波束方向；λ为载波波长，d为发射单元间距。显然，主波束方向和副瓣方向之间的角度差距与馈电相位差β无关。因此，当两个反馈单元与发射单元的角度满足
Figure PCTCN2019130500-appb-000025
能够保证无论波束方向指向何处，至少存在一路反馈单元能接收到较强的信号。
本实施例中，反馈信号综合器可采用图7(a)、7(b)和图7(c)所示的结构。在图7(a)中，反馈信号综合器包括射频合路器，将各路反馈单元的接收信号通过一个反馈链路发给反馈信号处理与主波束信号近似处理器。在图7(b)中，反馈信号综合器包括复用器，复用器依次将各路反馈单元的接收信号通过一个反馈链路发给反馈信号处理与主波束信号近似处理器。在图7(c)中，包括多个反馈链路，反馈信号综合器将各路反馈单元的接收信号通过各个反馈链路发给反馈信号处理与主波束信号近似处理器。
反馈信号处理与主波束信号近似处理器根据各路反馈单元的接收信号的强度，计算主波束信号。具体来说，
当各路反馈单元的接收信号的强度差异大时，将强度最大的一路反馈单元的接收信号近似为主波束信号，或将较大的几路反馈单元的接收信号相加，相加结果近似为主波束信号。
当各路反馈单元的接收信号的强度相仿时，将各路反馈单元的接收信号相加，相加结果近似为主波束信号。
本公开通过少量的反馈天线耦合阵列的发射信息，利用阵列和反馈单元的对称性恢复数字预失真线性化目标——主波束信号，解决了混合波束成形阵列DPD方案的反馈配置问题。值得注意的是，本公开中的反馈天线位置可灵活配置，即它可以是发射阵列中的某几个单元，也可以在空间中另行架设配置。
本公开实施例提供了一种基于空域反馈的面向混合大规模MIMO阵列的数字预失真方法，包括以下步骤：
数字域部分根据反馈信号对输入信号进行预失真处理，得到预失真信号；
模拟波束成形网络对预失真信号进行处理；
发射单元发射处理后的预失真信号；
近场反馈单元或远场反馈单元接收发射单元的发射信号，生成接收信 号；
反馈信号综合器接收近场反馈单元或远场反馈单元的接收信号，得到反馈信号并发送给数字域部分。
当N路天线单元中的一路天线单元作为近场反馈单元，剩余N-1路天线单元作为发射单元时，控制模拟波束成形网络的权重，近场反馈单元的接收信号由N-1路发射单元的功率放大器的输出信号的同相叠加构成，反馈信号综合器将近场反馈单元的接收信号作为反馈信号。
当N路天线单元均作为发射单元，另设一路位于子阵列远场区的反馈单元，作为远场反馈单元时，远场反馈单元的接收信号由N路发射单元的功率放大器的输出信号的同相叠加构成；反馈信号综合器将远场反馈单元的接收信号作为反馈信号。
当N路天线单元中的位置对称的多路天线单元作为近场反馈单元，剩余的天线单元作为发射单元时，通过模拟波束成形网络调节各路发射单元功率放大器输入信号的相位，使每路近场反馈单元的接收信号为各路发射单元功率放大器的输出信号的同相叠加；反馈信号综合器将各路近场反馈单元的接收信号相加，得到反馈信号。
当N路天线单元均作为发射单元，另设多路位于子阵列远场区的反馈单元，作为远场反馈单元时，无需调节模拟波束成形网络使波束指向反馈单元；反馈信号综合器将各路远场反馈单元的接收信号发给数字域部分的反馈信号处理与主波束信号近似处理器；当各路反馈单元的接收信号的强度差异大时，反馈信号处理与主波束信号近似处理器将强度最大的一路反馈单元的接收信号近似为主波束信号，或将较大的若干路反馈单元的接收信号相加，相加结果近似为主波束信号；当各路反馈单元的接收信号的强度相仿时，反馈信号处理与主波束信号近似处理器将各路反馈单元的接收信号相加，相加结果近似为主波束信号。
以下通过仿真及实验验证显示本公开技术方案的效果。
1)64单元阵列仿真
图8(a)为HFSS电磁仿真天线阵列，图8(b)显示了天线单元编号。针对图5(a)和图6(a)所示的近场空域反馈结构，本公开进行了基于HFSS电磁仿真平台及MATLAB平台的联合仿真验证，单元间距λ/2(半 波长)，中心频率3.5GHz。仿真信号为10MHz带宽LTE信号，功放模型为基于GaN Doherty功放提取的64个不同行为模型。表1和表2是选择不同反馈单元时，得到的总反馈信号和主波束信号的相似程度，以及预失真性能。仿真结果说明了近场空域反馈的有效性，分集反馈的性能优于单一反馈。
表1近场空域单一反馈仿真性能
Figure PCTCN2019130500-appb-000026
表2近场空域分集反馈仿真性能
Figure PCTCN2019130500-appb-000027
2)4单元阵列实验验证
针对图5(b)和图6(b)所示的远场空域反馈结构，搭建了4通道测试平台。单元间距15cm，中心频率3.5GHz，测试信号为20MHz带宽64QAM调制信号，被测功放为4个GaN Class AB功放。表3和表4不同波束方向时观察到的预失真性能，即主波束信号线性度。测试结果验证了远场空域反馈的有效性，且说明了分集反馈的方案性能优于单一反馈方案的性能。
表3主波束方向为90°时观察的主波束线性度
Figure PCTCN2019130500-appb-000028
表4主波束方向为79°时观察的主波束线性度
Figure PCTCN2019130500-appb-000029
以上的详细描述通过使用示意图、流程图和/或示例，已经阐述了上述空净一体机的众多实施例。在这种示意图、流程图和/或示例包含一个或多个功能和/或操作的情况下，本领域技术人员应理解，这种示意图、流程图或示例中的每一功能和/或操作可以通过各种结构、硬件、软件、固件或实质上它们的任意组合来单独和/或共同实现。
除非存在技术障碍或矛盾，本公开的上述各种实施例可以自由组合以形成另外的实施例，这些另外的实施例均在本公开的保护范围中。
虽然结合附图对本公开进行了说明，但是附图中公开的实施例旨在对本公开优选实施方式进行示例性说明，而不能理解为对本公开的一种限制。附图中的尺寸比例仅仅是示意性的，并不能理解为对本公开的限制。
虽然本公开总体构思的一些实施例已被显示和说明，本领域普通技术人员将理解，在不背离本公开公开构思的原则和精神的情况下，可对这些实施例做出改变，本公开的范围以权利要求和它们的等同物限定。

Claims (10)

  1. 一种基于空域反馈的面向混合大规模MIMO阵列的数字预失真结构，包括：多个子阵列；每个子阵列包括：
    数字域部分，根据反馈信号对输入信号进行预失真处理，得到预失真信号；
    模拟域部分，包括：模拟波束成形网络、发射单元、近场反馈单元或远场反馈单元、反馈信号综合器；
    模拟波束成形网络对预失真信号进行处理；
    发射单元发射处理后的预失真信号；
    近场反馈单元或远场反馈单元接收发射单元的发射信号，生成接收信号；
    反馈信号综合器接收近场反馈单元或远场反馈单元的接收信号，得到反馈信号并发送给数字域部分。
  2. 如权利要求1所述的数字预失真结构，模拟域部分包括：N路天线单元，N路天线单元中的一路天线单元作为近场反馈单元，剩余N-1路天线单元作为发射单元。
  3. 如权利要求2所述的数字预失真结构，通过控制模拟波束成形网络的权重；近场反馈单元的接收信号由N-1路发射单元的功率放大器的输出信号的同相叠加构成；
    反馈信号综合器将近场反馈单元的接收信号作为反馈信号。
  4. 如权利要求1所述的数字预失真结构，模拟域部分包括：N路天线单元，N路天线单元均作为发射单元，另设一路位于子阵列远场区的反馈单元，作为远场反馈单元。
  5. 如权利要求4所述的数字预失真结构，远场反馈单元的接收信号由N路发射单元的功率放大器的输出信号的同相叠加构成；
    反馈信号综合器将远场反馈单元的接收信号作为反馈信号。
  6. 如权利要求1所述的数字预失真结构，模拟域部分包括：N路天线单元，N路天线单元中的位置对称的多路天线单元作为近场反馈单元，剩余的天线单元作为发射单元。
  7. 如权利要求6所述的数字预失真结构，通过模拟波束成形网络调 节各路发射单元功率放大器输入信号的相位，使每路近场反馈单元的接收信号为各路发射单元功率放大器的输出信号的同相叠加；
    反馈信号综合器将各路近场反馈单元的接收信号相加，得到反馈信号。
  8. 如权利要求1所述的数字预失真结构，模拟域部分包括：N路天线单元，N路天线单元均作为发射单元，另设多路位于子阵列远场区的反馈单元，作为远场反馈单元。
  9. 如权利要求6所述的数字预失真结构，无需调节模拟波束成形网络使波束指向反馈单元；
    反馈信号综合器将各路远场反馈单元的接收信号发给数字域部分的反馈信号处理与主波束信号近似处理器；
    当各路反馈单元的接收信号的强度差异大时，反馈信号处理与主波束信号近似处理器将强度最大的一路反馈单元的接收信号近似为主波束信号，或将较大的若干路反馈单元的接收信号相加，相加结果近似为主波束信号；
    当各路反馈单元的接收信号的强度相仿时，反馈信号处理与主波束信号近似处理器将各路反馈单元的接收信号相加，相加结果近似为主波束信号。
  10. 一种基于空域反馈的面向混合大规模MIMO阵列的数字预失真方法，包括：
    数字域部分根据反馈信号对输入信号进行预失真处理，得到预失真信号；
    模拟波束成形网络对预失真信号进行处理；
    发射单元发射处理后的预失真信号；
    近场反馈单元或远场反馈单元接收发射单元的发射信号，生成接收信号；
    反馈信号综合器接收近场反馈单元或远场反馈单元的接收信号，得到反馈信号并发送给数字域部分。
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