WO2020232972A1 - 一种开关变换器及其控制方法 - Google Patents

一种开关变换器及其控制方法 Download PDF

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Publication number
WO2020232972A1
WO2020232972A1 PCT/CN2019/113678 CN2019113678W WO2020232972A1 WO 2020232972 A1 WO2020232972 A1 WO 2020232972A1 CN 2019113678 W CN2019113678 W CN 2019113678W WO 2020232972 A1 WO2020232972 A1 WO 2020232972A1
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inductor
switch
switching tube
voltage
current
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PCT/CN2019/113678
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English (en)
French (fr)
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卢鹏飞
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广州金升阳科技有限公司
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to a switching power supply, in particular to a switching converter circuit and a control method thereof.
  • FIG 1 shows the traditional Buck_Boost circuit.
  • the effective value of the current is larger when the circuit works in discontinuous mode.
  • the MOS tube Q1 is hard-switched, and the conduction loss of D1 is large.
  • Figure 2 shows the CUK circuit, which is a fourth-order or even higher-order circuit.
  • the dynamic process is complex and the output is easy to overshoot.
  • the circuit works in discontinuous mode and the current effective value is large.
  • the MOS transistor Q1 is hard-switched, and the conduction loss of D1 is large.
  • the traditional Buck_Boost circuit and CUK circuit are both reverse polarity circuits with inverted input and output voltages.
  • the current effective value of the circuit is large when the circuit works in the discontinuous mode, and the conduction loss is too large; the MOS transistor Q1 is hard-switched and the switching loss is large, so it is not suitable for high-voltage input and high-frequency applications.
  • the absolute value of the ratio between the intermittent mode and the input/output voltage is greater than 2, the demagnetization time of the inductor L1 is too long, and the demagnetization time is proportional to the size of the output current. It is difficult to compromise between large current output and high frequency. The problem.
  • the present invention proposes a switching converter circuit and its control method.
  • the circuit works in the discontinuous mode to reduce the effective value of the current and solve the problem of large conduction loss; all switches
  • the tube realizes the ZVS turn-on.
  • the absolute value of the ratio between the discontinuous mode and the input-output voltage is greater than 2, the problem of too long demagnetization time of the inductor L1 and difficulty in large current output and high frequency is solved.
  • a switching converter including positive input power supply, negative output voltage, power supply common ground, switching tube Q1, switching tube Q2, switching tube Q3, switching tube Q4, inductor L1 and capacitor C1; the drain and switch of switching tube Q1
  • the drain of the tube Q3 is connected to the input power supply
  • the source of the switching tube Q1 and the drain of the switching tube Q2 are connected to one end of the inductor L1
  • the source of the switching tube Q3 and the drain of the switching tube Q4 are connected to the inductor L1
  • the source of the switch Q4 is connected to one end of the capacitor C1
  • the source of the switch Q2 and the other end of the capacitor C1 are connected to the power supply common ground.
  • the switching tube Q1, the switching tube Q2, the switching tube Q3, and the switching tube Q4 are MOS tubes, triodes or IGBTs.
  • the first control method of the above switching converter is:
  • the switch Q1 is turned off at the end of the previous cycle. Since the current of the inductor L1 is negative, the current IL of the inductor L1 charges the output capacitor Coss1 of the switch Q1, the output capacitor Coss2 of the switch Q2 discharges, and one end of the inductor L1 The voltage drops from Vin to 0V, and the switch Q2 realizes ZVS turn-on; since the switch Q3 is in the conducting state, the voltage across the inductor L1 is Vin, the voltage of Vin excites the inductor L1, and the current IL of the inductor L1 rises, Turn off the switch Q3 according to the closed-loop control requirements; the current IL of the inductor L1 charges the output capacitor Coss3 of the switch Q3 and discharges the output capacitor Coss4 of the switch Q4, so that the voltage at the other end of the inductor L1 is reduced from Vin to Vo ,
  • the switching tube Q4 realizes the ZVS opening; the voltage across the inductor L1 is Vo,
  • the current IL of the inductor L1 charges the output capacitor Coss4 of the switch Q4 and charges the output capacitor Coss3 of the switch Q3 Discharging causes the voltage at the other end of the inductor L1 to rise from Vo to Vin, and the switching tube Q3 realizes ZVS opening; thus, the switching tube Q1, the switching tube Q2, the switching tube Q3 and the switching tube Q4 all realize the ZVS opening in one cycle. Then according to the closed-loop control requirement, the switch Q1 is turned off, and the next cycle is turned on.
  • Stage t1 ⁇ t2 After the switching tube Q3 is turned off, the current IL of the inductor L1 charges the output capacitance Coss3 of the switching tube Q3 and discharges the output capacitance Coss4 of the switching tube Q4. At t2, the voltage at the other end of the inductor L1 drops from Vin to Vo, and the switch Q4 realizes ZVS turn-on;
  • Stage t3 ⁇ t4 The current IL of the inductor L1 charges the output capacitor Coss2 of the switch Q2 and discharges the output capacitor Coss1 of the switch Q1. At t4, the voltage at one end of the inductor L1 rises from 0V to Vin, and the switch Q1 realizes ZVS opened;
  • the current IL of the inductor L1 charges the output capacitor Coss4 of the switch Q4 and discharges the output capacitor Coss3 of the switch Q3.
  • the voltage at the other end of the inductor L1 rises from Vo to Vin, and the switch Q3 Realize the opening of ZVS;
  • Stage t7 ⁇ t0+Tx The current IL of the inductor L1 charges the output capacitor Coss1 of the switch Q1, and the output capacitor Coss2 of the switch Q2 discharges.
  • the voltage at one end of the inductor L1 drops from Vin to 0V, and the switch Manage Q2 to realize ZVS opening;
  • the second control method of the above switching converter is:
  • the switch Q4 is turned off at the end of the previous cycle. Since the current of the inductor L1 is negative, the current IL of the inductor L1 charges the output capacitor Coss4 of the switch Q4, the output capacitor Coss3 of the switch Q3 discharges, and the inductor L1 is another The voltage at one end rises from Vo to Vin, the switch Q3 realizes ZVS turn-on; since the switch Q2 is in the on state, the voltage across the inductor L1 is Vin, the voltage of Vin excites the inductor L1, and the current IL of the inductor L1 rises According to the closed-loop control requirements, the switch Q3 is turned off; the current IL of the inductor L1 charges the output capacitor Coss3 of the switch Q3 and discharges the output capacitor Coss4 of the switch Q4, so that the voltage at the other end of the inductor L1 drops from Vin to Vo, the switch tube Q4 realizes ZVS turn-on; the voltage across the inductor L1 is Vo, the in
  • the switch Q1 When the current IL of the inductor L1 drops to zero, the switch Q1 is turned off, the output capacitor Coss1 of the switch Q1 starts to charge, the output capacitor Coss2 of the switch Q2 discharges, and the current IL of the inductor L1 Decreasing from zero to negative current, the voltage at one end of the inductor L1 drops from Vin to 0V, and the switching tube Q2 realizes ZVS opening; thus the switching tube Q1, the switching tube Q2, the switching tube Q3 and the switching tube Q4 are realized in one cycle ZVS opened. Then according to the closed-loop control requirements, the switch Q4 is turned off, and the next cycle is turned on.
  • Stage t0 ⁇ t1 At t0, the switch Q3 is turned on, the voltage across the inductor L1 is Vin, and the inductor L1 is excited, the current IL of the inductor L1 rises, and the switch Q3 is turned off at t1;
  • Stage t1 ⁇ t2 After the switching tube Q3 is turned off, the current IL of the inductor L1 charges the output capacitance Coss3 of the switching tube Q3 and discharges the output capacitance Coss4 of the switching tube Q4. At t2, the voltage at the other end of the inductor L1 drops from Vin to Vo, and the switch Q4 realizes ZVS turn-on;
  • Stage t3 ⁇ t4 The current IL of the inductor L1 charges the output capacitor Coss2 of the switch Q2 and discharges the output capacitor Coss1 of the switch Q1. At t4, the voltage at one end of the inductor L1 rises from 0V to Vin, and the switch Q1 realizes ZVS opened;
  • Stage t7 ⁇ t0+Tx The current of the inductor L1 discharges the output capacitor Coss3 of the switch Q3 and charges the output capacitor Coss4 of the switch Q4. At t0+Tx, the voltage at the other end of the inductor L1 rises from Vo to Vin. Switch Q3 realizes ZVS opening;
  • the second control method is characterized in that: when the load becomes lighter, the time from the switching tube Q3 on to the off phase of the switching tube Q1 begins to decrease, and the time from switching on to the switching tube Q4 off phase lengthen.
  • the present invention also provides another switching converter with the same inventive concept.
  • the technical solution is as follows:
  • a switching converter including positive input power supply, negative output voltage, power supply common ground, diode D1, switching tube Q2, switching tube Q3, switching tube Q4, inductor L1 and capacitor C1; the cathode of diode D1 and switching tube Q3
  • the drain is connected to the input power supply, the anode of the diode D1 and the drain of the switch Q2 are connected to one end of the inductor L1, the source of the switch Q3 and the drain of the switch Q4 are connected to the other end of the inductor L1, the switch The source of the tube Q4 is connected to one end of the capacitor C1, and the source of the switching tube Q2 and the other end of the capacitor C1 are connected to the power supply common ground.
  • the switching tube Q2, the switching tube Q3 and the switching tube Q4 are MOS tubes, triodes or IGBTs.
  • the control method of the above switching converter is:
  • the switch Q4 is turned off at the end of the previous cycle. Since the current of the inductor L1 is negative, the current IL of the inductor L1 charges the output capacitor Coss4 of the switch Q4, the output capacitor Coss3 of the switch Q3 discharges, and the inductor L1 is another The voltage at one end rises from Vo to Vin, the switch Q3 realizes ZVS turn-on; since the switch Q2 is in the on state, the voltage across the inductor L1 is Vin, the voltage of Vin excites the inductor L1, and the current IL of the inductor L1 rises According to the closed-loop control requirements, the switch Q3 is turned off; the current IL of the inductor L1 charges the output capacitor Coss3 of the switch Q3 and discharges the output capacitor Coss4 of the switch Q4, so that the voltage at the other end of the inductor L1 drops from Vin to Vo, the switch tube Q4 realizes ZVS turn-on; the voltage across the inductor L1 is Vo, the in
  • Stage t0 ⁇ t1 At t0, the switch Q3 is turned on, the voltage across the inductor L1 is Vin, and the inductor L1 is excited, the current IL of the inductor L1 rises, and the switch Q3 is turned off at t1;
  • Stage t1 ⁇ t2 After the switching tube Q3 is turned off, the current IL of the inductor L1 charges the output capacitance Coss3 of the switching tube Q3 and discharges the output capacitance Coss4 of the switching tube Q4. At t2, the voltage at the other end of the inductor L1 drops from Vin to Vo, and the switch Q4 realizes ZVS turn-on;
  • Stage t3 ⁇ t4 The current IL of the inductor L1 charges the output capacitor Coss2 of the switch Q2, the voltage at one end of the inductor L1 rises from 0V to Vin, and the voltage across the inductor L1 is clamped by Vin-Vo, when the current IL drops After reaching zero, the current IL reverses and the output capacitor Coss2 of the switching tube Q2 begins to discharge.
  • the voltage at one end of the inductor L1 drops from Vin to 0V, and the switching tube Q2 realizes ZVS opening;
  • Phase t5 ⁇ t0+Tx The current IL of the inductor L1 charges the output capacitor Coss4 of the switch Q4 and discharges the output capacitor Coss3 of the switch Q3. At t0+Tx, the voltage at the other end of the inductor L1 rises from Vo to Vin , The switch Q3 realizes the ZVS opening at t0+Tx;
  • the drain of the switching tube For MOS tube, it refers to the drain, for the triode, it is the collector, and for the IGBT, it is the drain.
  • Other switching tubes can correspond to the knowledge of those skilled in the art, not one by one. Enumerate
  • the source of the switching tube For MOS tube, it refers to the source, for triode, it is the emitter, and for IGBT, it is the source. Other switching tubes can correspond to each other according to the knowledge of those skilled in the art. Enumerate.
  • the present invention has the following beneficial effects:
  • FIG. 1 is a schematic diagram of the traditional Buck_Boost circuit
  • FIG. 2 is a schematic diagram of the CUK circuit
  • FIG. 3 is a schematic diagram of the circuit of the first embodiment of the present invention.
  • Figure 4 shows the relationship between the ratio of input and output voltage and the switching frequency
  • Figure 5 is a first working sequence diagram of the first embodiment of the present invention.
  • Figure 6 is a second working sequence diagram of the first embodiment of the present invention.
  • Fig. 7 is a schematic circuit diagram of the second embodiment of the present invention.
  • Fig. 8 is a working sequence diagram of the second embodiment of the present invention.
  • Fig. 3 is a schematic circuit diagram of the first embodiment of the present invention. Including input power supply positive Vin, output voltage negative Vo, power supply common ground GND, MOS tube Q1, MOS tube Q2, MOS tube Q3, MOS tube Q4, inductor L1 and capacitor C1; the drain of MOS tube Q1 and MOS tube Q3 The drain is connected to the input power positive Vin, the source of MOS transistor Q1 and the drain of MOS transistor Q2 are connected to one end of inductor L1, and the source of MOS transistor Q3 and the drain of MOS transistor Q4 are connected to the other of inductor L1. At one end, the source of the MOS transistor Q4 is connected to one end of the capacitor C1, and the source of the MOS transistor Q2 and the other end of the capacitor C1 are connected to the power supply common ground GND.
  • Coss1, Coss2, Coss3, and Coss4 in Figure 3 are the output capacitors of MOS transistors Q1, MOS transistors Q2, MOS transistors Q3 and MOS transistors Q4, respectively.
  • MOS transistors Q1, MOS transistors Q2, and MOS transistors are also shown in Figure 3. The body diode of the tube Q3 and MOS tube Q4.
  • MOS tube Q1, MOS tube Q2, MOS tube Q3, and MOS tube Q4 are common methods used by those skilled in the art.
  • Figure 4 shows the waveform of the inductor L1 current IL and the output current Io when the traditional Buck_Boost circuit works in discontinuous mode.
  • the rising slope of current IL is The decreasing slope of current IL is Therefore, the current IL rises and falls in the same time, and the corresponding duty cycle is T1.
  • the rising slope of current IL is The decreasing slope of current IL is Change the inductance of inductor L1 so that the rising slope of current IL and
  • the falling time of current IL is When the current IL falls time twice, the corresponding duty cycle is T2, which is greater than T1.
  • the rising slope of current IL is The decreasing slope of current IL is Change the inductance of inductor L1 so that the rising slope of current IL and
  • the current IL falling time is When the current IL falls 4 times, the duty cycle is T3, which is greater than T2.
  • Figure 5 shows the first working sequence of the first embodiment, which is specifically as follows:
  • Stage t0 ⁇ t1 At t0, the MOS transistor Q2 is turned on, the voltage across the inductor L1 is Vin, and the inductor L1 is excited, the current IL of the inductor L1 rises, and the MOS transistor Q3 is turned off at the time t1;
  • Phase t7 ⁇ t0+Tx The current IL of the inductor L1 charges the output capacitor Coss1 of the MOS transistor Q1, and the output capacitor Coss2 of the MOS transistor Q2 discharges.
  • the voltage at the circuit node SW1 ie one end of the inductor L1 Decrease Vin to 0V, MOS transistor Q2 realizes ZVS opening;
  • the first control method As an improvement of the above-mentioned first control method, it is characterized in that when the load becomes lighter, the period t0 to t1, the period t2 to t3 and the period t4 to t5 begin to decrease, and the period t6 to t7 becomes longer.
  • the reduced value or increased value of each stage is related to the input and output voltage value, the inductance of inductor L1, the setting of the optimal efficiency point, the switching frequency, etc., which is a trend described here.
  • Tx in the above t0+Tx represents a time length of X cycles.
  • FIG 6 shows the second working sequence of the first embodiment, which is specifically as follows:
  • Stage t0 ⁇ t1 At t0, the MOS transistor Q3 is turned on, the voltage across the inductor L1 is Vin, and the inductor L1 is excited, the current IL of the inductor L1 rises, and the MOS transistor Q3 is turned off at the time t1;
  • Stage t6 ⁇ t7 The voltage across the inductor L1 is Vo, and Vo reversely excites the inductor L1, and the MOS transistor Q4 is turned off at t7;
  • Stage t7 ⁇ t0+Tx The current of the inductor L1 discharges the output capacitor Coss3 of the MOS transistor Q3 and charges the output capacitor Coss4 of the MOS transistor Q4.
  • the circuit node SW2 that is, the other end of the inductor L1
  • the voltage rises from Vo to Vin, and the MOS transistor Q3 realizes ZVS opening;
  • the waveform of the current IL of the inductor L1 is also quadrilateral, which also achieves the purpose of the invention.
  • Fig. 7 is a schematic circuit diagram of the second embodiment of the present invention.
  • the MOS transistor Q1 is replaced by a diode D1
  • the cathode of the diode D1 is connected to the drain of the MOS transistor Q3 and the input power source is positive Vin
  • the anode of the diode D1 is connected to the drain and the inductor of the MOS transistor Q2 One end of L1.
  • the time for the diode D1 to flow through the current is relatively small. Compared with the MOS tube solution, the conduction loss will not increase too much, but it does not need a floating drive, which reduces the drive loss, simplifies the drive circuit, and is suitable for small and medium current output. Scenes.
  • the absolute value of the ratio of input to output voltage of this embodiment is greater than 2 to obtain better implementation effects.
  • the figure 8 shows the working sequence of the second embodiment, which is specifically as follows:
  • Stage t0 ⁇ t1 At t0, the MOS transistor Q3 is turned on, the voltage across the inductor L1 is Vin, and the inductor L1 is excited, the current IL of the inductor L1 rises, and the MOS transistor Q3 is turned off at the time t1;
  • Stage t3 ⁇ t4 The current IL of the inductor L1 charges the output capacitor Coss2 of the MOS transistor Q2, the voltage of the circuit node SW1 (ie the end of the inductor L1) rises from 0V to Vin, and the voltage across the inductor L1 is Vin-Vo Clamping. When the current IL drops to zero, the current IL reverses, and the output capacitor Coss2 of the MOS transistor Q2 begins to discharge. At t4, the voltage at the circuit node SW1 (ie the end of the inductor L1) drops from Vin to 0V, and the MOS transistor Q2 Achieve ZVS opening;
  • Stage t4 ⁇ t5 The voltage across the inductor L1 is Vo, and Vo reversely excites the inductor L1, and the MOS transistor Q4 is turned off at t5;
  • Phase t5 ⁇ t0+Tx The current IL of the inductor L1 charges the output capacitor Coss4 of the MOS transistor Q4 and discharges the output capacitor Coss3 of the MOS transistor Q3.
  • the circuit node SW2 ie the other end of the inductor L1
  • the voltage rises from Vo to Vin, and the MOS transistor Q3 realizes the ZVS turn-on at t0+Tx;
  • the waveform of the current IL of the inductor L1 is also quadrilateral, which also achieves the purpose of the invention.
  • switching converters with other parameters can also be selected, and the efficiency of the circuit can also be improved through the above-mentioned mode switching, which is not repeated here.

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Abstract

本发明公开了一种开关变换器及其控制方法，开关变换器包括输入电源正、输出电压负、电源公共地、开关管Q1、开关管Q2、开关管Q3、开关管Q4、电感器L1和电容器C1；开关管Q1的漏极和开关管Q3的漏极连接到输入电源正，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，开关管Q3的源极和开关管Q4的漏极连接到电感器L1的另一端，开关管Q4的源极连接到电容器C1的一端，开关管Q2的源极和电容器C1的另一端连接到电源公共地。本发明实现输入输出电压极性相反，所有开关管ZVS开通，效率高；在输入输出电压之比的绝对值较大时能实现电感器L1快速去磁，并将电感器L1的电流波形从三角形变为四边形，实现开关变换器高频高效工作。

Description

一种开关变换器及其控制方法 技术领域
本发明涉及开关电源，特别涉及开关变换器电路及其控制方法。
背景技术
图1为传统的Buck_Boost电路，电路工作在断续模式下电流有效值较大，MOS管Q1硬开关，D1的导通损耗大。
图2为CUK电路，是四阶甚至更高阶电路，动态过程复杂，输出容易超调，电路工作在断续模式下电流有效值较大，MOS管Q1硬开关，D1的导通损耗大。
传统的Buck_Boost电路和CUK电路都属于反极性电路，输入输出电压反相。但是都存在电路工作在断续模式下电流有效值较大，导通损耗偏大的问题；MOS管Q1硬开关，开关损耗较大，所以不适合高电压输入，高频化场景应用。在断续模式和输入输出电压之比的绝对值大于2时，电感器L1的去磁时间太长，且去磁时间与输出电流的大小成正比，存在大电流输出和高频化难以折中的问题。
发明内容
鉴于现有反极性电路的技术缺陷，本发明提出一种开关变换器电路及其控制方式，电路工作在断续模式下降低了电流有效值，解决了导通损耗偏大的问题；所有开关管实现ZVS开通，在断续模式和输入输出电压之比的绝对值大于2时，解决了电感器L1的去磁时间太长，难以大电流输出和高频化的问题。
为了实现上述发明目的，本发明采用以下技术方案：
一种开关变换器，包括输入电源正、输出电压负、电源公共地、开关管Q1、开关管Q2、开关管Q3、开关管Q4、电感器L1和电容器C1；开关管Q1的漏极和开关管Q3的漏极连接到输入电源正，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，开关管Q3的源极和开关管Q4的漏极连接到电感器L1的另一端，开关管Q4的源极连接到电容器C1的一端，开关管Q2的源极和电容器C1的另一端连接到电源公共地。
优选地，所述的开关管Q1、开关管Q2、开关管Q3和开关管Q4为MOS管、三极管或者IGBT。
上述开关变换器的第一种控制方法为：
在上一个周期结束时关断开关管Q1，由于电感器L1的电流为负，电感器L1的电流IL给开关管Q1的输出电容Coss1充电，开关管Q2的输出电容Coss2放电，电感器L1一端的电压由Vin降到0V，开关管Q2实现ZVS开通；由于开关管Q3处于导通状态，所以电感器L1两端的电压为Vin，Vin电压对电感器L1励磁，电感器L1的电流IL上升，根据闭环控制要求再关断开关管Q3；电感器L1的电流IL给开关管Q3的输出电容Coss3充电，给开关管Q4的输出电容Coss4放电，使电感器L1另一端的电压由Vin降为Vo，开关管Q4实现ZVS开通；电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，然后根据闭环控制要求再关断开关管Q2，电感器L1的电流IL给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，使电感器L1一端的电压从0V上升到Vin，开关管Q1实现ZVS开通；电感器L1两端的电压为Vin-Vo，Vin-Vo电压对电感器L1去磁，电感器L1的电流IL下降到负电流，然后关断开关管Q4，电感器L1的电流IL给开关管Q4的输出电容Coss4充电，给开关管Q3的输出电容Coss3放电，使电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；从而开关管Q1、开关管Q2、开关管Q3和开关管Q4在一个周期里均实现了ZVS开通。再根据闭环控制要求关断开关管Q1，下一个周期开启。
结合图5描述工作过程如下：
t0～t1阶段：在t0时刻开关管Q2导通，电感器L1两端的电压为Vin，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断开关管Q3；
t1～t2阶段：开关管Q3关断后，电感器L1的电流IL给开关管Q3的输出电容Coss3充电，给开关管Q4的输出电容Coss4放电。在t2时刻电感器L1另一端的电压由Vin降为Vo，开关管Q4实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断开关管Q2；
t3～t4阶段：电感器L1的电流IL给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，在t4时刻电感器L1一端的电压从0V上升到Vin，开关管Q1实现ZVS开通；
t4～t5阶段：电感器L1的电流IL存在一次换相，由正转负，在t5时刻关断开关管Q4；
t5～t6阶段：电感器L1的电流IL给开关管Q4的输出电容Coss4充电，给开关管Q3的输出电容Coss3放电，在t6时刻电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；
t6～t7阶段：电感器L1两端的电压均为Vin，电压差为零，所以电感器L1的电流IL保持不变，t7时刻关断开关管Q1；
t7～t0+Tx阶段：电感器L1的电流IL给开关管Q1的输出电容Coss1充电，开关管Q2的输出电容Coss2放电，在t0+Tx时刻电感器L1一端的电压由Vin降到0V，开关管Q2实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
作为上述第一种控制方法的改进，当负载变轻时，开关管Q2开通到开关管Q4关断阶段的时间开始减小，开关管Q3开通到开关管Q1关断阶段的时间变长。
上述开关变换器的第二种控制方法为：
在上一个周期结束时关断开关管Q4，由于电感器L1的电流为负，电感器L1的电流IL给开关管Q4的输出电容Coss4充电，开关管Q3的输出电容Coss3放电，电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；由于开关管Q2处于导通状态，所以电感器L1两端的电压为Vin，Vin电压对电感器L1励磁，电感器L1的电流IL上升，根据闭环控制要求再关断开关管Q3；电感器L1的电流IL给开关管Q3的输出电容Coss3充电，给开关管Q4的输出电容Coss4放电，使电感器L1另一端的电压从Vin降为Vo，开关管Q4实现ZVS开通；电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，然后根据闭环控制要求再关断开关管Q2，电感器L1的电流IL给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，使电感器L1一端的电压从0V上升到Vin，开关管Q1实现ZVS开通；电感器L1两端的电压为Vin-Vo，Vin-Vo电压对电感器L1去磁，电感器L1的电流IL下降到零时关断开关管Q1，开关管Q1的输出电容Coss1开始充电，开关管Q2的输出电容Coss2放电，电感器L1的电流IL从零下降为负向电流，电感器L1一端的电压由Vin下降到0V，开关管Q2实现ZVS开通；从而开关管Q1、开关管Q2、开关管Q3和开关管Q4在一个周期里均实现了ZVS开通。再根据闭环控制要求关断开关管Q4，下一个周期开启。
结合图6描述工作过程如下：
t0～t1阶段：在t0时刻开关管Q3导通，电感器L1两端的电压为Vin，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断开关管Q3；
t1～t2阶段：开关管Q3关断后，电感器L1的电流IL给开关管Q3的输出电容Coss3充电，给开关管Q4的输出电容Coss4放电。在t2时刻电感器L1另一端的电压由Vin降为Vo，开关管Q4实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断开关管Q2；
t3～t4阶段：电感器L1的电流IL给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，在t4时刻电感器L1一端的电压从0V上升到Vin，开关管Q1实现ZVS开通；
t4～t5阶段：电感器L1的电流IL在t5时刻下降到零，此时关断开关管Q1；
t5～t6阶段：开关管Q1的输出电容Coss1充电，开关管Q2的输出电容Coss2放电，电感器L1的电流IL从零下降为负向电流，在t6时刻电感器L1一端的电压由Vin下降到0V，开关管Q2实现ZVS开通；
t6～t7阶段：电感器L1两端的电压为Vo，Vo对电感器L1反向励磁，在t7时刻关断开关管Q4；
t7～t0+Tx阶段：电感器L1的电流给开关管Q3的输出电容Coss3放电，给开关管Q4的输出电容Coss4充电，在t0+Tx时刻电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
作为上述第二种控制方法的改进，其特征在于：当负载变轻时，开关管Q3开通到开关管Q1关断阶段的时间开始减小，开关管Q2开通到开关管Q4关断阶段的时间变长。
本发明还提供另外一种相同发明构思的开关变换器，技术方案如下：
一种开关变换器，包括输入电源正、输出电压负、电源公共地、二极管D1、开关管Q2、开关管Q3、开关管Q4、电感器L1和电容器C1；二极管D1的阴极和开关管Q3的漏极连接到输入电源正，二极管D1的阳极和开关管Q2的漏极连 接到电感器L1的一端，开关管Q3的源极和开关管Q4的漏极连接到电感器L1的另一端，开关管Q4的源极连接到电容器C1的一端，开关管Q2的源极和电容器C1的另一端连接到电源公共地。
优选地，所述的开关管Q2、开关管Q3和开关管Q4为MOS管、三极管或者IGBT。
上述开关变换器的控制方法为：
在上一个周期结束时关断开关管Q4，由于电感器L1的电流为负，电感器L1的电流IL给开关管Q4的输出电容Coss4充电，开关管Q3的输出电容Coss3放电，电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；由于开关管Q2处于导通状态，所以电感器L1两端的电压为Vin，Vin电压对电感器L1励磁，电感器L1的电流IL上升，根据闭环控制要求再关断开关管Q3；电感器L1的电流IL给开关管Q3的输出电容Coss3充电，给开关管Q4的输出电容Coss4放电，使电感器L1另一端的电压从Vin降为Vo，开关管Q4实现ZVS开通；电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，然后根据闭环控制要求再关断开关管Q2，电感器L1的电流IL给开关管Q2的输出电容Coss2充电，使电感器L1一端的电压从0V上升到Vin，电感器L1两端的电压被Vin-Vo钳位，当电流IL下降到零后，电流IL反向，开关管Q2的输出电容Coss2开始放电，电感器L1一端的电压从Vin下降到0V，开关管Q2实现ZVS开通；从而开关管Q1、开关管Q2、开关管Q3和开关管Q4在一个周期里均实现了ZVS开通。再根据闭环控制要求关断开关管Q4，下一个周期开启。
结合图8描述工作过程如下：
t0～t1阶段：在t0时刻开关管Q3导通，电感器L1两端的电压为Vin，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断开关管Q3；
t1～t2阶段：开关管Q3关断后，电感器L1的电流IL给开关管Q3的输出电容Coss3充电，给开关管Q4的输出电容Coss4放电。在t2时刻电感器L1另一端的电压由Vin降为Vo，开关管Q4实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断开关管Q2；
t3～t4阶段：电感器L1的电流IL给开关管Q2的输出电容Coss2充电， 电感器L1一端的电压从0V上升到Vin，电感器L1两端的电压被Vin-Vo钳位，当电流IL下降到零后，电流IL反向，开关管Q2的输出电容Coss2开始放电，在t4时刻电感器L1一端的电压从Vin下降到0V，开关管Q2实现ZVS开通；
t4～t5阶段：电感器L1两端的电压为Vo，Vo对电感器L1反向励磁，在t5时刻关断开关管Q4；
t5～t0+Tx阶段：电感器L1的电流IL给开关管Q4的输出电容Coss4充电，给开关管Q3的输出电容Coss3放电，在t0+Tx时刻电感器L1另一端的电压从Vo上升到Vin，开关管Q3在t0+Tx时刻实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
术语含义说明：
开关管的漏极：对于MOS管指的是漏极、对于三极管指的是集电极、对于IGBT指的是漏极，其它开关管依据本领域的技术人员的知识可以自行对应，不再一一列举；
开关管的源极：对于MOS管指的是源极、对于三极管指的是发射极、对于IGBT指的是源极，其它开关管依据本领域的技术人员的知识可以自行对应，不再一一列举。
与现有技术相比，本发明具有如下有益效果：
1)电路工作在断续模式，实现了所有MOS管的ZVS开通；
2)电感器电流波形从三角形变为四边形，在相同输出功率下电感器电流的有效值降低，导通损耗减小，效率提高，容易实现大电流输出；
3)输入输出电压之比的绝对值较大时，MOS管Q2的关断大大缩短了电感器L1的去磁时间，实现了高频化，而高频化使电感感值和电容容值降低，从而减小了电源尺寸，降低了成本。
附图说明
图1为传统的Buck_Boost电路原理图；
图2为CUK电路原理图；
图3为本发明第一实施例电路原理图；
图4为输入输出电压之比和开关频率的关系图；
图5为本发明第一实施例第一种工作时序图；
图6为本发明第一实施例第二种工作时序图；
图7为本发明第二实施例电路原理图；
图8为本发明第二实施例工作时序图。
具体实施方式
第一实施例
图3为本发明的第一实施例的电路原理图。包括输入电源正Vin、输出电压负Vo、电源公共地GND、MOS管Q1、MOS管Q2、MOS管Q3、MOS管Q4、电感器L1和电容器C1；MOS管Q1的漏极和MOS管Q3的漏极连接到输入电源正Vin，MOS管Q1的源极和MOS管Q2的漏极连接到电感器L1的一端，MOS管Q3的源极和MOS管Q4的漏极连接到电感器L1的另一端，MOS管Q4的源极连接到电容器C1的一端，MOS管Q2的源极和电容器C1的另一端连接到电源公共地GND。
图3中的Coss1、Coss2、Coss3和Coss4分别为MOS管Q1、MOS管Q2、MOS管Q3和MOS管Q4的输出电容，此外，图3中还画出了MOS管Q1、MOS管Q2、MOS管Q3和MOS管Q4的体二极管。
需要说明的是：将MOS管Q1、MOS管Q2、MOS管Q3和MOS管Q4替换为三极管和IGBT等其它类型的开关管为本领域技术人员的惯用手段。
图4为传统的Buck_Boost电路工作在断续模式时，电感器L1电流IL的波形和输出电流Io，
Figure PCTCN2019113678-appb-000001
时，根据公式
Figure PCTCN2019113678-appb-000002
电流IL的上升斜率为
Figure PCTCN2019113678-appb-000003
电流IL的下降斜率为
Figure PCTCN2019113678-appb-000004
所以电流IL上升和下降的时间相同，对应的工作周期为T1。
当
Figure PCTCN2019113678-appb-000005
时，电流IL的上升斜率为
Figure PCTCN2019113678-appb-000006
电流IL的下降斜率为
Figure PCTCN2019113678-appb-000007
改变电感器L1的感量，使电流IL的上升斜率和
Figure PCTCN2019113678-appb-000008
时的上升斜率相同，则电流IL的下降时间为
Figure PCTCN2019113678-appb-000009
时电流IL下降时间的2倍，对应的工作周期为T2，大于T1。
当
Figure PCTCN2019113678-appb-000010
时，电流IL的上升斜率为
Figure PCTCN2019113678-appb-000011
电流IL的下降斜率为
Figure PCTCN2019113678-appb-000012
改变电感器L1的感量，使电流IL的上升斜率和
Figure PCTCN2019113678-appb-000013
时的上升斜率相同，则电流IL下降时间为
Figure PCTCN2019113678-appb-000014
时电流IL下降时间的4倍，工 作周期为T3，大于T2。
因此在断续模式下，输入输出电压之比的绝对值越大，对应的开关周期越大，频率越小，越难以实现高频化，输入输出电压之比的绝对值大于2时，才能保证本发明获得较好的有益效果。
针对Vin电压为75V，Vo电压为负12V，电感器L1为1uH，输出电流为20A的开关变换器，图5所示为第一实施例第一种工作时序，具体如下：
t0～t1阶段：在t0时刻MOS管Q2导通，电感器L1两端的电压为Vin，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断MOS管Q3；
t1～t2阶段：MOS管Q3关断后，电感器L1的电流IL给MOS管Q3的输出电容Coss3充电，给MOS管Q4的输出电容Coss4放电。在t2时刻电路结点SW2(即电感器L1另一端)的电压由Vin降为Vo，MOS管Q4实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断MOS管Q2；
t3～t4阶段：电感器L1的电流IL给MOS管Q2的输出电容Coss2充电，给MOS管Q1的输出电容Coss1放电，在t4时刻电路结点SW1(即电感器L1一端)的电压从0V上升到Vin，MOS管Q1实现ZVS开通；
t4～t5阶段：电感器L1的电流IL存在一次换相，由正转负，在t5时刻关断MOS管Q4；
t5～t6阶段：电感器L1的电流IL给MOS管Q4的输出电容Coss4充电，给MOS管Q3的输出电容Coss3放电，在t6时刻电路结点SW2(即电感器L1另一端)的电压从Vo上升到Vin，MOS管Q3实现ZVS开通；
t6～t7阶段：电感器L1两端的电压均为Vin，电压差为零，所以电感器L1的电流IL保持不变，t7时刻关断MOS管Q1；
t7～t0+Tx阶段：电感器L1的电流IL给MOS管Q1的输出电容Coss1充电，MOS管Q2的输出电容Coss2放电，在t0+Tx时刻电路结点SW1(即电感器L1一端)的电压由Vin降到0V，MOS管Q2实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
作为上述第一种控制方法的改进，其特征在于：当负载变轻时，t0～t1阶段，t2～t3阶段和t4～t5阶段开始减小，t6～t7阶段变长。每个阶段减小的 值或增加的值与输入输出电压值，电感器L1的感量，最优效率点的设置，开关频率等均相关，这里说明的是一种趋势。
本周期结束，下一个工作周期开始，重复上面的阶段。
由于电路为周期性的工作，上述t0+Tx中的Tx代表的含义为X个周期的时间长度。
从图5中可以看出电感器L1的电流IL的波形为四边形，在相同输出功率下，较现有技术的三角形波形而言，电感电流的峰值降低，有效值降低，所以导通损耗减小，效率提高，由公式
Figure PCTCN2019113678-appb-000015
得L*di＝N*dB*Ae，电流峰值降低使di减小，在电感器感量L，圈数N和磁芯dB不变的条件下，电感器磁芯的有效截面积Ae减小，则磁芯尺寸变小；在相同输出纹波要求下di减小，则所需的滤波电容器容值减小，电容尺寸变小；MOS管Q2的关断大大缩短了电感器L1的去磁时间，实现了高频化，而高频化使电感感值和电容容值进一步降低；使电源尺寸减小，降低了成本。
图6所示为第一实施例第二种工作时序，具体如下：
t0～t1阶段：在t0时刻MOS管Q3导通，电感器L1两端的电压为Vin，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断MOS管Q3；
t1～t2阶段：MOS管Q3关断后，电感器L1的电流IL给MOS管Q3的输出电容Coss3充电，给MOS管Q4的输出电容Coss4放电。在t2时刻电路结点SW2(即电感器L1另一端)的电压由Vin降为Vo，MOS管Q4实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断MOS管Q2；
t3～t4阶段：电感器L1的电流IL给MOS管Q2的输出电容Coss2充电，给MOS管Q1的输出电容Coss1放电，在t4时刻电路结点SW1(即电感器L1一端)的电压从0V上升到Vin，MOS管Q1实现ZVS开通；
t4～t5阶段：电感器L1的电流IL在t5时刻下降到零，此时关断MOS管Q1；
t5～t6阶段：MOS管Q1的输出电容Coss1充电，MOS管Q2的输出电容Coss2放电，电感器L1的电流IL从零下降为负向电流，在t6时刻电路结点SW1(即电感器L1一端)的电压由Vin下降到0V，MOS管Q2实现ZVS开通；
t6～t7阶段：电感器L1两端的电压为Vo，Vo对电感器L1反向励磁，在t7时刻关断MOS管Q4；
t7～t0+Tx阶段：电感器L1的电流给MOS管Q3的输出电容Coss3放电，给MOS管Q4的输出电容Coss4充电，在t0+Tx时刻电路结点SW2(即电感器L1另一端)的电压从Vo上升到Vin，MOS管Q3实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
从图6中可以看出电感器L1的电流IL的波形也为四边形，同样实现发明目的。
需要说明的是，除了Vin电压为75V，Vo电压为负12V，电感器L1为1uH，输出电流为20A的开关变换器，选择其他参数的开关变换器也具有类同的工作时序图，电感器L1的电流IL的波形也为四边形，只是各时间点的幅值有所区别。
另外，上述两种工作时序都是针对负载为满载时的应用场景，在实际的应用场合中，经常有负载变轻的情况出现，这时可以通过模式切换来改善轻载时电路的效率，改善方法如下：
1.当负载变轻时(即输出电流减少到一定值时)，关闭Q1的驱动，减小驱动损耗，提高效率；
2.当负载进一步变轻时，(即输出电流进一步减小到一定值时)，t0～t1阶段，t2～t3阶段和t4～t5阶段减小较大，t6～t7阶段变长较多，使总的开关周期基本维持不变。但有效的电流IL时间减小使得电流IL的有效值偏大，导通损耗偏大，这时使Q2处于持续导通状态，进一步降低驱动损耗，电路变为传统的Buck_Boost电路，且MOS管Q4实现了同步整流功能，电感器L1的电流IL的波形从四边形加较长的t6～t7阶段变为普通的三角形，在同等输出电流下电流IL的有效值降低，效率提升。
第二实施例
图7为本发明的第二实施例的电路原理图。在第一实施例的基础上，将MOS管Q1换为二极管D1，二极管D1的阴极连接到MOS管Q3的漏极和输入电源正Vin，二极管D1的阳极连接到MOS管Q2的漏极和电感器L1的一端。
二极管D1流过电流的时间相对较小，和MOS管方案相比，导通损耗不会增 加太多，却省去了一路浮地驱动，降低了驱动损耗，简化了驱动电路，适合中小电流输出场景。
本实施例输入输出电压之比的绝对值大于2同样能获得较佳的实施效果，针对Vin电压为75V，Vo电压为负12V，电感器L1为1uH，输出电流为20A的开关变换器，图8所示为第二实施例工作时序，具体如下：
t0～t1阶段：在t0时刻MOS管Q3导通，电感器L1两端的电压为Vin，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断MOS管Q3；
t1～t2阶段：MOS管Q3关断后，电感器L1的电流IL给MOS管Q3的输出电容Coss3充电，给MOS管Q4的输出电容Coss4放电。在t2时刻电路结点SW2(即电感器L1另一端)的电压由Vin降为Vo，MOS管Q4实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断MOS管Q2；
t3～t4阶段：电感器L1的电流IL给MOS管Q2的输出电容Coss2充电，电路结点SW1(即电感器L1一端)的电压从0V上升到Vin，电感器L1两端的电压被Vin-Vo钳位，当电流IL下降到零时，电流IL反向，MOS管Q2的输出电容Coss2开始放电，在t4时刻电路结点SW1(即电感器L1一端)的电压从Vin下降到0V，MOS管Q2实现ZVS开通；
t4～t5阶段：电感器L1两端的电压为Vo，Vo对电感器L1反向励磁，在t5时刻关断MOS管Q4；
t5～t0+Tx阶段：电感器L1的电流IL给MOS管Q4的输出电容Coss4充电，给MOS管Q3的输出电容Coss3放电，在t0+Tx时刻电路结点SW2(即电感器L1另一端)的电压从Vo上升到Vin，MOS管Q3在t0+Tx时刻实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
从图8中可以看出电感器L1的电流IL的波形也为四边形，同样实现发明目的。
本实施例同样可以选择其他参数的开关变换器，也可以通过上述的模式切换来改善电路的效率，在此不赘述。
上述实施方式不应视为对本发明的限制，本发明的保护范围应当以权利要求所限定的范围为准。对于本技术领域的普通技术人员来说，在不脱离本发明的精 神和范围内，还可以做出若干等同替换、改进和润饰，如根据应用场合的不同，通过器件的简单串并联等手段对电路微调，这些改进和润饰也应视为本发明的保护范围。

Claims (8)

  1. 一种开关变换器的控制方法，用于控制一种开关变换器，所述的开关变换器包括输入电源正、输出电压负、电源公共地、开关管Q1、开关管Q2、开关管Q3、开关管Q4、电感器L1和电容器C1；开关管Q1的漏极和开关管Q3的漏极连接到输入电源正，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，开关管Q3的源极和开关管Q4的漏极连接到电感器L1的另一端，开关管Q4的源极连接到电容器C1的一端，开关管Q2的源极和电容器C1的另一端连接到电源公共地；
    其特征在于：在上一个周期结束时关断开关管Q1，由于电感器L1的电流为负，电感器L1的电流IL给开关管Q1的输出电容Coss1充电，给开关管Q2的输出电容Coss2放电，电感器L1一端的电压由Vin降到0V，开关管Q2实现ZVS开通；由于开关管Q3处于导通状态，所以电感器L1两端的电压为Vin，Vin电压对电感器L1励磁，电感器L1的电流IL上升，根据闭环控制要求再关断开关管Q3；电感器L1的电流IL给开关管Q3的输出电容Coss3充电，给开关管Q4的输出电容Coss4放电，使电感器L1另一端的电压由Vin降为Vo，开关管Q4实现ZVS开通；电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，然后根据闭环控制要求再关断开关管Q2，电感器L1的电流IL给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，使电感器L1一端的电压从0V上升到Vin，开关管Q1实现ZVS开通；电感器L1两端的电压为Vin-Vo，Vin-Vo电压对电感器L1去磁，电感器L1的电流IL下降到负电流，然后关断开关管Q4，电感器L1的电流IL给开关管Q4的输出电容Coss4充电，给开关管Q3的输出电容Coss3放电，使电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；从而开关管Q1、开关管Q2、开关管Q3和开关管Q4在一个周期里均实现了ZVS开通；再根据闭环控制要求关断开关管Q1，下一个周期开启。
  2. 根据权利要求1所述的开关变换器的控制方法，其特征在于：所述的开关管Q1、开关管Q2、开关管Q3和开关管Q4为MOS管、三极管或者IGBT。
  3. 根据权利要求1或2所述的开关变换器的控制方法，其特征在于：当负载变轻时，开关管Q2开通到开关管Q4关断阶段的时间开始减小，开关管Q3开通到开关管Q1关断阶段的时间变长。
  4. 一种开关变换器的控制方法，用于控制一种开关变换器，所述的开关变换器包括输入电源正、输出电压负、电源公共地、开关管Q1、开关管Q2、开关管Q3、开关管Q4、电感器L1和电容器C1；开关管Q1的漏极和开关管Q3的漏极连接到输入电源正，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，开关管Q3的源极和开关管Q4的漏极连接到电感器L1的另一端，开关管Q4的源极连接到电容器C1的一端，开关管Q2的源极和电容器C1的另一端连接到电源公共地；
    其特征在于：在上一个周期结束时关断开关管Q4，由于电感器L1的电流为负，电感器L1的电流IL给开关管Q4的输出电容Coss4充电，开关管Q3的输出电容Coss3放电，电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；由于开关管Q2处于导通状态，所以电感器L1两端的电压为Vin，Vin电压对电感器L1励磁，电感器L1的电流IL上升，根据闭环控制要求再关断开关管Q3；电感器L1的电流IL给开关管Q3的输出电容Coss3充电，给开关管Q4的输出电容Coss4放电，使电感器L1另一端的电压从Vin降为Vo，开关管Q4实现ZVS开通；电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，然后根据闭环控制要求再关断开关管Q2，电感器L1的电流IL给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，使电感器L1一端的电压从0V上升到Vin，开关管Q1实现ZVS开通；电感器L1两端的电压为Vin-Vo，Vin-Vo电压对电感器L1去磁，电感器L1的电流IL下降到零时关断开关管Q1，开关管Q1的输出电容Coss1开始充电，开关管Q2的输出电容Coss2放电，电感器L1的电流IL从零下降为负向电流，电感器L1一端的电压由Vin下降到0V，开关管Q2实现ZVS开通；从而开关管Q1、开关管Q2、开关管Q3和开关管Q4在一个周期里均实现了ZVS开通；再根据闭环控制要求关断开关管Q4，下一个周期开启。
  5. 根据权利要求4所述的开关变换器的控制方法，其特征在于：当负载变轻时，开关管Q3开通到开关管Q1关断阶段的时间开始减小，开关管Q2开通到开关管Q4关断阶段的时间变长。
  6. 一种开关变换器，其特征在于：包括输入电源正、输出电压负、电源公共地、二极管D1、开关管Q2、开关管Q3、开关管Q4、电感器L1和电容器C1；二极管D1的阴极和开关管Q3的漏极连接到输入电源正，二极管D1的阳极和开关管Q2的漏极连接到电感器L1的一端，开关管Q3的源极和开关管Q4的漏极连接到电感器L1的另一端，开关管Q4的源极连接到电容器C1的一端，开关管Q2的源极和电容器C1的另一端连接到电源公共地。
  7. 根据权利要求6所述的开关变换器，其特征在于：所述的开关管Q2、开关管Q3和开关管Q4为MOS管、三极管或者IGBT。
  8. 权利要求6或7所述的开关变换器的控制方法，其特征在于：在上一个周期结束时关断开关管Q4，由于电感器L1的电流为负，电感器L1的电流IL给开关管Q4的输出电容Coss4充电，开关管Q3的输出电容Coss3放电，电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；由于开关管Q2处于导通状态，所以电感器L1两端的电压为Vin，Vin电压对电感器L1励磁，电感器L1的电流IL上升，根据闭环控制要求再关断开关管Q3；电感器L1的电流IL给开关管Q3的输出电容Coss3充电，给开关管Q4的输出电容Coss4放电，使电感器L1另一端的电压从Vin降为Vo，开关管Q4实现ZVS开通；电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，然后根据闭环控制要求再关断开关管Q2，电感器L1的电流IL给开关管Q2的输出电容Coss2充电，使电感器L1一端的电压从0V上升到Vin，电感器L1两端的电压被Vin-Vo钳位，当电流IL下降到零后，电流IL反向，开关管Q2的输出电容Coss2开始放电，电感器L1一端的电压从Vin下降到0V，开关管Q2实现ZVS开通；从而开关管Q1、开关管Q2、开关管Q3和开关管Q4在一个周期里均实现了ZVS开通；再根据闭环控制要求关断开关管Q4，下一个周期开启。
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