WO2020227891A1 - 电机控制方法、控制器、存储介质及电机驱动系统 - Google Patents

电机控制方法、控制器、存储介质及电机驱动系统 Download PDF

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Publication number
WO2020227891A1
WO2020227891A1 PCT/CN2019/086653 CN2019086653W WO2020227891A1 WO 2020227891 A1 WO2020227891 A1 WO 2020227891A1 CN 2019086653 W CN2019086653 W CN 2019086653W WO 2020227891 A1 WO2020227891 A1 WO 2020227891A1
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Prior art keywords
current
motor
value
voltage vector
axis
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French (fr)
Inventor
孙天夫
谢刚
李慧云
梁嘉宁
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Shenzhen Institute of Advanced Technology of CAS
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Shenzhen Institute of Advanced Technology of CAS
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage

Definitions

  • the invention belongs to the field of power electronics, and in particular relates to a motor control method, a controller, a storage medium and a motor drive system.
  • Motors are widely used in aerospace, industrial automation, and electric vehicles, and the pros and cons of motor drive technology directly determine the overall reliability, stability, and efficiency of the motor drive system.
  • the currently widely used motor drive technology is mainly based on the Proportional Integral (PI) controller space vector control (Filed Oriented Control, FOC).
  • the basic control principle is shown in Figure 1.
  • the existing FOC technology mainly adopts Two PI current controllers adjust the d-axis and q-axis currents of the motor respectively, and generate corresponding voltage commands to act on the Space Vector Pulse Width Modulation (SVPWM) module to generate modulation waves to drive the inverter , So as to realize the control of the motor.
  • SVPWM Space Vector Pulse Width Modulation
  • the existing FOC control technology also needs to realize the decoupling of the d-axis and q-axis current through the current decoupling module.
  • the purpose of the present invention is to provide a motor control method, a controller, a storage medium, and a motor drive system, which aims to solve the problem that the performance of the motor and its drive system cannot be effectively improved due to the use of PI controllers in the prior art. problem.
  • the present invention provides a motor control method, including:
  • the next voltage vector currently output by the inverter is obtained, and it is decomposed into the next d-axis voltage value and the next q-axis voltage value .
  • the observed value of the current controlled quantity is: the current d-axis current value and the current q-axis current value transformed from the current measurement values of the current phases on the stator side of the motor, and the current controlled quantity predicted value Is: the current d-axis predicted current value and the current q-axis predicted current value,
  • the current controlled quantity observation value is: based on the current stator flux linkage amplitude and the current stator flux linkage angle value observed by the flux observer
  • the current controlled quantity predicted value is: current stator flux linkage amplitude prediction Value and the predicted value of the current stator flux linkage angle.
  • the method further includes:
  • the predictive control period is adjusted.
  • one of the candidate voltage vector groups corresponds to a stationary coordinate system sector, and the sector corresponds to two of the basic voltage vectors,
  • the loss function is constructed in the following manner: when the action time corresponding to each of the basic voltage vectors in the action time group is valid, use the candidate sector number corresponding to the candidate voltage vector group, and , The current sector number corresponding to the current voltage sum vector synthesized by the current d-axis voltage component and the current q-axis voltage component to determine the loss function value.
  • the method further includes:
  • the action time corresponding to each of the basic voltage vectors in the action time group is equalized Scale down.
  • the next voltage vector currently output by the inverter is obtained, and it is decomposed into the next d-axis voltage value and the next q Shaft voltage value, including:
  • next ⁇ -axis voltage value and the next ⁇ -axis voltage value From the next ⁇ -axis voltage value and the next ⁇ -axis voltage value, the next d-axis voltage value and the next q-axis voltage value currently output by the inverter in the rotating coordinate system are obtained.
  • the motor parameters are called from a data table or obtained through online parameter identification technology, and the motor parameters include one or a combination of the following parameters: d-axis inductance, q-axis inductance, permanent magnet flux, stator Resistance and number of motor pole pairs.
  • the present invention also provides a motor controller including a memory and a processor, and the processor implements the steps in the above method when the processor executes the program stored in the memory.
  • the present invention also provides a readable storage medium, the readable storage medium stores a program, and the program is executed by a processor to implement the steps in the above method.
  • the present invention also provides a motor drive system, including: an inverter, a space vector pulse width modulation module, and the motor controller according to claim 8, wherein the space vector pulse width modulation module combines the The next d-axis voltage value and the next q-axis voltage value are converted into state control commands of the inverter, so as to realize drive control of the motor.
  • the present invention constructs the predicted controlled variable value and the aforementioned candidate voltage vector when different candidate voltage vector groups act
  • the relationship model between the action time of the voltage vector in the group, the combination of the above model and the candidate voltage vector inversely deduces that after a predictive control cycle, if the predicted controlled value is the same as the control command value, each candidate
  • the voltage vector group corresponds to each action time group, and then the optimal candidate voltage vector group and the corresponding action time group are screened out, and then based on the principle of equivalent vector synthesis, the selected candidate voltage vector group is selected according to each voltage
  • the action time corresponding to the vector synthesizes the voltage vector command and decomposes it into the corresponding coordinate system to obtain the voltage command.
  • the space pulse width modulation technology is used to apply the above voltage command to the inverter and the motor to drive the motor to run.
  • no PI controller is needed to control the motor, avoid the current coupling, difficult parameter tuning, integral saturation, fast response and overshoot contradictions in the existing motor control technology, and effectively improve the performance of the motor and its drive system .
  • FIG. 1 is a schematic diagram of FOC in the prior art
  • FIG. 2 is an implementation flowchart of a motor control method provided by Embodiment 1 of the present invention
  • Fig. 3 is a schematic structural diagram of a motor controller provided by a fourth embodiment of the present invention.
  • FIG. 4 is a schematic diagram of the structure of a motor drive system provided by the sixth embodiment of the present invention.
  • Fig. 5 is a schematic diagram of a PI-free motor control strategy provided by a specific application example 1 of the present invention.
  • FIG. 6 is a schematic diagram of the circuit topology of the three-phase two-level inverter in specific application example 1 of the present invention.
  • FIG. 7 is a schematic diagram of six effective basic voltage vectors and two zero voltage vectors in specific application example 1 of the present invention.
  • FIG. 8 is a schematic diagram of synthesizing arbitrary voltage vectors through basic vectors in specific application example 1 of the present invention.
  • Fig. 9 is an experimental test result 1 of the motor drive technology invented in the corresponding experiment and simulation data of specific application example 1 of the present invention.
  • FIG. 10 is a schematic diagram of the three-phase current of the motor corresponding to the working condition shown in FIG. 9 in the corresponding experiment and simulation data of the specific application example 1 of the present invention.
  • FIG. 11 is a schematic diagram of computer simulation and comparison between the motor drive technology invented in the corresponding experiment and simulation data of specific application example 1 of the present invention and the existing space vector control technology based on PI controller;
  • FIG. 13 is a schematic diagram of a PI-free motor control strategy provided by the second specific application example of the present invention.
  • FIG. 14 is the control result of the flux linkage angle of the motor drive technology invented in the corresponding experiment and simulation data of the specific application example 2 of the present invention.
  • FIG. 15 is a schematic diagram of the motor flux amplitude control results corresponding to the operating conditions shown in FIG. 14 in the corresponding experiment and simulation data of the specific application example 2 of the present invention
  • FIG. 16 is a schematic diagram of the motor torque control results corresponding to the operating conditions shown in FIG. 14 in the corresponding experiment and simulation data of the specific application example 2 of the present invention
  • FIG. 2 shows the implementation process of the motor control method provided in the first embodiment of the present invention.
  • FIG. 2 shows the implementation process of the motor control method provided in the first embodiment of the present invention.
  • the parts related to the embodiment of the present invention are shown, which are detailed as follows:
  • step S201 obtain the observed values of several current controlled quantities on the stator side of the motor at the current moment, the current rotor electrical angular velocity of the motor, and the motor parameters under the motor operating conditions in the predicted control period.
  • the inverter controls the variable a, b, and c three-phase currents at the stator side of the motor through the state change of its switch tube, thereby controlling the operation of the motor.
  • the three-phase current can be collected by Hall sensors or resistors.
  • the three-phase current can be obtained through clark transformation and park transformation to obtain the corresponding direct-axis current in the rotating coordinate system, namely the d-axis current, and the quadrature-axis current, namely the q-axis current.
  • the motor flux observer can also be used to observe the stator side of the motor to obtain the corresponding stator flux linkage amplitude and stator flux angle value in the f-t coordinate system (or called f-m coordinate system).
  • the above-mentioned d and q-axis current values, or the stator flux linkage amplitude and angle values are the observed values of the above-mentioned current controlled variables.
  • the current observed value of the controlled quantity can also be other types of values in other coordinate systems.
  • the electrical angular velocity can be acquired through position sensor or non-position sensing technology.
  • the corresponding motor parameters are also needed, such as: d-axis inductance, q-axis inductance, permanent magnet flux linkage, stator resistance, and motor pole pairs. All or part of these motor parameters can be fixed in advance, or called from a data sheet, or obtained through online parameter identification technology.
  • a model of the relationship between the action time of the basic voltage vector and the zero voltage vector output by the inverter and the predicted controlled value that is, the prediction model.
  • a prediction model under the corresponding coordinate system ie, dq coordinate system or ft coordinate system, etc.
  • the predicted control period, the current observed value of the controlled variable, the motor parameters, and the current rotor electrical angular velocity are used as inputs to obtain each voltage The relationship between the vector group and its action time and the predicted controlled value.
  • step S202 the predicted control period, the observed value of the current controlled variable, the motor parameters, the current rotor electrical angular velocity, and a number of candidate voltage vector groups are input into a relational model to obtain each current corresponding to each candidate voltage vector group.
  • the candidate voltage vector group includes the basic voltage vector and zero voltage vector output by the inverter.
  • the current predicted controlled quantity group includes a number of current controlled quantity predicted values, and the current controlled quantity is predicted The value is equal to the current control command value, and each action time group corresponding to each candidate voltage vector group is obtained.
  • the action time group includes the action time of the basic voltage vector and the zero voltage vector.
  • the switching state of the switch tube of the inverter can correspond to a number of basic voltage vectors and zero voltage vectors.
  • the candidate voltage vector group can be composed of basic voltage vector and zero voltage vector. These basic voltage vectors and zero voltage vectors can be projected onto the space voltage vector distribution diagram in the stationary coordinate system (that is, the ⁇ - ⁇ coordinate system). There are several sectors on the space voltage vector distribution diagram, and the basic voltage vector corresponds to the sector. At the edge position of, the zero-voltage vector corresponds to the origin of the stationary coordinate system, then each sector corresponds to a zero-voltage vector and two basic voltage vectors, and a candidate voltage vector group corresponds to one sector.
  • Each candidate voltage vector group corresponds to an action time group, which includes the action time of the corresponding basic voltage vector and the action time of the zero voltage vector.
  • step S203 the pre-established loss function based on the action time group is used to select the action time group that causes the smallest loss function value and the corresponding candidate voltage vector group.
  • each candidate voltage vector group corresponding to each sector and each action time group corresponding to each candidate voltage vector group are screened, and the one that can achieve fast transient response and minimize the fluctuation of the controlled quantity is selected.
  • Alternative voltage vector group and action time group are selected.
  • the loss function can be judged based on whether the calculated action time is valid or not.
  • Each candidate voltage vector group and each action time group can correspond to a loss function value. According to the comparison of the loss function values, the minimum loss function value is obtained, and the minimum loss is determined The candidate voltage vector group and action time group corresponding to the function value.
  • step S204 from the selected action time group and the corresponding candidate voltage vector group, the next voltage vector currently output by the inverter is obtained, and it is decomposed into the next d-axis voltage value and the next q-axis voltage value.
  • the candidate voltage vector group and the action time group corresponding to the minimum loss function value can be used to obtain the ⁇ -axis voltage component and the voltage component in the stationary coordinate system.
  • the ⁇ -axis voltage component is further converted from the ⁇ -axis voltage component and the ⁇ -axis voltage component to obtain the d-axis voltage component and the q-axis voltage component.
  • the obtained d-axis voltage component and q-axis voltage component are output to the inverter for the next stage of motor control.
  • the predicted controlled value and the aforementioned candidate voltage are constructed.
  • the relationship model between the action time of the voltage vector in the vector group, the combination of the above model and the alternative voltage vector, inversely deduces that after a predictive control period, if the predicted controlled value is the same as the control command value, each standby Select each action time group corresponding to the voltage vector group, and then filter out the optimal candidate voltage vector group and the corresponding action time group, and then based on the principle of equivalent vector synthesis, the selected candidate voltage vector group is based on each of them
  • the action time corresponding to the voltage vector synthesizes the voltage vector command and decomposes it into the corresponding coordinate system to obtain the voltage command.
  • the space pulse width modulation technology is used to apply the above voltage command to the motor to drive the motor to run. In this way, no PI controller is needed to control the motor, avoid the current coupling, difficult parameter tuning, integral saturation, fast response and overshoot contradictions in the existing motor control technology, and effectively improve the performance of the motor and its drive system .
  • this embodiment further provides the following content:
  • the loss function involved in step S203 is constructed in the following manner:
  • the current voltage combined vector composed of the candidate sector number corresponding to the candidate voltage vector group, the current d-axis voltage component and the current q-axis voltage component The corresponding current sector number determines the loss function value.
  • step S203 after calculating the candidate voltage vector groups and the action time groups corresponding to each sector, if the action time corresponding to a certain basic voltage vector in the action time group corresponding to a certain sector is less than 0, the action time is considered Invalid, the candidate voltage vector group and action time group will not be selected, but only when the action time corresponding to all the basic voltage vectors in the action time group corresponding to the sector is greater than 0, the action time is valid. Only the voltage vector group and the action time group can be selected.
  • the loss function value can be calculated by the candidate sector number corresponding to the candidate voltage vector group and the above current sector number, and then proceed Comparison of loss function values.
  • this embodiment further provides the following content:
  • step S202 and before step S203 the following processing may be performed:
  • the action time corresponding to each basic voltage vector in the action time group is reduced proportionally.
  • step S202 when the action time corresponding to each basic voltage vector is greater than or equal to 0 and the sum value is greater than the predictive control period, the operability requirement cannot actually be met. At this time, the action The action time corresponding to each basic voltage vector in the time group is reduced proportionally, so that the sum of the action time corresponding to each basic voltage vector in the action time group is not greater than the predictive control period.
  • FIG. 3 shows the structure of the motor controller provided in the fourth embodiment of the present invention. For ease of description, only the parts related to the embodiment of the present invention are shown.
  • the motor controller of the embodiment of the present invention includes a processor 301 and a memory 302 (including the integration of the two, for example: some special chips have both storage and processing functions.
  • the control program is solidified into a logic circuit of hardware in these chips. These logic circuits themselves record the control program, and no additional memory is needed.
  • the processor 301 executes the program 303 stored in the memory 302, the steps in the foregoing method embodiments are implemented, such as steps S201 to S204 shown in FIG. .
  • the program is generally an embedded system program, of course, non-embedded system programs can also be used in some cases.
  • the inverter and the motor controller of the embodiment of the present invention may be a Micro Controller Unit (MCU), a Digital Signal Processor (Digital Signal Processing, DSP), a Field-Programmable Gate Array (Field-Programmable Gate Array, FPGA) ), programmable logic controllers (Programmable Logic Controller, PLC), chipsets, application specific integrated circuits (Application Specific Integrated Circuits, ASICs) and other devices with signal processing capabilities, or even separate computers or computer networking.
  • MCU Micro Controller Unit
  • DSP Digital Signal Processing
  • FPGA Field-Programmable Gate Array
  • PLC programmable logic controllers
  • chipsets application specific integrated circuits (Application Specific Integrated Circuits, ASICs) and other devices with signal processing capabilities, or even separate computers or computer networking.
  • the motor current controller in the embodiment of the present invention is not a PI controller.
  • the steps implemented when the processor 301 executes the program 303 in the motor controller to implement the above methods please refer to the description of the above method embodiments, which will not be repeated here.
  • a readable storage medium stores a program, and when the program is executed by a processor, the steps in the foregoing method embodiments are implemented, for example, the steps shown in FIG. 2 S201 to S204.
  • the readable storage medium in the embodiment of the present invention may include any entity or device or recording medium capable of carrying and storing program code, such as flash memory chips, field programmable gate arrays, ROM/RAM, magnetic disks, optical disks and other memories.
  • Embodiment 6 is a diagrammatic representation of Embodiment 6
  • Fig. 4 shows the structure of the motor drive system provided by the sixth embodiment of the present invention. For ease of description, only the parts related to the embodiment of the present invention are shown.
  • the motor drive system of the embodiment of the present invention includes the following devices that control the motor 401: an inverter 402, a space vector pulse width modulation module 403, and the motor controller 404 described in the fourth embodiment, wherein the space vector pulse width modulation
  • the module 403 converts the next d-axis voltage value and the next q-axis voltage value obtained by the above processing into the state control command of the inverter 402 to realize the drive control of the motor 401.
  • the space vector pulse width modulation module 403 and the motor controller 404 can be integrated in the same chip or embedded system.
  • the motor drive system may also include other electronic circuits, such as current acquisition circuits, rotor position (rotation speed) acquisition circuits, protection circuits, etc.
  • the motor can be any motor, such as: built-in permanent magnet synchronous motor, surface mount permanent magnet synchronous motor, switched reluctance motor, linear motor, magnetic field memory motor, rotor excitation motor, induction motor, etc.
  • the three-phase current output device is a two-stage three-phase inverter.
  • the power electronic devices in the inverter can be insulated gate bipolar transistors (IGBT), metal-oxide semiconductor field effect transistors (MOSFET), silicon carbide, etc. Electronic devices.
  • the motor drive control technology based on PI current controller has the contradiction between fast response and overshoot.
  • the motor drive control technology based on the PI current controller has the problem of AC-DC axis current coupling.
  • this example proposes a PI-free motor control strategy to realize the control and regulation of the motor current and avoid the current coupling existing in the existing motor drive technology , Parameter is difficult to set, integral saturation, and the contradiction between fast response and overshoot.
  • This example first derives the influence of the switching state of each leg of the inverter on the dq-axis current according to the formula, and builds a prediction model based on this, and then calculates based on the prediction model within the specified prediction time step (T s ), If the predicted d-axis and q-axis currents are equal to the d-axis and q-axis current command values, the action time of the basic voltage vector corresponding to the different switch state combinations of the inverter, and then combined with the current motor voltage vector command position, select one The optimal basic vector combination and its corresponding action time are used to calculate the synthesized new voltage vector command based on the selected optimal basic vector combination.
  • the synthesized motor vector command is applied to the motor to realize the drive control of the motor.
  • This technology can easily adjust the bandwidth of the current controller by changing the predictive control step length (that is, the above-mentioned T s ) without overshoot and overshoot, and at the same time avoids the problem of the traditional method of d-axis and q-axis current coupling.
  • the controlled current is used to control the motor torque, and then the motor speed is controlled by controlling the motor torque.
  • the d-axis inductance, q-axis inductance and permanent magnet flux linkage of the motor used in the proposed control technology can be obtained through table lookup or online parameter identification.
  • the online parameter identification strategies involved include but are not limited to: least square method, segmented affine projection, particle swarm and other parameter identification strategies.
  • the three bridge arms of the three-phase two-level inverter have a total of eight switching states, which correspond to the six effective basic voltage vectors and two zero voltage vectors as shown in Figure 7, and can be reversed
  • the clockwise rotation direction divides it into six sectors (sector I-sector VI).
  • S a (k), S b (k), S c (k) are the switching states of the upper arm of the inverter (the upper arm is turned on as 1, otherwise it is 0), and ⁇ r (k) is the motor Rotor position, V dc is the DC bus voltage, Is the d-axis voltage, Is the q-axis voltage, and k is the sampling time.
  • T s is the time step of the voltage vector action, They are the projections of a certain voltage vector on the dq coordinate axis in Fig. 7, L d , L q , and ⁇ m are the d-axis inductance, q-axis inductance and permanent magnet flux of the motor.
  • R or the following R s is the stator Resistance
  • p is the number of pole pairs of the motor
  • Is the mechanical angular velocity of the motor rotor, from p and The electrical angular velocity of the motor rotor can be obtained.
  • the template voltage vector U a can be synthesized. If the total action time of U x , U x+1 and the zero vector are respectively t 1 , t 2 and t 0 , then based on formulas (2) and (3), the predicted dq axis current With t 1 , t 2 and t 0 and the current dq axis current The relationship can be expressed as:
  • C d0 , C q0 , C d_x , C q_x , C d_x+1 , C q_x+1 are the changes of the d and q axis currents when the zero vector, voltage vector U x , and voltage vector U x+1 act, respectively.
  • u d_x , u q_x , u d_x+1 , u q_x+1 are the components of the first voltage vector (U x ) and the second voltage vector (U x+1 ) on the d and q axes, respectively.
  • the coordinate transformation is obtained by (1), where S a (k), S b (k), and S c (k) in (1) correspond to the corresponding U x and U x+1 .
  • (4)-(11) is the relationship model between the action time of the inverter output vector and the predicted d-axis and q-axis current values, that is, the prediction model.
  • the d and q axis current values at the k+N p time step can be predicted
  • the d and q axis current values at the k+N p time step are equal to the current command Have:
  • t 1 , t 2 are scaled proportionally according to (16) and (17) :
  • the voltage vector command can be synthesized based on its corresponding U x and U x+1 for controlling the motor. Will minimize the loss function value (J x ) Mark as Is based on And its corresponding U x , U x+1 , the synthesized voltage vector command can be calculated through (19)-(22) Projection on the d-axis and q-axis.
  • ⁇ x and ⁇ x+1 are the spatial phase angles between U x and U x+1 and the ⁇ axis, respectively, and ⁇ r is the electrical angle between the d axis and the ⁇ axis.
  • the inverter switching state control commands can be generated to realize the drive control of the motor.
  • the content shown in Figure 5 can realize the control of the motor d-axis and q-axis current, and then realize the control of the motor torque. And by controlling the motor torque, the speed of the motor can be further controlled. In addition, by adjusting the size of T s , the current control bandwidth can be easily adjusted.
  • control strategy of this example has the following advantages:
  • the existing PI controller-based motor control technology has the problem of difficulty in tuning the PI controller parameters.
  • the PI-free control technology proposed in this example has only one parameter that needs to be adjusted, that is, the predicted time step (T s ), and the motor current control bandwidth can be adjusted by adjusting the size of T s , and there is no overshoot and overshoot. . Therefore, the problem of difficulty in tuning the technical parameters of the existing motor control based on the PI controller is avoided.
  • Fig. 10 The three-phase current and q-axis current of the motor corresponding to the working conditions shown in Fig. 9 are shown in Fig. 10. It can be seen from Figure 10 that the proposed method can quickly and accurately control the motor current with small harmonics, no overshoot and overshoot.
  • the inventors respectively compared the invented motor drive technology with the existing PI based on the nonlinear motor system model.
  • the space vector control technology of the controller is simulated and compared by computer.
  • the response of the torque generated by the two control techniques to the step torque command is shown in Figure 11.
  • the torque response speed generated by the traditional motor control technology is slow, and there are obvious overshoot and overshoot.
  • the invented motor drive technology produces a fast torque response without overshoot and overshoot.
  • the invented motor control technology can control the response bandwidth by adjusting the predicted time step (T s ), the inventor performed computer simulations for 0.01 second, 0.001 second, and 0.0001 second based on the nonlinear motor system model.
  • the response effect of the motor control technology to the torque step command is shown in Figure 12.
  • the direct flux linkage vector (DFVC) control technology in the prior art mainly controls the motor flux linkage amplitude and the t-axis current through two PI controllers, and generates the f-axis and t-axis voltage commands, and generates the corresponding
  • the voltage commands of the d-axis and q-axis are input to the inverter to generate a control signal to act on the inverter, thereby realizing the control of the motor.
  • the disadvantages are mainly listed as follows:
  • the existing PI controller-based motor flux linkage amplitude and t-axis current control technology has the contradiction between fast response and overshoot.
  • this example proposes a PI-free direct flux vector control strategy for the motor to realize the control and regulation of the motor torque and flux linkage, avoiding the existing motor
  • PI parameters difficult to tune, integral saturation, fast response and overshoot in control technology.
  • This example first calculates the influence of the switching state of each bridge arm of the inverter on the motor stator flux linkage amplitude ( ⁇ s ) and the flux linkage angle, that is, the angle between the f-axis and the d-axis, ⁇ , according to the formula.
  • the corresponding basic voltage vector action time group combined with the current voltage vector command position of the motor, selects an optimal basic voltage vector combination and its corresponding action time group, and based on the selected optimal basic voltage vector combination and its
  • the corresponding action time group calculates the synthesized new voltage vector command, and uses the space pulse width modulation technology (SVPWM) to apply the synthesized motor vector command to the inverter to realize the control of the motor torque.
  • SVPWM space pulse width modulation technology
  • This technology can easily adjust the bandwidth of torque control by changing T s without overshoot and overshoot.
  • control the motor speed by controlling the motor torque.
  • the current motor stator flux linkage amplitude and the feedback value of the flux linkage angle used by the proposed control technology can be obtained by the flux linkage observer.
  • the three legs of the three-phase two-level inverter have a total of eight switching states, corresponding to the six effective basic voltage vectors and two zero voltage vectors as shown in Figure 7, and can be The counterclockwise rotation direction divides it into six sectors (sector I-sector VI).
  • S a (k), S b (k), and Sc (k) are the switching states of the upper arm of the inverter (the upper arm is turned on as 1, otherwise it is 0), and ⁇ f (k) is the motor
  • ⁇ f (k) is the motor
  • V dc is the DC bus voltage
  • k is the sampling time
  • t s is the time step of the voltage vector action, They are the t-axis current and the f-axis current, R is the stator resistance, p is the number of motor pole pairs, and ⁇ m is the mechanical angular velocity of the motor rotor. From p and ⁇ m, the electrical angular velocity of the motor rotor can be obtained.
  • current f t axis stator flux linkage amplitude And the flux linkage angle ⁇ k
  • the template voltage vector U a can be synthesized. If the total action time of U x , U x+1 and the zero vector are respectively t 1 , t 2 and t 0 , based on formulas (2) and (3), the stator flux linkage amplitude ⁇ s and the flux linkage angle ⁇
  • the relationship with t 1 , t 2 and t 0 can be expressed as:
  • ⁇ k+1 ⁇ k +C t0 t 0 +C t_x t 1 +C t_x+1 t 2 (5)
  • ⁇ k are the current stator flux amplitude and angle of the motor observed by the flux observer.
  • u f_x, u t_x, u f_x + 1, u t_x + 1 are a first voltage vector (U x) and a second voltage vector (U x + 1) in F, the t-axis component, according to The coordinate transformation is obtained by (1), where S a (k), S b (k), and S c (k) in (1) correspond to the corresponding U x and U x+1 .
  • t 1 , t 2 are scaled proportionally according to (16) and (17) :
  • the voltage vector command can be synthesized based on the corresponding U x and U x+1 to control the inverter to control the motor. Will minimize the loss function value (J x ) Mark as Is based on And its corresponding U x , U x+1 , the synthesized voltage vector command can be calculated through (19)-(22) Projection on the d-axis and q-axis.
  • ⁇ x and ⁇ x+1 are the spatial phase angles between U x and U x+1 and the ⁇ axis, respectively, and ⁇ e is the electrical angle of the motor rotor.
  • the inverter switching state control commands can be generated to realize the drive control of the inverter and the motor.
  • the content shown in Figure 13 can realize the control of the amplitude and angle of the motor stator flux linkage, thereby realizing the control of the motor torque.
  • the speed of the motor can be further controlled.
  • the current control bandwidth can be easily adjusted.
  • the traditional direct flux vector control strategy based on the ft coordinate system is designed based on the PI controller.
  • Problems such as contradictions have seriously affected the performance of the existing motor control technology.
  • the PI-free direct flux vector control technology of the motor proposed in this example solves the integrator saturation, direct-axis quadrature-axis current coupling, system constraints that are not easy to handle and the traditional motor control technology mentioned above.
  • problems such as the contradiction between fast response and overshoot. It has the advantages of no overshoot, fast current response, etc., and avoids the problem of AC-DC axis current coupling.
  • the existing PI controller-based motor control technology has the problem of difficulty in tuning the PI controller parameters.
  • the PI-free direct flux vector control technology proposed in this example has only one parameter that needs to be adjusted, that is, the predicted time step (T s ), and the motor current control bandwidth can be adjusted by adjusting the size of T s , and there is no Overshoot and overshoot. Therefore, the problem of difficulty in tuning the technical parameters of the existing motor control based on the PI controller is avoided.
  • the motor control method proposed in this example has only one parameter that needs to be tuned, it is easy to implement and has stronger adaptability to motors with different parameters. There is no need to re-tune the controller parameters according to the different motor parameters. At the same time, as the motor runs, the motor parameters will change, reducing the accuracy and accuracy of the motor control strategy.
  • the PI-free direct flux vector control technology of the motor proposed in this example does not participate in the feedback calculation because of the motor parameters. Therefore, the influence of motor parameter changes on the motor control strategy is avoided, and the realization and requirements of higher-precision control strategies are easier to achieve.

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Abstract

一种电机控制方法、控制器、存储介质及电机驱动系统，可基于预先构建的关系模型，得到不同备选电压矢量组作用时所预测的被控量的值，再由该关系模型及备选电压矢量组，反推出当一个预测控制周期后，若所预测的被控量的值与控制命令值相同时，各备选电压矢量组所对应的作用时间组，然后筛选出最优的备选电压矢量组及对应的作用时间组，再基于等效矢量合成原理，由最优备选电压矢量组及作用时间组合成电压矢量命令，并将其分解到相应坐标系，得到电压命令，进而可将该电压命令作用于电机或逆变器，驱动电机运转，从而无需PI控制器进行电机的控制，有效提高电机及其驱动系统性能。

Description

电机控制方法、控制器、存储介质及电机驱动系统 技术领域
本发明属于电力电子领域，尤其涉及一种电机控制方法、控制器、存储介质及电机驱动系统。
背景技术
电机被广泛应用于航空航天、工业自动化以及电动汽车等领域而电机驱动技术的优劣直接决定了电机驱动系统整体的可靠性、稳定性、效率等诸多方面。
目前被广泛采用的电机驱动技术主要是基于比例积分(Proportional Integral，PI)控制器的空间矢量控制(Filed Oriented Control，FOC)，其基本控制原理如图1所示，现有的FOC技术主要通过两个PI电流控制器分别调节电机的d-轴和q-轴电流，并产生相应的电压命令作用于空间矢量脉宽调制(Space Vector Pulse Width Modulation，SVPWM)模块生成调制波来驱动逆变器，从而实现对电机的控制。为了避免d-轴和q-轴电流耦合，现有FOC控制技术还需要通过电流解耦模块实现对d-轴和q-轴电流的解耦。
但是，由于上述FOC技术的电流控制一般基于PI控制器，因此存在积分饱和、d-q轴电流耦合、不易处理系统约束、动态响应速度慢、PI参数整定困难以及超调和过冲等问题，严重影响电机及其驱动系统性能的进一步提高。
发明内容
本发明的目的在于提供一种电机控制方法、控制器、存储介质及电机驱动系统，旨在解决现有技术所存在的、因采用PI控制器而存在的电机及其驱动系统性能无法有效提高的问题。
一方面，本发明提供了一种电机控制方法，包括：
获得当前时刻电机定子侧的若干当前被控量的观测值、所述电机的当前转子电角速度，以及预测控制周期内所述电机运行工况下的电机参数；
将预测控制周期、所述当前被控量的观测值、所述电机参数、所述当前转子电角速度，以及若干备选电压矢量组输入至一关系模型，得到各所述备选电压矢量组对应的各当前预测被控量值组，所述备选电压矢量组包括逆变器所输出的基本电压矢量及零电压矢量，所述当前预测被控量值组包括若干当前被控量预测值，并令所述当前被控量预测值等于当前控制命令值，得到与各所述备选电压矢量组对应的各作用时间组，所述作用时间组包括所述基本电压矢量及所述零电压矢量的作用时间；
利用预先建立的、以所述作用时间组为条件的损失函数，选择造成损失函数值最小的所述作用时间组及对应的所述备选电压矢量组；
由所选择的所述作用时间组及所述备选电压矢量组，得到所述逆变器当前输出的下一电压矢量，并将其分解为下一d轴电压值及下一q轴电压值。
进一步的，所述当前被控量观测值为：由所述电机定子侧的当前各相电流测量值变换得来的当前d轴电流值与当前q轴电流值，所述当前被控量预测值为：当前d轴预测电流值和当前q轴预测电流值，
或者，所述当前被控量观测值为：基于磁链观测器观测得到的当前定子磁链幅值与当前定子磁链角度值，所述当前被控量预测值为：当前定子磁链幅度预测值和当前定子磁链角度预测值。
进一步的，所述方法还包括：
对所述预测控制周期进行调节。
进一步的，一所述备选电压矢量组对应一静止坐标系扇区，所述扇区对应两个所述基本电压矢量，
所述损失函数通过如下方式构建：当所述作用时间组中各所述基本电压矢量对应的所述作用时间均有效时，以所述备选电压矢量组所对应的备选扇区编号，以及，当前d轴电压分量与当前q轴电压分量所合成的当前电压合矢量对 应的当前扇区编号，确定损失函数值。
进一步的，所述方法还包括：
当所述作用时间组中各所述基本电压矢量对应的所述作用时间之和大于所述预测控制周期时，对所述作用时间组中各所述基本电压矢量对应的所述作用时间进行等比例缩小。
进一步的，由所选择的所述作用时间组及所述备选电压矢量组，得到所述逆变器当前输出的下一电压矢量，并将其分解为下一d轴电压值及下一q轴电压值，具体包括：
由所选择的所述作用时间组及所述备选电压矢量组，得到静止坐标系下当前输出的下一α轴电压值及下一β轴电压值；
由所述下一α轴电压值及所述下一β轴电压值，得到旋转坐标系下所述逆变器当前输出的所述下一d轴电压值及所述下一q轴电压值。
进一步的，所述电机参数从数据表调用或通过在线参数辨识技术获得，所述电机参数包括如下参数中的一种或多种的组合：d轴电感、q轴电感、永磁体磁链、定子电阻及电机极对数。
另一方面，本发明还提供了一种电机控制器，包括存储器及处理器，所述处理器执行所述存储器中存储的程序时实现如上述方法中的步骤。
另一方面，本发明还提供了一种可读存储介质，所述可读存储介质存储有程序，所述程序被处理器执行时实现如上述方法中的步骤。
另一方面，本发明还提供了一种电机驱动系统，包括：逆变器、空间矢量脉宽调制模块以及如权利要求8所述的电机控制器，所述空间矢量脉宽调制模块将所述下一d轴电压值及所述下一q轴电压值转换成所述逆变器的状态控制命令，以实现对所述电机的驱动控制。
本发明根据拟设置的预测控制周期、当前被控量的观测值、电机参数以及转子电角速度，构建当不同备选电压矢量组作用时，所预测的被控量的值与上述备选电压矢量组中电压矢量的作用时间之间的关系模型，由上述模型及备选 电压矢量组合，反推出当一个预测控制周期后，若所预测的被控量值与控制命令值相同时，各备选电压矢量组所对应的各作用时间组，然后筛选出最优的备选电压矢量组及对应的作用时间组，再基于等效矢量合成原理，将所选择的备选电压矢量组根据其中各个电压矢量所对应的作用时间合成电压矢量命令，并将其分解到相应坐标系，得到电压命令，最后利用空间脉宽调制技术将上述电压命令作用于逆变器及电机，驱动电机运转。这样，无需PI控制器进行电机的控制，避免现有电机控制技术所存在的电流耦合、参数难以整定、积分饱和以及快速响应和超调过冲之间的矛盾，有效提高电机及其驱动系统性能。
附图说明
图1是现有技术的FOC原理图；
图2是本发明实施例一提供的电机控制方法的实现流程图；
图3是本发明实施例四提供的电机控制器的结构示意图；
图4是本发明实施例六提供的电机驱动系统的结构示意图；
图5是本发明具体应用例一提供的无PI电机控制策略示意图；
图6是本发明具体应用例一中三相两级逆变器的电路拓扑示意图；
图7是本发明具体应用例一中六个有效基本电压矢量和两个零电压矢量示意图；
图8是本发明具体应用例一中通过基本矢量合成任意电压矢量示意图；
图9是本发明具体应用例一对应实验及仿真数据中所发明的电机驱动技术的实验测试结果一；
图10是本发明具体应用例一对应实验及仿真数据中与图9所示工况对应的电机三相电流示意图；
图11是本发明具体应用例一对应实验及仿真数据中所发明的电机驱动技术与现有基于PI控制器的空间矢量控制技术进行的计算机仿真和对比示意图；
图12是本发明具体应用例一对应实验及仿真数据中T s＝0.01秒、0.001秒、 0.0001秒时电机驱动技术对转矩阶跃命令响应效果的计算机仿真对比图。
图13是本发明具体应用例二提供的无PI电机控制策略示意图；
图14是本发明具体应用例二对应实验及仿真数据中所发明的电机驱动技术的对磁链角度的控制结果；
图15是本发明具体应用例二对应实验及仿真数据中与图14所示工况对应的电机磁链幅值控制结果示意图；
图16是本发明具体应用例二对应实验及仿真数据中与图14所示工况相对应的电机转矩控制结果示意图；
图17是本发明具体应用例二对应实验及仿真数据中T s＝0.001秒、0.0001秒时电机驱动技术对转矩阶跃命令响应效果的计算机仿真对比图。
具体实施方式
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本发明进行进一步详细说明。应当理解，此处所描述的具体实施例仅仅用以解释本发明，并不用于限定本发明。
以下结合具体实施例对本发明的具体实现进行详细描述：
实施例一：
图2示出了本发明实施例一提供的电机控制方法的实现流程，为了便于说明，仅示出了与本发明实施例相关的部分，详述如下：
在步骤S201中，获得当前时刻电机定子侧的若干当前被控量的观测值、所述电机的当前转子电角速度，以及预测控制周期内电机运行工况下的电机参数。
[根据细则91更正 14.05.2019]　
本实施例中，逆变器通过其开关管的状态变化，对电机定子侧输出变化的a、b、c三相电流进行控制，从而控制电机工作。
三相电流可由霍尔传感器或电阻采集。三相电流可通过clark变换和park变换，得到相应的、旋转坐标系下的直轴电流即d轴电流，以及交轴电流即q 轴电流。另外，还可以通过电机磁链观测器，对电机定子侧进行观测，得到相应的、f-t坐标系下(或称为f-m坐标系)的定子磁链幅值与定子磁链角度值。上述d、q轴电流值，或定子磁链幅值、角度值即为上述当前被控量观测值。当前被控量观测值还可以是在其他坐标系下的其他类型的数值。
电角速度可通过位置传感器或无位置传感技术采集得到。
在后续关系模型中，还需要借助相应的电机参数，例如：d轴电感、q轴电感、永磁体磁链、定子电阻及电机极对数等。这些电机参数中的全部或部分都可以预先固定设置，或者从数据表调用，或者通过在线参数辨识技术获得。
需预先或实时构建一逆变器所输出的基本电压矢量及零电压矢量的作用时间与所预测的被控量值的关系模型，即预测模型。本实施例中，建立相应坐标系(即d-q坐标系或f-t坐标系等)下的预测模型，以预测控制周期、当前被控量观测值、电机参数及当前转子电角速度作为输入，得到各电压矢量组及其作用时间与所预测的被控量值的关系。
在步骤S202中，将预测控制周期、当前被控量的观测值、电机参数、当前转子电角速度，以及若干备选电压矢量组输入至一关系模型，得到各备选电压矢量组对应的各当前预测被控量值组，备选电压矢量组包括逆变器所输出的基本电压矢量及零电压矢量，当前预测被控量值组包括若干当前被控量预测值，并令当前被控量预测值等于当前控制命令值，得到与各备选电压矢量组对应的各作用时间组，作用时间组包括基本电压矢量及零电压矢量的作用时间。
本实施例中，对于逆变器而言，逆变器开关管的开关状态可对应若干数量的基本电压矢量及零电压矢量。备选电压矢量组可由基本电压矢量及零电压矢量组成。这些基本电压矢量及零电压矢量可投射到静止坐标系(即α-β坐标系)下的空间电压矢量分布图上，空间电压矢量分布图上分布有若干扇区，基本电压矢量对应到扇区的边位置上，零电压矢量对应到静止坐标系的原点，那么，每个扇区对应有零电压矢量及两个基本电压矢量，而一个备选电压矢量组对应一个扇区。
每一备选电压矢量组对应一个作用时间组，该作用时间组包括相应的基本电压矢量的作用时间及零电压矢量的作用时间。
在步骤S203中，利用预先建立的、以作用时间组为条件的损失函数，选择造成损失函数值最小的作用时间组及对应的备选电压矢量组。
本实施例中，对各扇区所对应的各备选电压矢量组以及与各备选电压矢量组对应的各作用时间组进行筛选，从中选择可实现快速瞬态响应且被控量波动最小的备选电压矢量组及作用时间组。
损失函数可基于计算得到的作用时间是否有效进行判断输出，每备选电压矢量组及每作用时间组可对应一损失函数值，根据各损失函数值比较，得到最小损失函数值，并确定最小损失函数值所对应的备选电压矢量组及作用时间组。
在步骤S204中，由所选择的作用时间组及对应的备选电压矢量组，得到逆变器当前输出的下一电压矢量，并将其分解为下一d轴电压值及下一q轴电压值。
本实施例中，最小损失函数值所对应的备选电压矢量组及作用时间组被确定后，则可通过该备选电压矢量组及作用时间组，得到静止坐标系下的α轴电压分量及β轴电压分量，进而由α轴电压分量及β轴电压分量转换得到d轴电压分量及q轴电压分量。所得到的d轴电压分量及q轴电压分量则被输出到逆变器，进行下一阶段电机的控制。
实施本实施例，根据拟设置的预测控制周期、当前被控量观测值、电机参数以及转子电角速度，构建当不同备选电压矢量组作用时，所预测的被控量值与上述备选电压矢量组中电压矢量的作用时间之间的关系模型，由上述模型及备选电压矢量组合，反推出当一个预测控制周期后，若所预测的被控量值与控制命令值相同时，各备选电压矢量组所对应的各作用时间组，然后筛选出最优的备选电压矢量组及对应的作用时间组，再基于等效矢量合成原理，将所选择的备选电压矢量组根据其中各个电压矢量所对应的作用时间合成电压矢量命令，并将其分解到相应坐标系，得到电压命令，最后利用空间脉宽调制技术将上述 电压命令作用于电机，驱动电机运转。这样，无需PI控制器进行电机的控制，避免现有电机控制技术所存在的电流耦合、参数难以整定、积分饱和以及快速响应和超调过冲之间的矛盾，有效提高电机及其驱动系统性能。
实施例二：
本实施例在实施例一基础上，进一步提供了如下内容：
本实施例中，步骤S203所涉及的损失函数通过如下方式构建：
当作用时间组中各基本电压矢量对应的作用时间均有效时，以备选电压矢量组所对应的备选扇区编号、当前d轴电压分量与当前q轴电压分量所组成的当前电压合矢量所对应的当前扇区编号确定损失函数值。
步骤S203计算获得各扇区对应的各备选电压矢量组及作用时间组后，如果其中某扇区对应的作用时间组中某一基本电压矢量对应的作用时间小于0时，则认为该作用时间无效，该备选电压矢量组及作用时间组不会被选择，而只有当扇区对应的作用时间组中所有基本电压矢量对应的作用时间均大于0时，该作用时间才有效，该备选电压矢量组及作用时间组才可能被选择。
当存在多组作用时间均有效的备选电压矢量组及作用时间组时，即可通过备选电压矢量组对应的备选扇区编号以及上述当前扇区编号，来计算损失函数值，进而进行损失函数值的比较。
当然，在其他实施例中，还可以构建其他形式的、符合快速瞬态响应要求和限制条件的损失函数。
实施例三：
本实施例在实施例一或二基础上，进一步提供了如下内容：
本实施例中，在步骤S202之后，S203之前，可进行如下处理：
当作用时间组中各基本电压矢量对应的作用时间之和大于预测控制周期时，对作用时间组中各基本电压矢量对应的作用时间进行等比例缩小。
步骤S202计算所得的作用时间组中，每基本电压矢量所对应的作用时间均大于或等于0且和值大于预测控制周期时，实际上还是无法满足可操作性要求， 此时，则需要将作用时间组中各基本电压矢量所对应的作用时间进行等比例缩小，从而使得作用时间组中各基本电压矢量对应的作用时间之和不大于预测控制周期。
实施例四：
图3示出了本发明实施例四提供的电机控制器的结构，为了便于说明，仅示出了与本发明实施例相关的部分。
本发明实施例的电机控制器包括处理器301及存储器302(包含两者集成的情况，例如：有些特制芯片，即有存储功能又有处理功能，这些芯片内把控制程序固化成硬件的逻辑电路，这些逻辑电路本身记录了控制程序，而不需要额外的存储器)，处理器301执行存储器302中存储的程序303时实现上述各个方法实施例中的步骤，例如图2所示的步骤S201至S204。
程序一般为嵌入式系统程序，当然某些情况下也可以采用非嵌入式系统程序。
本发明实施例的逆变器及电机控制器可以为微控制单元(Micro Controller Unit，MCU)、数字信号处理器(Digital Signal Processing，DSP)、现场可编程门阵列(Field-Programmable Gate Array，FPGA)、可编程逻辑控制器(Programmable Logic Controller，PLC)、芯片组、专用集成电路(Application Specific Integrated Circuit，ASIC)等具有信号处理能力的设备，甚至还可以是单独的计算机或计算机组网等。
本发明实施例的电机电流控制器不是PI控制器，该电机控制器中处理器301执行程序303时实现上述各方法时实现的步骤，可参考前述方法实施例的描述，在此不再赘述。
实施例五：
在本发明实施例中，提供了一种可读存储介质，该可读存储介质存储有程序，该程序被处理器执行时实现上述各方法实施例中的步骤，例如，图2所示的步骤S201至S204。
本发明实施例的可读存储介质可以包括能够携带和储存程序代码的任何实体或装置、记录介质，例如，闪存芯片、现场可编程门阵列、ROM/RAM、磁盘、光盘等存储器。
实施例六：
图4示出了本发明实施例六提供的电机驱动系统的结构，为了便于说明，仅示出了与本发明实施例相关的部分。
本发明实施例的电机驱动系统包括对电机401执行控制的如下器件：逆变器402、空间矢量脉宽调制模块403以及如实施例四所述的电机控制器404，其中，空间矢量脉宽调制模块403将上述处理所得下一d轴电压值及下一q轴电压值转换成逆变器402的状态控制命令，以实现对电机401的驱动控制。
空间矢量脉宽调制模块403可以与电机控制器404集成于同一芯片或嵌入式系统中。
电机驱动系统中还可以包含其他电子电路，例如：电流采集电路、转子位置(转速)采集电路、保护电路等。
电机驱动系统中，电机可以为任何电机，例如：内置式永磁同步电机、表贴式永磁同步电机、开关磁阻电机、直线电机、磁场记忆电机、转子励磁电机、感应电机等。三相电流输出器件为两级三相逆变器，逆变器中的电力电子器件可以为绝缘栅双极型晶体管(IGBT)、金属-氧化物半导体场效应晶体管(MOSFET)、碳化硅等电力电子器件。
具体应用例一：
下面通过具体应用例一，对上述各实施例的内容进行示例性说明。
现有技术中基于PI电流控制器的电机驱动技术的缺点主要列举如下：
1、基于PI电流控制器的电机驱动控制技术存在着快速响应和超调过冲之间的矛盾。
2、基于PI电流控制器的电机驱动控制技术存在交直轴电流耦合问题。
3、由于电机参数随运行工况变化，现有基于PI电流控制器的电机驱动控 制技术的PI控制器参数整定困难。
4、传统PI电流控制不易处理系统约束，存在积分饱和的现象。
针对上述现有基于PI电流控制器的电机驱动技术所存在的缺点，本例提出一种无PI电机控制策略，来实现对电机电流的控制和调节，避免现有电机驱动技术所存在的电流耦合、参数难以整定、积分饱和以及快速响应和超调过冲之间的矛盾。
本例首先根据公式推导出逆变器各个桥臂的开关状态对d-q轴电流的影响，并以此构建预测模型，然后基于预测模型计算出在所指定的预测时间步长(T s)内，若所预测的d轴及q轴电流等于d轴及q轴电流命令值时，逆变器不同开关状态组合所对应的基本电压矢量的作用时间，接着结合电机当前电压矢量命令的位置，选择一个最优的基本矢量组合以及其对应的作用时间，并基于所选择的最优基本矢量组合计算出所合成的新的电压矢量命令。通过空间脉宽调制技术(SVPWM)将所合成的电机矢量命令作用于电机，实现对电机驱动控制。本技术可以通过改变预测控制步长(即上述T s)轻易调节电流控制器的带宽，且无超调和过冲，同时也避免了传统方法d-轴和q-轴电流耦合的问题。最后利用所控制的电流实现对电机转矩的控制，再通过控制电机转矩实现对电机转速的控制。所提出的控制技术用到的电机的d轴电感、q轴电感以及永磁体磁链可通过查表或在线参数辨识的方法获得。涉及的在线参数辨识策略包括但不限于：最小二乘法、分段仿射投影以及粒子群等参数辨识策略等。
本例提出了一种无PI永磁同步电机控制策略，其控制原理框图如图5所示，具体细节如下：
如图6所示，三相两级逆变器的三个桥臂一共有八个开关状态，对应于如图7所示的六个有效基本电压矢量和两个零电压矢量，并可依逆时针旋转方向将其划分为六个扇区(扇区Ⅰ-扇区Ⅵ)。
根据坐标变换，图6中的开关状态与d、q轴电压的关系为：
Figure PCTCN2019086653-appb-000001
其中，S a(k)、S b(k)、S c(k)分别为逆变器上桥臂开关状态(上桥臂导通为1，否则为0)，θ r(k)为电机转子位置，V dc为直流母线电压，
Figure PCTCN2019086653-appb-000002
为d轴电压，
Figure PCTCN2019086653-appb-000003
为q轴电压，k为采样时刻。
永磁同步电机在d-q轴下的数学模型为：
Figure PCTCN2019086653-appb-000004
Figure PCTCN2019086653-appb-000005
其中，T s为电压矢量作用的时间步长，
Figure PCTCN2019086653-appb-000006
分别为图7中某一个电压矢量在d-q坐标轴上的投影，L d、L q、Ψ m分别为电机的d轴电感、q轴电感以及永磁体磁链，R或下述R s为定子电阻，p为电机极对数，
Figure PCTCN2019086653-appb-000007
为电机转子机械角速度，由p及
Figure PCTCN2019086653-appb-000008
可得到电机转子电角速度。
基于当前d、q轴电流
Figure PCTCN2019086653-appb-000009
即可利用(2)、(3)预测下一时间步长的d、q轴电流
Figure PCTCN2019086653-appb-000010
通过反复迭代(2)、(3)则可实现对未来N p个时间步长后d、q轴电流的预测。
如图8所示，根据空间矢量脉宽调制(SVPWM)的原理，通过快速变换图8中两个相邻的基本电压矢量(U x、U x+1)以及零矢量并调整它们的作用时间(t 1，t 2，t 0)，可以合成在电压限值以内的任意电压矢量U a，其中x为两个电压矢量所在图7中所示扇区的编号(当x＝6时，x+1被赋值为1)。
图8中的U x、U x+1是两个属于编号为x的扇区的相邻的基本电压矢量，其中x＝1,2,…,6。通过在U x、U x+1以及零矢量间切换，可以合成模板电压矢量U a。若令U x、U x+1以及零矢量的作用总时间分别为t 1、t 2和t 0，则基于公式(2)、 (3)，所预测的d-q轴电流
Figure PCTCN2019086653-appb-000011
与t 1、t 2和t 0及当前d-q轴电流
Figure PCTCN2019086653-appb-000012
的关系可表示为：
Figure PCTCN2019086653-appb-000013
Figure PCTCN2019086653-appb-000014
其中，C d0、C q0、C d_x、C q_x、C d_x+1、C q_x+1分别为当零矢量、电压矢量U x、电压矢量U x+1作用时，d、q轴电流的变化率，这些变化率可根据(2)、(3)由下式给出：
Figure PCTCN2019086653-appb-000015
Figure PCTCN2019086653-appb-000016
Figure PCTCN2019086653-appb-000017
Figure PCTCN2019086653-appb-000018
Figure PCTCN2019086653-appb-000019
Figure PCTCN2019086653-appb-000020
其中，u d_x、u q_x、u d_x+1、u q_x+1分别为第一个电压矢量(U x)和第二个电压矢量(U x+1)在d、q轴的分量，可根据坐标变换由(1)求得，其中，(1)中S a(k)、S b(k)、S c(k)对应相应的U x和U x+1。(4)-(11)即为逆变器输出矢量的作用时间与所预测的d轴及q轴电流值的关系模型，即预测模型。
通过反复迭代(4)、(5)可预测第k+N p时间步长时刻的d、q轴电流值
Figure PCTCN2019086653-appb-000021
Figure PCTCN2019086653-appb-000022
为了消除电流控制的稳态误差，令第k+N p时间步长时刻的d、q轴电流值等于电流命令
Figure PCTCN2019086653-appb-000023
有：
Figure PCTCN2019086653-appb-000024
为方便描述，本例说明中只选取预测步长为1的情况进行说明，即N p＝1。N p等于其他值的情况也类似推导。联立(4)-(12)可求得三个矢量(两个基本电压矢量及零电压矢量)的作用时间为：
Figure PCTCN2019086653-appb-000025
Figure PCTCN2019086653-appb-000026
其中，
Q＝C q0C d_x+1+C q_xC d0+C q_x+1C d_x-C q_xC d_x+1-C q_x+1C d0-C q0k d_x
                                 (15)
若根据所选择的U x、U x+1基本电压矢量组合计算出的t 1、t 2之和大于T s，则根据(16)、(17)对t 1、t 2进行等比例放缩：
Figure PCTCN2019086653-appb-000027
Figure PCTCN2019086653-appb-000028
依次将不同扇区的U x、U x+1代入(13)-(15)，可得到若干组(t 1,t 2,t 0)，其中t 0＝T s-t 1-t 2。为方便表述，将它们写作
Figure PCTCN2019086653-appb-000029
其中x为1,2,…,6或与当前电压矢量所在扇区相邻的两个扇区的编号。然后将上述
Figure PCTCN2019086653-appb-000030
分别代入损失函数(18)，其中x为
Figure PCTCN2019086653-appb-000031
所对应的电压矢量扇区编号，x present为当前电压矢量命令所对应的扇区编号。M为一个很大的值，以保证当
Figure PCTCN2019086653-appb-000032
或
Figure PCTCN2019086653-appb-000033
小于0时，J x会大于任意在
Figure PCTCN2019086653-appb-000034
且
Figure PCTCN2019086653-appb-000035
大于等于0时所对应的损失函数值。
Figure PCTCN2019086653-appb-000036
通过选择使得损失函数(J x)最小的
Figure PCTCN2019086653-appb-000037
可以基于其所对应的U x、U x+1合成电压矢量命令，用于控制电机。将使得损失函数值(J x)最小的
Figure PCTCN2019086653-appb-000038
标记为
Figure PCTCN2019086653-appb-000039
则基于
Figure PCTCN2019086653-appb-000040
以及其所对应的U x、U x+1，可以通过(19)-(22)计算出所合成的电压矢量命令
Figure PCTCN2019086653-appb-000041
在d-轴和q-轴上的投影。
Figure PCTCN2019086653-appb-000042
Figure PCTCN2019086653-appb-000043
Figure PCTCN2019086653-appb-000044
Figure PCTCN2019086653-appb-000045
其中，θ x、θ x+1分别为U x、U x+1与α轴之间的空间相位角，θ r为d轴与α轴之间的电角度。
然后将所计算的
Figure PCTCN2019086653-appb-000046
分别作为d-轴和q-轴电压命令输送给图5中的SVPWM模块，即可产生逆变器开关状态控制命令实现对电机的驱动控制。图5所示的内容可以实现对电机d-轴和q-轴电流的控制，进而实现对电机转矩的控制。而通过控制电机转矩可以进一步实现对电机转速的控制。此外，通过调节T s的大小，可以轻易实现对电流控制带宽的调节。
本例的控制策略相比现有被广泛采用的空间矢量控制策略而言具有以下优点：
1、传统控制策略由于基于PI控制器进行设计，存在着动态响应速度较低、积分器饱和、直轴交轴电流耦合、系统约束不易处理以及存在快速响应与超调和过冲之间的矛盾等问题，严重影响了现有电机控制技术的性能。针对现有电机控制技术的以上不足，本例所提出的无PI永磁同步电机控制技术解决了上述传统电机控制技术所面临的积分器饱和、直轴交轴电流耦合、系统约束不易处理以及存在快速响应与超调之间的矛盾等问题，具有无超调、电流响应速度快等优点，并避免了交直轴电流耦合的问题。
2、由于电机参数会随运行工况改变，因此现有基于PI控制器的电机控制 技术存在PI控制器参数整定困难的问题。而本例所提出的无PI控制技术只有一个需要调节的参数，即预测时间步长(T s)，并可以通过调节T s的大小对电机电流控制带宽进行调节，且不存在超调和过冲。因此避免了现有基于PI控制器的电机控制技术参数难以整定的问题。
3、由于本例所提出的电机控制方法只有一个需要整定的参数，因此实现起来较为容易，并对不同参数的电机具有更强的适应性，不需要根据电机参数的不同而重新整定控制器参数。
具体应用例一对应的实验及仿真数据：
实验及仿真数据一：
为了论证本专利提出的控制策略的有效性以及正确性，发明人基于一台永磁同步伺服电机驱动系统对本发明所提出的控制技术进行了实验验证。所测量的电机转矩、d-轴电流、q-轴电流以及其命令值如图9所示。可以发现：所提出的控制方法可以准确和快速地控制电机电流。图9中所测量的转矩相对于q轴电流约有0.5秒的时延，这是由于电机转矩传感器的采样时间为0.5秒，因此所测量的电机转矩照比电机实际产生的转矩约有0.5秒的时延。
与图9所示工况相对应的电机三相电流及q轴电流如图10所示。由图10可以看出，所提出的方法可以快速准确地控制电机电流且谐波较小、没有超调和过冲。
实验及仿真数据二：
为了验证所发明的电机驱动控制方法照比现有方法具有更快的电流响应速率且无过冲和超调，发明人基于非线性电机系统模型分别对所发明的电机驱动技术与现有基于PI控制器的空间矢量控制技术进行了计算机仿真和对比。两种控制技术所产生的转矩对阶跃转矩命令的响应如图11所示。
如图11所示，传统电机控制技术所产生的转矩响应速度较慢，且存在明显的超调和过冲。而所发明的电机驱动技术所产生的转矩响应速度快，且无超调和过冲。
实验及仿真数据三：
为了证明所发明的电机控制技术可以通过调节预测时间步长(T s)控制响应带宽，发明人基于非线性电机系统模型分别对0.01秒、0.001秒、0.0001秒的情况进行了计算机仿真。电机控制技术对转矩阶跃命令的响应效果如图12所示。
如图12所示，随着预测时间步长(T s)的变化，电机的转矩响应带宽也随之改变。但是所产生的转矩均无超调和过冲。因此可以通过调节T s方便地控制电机转矩或电流的响应带宽。因此对电机控制技术的参数整定非常容易，只需要调节T s即可。
具体应用例二：
下面通过具体应用例二，对上述各实施例的内容进行示例性说明。
现有技术中直接磁链矢量(DFVC)控制技术主要通过两个PI控制器分别控制电机磁链幅值和t轴电流，并产生f轴和t轴的电压命令，经坐标变换后产生相应的d轴和q轴的电压命令输入到逆变器产生控制信号作用于逆变器，从而实现对电机的控制，其缺点主要列举如下：
1、现有基于PI控制器的电机磁链幅值及t轴电流控制技术存在着快速响应和超调过冲之间的矛盾。
2、由于电机参数随运行工况变化，现有基于PI控制器的电机磁链幅值及t轴电流控制技术的PI控制器参数整定存在困难。
3、传统基于PI控制器的电机磁链幅值及t轴电流控制技术不易处理系统约束，存在积分饱和的现象。
针对上述现有基于PI控制器的电机驱动控制技术所存在的问题，本例提出一种电机无PI直接磁链矢量控制策略来实现对电机转矩以及磁链的控制和调节，避免现有电机控制技术所存在的PI参数难以整定、积分饱和以及快速响应和超调过冲之间的矛盾。
本例首先根据公式计算出逆变器各个桥臂的开关状态对电机定子磁链幅值 (ψ s)以及磁链角度，即f轴与d轴夹角δ，的影响，并以此构造预测模型，然后基于该预测模型计算出在所指定的预测时间步长(T s)内，当所预测的定子磁链幅值和角度分别等于它们的命令值时，逆变器不同备选开关状态组合所对应的基本电压矢量的作用时间组，结合电机当前电压矢量命令的位置，选择一个最优的基本电压矢量组合以及其对应的作用时间组，并基于所选择的最优基本电压矢量组合以及其对应的作用时间组计算出所合成的新的电压矢量命令，通过空间脉宽调制技术(SVPWM)将所合成的电机矢量命令作用于逆变器，实现对电机转矩的控制。本技术可以通过改变T s轻易调节转矩控制的带宽，且无超调和过冲，最后通过控制电机转矩实现对电机转速的控制。所提出的控制技术用到的当前电机定子磁链幅值及磁链角度的反馈值可通磁链观测器获得。
本例提出了一种电机无PI直接磁链矢量控制技术，其控制原理框图如图13所示，具体细节如下：
仍如图6所示，三相两级逆变器的三个桥臂一共有八个开关状态，对应于如图7所示的六个有效基本电压矢量和两个零电压矢量，并可依逆时针旋转方向将其划分为六个扇区(扇区Ⅰ-扇区Ⅵ)。
根据坐标变换，图6中的开关状态与f、t轴电压的关系为：
Figure PCTCN2019086653-appb-000047
其中，S a(k)、S b(k)、S c(k)分别为逆变器上桥臂开关状态(上桥臂导通为1，否则为0)，θ f(k)为电机定子磁链与α轴的夹角，即θ f(k)＝δ(k)+θ e(k)，其中，δ(k)为f轴与d轴夹角，θ e(k)为电机转子电角度。V dc为直流母线电压，
Figure PCTCN2019086653-appb-000048
为电压矢量在f轴上的投影，即f轴电压，
Figure PCTCN2019086653-appb-000049
为电压矢量在t轴上的投影，即t轴电压， k为采样时刻。
电机在f-t轴下的数学模型为：
Figure PCTCN2019086653-appb-000050
Figure PCTCN2019086653-appb-000051
其中，t s为电压矢量作用的时间步长，
Figure PCTCN2019086653-appb-000052
分别为t轴电流和f轴电流，R为定子电阻，p为电机极对数，ω m为电机转子机械角速度，由p及ω m可得到电机转子电角速度。基于当前f、t轴定子磁链幅值
Figure PCTCN2019086653-appb-000053
以及磁链角度δ k，即可利用(2)、(3)预测下一时间步长后定子磁链幅值以及角度，即
Figure PCTCN2019086653-appb-000054
和δ k+1。通过反复迭代(2)、(3)，则可实现对未来N p个时间步长后定子磁链幅值以及磁链角度的预测。
如图8所示，根据空间矢量脉宽调制(SVPWM)的原理，通过快速变换图8中两个相邻的基本电压矢量(U x、U x+1)以及零矢量并调整它们的作用时间(t 1，t 2，t 0)，可以合成在电压限值以内的任意电压矢量U a，其中x为两个电压矢量所在图7中所示扇区的编号(当x＝6时，x+1被赋值为1)。
图8中的U x、U x+1是两个属于编号为x的扇区的相邻的基本电压矢量，其中x＝1,2,…,6。通过在U x、U x+1以及零矢量间切换，可以合成模板电压矢量U a。若令U x、U x+1以及零矢量的作用总时间分别为t 1、t 2和t 0，则基于公式(2)、(3)，定子磁链幅值ψ s以及磁链角度δ与t 1、t 2和t 0的关系可表示为：
Figure PCTCN2019086653-appb-000055
δ k+1＝δ k+C t0t 0+C t_xt 1+C t_x+1t 2     (5)
其中，
Figure PCTCN2019086653-appb-000056
和δ k为由磁链观测器所观测的电机当前定子磁链幅值及角度。C f0、C t0、C f_x、C t_x、C f_x+1、C t_x+1分别为当零矢量、电压矢量U x、电压矢量U x+1作用时，f、t轴坐标系下定子磁链幅值以及磁链夹角的变化率。这些变化率可根据(2)、(3)由下式给出：
Figure PCTCN2019086653-appb-000057
Figure PCTCN2019086653-appb-000058
Figure PCTCN2019086653-appb-000059
Figure PCTCN2019086653-appb-000060
Figure PCTCN2019086653-appb-000061
Figure PCTCN2019086653-appb-000062
其中，u f_x、u t_x、u f_x+1、u t_x+1分别为第一个电压矢量(U x)和第二个电压矢量(U x+1)在f、t轴的分量，可根据坐标变换由(1)求得，其中，(1)中S a(k)、S b(k)、S c(k)对应相应的U x和U x+1。
[根据细则91更正 14.05.2019]　
通过反复迭代(4)、(5)可预测第k+N p时间步长时刻的定子磁链
Figure PCTCN2019086653-appb-000063
以及磁链角度
Figure PCTCN2019086653-appb-000064
为了消除电流控制的稳态误差，令第k+N p时间步长时刻定子磁链
Figure PCTCN2019086653-appb-000065
以及磁链角度
Figure PCTCN2019086653-appb-000066
等于参考命令值（ψ*s, δ*)，有：
Figure PCTCN2019086653-appb-000067
为方便描述，本例说明中只选取预测步长为1的情况进行说明，即N p＝1。N p等于其他值的情况也类似推导。联立(4)-(12)可求得三个矢量(两个基本电压矢量及零电压矢量)的作用时间为：
Figure PCTCN2019086653-appb-000068
Figure PCTCN2019086653-appb-000069
其中，
Q＝C t0C f_x+1+C t_xC f0+C t_x+1C f_x-C t_xC f_x+1-C t_x+1C f0-C t0C f_x
                             (15)
若根据所选择的U x、U x+1基本电压矢量组合计算出的t 1、t 2之和大于T s，则根据(16)、(17)对t 1、t 2进行等比例放缩：
Figure PCTCN2019086653-appb-000070
Figure PCTCN2019086653-appb-000071
依次将不同扇区的U x、U x+1代入(13)-(15)，可得到若干组(t 1,t 2,t 0)，其中t 0＝T s-t 1-t 2。为方便表述，将它们写作
Figure PCTCN2019086653-appb-000072
其中x为1,2,…,6或与当前电压矢量所在扇区相邻的两个扇区的编号。然后将上述
Figure PCTCN2019086653-appb-000073
分别代入损失函数(18)，其中x为
Figure PCTCN2019086653-appb-000074
所对应的电压矢量扇区编号，x present为当前电压矢量命令所对应的扇区编号。M为一个很大的值，以保证当
Figure PCTCN2019086653-appb-000075
或
Figure PCTCN2019086653-appb-000076
小于0时，J x会大于任意在
Figure PCTCN2019086653-appb-000077
且
Figure PCTCN2019086653-appb-000078
大于等于0时所对应的损失函数值。
Figure PCTCN2019086653-appb-000079
通过选择使得损失函数(J x)最小的
Figure PCTCN2019086653-appb-000080
可以基于其所对应的U x、U x+1合成电压矢量命令，用于控制逆变器实现对电机控制。将使得损失函数值(J x)最小的
Figure PCTCN2019086653-appb-000081
标记为
Figure PCTCN2019086653-appb-000082
则基于
Figure PCTCN2019086653-appb-000083
以及其所对应的U x、U x+1，可以通过(19)-(22)计算出所合成的电压矢量命令
Figure PCTCN2019086653-appb-000084
在d-轴和q-轴上的投影。
Figure PCTCN2019086653-appb-000085
Figure PCTCN2019086653-appb-000086
Figure PCTCN2019086653-appb-000087
Figure PCTCN2019086653-appb-000088
其中，θ x、θ x+1分别为U x、U x+1与α轴之间的空间相位角，θ e为电机转子电角度。
然后将所计算的
Figure PCTCN2019086653-appb-000089
分别作为d轴和q轴电压命令输送给图13中的SVPWM模块，即可产生逆变器开关状态控制命令实现对逆变器及电机的驱动控制。图13所示的内容可以实现对电机定子磁链幅值和角度的控制，进而实现对电机转矩的控制。而通过控制电机转矩可以进一步实现对电机转速的控制。此外，通过调节T s的大小，可以轻易实现对电流控制带宽的调节。
本例的控制策略相比现有被广泛采用的DFVC控制策略而言具有以下优点：
1、传统基于f-t坐标系的直接磁链矢量控制策略由于基于PI控制器进行设计，存在着动态响应速度较低、积分器饱和、系统约束不易处理以及存在快速响应与超调和过冲之间的矛盾等问题，严重影响了现有电机控制技术的性能。针对现有电机控制技术的以上不足，本例所提出的电机无PI直接磁链矢量控制技术解决了上述传统电机控制技术所面临的积分器饱和、直轴交轴电流耦合、系统约束不易处理以及存在快速响应与超调之间的矛盾等问题，具有无超调、电流响应速度快等优点，并避免了交直轴电流耦合的问题。
2、由于电机参数会随运行工况改变，因此现有基于PI控制器的电机控制技术存在PI控制器参数整定困难的问题。而本例所提出的无PI直接磁链矢量控制技术只有一个需要调节的参数，即预测时间步长(T s)，并可以通过调节T s的大小对电机电流控制带宽进行调节，且不存在超调和过冲。因此避免了现有基于PI控制器的电机控制技术参数难以整定的问题。
3、由于本例所提出的电机控制方法只有一个需要整定的参数，因此实现起来容易，并对不同参数的电机具有更强的适应性。不需要根据电机参数的不同而重新整定控制器参数。同时随着电机的运行，电机参数会发生变化，降低了电机控制策略的精确度与准确度。本例所提出的电机无PI直接磁链矢量控制技术由于电机参数没有参与反馈量计算。因此避免了电机参数变化给电机控制策略带来的影响，更容易实现更高精度的控制策略的实现与要求。
具体应用例二对应的实验及仿真数据：
实验及仿真数据一：
为了论证本专利提出的控制策略的有效性以及正确性，申请人基于一台非线性永磁同步伺服电机驱动系统模型对本发明所提出的控制技术进行了计算机仿真验证。磁链角度命令以及磁链角度控制结果如图14所示。可以发现所提出的控制方法可以准确和快速地控制电机磁链角度。
与图14所示工况相对应的电机磁链幅值控制效果图15所示。由图15可以看出，所提出的方法可以快速准确地控制电机磁链幅值且没有超调和过冲。
与图14所示工况相对应的电机磁转矩控制效果图16所示。可以看出，电机的转矩控制也没有超调和过冲。
实验及仿真数据二：
为了证明本专利所发明的电机控制技术可以通过调节预测时间步长(T s)控制响应带宽，发明人基于非线性电机系统模型分别对0.001秒，0.0001秒的情况进行了仿真。电机控制技术对转矩阶跃命令的响应效果如图17所示。
如图17所示，随着预测时间步长(T s)的变化，电机的转矩响应带宽也随之改变。但是所产生的转矩均无超调和过冲。因此可以通过调节T s方便地控制电机转矩或电流的响应带宽。因此对电机控制技术的参数整定非常容易，只需要调节T s即可。以上所述仅为本发明的较佳实施例而已，并不用以限制本发明，凡在本发明的精神和原则之内所作的任何修改、等同替换和改进等，均应包含在本发明的保护范围之内。

Claims (10)

  1. 一种电机控制方法，其特征在于，包括：
    获得当前时刻电机定子侧的若干当前被控量的观测值、所述电机的当前转子电角速度，以及预测控制周期内所述电机运行工况下的电机参数；
    将预测控制周期、所述当前被控量的观测值、所述电机参数、所述当前转子电角速度，以及若干备选电压矢量组输入至一关系模型，得到各所述备选电压矢量组对应的各当前预测被控量值组，所述备选电压矢量组包括逆变器所输出的基本电压矢量及零电压矢量，所述当前预测被控量值组包括若干当前被控量预测值，并令所述当前被控量预测值等于当前控制命令值，得到与各所述备选电压矢量组对应的各作用时间组，所述作用时间组包括所述基本电压矢量及所述零电压矢量的作用时间；
    利用预先建立的、以所述作用时间组为条件的损失函数，选择造成损失函数值最小的所述作用时间组及对应的所述备选电压矢量组；
    由所选择的所述作用时间组及对应的所述备选电压矢量组，得到所述逆变器当前输出的下一电压矢量，并将其分解为下一d轴电压值及下一q轴电压值。
  2. 如权利要求1所述的方法，其特征在于，所述当前被控量观测值为：由所述电机定子侧的当前各相电流测量值变换得来的当前d轴电流值与当前q轴电流值，所述当前被控量预测值为：当前d轴预测电流值和当前q轴预测电流值，
    或者，所述当前被控量观测值为：基于磁链观测器观测得到的当前定子磁链幅值与当前定子磁链角度值，所述当前被控量预测值为：当前定子磁链幅度预测值和当前定子磁链角度预测值。
  3. 如权利要求1所述的方法，其特征在于，所述方法还包括：
    对所述预测控制周期进行调节。
  4. 如权利要求1所述的方法，其特征在于，一所述备选电压矢量组对应一静止坐标系扇区，所述扇区对应两个所述基本电压矢量，
    所述损失函数通过如下方式构建：当所述作用时间组中各所述基本电压矢量对应的所述作用时间均有效时，以所述备选电压矢量组所对应的备选扇区编号，以及，当前d轴电压分量与当前q轴电压分量所合成的当前电压合矢量对应的当前扇区编号，确定损失函数值。
  5. 如权利要求1所述的方法，其特征在于，所述方法还包括：
    当所述作用时间组中各所述基本电压矢量对应的所述作用时间之和大于所述预测控制周期时，对所述作用时间组中各所述基本电压矢量对应的所述作用时间进行等比例缩小。
  6. 如权利要求1所述的方法，其特征在于，由所选择的所述作用时间组及所述备选电压矢量组，得到所述逆变器当前输出的下一电压矢量，并将其分解为下一d轴电压值及下一q轴电压值，具体包括：
    由所选择的所述作用时间组及所述备选电压矢量组，得到静止坐标系下当前输出的下一α轴电压值及下一β轴电压值；
    由所述下一α轴电压值及所述下一β轴电压值，得到旋转坐标系下所述逆变器当前输出的所述下一d轴电压值及所述下一q轴电压值。
  7. 如权利要求1所述的方法，其特征在于，所述电机参数从数据表调用或通过在线参数辨识技术获得，所述电机参数包括如下参数中的一种或多种的组合：d轴电感、q轴电感、永磁体磁链、定子电阻及电机极对数。
  8. 一种电机控制器，包括存储器及处理器，其特征在于，所述处理器执行所述存储器中存储的程序时实现如权利要求1至7任一项所述方法中的步骤。
  9. 一种可读存储介质，所述可读存储介质存储有程序，其特征在于，所述程序被处理器执行时实现如权利要求1至7任一项所述方法中的步骤。
  10. 一种电机驱动系统，其特征在于，包括：逆变器、空间矢量脉宽调制模块以及如权利要求8所述的电机控制器，所述空间矢量脉宽调制模块将所述下一d轴电压值及所述下一q轴电压值转换成所述逆变器的状态控制命令，以实现对所述逆变器及所述电机的驱动控制。
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