WO2020211424A1 - 一种功率变换电路、逆变器及控制方法 - Google Patents

一种功率变换电路、逆变器及控制方法 Download PDF

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Publication number
WO2020211424A1
WO2020211424A1 PCT/CN2019/126910 CN2019126910W WO2020211424A1 WO 2020211424 A1 WO2020211424 A1 WO 2020211424A1 CN 2019126910 W CN2019126910 W CN 2019126910W WO 2020211424 A1 WO2020211424 A1 WO 2020211424A1
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WIPO (PCT)
Prior art keywords
circuit
winding
common mode
filter capacitor
mode filter
Prior art date
Application number
PCT/CN2019/126910
Other languages
English (en)
French (fr)
Inventor
王朝辉
石磊
陈东
Original Assignee
华为技术有限公司
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
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Publication date
Application filed by 华为技术有限公司 filed Critical 华为技术有限公司
Priority to EP19925326.1A priority Critical patent/EP3934051B1/en
Publication of WO2020211424A1 publication Critical patent/WO2020211424A1/zh
Priority to US17/495,117 priority patent/US11984819B2/en

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    • H02J3/383—
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H02M1/123—Suppression of common mode voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J2300/00—Systems for supplying or distributing electric power characterised by decentralized, dispersed, or local generation
    • H02J2300/20—The dispersed energy generation being of renewable origin
    • H02J2300/22—The renewable source being solar energy
    • H02J2300/24—The renewable source being solar energy of photovoltaic origin
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H02J3/381—Dispersed generators
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E10/00—Energy generation through renewable energy sources
    • Y02E10/50—Photovoltaic [PV] energy
    • Y02E10/56—Power conversion systems, e.g. maximum power point trackers

Definitions

  • This application relates to the field of photovoltaic technology, and in particular to a control method of a power conversion circuit, an inverter and a hybrid modulation.
  • the inverter In the photovoltaic grid-connected power generation system, the inverter is the key equipment, and its conversion efficiency and performance directly determine the revenue of the power generation system.
  • the current inverters In order to reduce the cost of the inverter, the current inverters mostly adopt a non-isolated architecture, and achieve direct grid-connected photovoltaic power generation through two-stage relays.
  • a single-phase power grid system usually one end of the power grid is grounded at the remote end through a transmission line. At this time, photovoltaic panels, inverters, and the power grid form a current loop through the ground, which will cause serious leakage current problems.
  • the embodiment of the application discloses a power conversion circuit, which can improve the conversion efficiency of the inverter and can effectively reduce the common mode leakage current.
  • the embodiments of the present application also provide corresponding inverters and hybrid modulation control methods.
  • the first aspect of the present application provides a power conversion circuit, which may include:
  • Switching network control circuit, filter circuit, DC side circuit and AC side circuit
  • the switch network is connected to the DC side circuit, the switch network is connected to the control circuit, the switch network is connected to the filter circuit, and the filter circuit is connected to the AC side circuit;
  • the control circuit is used to control the switch network
  • the filter circuit includes a first power inductor, a common mode inductor, a first differential mode filter capacitor, a first common mode filter capacitor, and a second common mode filter capacitor;
  • the first power inductor includes a first winding and a second winding
  • the common mode inductor includes a third winding and a fourth winding
  • the first end of the first winding and the first end of the second winding are respectively connected to the switch network, the second end of the first winding is connected to the first end of the third winding, so The second end of the second winding is connected to the first end of the fourth winding;
  • Two ends of the first differential mode filter capacitor are respectively connected to the second ends of the first winding and the second winding;
  • the first end of the first common mode filter capacitor is connected to the second end of the third winding, and the second end of the first common mode filter capacitor is connected to the DC side circuit through a low impedance circuit;
  • the first end of the second common mode filter capacitor is connected to the second end of the quad winding, and the second end of the second common mode filter capacitor is connected to the DC side circuit through a low impedance circuit.
  • the second end of the first common-mode filter capacitor is connected to the DC side circuit through a low-impedance circuit.
  • the low-impedance circuit can be called a first low-impedance circuit, and the second end of the second common-mode filter capacitor is a low-impedance circuit.
  • the low impedance circuit in which the circuit is connected to the DC side circuit may be referred to as a second low impedance circuit.
  • the first low-impedance circuit and the second low-impedance circuit may be the same low-impedance circuit or different low-impedance circuits.
  • the first differential mode filter capacitor is connected to the filter circuit before the common mode inductor, which can effectively avoid the loss caused by the high frequency current component flowing into the common mode inductor, thereby improving the conversion efficiency of the power conversion circuit.
  • the second ends of the first common mode filter capacitor and the second common mode filter capacitor are both connected to the DC side circuit, which can provide a common mode current low impedance loop, thereby effectively reducing the common mode leakage current of the power conversion circuit to the output port.
  • the low impedance circuit is a zero impedance circuit, or the low impedance circuit includes one resistor or at least two series resistors.
  • the filter circuit further includes a second power inductor, and the second power inductor It includes a fifth winding and a sixth winding, the second end of the third winding is connected to the first end of the fifth winding, and the second end of the fourth winding is connected to the first end of the sixth winding , The second end of the fifth winding and the second end of the sixth winding are connected to the AC side circuit.
  • the DC side circuit includes a positive bus, a bus capacitor, and a negative The bus bar, the two ends of the bus bar capacitor are respectively connected with the positive bus bar and the negative bus bar.
  • a possible implementation manner combined with the third possible implementation manner of the first aspect, in the fifth possible implementation manner, the second end of the first common-mode filter capacitor and the second common-mode filter capacitor The second ends are respectively connected to the negative bus.
  • the bus capacitance includes a positive bus capacitance and a negative bus capacitance, and the first of the positive bus capacitance Terminal is connected to the positive bus, the second terminal of the positive bus capacitor is connected to the first terminal of the negative bus capacitor, and the second terminal of the negative bus capacitor is connected to the negative bus; the first common The second end of the mode filter capacitor and the second end of the second common mode filter capacitor are connected to the midpoint of the positive bus capacitor and the negative bus capacitor.
  • a possible implementation manner combining the first aspect and any one of the first to seventh possible implementation manners of the first aspect, in the eighth possible implementation manner,
  • the switching network includes a first converter bridge arm and a second converter bridge arm, the first converter bridge arm includes a first switching device and a second switching device, and the second converter bridge arm includes a third switch Device and the fourth switching device;
  • the control circuit controls the first switching device and the second switching device of the first converter bridge arm through a first sinusoidal modulation wave and a first carrier;
  • the control circuit controls the third switching device and the fourth switching device of the second converter bridge arm through a second sinusoidal modulation wave and a second carrier;
  • the first carrier and the second carrier adopt bipolar modulation in a first preset angle range, and adopt unipolar modulation in a second preset angle range
  • the first preset angle range is based on the Set at the DC offset of the first sine modulation wave or the second sine modulation wave
  • the second preset angle range is the sine wave period of the first sine modulation wave or the second sine modulation wave
  • the switching frequency of the bipolar modulation is higher than the switching frequency of the unipolar modulation.
  • the DC bias refers to the presence of a DC component in the alternating current.
  • the sinusoidal modulation wave may be under a DC bias. If the DC bias is 0, it is a zero-crossing point; if the DC bias is If it is not zero, it is the DC offset value. It can be seen from the above eighth possible implementation that the use of unipolar and bipolar mixed modulation can further improve the conversion efficiency of the power conversion circuit and further reduce the leakage current.
  • the first preset angle range includes (- ⁇ , ⁇ ), the values of - ⁇ and ⁇ are adjusted according to state information, and the state information includes the voltage of the positive bus, the voltage of the negative bus, and the The voltage of the AC side circuit.
  • a second aspect of the embodiments of the present application provides a power conversion circuit, which may include:
  • Switching network control circuit, filter circuit, DC side circuit and AC side circuit
  • the switch network is connected to the DC side circuit, the switch network is connected to the control circuit, the switch network is connected to the filter circuit, and the filter circuit is connected to the AC side circuit;
  • the control circuit is used to control the switch network
  • the filter circuit includes a first power inductor, a common mode inductor, a first differential mode filter capacitor, a first common mode filter capacitor, and a second common mode filter capacitor;
  • the common mode inductor includes a third winding and a fourth winding
  • the first end of the first power inductor is connected to the switch network, the second end of the first power inductor is connected to the first end of the third winding, and the first end of the fourth winding is connected to the switch network.
  • the first end of the first differential mode filter capacitor is connected to the second end of the first power inductor, and the second end of the first differential mode filter capacitor is connected to the first end of the third winding;
  • the first end of the first common mode filter capacitor is connected to the second end of the third winding, and the second end of the first common mode filter capacitor is connected to the DC side circuit through a low impedance circuit;
  • the first end of the second common mode filter capacitor is connected to the second end of the quad winding, and the second end of the second common mode filter capacitor is connected to the DC side circuit through a low impedance circuit.
  • the second end of the first common-mode filter capacitor is connected to the DC side circuit through a low-impedance circuit.
  • the low-impedance circuit can be called a first low-impedance circuit, and the second end of the second common-mode filter capacitor is a low-impedance circuit.
  • the low impedance circuit in which the circuit is connected to the DC side circuit may be referred to as a second low impedance circuit.
  • the first low impedance circuit and the second low impedance circuit may be the same low impedance circuit or different low impedance circuits.
  • the first differential mode filter capacitor is connected before the common mode inductor on the filter circuit, which can effectively avoid the loss caused by the high frequency current component flowing into the common mode inductor, thereby improving the conversion of the power conversion circuit.
  • the second end of the first common mode filter capacitor and the second common mode filter capacitor are connected to the DC side circuit, which can provide a common mode current low impedance loop, thereby effectively reducing the common mode leakage current of the power conversion circuit to the output port .
  • the low impedance circuit is a zero impedance circuit, or the low impedance circuit includes one resistor or at least two series resistors.
  • the filter circuit further includes a second power inductor, and the second power inductor It includes a fifth winding and a sixth winding, the second end of the third winding is connected to the first end of the fifth winding, and the second end of the fourth winding is connected to the first end of the sixth winding , The second end of the fifth winding and the second end of the sixth winding are connected to the AC side circuit.
  • the DC side circuit includes a positive bus, a bus capacitor, and a negative The bus bar, the two ends of the bus bar capacitor are respectively connected with the positive bus bar and the negative bus bar.
  • a possible implementation manner combined with the third possible implementation manner of the second aspect, in a fourth possible implementation manner, the second end of the first common-mode filter capacitor and the second common-mode filter capacitor The second ends of the are respectively connected to the positive bus.
  • the bus capacitance includes a positive bus capacitance and a negative bus capacitance, and the first of the positive bus capacitance Terminal is connected to the positive bus, the second terminal of the positive bus capacitor is connected to the first terminal of the negative bus capacitor, and the second terminal of the negative bus capacitor is connected to the negative bus; the first common The second end of the mode filter capacitor and the second end of the second common mode filter capacitor are connected to the midpoint of the positive bus capacitor and the negative bus capacitor.
  • a possible implementation manner combining the second aspect and any one of the first to seventh possible implementation manners of the second aspect, in the eighth possible implementation manner,
  • the switching network includes a first converter bridge arm and a second converter bridge arm, the first converter bridge arm includes a first switching device and a second switching device, and the second converter bridge arm includes a third switch Device and the fourth switching device;
  • the control circuit controls the first switching device and the second switching device of the first converter bridge arm through a first sinusoidal modulation wave and a first carrier;
  • the control circuit controls the third switching device and the fourth switching device of the second converter bridge arm through a second sinusoidal modulation wave and a second carrier;
  • the first carrier and the second carrier adopt bipolar modulation in a first preset angle range, and adopt unipolar modulation in a second preset angle range
  • the first preset angle range is based on the Set at the DC offset of the first sine modulation wave or the second sine modulation wave
  • the second preset angle range is the sine wave period of the first sine modulation wave or the second sine modulation wave
  • the switching frequency of the bipolar modulation is higher than the switching frequency of the unipolar modulation.
  • the DC bias refers to the presence of a DC component in the alternating current.
  • the sinusoidal modulation wave may be under a DC bias. If the DC bias is 0, it is a zero-crossing point; if the DC bias is If it is not zero, it is the DC offset value. It can be seen from the above eighth possible implementation that the use of unipolar and bipolar mixed modulation can further improve the conversion efficiency of the power conversion circuit and further reduce the leakage current.
  • the first preset angle range includes (- ⁇ , ⁇ ), the values of - ⁇ and ⁇ are adjusted according to state information, and the state information includes the voltage of the positive bus, the voltage of the negative bus, and the The voltage of the AC side circuit.
  • the third aspect of the present application provides a power conversion circuit, which may include:
  • Switching network control circuit, filter circuit, DC side circuit and AC side circuit
  • the switch network is connected to the DC side circuit, the switch network is connected to the control circuit, the switch network is connected to the filter circuit, and the filter circuit is connected to the AC side circuit;
  • the control circuit is used to control the switch network
  • the filter circuit includes a third power inductor and a fourth power inductor, a common mode inductor, a first differential mode filter capacitor, a first common mode filter capacitor, and a second common mode filter capacitor;
  • the common mode inductor includes a third winding and a fourth winding
  • the first end of the third power inductor and the first end of the fourth power inductor are respectively connected to the switch network, and the second end of the third power inductor is connected to the first end of the third winding Connected, the second end of the fourth power inductor is connected to the first end of the fourth winding;
  • Two ends of the first differential mode filter capacitor are respectively connected to the second ends of the third power inductor and the fourth power inductor;
  • the first end of the first common mode filter capacitor is connected to the second end of the third winding, and the second end of the first common mode filter capacitor is connected to the DC side circuit through a low impedance circuit;
  • the first end of the second common mode filter capacitor is connected to the second end of the quad winding, and the second end of the second common mode filter capacitor is connected to the DC side circuit through a low impedance circuit.
  • the second end of the first common-mode filter capacitor is connected to the DC side circuit through a low-impedance circuit.
  • the low-impedance circuit can be called a first low-impedance circuit, and the second end of the second common-mode filter capacitor is a low-impedance circuit.
  • the low impedance circuit in which the circuit is connected to the DC side circuit may be referred to as a second low impedance circuit.
  • the first low impedance circuit and the second low impedance circuit may be the same low impedance circuit or different low impedance circuits.
  • the first differential mode filter capacitor is connected to the filter circuit before the common mode inductor, which can effectively avoid the loss caused by high-frequency current components flowing into the common mode inductor, thereby improving the conversion of the power conversion circuit.
  • the second end of the first common mode filter capacitor and the second common mode filter capacitor are connected to the DC side circuit, which can provide a common mode current low impedance loop, thereby effectively reducing the common mode leakage current of the power conversion circuit to the output port .
  • the third aspect of this application is different from the first aspect in that the functions of the first winding and the second winding are realized by the third power inductor and the fourth power inductor.
  • the first winding and the second winding can also be understood as For the two power inductors, the essence of the solutions provided by the first and third aspects is still the same.
  • a fourth aspect of the present application provides a hybrid modulation control method, which is applied to a power conversion circuit, the power conversion circuit including a switch network, a control circuit, a filter circuit, a DC side circuit, and an AC side circuit;
  • the switch network is connected to the DC side circuit, the switch network is connected to the control circuit, the switch network is connected to the filter circuit, the filter circuit is connected to the AC side circuit, and the switch network It includes a first converter bridge arm and a second converter bridge arm.
  • the first converter bridge arm includes a first switching device and a second switching device.
  • the second converter bridge arm includes a third switching device and a fourth switching device. Switching device
  • the method includes:
  • the control circuit controls the first switching device and the second switching device of the first converter bridge arm through a first sinusoidal modulation wave and a first carrier;
  • the control circuit controls the third switching device and the fourth switching device of the second converter bridge arm through a second sinusoidal modulation wave and a second carrier;
  • the first carrier and the second carrier adopt bipolar modulation in a first preset angle range, and adopt unipolar modulation in a second preset angle range
  • the first preset angle range is based on the Set at the DC offset of the first sine modulation wave or the second sine modulation wave
  • the second preset angle range is the sine wave period of the first sine modulation wave or the second sine modulation wave
  • the switching frequency of the bipolar modulation is higher than the switching frequency of the unipolar modulation.
  • the first preset angle range includes (- ⁇ , ⁇ ), and the values of - ⁇ and ⁇ are adjusted according to state information, and the state information includes all The voltage of the positive bus, the voltage of the negative bus, and the voltage of the AC side circuit.
  • the fifth aspect of the present application provides an inverter, including any possible implementation manner of the first aspect and the first aspect, or any possible implementation manner of the second and second aspects, or the third and third aspects The power conversion circuit described in any possible implementation manner.
  • the sixth aspect of the present application provides a photovoltaic power generation system, which may include:
  • the photovoltaic panel is connected to the inverter, and the inverter is connected to the AC network;
  • the photovoltaic panel is used to convert the light energy into direct current
  • the inverter includes any possible implementation manner of the first aspect or the first aspect, or any possible implementation manner of the second aspect or the second aspect, or any possible implementation manner of the third aspect or the third aspect.
  • the power conversion circuit described is used to convert the direct current into alternating current;
  • the AC network is used to transmit the AC power.
  • the seventh aspect of the present application provides a computer-readable storage medium having instructions stored in the computer-readable storage medium, which when run on a computer, cause the computer to execute the control method described in the fourth aspect.
  • the first differential mode filter capacitor is connected to the filter circuit before the common mode inductor, which can effectively avoid the loss caused by the high-frequency current component flowing into the common mode inductor, thereby improving the power conversion circuit.
  • the second ends of the first common-mode filter capacitor and the second common-mode filter capacitor are connected to the DC side circuit, which can provide a common-mode current low-impedance loop, thereby effectively reducing the common-mode leakage of the power conversion circuit to the output port Current.
  • Fig. 1 is a schematic diagram of a scenario applied by an embodiment of the present application
  • FIG. 2 is a schematic diagram of another scenario applied in an embodiment of the present application.
  • FIG. 3 is a schematic diagram of a power conversion circuit provided by an embodiment of the present application.
  • Fig. 4 is a structural diagram of a power conversion circuit provided by an embodiment of the present application.
  • FIG. 5 is a structural diagram of another power conversion circuit provided by an embodiment of the present application.
  • FIG. 6 is a structural diagram of another power conversion circuit provided by an embodiment of the present application.
  • FIG. 7 is a structural diagram of another power conversion circuit provided by an embodiment of the present application.
  • FIG. 8 is a structural diagram of another power conversion circuit provided by an embodiment of the present application.
  • FIG. 9 is a structural diagram of another power conversion circuit provided by an embodiment of the present application.
  • FIG. 10 is a structural diagram of another power conversion circuit provided by an embodiment of the present application.
  • FIG. 11 is a circuit waveform diagram based on unipolar modulation provided by an embodiment of the present application.
  • FIG. 12 is a schematic diagram of a waveform of a hybrid modulation control method provided by an embodiment of the present application.
  • FIG. 13 is a circuit waveform diagram based on hybrid modulation provided by an embodiment of the present application.
  • FIG. 14 is a comparison diagram of variable flow efficiency under different control modes provided by the embodiments of the present application.
  • FIG. 15 is a waveform diagram of inverter voltage and common mode inductor current provided by an embodiment of the present application.
  • the embodiment of the application discloses a power conversion circuit, which can improve the conversion efficiency of the inverter and can effectively reduce the common mode leakage current.
  • the embodiments of the present application also provide corresponding inverters and hybrid modulation control methods.
  • This application proposes a control method for a power conversion circuit, an inverter and a hybrid modulation.
  • the power conversion circuit, inverter, and hybrid modulation control method can be applied to the scene architecture shown in FIG. 1.
  • Fig. 1 is a schematic diagram of a scenario applied in an embodiment of the present application.
  • Figure 1 shows the architecture of an uninterruptible power system (UPS) power supply system.
  • the mains supply power directly to the load, and the power frequency AC mains can pass through direct current (DC)/alternating current (AC)
  • the inverter converts alternating current to direct current and charges the battery through the DC/DC inverter; when the mains fails, the battery passes through the DC/DC inverter and the DC/AC inverter to convert the direct current into alternating current.
  • the load is supplied with power, wherein the power conversion circuit and hybrid modulation control method provided in the present application can be applied to the operation process of an AC/DC rectifier or a DC/AC inverter.
  • the power conversion circuit, inverter, and control method proposed in this application can be applied to a photovoltaic grid-connected power generation system.
  • Fig. 2 is a schematic diagram of another scenario applied in an embodiment of the present application.
  • the direct current output by the photovoltaic panel is converted into alternating current through a DC/AC inverter to realize grid-connected power generation of the photovoltaic panel.
  • the power conversion circuit and hybrid modulation control method provided by this application can be applied During the operation of the DC/AC inverter.
  • Fig. 3 is a schematic diagram of a power conversion circuit provided by an embodiment of the present application.
  • the power conversion circuit may include a DC side circuit 701, a switch network 702, a control circuit 703, a filter circuit 704, and an AC side circuit 705.
  • the switching network 702 is connected to the DC side circuit 701, the switching network 702 is connected to the control circuit 703, the switching network 702 is connected to the filter circuit 704, and the filter circuit 704 is connected to the AC side circuit 705.
  • the control circuit 703 is used to control the switch network 702 to convert the direct current input from the DC side circuit 701 into alternating current according to a preset modulation scheme, and the filter circuit 704 is used to filter out high frequency ripples generated by the switch network 702 during the modulation process. Wave and transmit the processed AC power to the AC side circuit 705.
  • the power conversion circuit shown in FIG. 3 may adopt the structure diagram shown in FIG. 4.
  • Fig. 4 is a structural diagram of a power conversion circuit provided by an embodiment of the present application.
  • the DC side circuit shown in Figure 4 includes a positive bus, a negative bus, and a bus capacitor.
  • the switching network may include a first converter bridge arm and a second converter bridge arm.
  • the first converter bridge arm includes a first switching device T1 and a second switching device T2
  • the second converter bridge arm includes a third switching device T3 and The fourth switching device T4.
  • the first switching device T1, the second switching device T2, the third switching device T3, and the fourth switching device T4 can be insulated gate bipolar transistors (IGBT), gallium nitride (gallium nitride) , GaN), metal-oxide-semiconductor field effect transistor (MOSFET) or other power semiconductor devices.
  • IGBT insulated gate bipolar transistors
  • gallium nitride gallium nitride
  • GaN gallium nitride
  • MOSFET metal-oxide-semiconductor field effect transistor
  • the filter circuit may include a first power inductor La1, a common mode inductor Lcm, a first differential mode filter capacitor Cdm, and a first common mode filter capacitor Ccm1.
  • the filter circuit may further include a second common mode filter capacitor Ccm2.
  • the first power inductor La1 includes a first winding La11 and a second winding La12. It can be understood that the ratio of the number of coils of the first winding La11 and the second winding La12 can be adjusted according to requirements. In a possible scenario, For the same number of coils, of course the specific number of coils depends on the actual scene, which is not limited in the embodiment of the present application.
  • the common mode inductance Lcm includes a third winding Lcm11 and a fourth winding Lcm12.
  • the first end of the first winding La11 and the first end of the second winding La12 are respectively connected to point A of the switch network, and the second end of the first winding La11 is connected to the first end of the third winding Lcm11, The second end of the second winding La12 is connected to the first end of the fourth winding Lcm12.
  • the first end of the first differential mode filter capacitor Cdm is connected to the second end of the first winding La11, and the second end of the first differential mode filter capacitor Cdm is connected to the second end of the second winding La12.
  • the first end of the first common mode filter capacitor Ccm1 is connected to the second end of the third winding Lcm11, and the second end of the first common mode filter capacitor Ccm1 is connected to the DC side circuit through a low impedance circuit.
  • the low impedance circuit can be understood as a wire connecting the second end of the first common mode filter capacitor Ccm1 directly to the DC side circuit.
  • the low impedance circuit is a zero impedance circuit.
  • the first end of the second common mode filter capacitor Ccm2 is connected to the second end of the quad winding Lcm12, and the second end of the second common mode filter capacitor Ccm2 is connected to the DC side circuit through a low impedance circuit.
  • the low-impedance circuit can be understood as a wire connecting the second end of the second common-mode filter capacitor Ccm2 directly to the DC side circuit. In this scenario, the low-impedance circuit is a zero-impedance circuit.
  • the common mode leakage current is drawn back to the DC side circuit through the first common mode filter capacitor Ccm1 and the second common mode filter capacitor Ccm2, thereby providing a common mode leakage current low-impedance loop, which can significantly reduce the inverter's ground leakage current;
  • the first power inductor La1 is composed of two windings La11 and La12, which can provide common mode impedance, reduce the influence of the inverter common mode voltage, and reduce the required common mode inductance.
  • the first power inductor La1 in the above solution can also be just one winding, and does not need to be divided into a first winding and a second winding.
  • the first end of the first power inductor and the switching network Connected the second end of the first power inductor is connected to the first end of the third winding, the first end of the fourth winding is connected to the switch network; the first end of the first differential mode filter capacitor One end is connected to the second end of the first power inductor, and the second end of the first differential mode filter capacitor is connected to the first end of the third winding.
  • the first power inductor La1 may also be composed of two independent power inductors, which can be understood with reference to FIG. 5.
  • Fig. 5 is a structural diagram of another power conversion circuit provided by an embodiment of the present application.
  • the first power inductor La1 in FIG. 4 is replaced with a third power inductor La3 and a fourth power inductor La4.
  • the third power inductor La3 and The number of coils of the fourth power inductor La4 can be selected with reference to the first winding La11 and the second winding La12, or can be adjusted according to requirements, and the specific number of coils depends on actual scenarios.
  • a second power inductor La2 can also be added between the filter circuit and the AC side circuit to provide impedance and reduce the impact of the power conversion circuit on the AC port.
  • the impact of this can be understood by referring to Figure 6.
  • Fig. 6 is a structural diagram of another power conversion circuit provided by an embodiment of the present application.
  • the second power inductor La2 includes a fifth winding La21 and a sixth winding La22.
  • the second end of the third winding Lcm11 is connected to the first end of the fifth winding La21
  • the second end of the fourth winding Lcm12 is connected to the The first end of the sixth winding La22 is connected
  • the second end of the fifth winding La21 and the second end of the sixth winding La21 are connected to the AC side circuit. It is understandable that the ratio of the number of coils of the fifth winding La21 and the sixth winding La22 can be adjusted according to requirements. In a possible scenario, the same number of coils can be used.
  • the embodiment of the application uses the first common mode filter capacitor Ccm1 and the second common mode filter capacitor Ccm2 to draw the common mode leakage current back to the DC side circuit, thereby providing a common mode leakage current low impedance loop, which can significantly reduce The earth leakage current of the inverter.
  • the DC side circuit that is the target of the common mode leakage current drawback includes a positive bus and a negative bus. It is understood that the common mode leakage current drawback target can include the following multiple situations, which will be described below with reference to the accompanying drawings.
  • FIG. 7 reference may be made to the structure diagram of another power conversion circuit shown in FIG. 7. As shown in Figure 7, the second end of the first common-mode filter capacitor Ccm1 is connected to the positive bus, and the second end of the second common-mode filter capacitor Ccm2 is connected to the positive bus. For the rest of the description, please refer to the correlation in Figure 4-10. To understand the content, I won’t repeat it here.
  • the second end of the first common-mode filter capacitor Ccm1 is connected to the positive bus, and the second end of the second common-mode filter capacitor Ccm2 is connected to the negative bus, so as to provide completeness for the two windings of the common-mode inductance Lcm.
  • Equivalent impedance circuit, the common mode inductor is not prone to unilateral saturation. The rest of the description can be understood with reference to the relevant content in Figures 4-10, and will not be repeated here.
  • the bus capacitance Cdc in this embodiment includes a positive bus capacitance Cdcp and a negative bus capacitance Cdcn.
  • the first end of the positive bus capacitor Cdcp is connected to the positive bus
  • the second end of the positive bus capacitor Cdcp is connected to the first end of the negative bus capacitor Cdcn
  • the second end of the negative bus capacitor Cdcn is connected to the negative bus.
  • the positive bus capacitor Cdcp There is a bus midpoint M between it and the negative bus capacitor Cdcn.
  • This filtering method can also provide a low-impedance circuit with common-mode leakage current, significantly reducing the output common-mode leakage current of the inverter.
  • this filtering method can also provide completely equivalent impedance loops for the two windings of the common mode inductor Lcm, and the common mode inductor Lcm is not prone to unilateral saturation.
  • the rest of the description can be understood with reference to the relevant content in Figures 4-10, and will not be repeated here.
  • a low-impedance resistor R1 can be connected as a low-impedance circuit on the wire of the common-mode leakage current drawn back to the DC side circuit, as shown in the structure diagram of another power conversion circuit in FIG.
  • a low-impedance resistor R1 is connected between the midpoint M and the filter circuit, that is, there is a resistor R1 on the low-impedance circuit.
  • the low-impedance resistor R1 can provide damping in the common-mode loop and effectively suppress the oscillation in the common-mode loop, and
  • the low impedance resistor R1 can be adjusted according to the actual measurement conditions to meet different circuit conditions.
  • this embodiment only takes the addition of a low-impedance resistor R1 as an example for description. In fact, R1 can also be replaced with two or more resistors connected in series.
  • connection of the low-impedance resistor R1 can also be applied to the above-mentioned solutions of one to four busbar connection points.
  • connection method refer to the relevant content of FIG. 10, and will not be repeated here.
  • the control circuit in the embodiment of this application needs to control the switching frequency in the switching network based on the control strategy, and draw the common mode leakage current back to the DC side circuit, thereby providing a common mode leakage current low impedance circuit, which can significantly reduce the inverter's Leakage current to ground.
  • the control strategy can be based on unipolar modulation, or mixed unipolar and bipolar modulation.
  • the waveform shown in FIG. 11 can be measured based on the circuit shown in FIG. 6, which is a circuit waveform based on unipolar modulation provided by an embodiment of the present application Figure. It can be seen from Fig. 11 that although the control strategy of the above-mentioned circuit based on unipolar modulation can reduce the output power inductance, unipolar modulation causes extremely serious common-mode voltage at the voltage zero crossing. On the one hand, the common-mode voltage leads to high common-mode inductance, which requires a large volume and high cost; on the other hand, the common-mode voltage causes extremely serious high-frequency loss in the common-mode inductance, which significantly reduces the unit The advantages of polar modulation.
  • an embodiment of the present application proposes a hybrid modulation control method, which is implemented by the control circuit 703. Specifically, the control The circuit 703 controls the first switching device T1 and the second switching device T2 of the first converter bridge arm through the first sinusoidal modulation wave and the first carrier; the control circuit controls the The third switching device T3 and the fourth switching device T4 of the second converter bridge arm.
  • the switching frequency of the switching network can be controlled by the unipolar and bipolar hybrid modulation method shown in FIG. 12.
  • FIG. 12 is a schematic diagram of a waveform of a hybrid modulation control method provided by an embodiment of the present application.
  • the first carrier and the second carrier shown in Fig. 12 adopt bipolar modulation in the first preset angle range (- ⁇ , ⁇ ), and remove ( - ⁇ , ⁇ ) adopt unipolar modulation within the angle range.
  • a 360-degree period it can also be understood as the angle (0 ⁇ ), (( ⁇ - ⁇ ) ⁇ ), ( ⁇ ( ⁇ + ⁇ )), ((2 ⁇ - ⁇ ) ⁇ 2 ⁇ )
  • the modulation method adopts bipolar modulation; in the angle ( ⁇ ( ⁇ - ⁇ )) and (( ⁇ + ⁇ ) ⁇ (2 ⁇ - ⁇ )), adopt unipolar modulation.
  • the first preset angle range is set based on the DC offset of the first sinusoidal modulation wave or the second sinusoidal modulation wave
  • the second preset angle range is set at the first sinusoidal modulation wave or the second sinusoidal modulation wave
  • the switching frequency of the bipolar modulation is higher than the switching frequency of unipolar modulation
  • the switching frequency of the bipolar modulation is unipolar modulation switching Double the frequency.
  • the frequency of bipolar modulation can be set to several times the switching frequency of unipolar modulation according to the current ripple of the first power inductor La1.
  • the possible range is 15 to 3 times between.
  • This modulation method can achieve a smooth transition between unipolar modulation and bipolar modulation, and the filter inductor current will not be distorted, thereby suppressing circuit oscillation and electromagnetic compatibility (EMC) problems caused by current distortion.
  • EMC electromagnetic compatibility
  • the above DC offset can be any value.
  • the zero point is used as the DC offset for the description of the experimental measurement scenario, that is, different DC offsets can be obtained according to different experimental scenarios, which is not limited here.
  • the first sine modulated wave or the second sine modulated wave set at the DC bias can be symmetrical, and it can be one or more in number. The specific number depends on the actual scene. Here Not limited.
  • the switching of unipolar modulation and bipolar modulation can be realized by adjusting the counting frequency of the carrier and the count change of the counter.
  • ⁇ and ⁇ are adjusted according to state information, that is, ⁇ and ⁇ may be the same or different, and the state information includes the voltage of the positive bus, the voltage of the negative bus, and the voltage of the AC side circuit.
  • ⁇ can be set to 30° and ⁇ to 30°.
  • the waveform shown in FIG. 13 can be measured.
  • FIG. 13 it is a kind of Comparing the waveform diagram of the hybrid modulation circuit with the results shown in Figure 11, it can be found that the amplitude of the bus current at the zero crossing of the AC voltage drops significantly without any saturation problem, that is, the same common mode inductance is under unipolar modulation
  • the normal operation requirements of the inverter cannot be met, but the normal operation of the inverter can be guaranteed under the control method proposed in this application.
  • the efficiency of the inverter can be significantly improved, the volume of passive devices can be reduced, and the effective control of common-mode leakage current can be realized at the same time.
  • FIG. 14 is a comparison diagram of converter efficiency under different control modes provided by the embodiments of the present application.
  • unipolar modulation is at the zero crossing of power frequency voltage.
  • the common-mode voltage is serious, which causes extremely serious high-frequency loss of the common-mode inductor under low load, and the efficiency is low.
  • the output power increases, the advantage of halving the switching loss of the unipolar modulation semiconductor is reflected, and the inverter efficiency increases.
  • the hybrid modulation control strategy provided by the embodiments of the present application can make full use of the advantages of unipolar modulation and bipolar modulation, and on the one hand, it can reduce semiconductor loss. On the other hand, reduce the required common mode inductance.
  • the data diagram shown in FIG. 15 can be obtained.
  • FIG. 15 it is an inverter voltage and common mode inductance provided by an embodiment of the present application. From the waveform diagram of the current, it can be clearly seen that there is no high-frequency ripple current on the common-mode inductor winding of the circuit proposed in this application, so that the high-frequency loss on the common-mode inductor can be reduced, thereby improving the performance of the inverter;
  • the first power inductor La1 and the second power inductor La2 of the circuit structure proposed in this application can be composed of two windings to provide common mode impedance, reduce the influence of the inverter common mode voltage, and reduce the impact on the common mode inductance. demand.
  • the inverter provided in the embodiment of the present application refers to a device including the power conversion circuit described above, and can be understood with reference to the foregoing description of the power conversion circuit.
  • the program can be stored in a computer-readable storage medium, and the storage medium can include: U disk, mobile hard disk, read-only memory (read-only memory, ROM), random access memory (random access memory, RAM), magnetic disk or optical disk and other media that can store program codes.

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Abstract

一种功率变换电路、逆变器以及控制方法，该功率变换电路包括开关网络（702）、控制电路（703）、滤波电路（704）和直流侧电路（701）；滤波电路（704）包括第一功率电感、共模电感、第一差模滤波电容（Cdm）、第一共模滤波电容（Ccm1）和第二共模滤波电容（Ccm2）；第一功率电感包括第一绕组（La11）和第二绕组（La12），共模电感包括第三绕组（Lcm11）和第四绕组（Lcm12）；第一绕组（La11）和第二绕组（La12）的第一端都分别与开关网络（702）相连，第一绕组（La11）和第二绕组（La12）的第二端分别与第三绕组（Lcm11）和第四绕组（Lcm12）的第一端连接，第一差模滤波电容（Cdm）的两端分别连接到第一绕组（La11）和第二绕组（La12）的第二端；第一共模滤波电容（Ccm1）和第二共模滤波电容（Ccm2）各有一端通过低阻抗电路连接到直流侧电路（701）。可以提高逆变器的转换效率，可以有效减少共模漏电流。

Description

一种功率变换电路、逆变器及控制方法
本申请要求于2019年4月17日提交中国专利局、申请号为201910310890.9、发明名称为“一种功率变换电路、逆变器及控制方法”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本申请涉及光伏技术领域，尤其涉及一种功率变换电路、逆变器及混合调制的控制方法。
背景技术
在光伏并网发电系统中，逆变器作为其中的关键设备，其变换效率和性能直接决定了发电系统的收益。
为了降低逆变器的成本，当前的逆变器多采用非隔离架构，通过双级继电器实现光伏的直接并网发电。而单相电网系统中，通常电网的一端通过输电线在远端接地，此时光伏板、逆变器和电网通过大地形成通流回路，该回路会导致严重的漏电流问题。
该漏电流一方面会导致人身安全问题，另一方面会增加逆变器损耗，降低逆变器的转换效率。
发明内容
本申请实施例公开了一种功率变换电路，可以提高逆变器的转换效率，可以有效减少共模漏电流。本申请实施例还提供了相应的逆变器以及混合调制的控制方法。
本申请第一方面提供一种功率变换电路，可以包括：
开关网络、控制电路、滤波电路、直流侧电路和交流侧电路；
所述开关网络与所述直流侧电路相连，所述开关网络与所述控制电路相连，所述开关网络与所述滤波电路相连，所述滤波电路与所述交流侧电路相连；
所述控制电路用于控制所述开关网络；
所述滤波电路包括第一功率电感、共模电感、第一差模滤波电容、第一共模滤波电容和第二共模滤波电容；
所述第一功率电感包括第一绕组和第二绕组，所述共模电感包括第三绕组和第四绕组；
所述第一绕组的第一端和所述第二绕组的第一端都分别与所述开关网络相连，所述第一绕组的第二端与所述第三绕组的第一端连接，所述第二绕组的第二端与所述第四绕组的第一端连接；
所述第一差模滤波电容的两端分别连接到所述第一绕组和所述第二绕组的第二端；
所述第一共模滤波电容的第一端连接到所述第三绕组的第二端，所述第一共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接；
所述第二共模滤波电容的第一端连接到所述四绕组的第二端，所述第二共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接。
所述第一共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接中的低阻抗电路可以称为第一低阻抗电路，第二共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接中的低阻抗电路可以称为第二低阻抗电路。该第一低阻抗电路与第二低阻抗电路可 以是同一低阻抗电路，也可以是不同的低阻抗电路。
由上述第一方面可知，第一差模滤波电容在滤波电路上连接在共模电感之前，可以有效避免高频电流分量流入到共模电感而造成的损耗，从而提高了功率变换电路的转换效率，另外第一共模滤波电容和第二共模滤波电容的第二端都连接到直流侧电路，可以提供共模电流低阻抗回路，从而有效减少功率变换电路对输出端口的共模漏电流。
一种可能的实现方式，结合第一方面，在第一种可能的实现方式中，所述低阻抗电路为零阻抗电路，或所述低阻抗电路中包括一个电阻或至少两个串联电阻。
一种可能的实现方式，结合第一方面或第一方面第一种可能的实现方式，在第二种可能的实现方式中，所述滤波电路还包括第二功率电感，所述第二功率电感包括第五绕组和第六绕组，所述第三绕组的第二端与所述第五绕组的第一端连接，所述第四绕组的第二端与所述第六绕组的第一端连接，所述第五绕组的第二端和所述第六绕组的第二端与所述交流侧电路连接。
一种可能的实现方式，结合第一方面或第一方面第一种或第二种可能的实现方式，在第三种可能的实现方式中，所述直流侧电路包括正母线、母线电容和负母线，所述母线电容的两端分别与所述正母线和所述负母线连接。
一种可能的实现方式，结合第一方面第三种可能的实现方式，在第四种可能的实现方式中，所述第一共模滤波电容的第二端和所述第二共模滤波电容的第二端都分别连接到所述正母线。
一种可能的实现方式，结合第一方面第三种可能的实现方式，在第五种可能的实现方式中，所述第一共模滤波电容的第二端和所述第二共模滤波电容的第二端都分别连接到所述负母线。
一种可能的实现方式，结合第一方面第三种可能的实现方式，在第六种可能的实现方式中，所述第一共模滤波电容的第二端连接到所述正母线，所述第二共模滤波电容的第二端连接到所述负母线。
一种可能的实现方式，结合第一方面第三种可能的实现方式，在第七种可能的实现方式中，所述母线电容包括正母线电容和负母线电容，所述正母线电容的第一端与所述正母线连接，所述正母线电容的第二端与所述负母线电容的第一端连接，所述负母线电容的第二端与所述负母线连接；所述第一共模滤波电容的第二端和所述第二共模滤波电容的第二端连接到所述正母线电容和所述负母线电容的中点。
一种可能的实现方式，结合第一方面、第一方面第一种至第七种中任一种可能的实现方式，在第八种可能的实现方式中，
所述开关网络包括第一换流桥臂和第二换流桥臂，所述第一换流桥臂包括第一开关器件和第二开关器件，所述第二换流桥臂包括第三开关器件和第四开关器件；
所述控制电路通过第一正弦调制波和第一载波控制所述第一换流桥臂的所述第一开关器件和所述第二开关器件；
所述控制电路通过第二正弦调制波和第二载波控制所述第二换流桥臂的所述第三开关器件和所述第四开关器件；
其中，所述第一载波和所述第二载波在第一预设角度范围采用双极性调制，在第二预设角度范围采用单极性调制，所述第一预设角度范围基于所述第一正弦调制波或所述第二正弦调制波的直流偏置处设定，所述第二预设角度范围为在所述第一正弦调制波或所述第二正弦调制波的正弦波周期内除所述第一预设角度范围的其他角度，所述双极性调制的开关频率高于单极性调制的开关频率。
上述第八种可能的实现方式中，直流偏置指的是在交流电中存在直流成分，正弦调制波可能在一个直流偏置下，如果直流偏置为0，则是过零点；如果直流偏置不为零，则为直流偏置值。由上述第八种可能的实现方式可知，采用单极性和双极性混合调制的方式，可以进一步提升功率变换电路的转换效率，以及进一步减小漏电流。
一种可能的实现方式，结合第一方面第八种可能的实现方式，在第九种可能的实现方式中，
所述第一预设角度范围包括(-α，β)，所述-α和β的值根据状态信息调整，所述状态信息包括所述正母线的电压、所述负母线的电压、所述交流侧电路的电压。
本申请实施例第二方面提供一种功率变换电路，可以包括：
开关网络、控制电路、滤波电路、直流侧电路和交流侧电路；
所述开关网络与所述直流侧电路相连，所述开关网络与所述控制电路相连，所述开关网络与所述滤波电路相连，所述滤波电路与所述交流侧电路相连；
所述控制电路用于控制所述开关网络；
所述滤波电路包括第一功率电感、共模电感、第一差模滤波电容、第一共模滤波电容和第二共模滤波电容；
所述共模电感包括第三绕组和第四绕组；
所述第一功率电感的第一端与所述开关网络相连，所述第一功率电感的第二端与所述第三绕组的第一端连接，所述第四绕组的第一端与所述开关网络连接；
所述第一差模滤波电容的第一端连接到所述第一功率电感的第二端，所述第一差模滤波电容的第二端连接到所述第三绕组的第一端；
所述第一共模滤波电容的第一端连接到所述第三绕组的第二端，所述第一共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接；
所述第二共模滤波电容的第一端连接到所述四绕组的第二端，所述第二共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接。
所述第一共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接中的低阻抗电路可以称为第一低阻抗电路，第二共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接中的低阻抗电路可以称为第二低阻抗电路。该第一低阻抗电路与第二低阻抗电路可以是同一低阻抗电路，也可以是不同的低阻抗电路。
由上述第二方面可知，第一差模滤波电容在滤波电路上连接在共模电感之前，可以有效避免高频电流分量流入到共模电感而造成的损耗，从而提高了提高功率变换电路的转换效率，另外第一共模滤波电容和第二共模滤波电容的第二端都连接到直流侧电路，可以提供共模电流低阻抗回路，从而有效减少功率变换电路对输出端口的共模漏电流。
一种可能的实现方式，结合第二方面，在第一种可能的实现方式中，所述低阻抗电路为零阻抗电路，或所述低阻抗电路中包括一个电阻或至少两个串联电阻。
一种可能的实现方式，结合第二方面或第二方面第一种可能的实现方式，在第二种可能的实现方式中，所述滤波电路还包括第二功率电感，所述第二功率电感包括第五绕组和第六绕组，所述第三绕组的第二端与所述第五绕组的第一端连接，所述第四绕组的第二端与所述第六绕组的第一端连接，所述第五绕组的第二端和所述第六绕组的第二端与所述交流侧电路连接。
一种可能的实现方式，结合第二方面、第二方面第一种或第二种可能的实现方式，在第三种可能的实现方式中，所述直流侧电路包括正母线、母线电容和负母线，所述母线电容的两端分别与所述正母线和所述负母线连接。
一种可能的实现方式，结合第二方面第三种可能的实现方式，在第四种可能的实现方式中，所述第一共模滤波电容的第二端和所述第二共模滤波电容的第二端都分别连接到所述正母线。
一种可能的实现方式，结合第二方面第三种可能的实现方式，在第五种可能的实现方式中，所述第一共模滤波电容的第二端和所述第二共模滤波电容的第二端都分别连接到所述负母线。
一种可能的实现方式，结合第二方面第三种可能的实现方式，在第六种可能的实现方式中，所述第一共模滤波电容的第二端连接到所述正母线，所述第二共模滤波电容的第二端连接到所述负母线。
一种可能的实现方式，结合第二方面第三种可能的实现方式，在第七种可能的实现方式中，所述母线电容包括正母线电容和负母线电容，所述正母线电容的第一端与所述正母线连接，所述正母线电容的第二端与所述负母线电容的第一端连接，所述负母线电容的第二端与所述负母线连接；所述第一共模滤波电容的第二端和所述第二共模滤波电容的第二端连接到所述正母线电容和所述负母线电容的中点。
一种可能的实现方式，结合第二方面、第二方面第一种至第七种中任一种可能的实现方式，在第八种可能的实现方式中，
所述开关网络包括第一换流桥臂和第二换流桥臂，所述第一换流桥臂包括第一开关器件和第二开关器件，所述第二换流桥臂包括第三开关器件和第四开关器件；
所述控制电路通过第一正弦调制波和第一载波控制所述第一换流桥臂的所述第一开关器件和所述第二开关器件；
所述控制电路通过第二正弦调制波和第二载波控制所述第二换流桥臂的所述第三开关器件和所述第四开关器件；
其中，所述第一载波和所述第二载波在第一预设角度范围采用双极性调制，在第二预设角度范围采用单极性调制，所述第一预设角度范围基于所述第一正弦调制波或所述第二正弦调制波的直流偏置处设定，所述第二预设角度范围为在所述第一正弦调制波或所述第二正弦调制波的正弦波周期内除所述第一预设角度范围的其他角度，所述双极性调制的开关频率高于单极性调制的开关频率。
上述第八种可能的实现方式中，直流偏置指的是在交流电中存在直流成分，正弦调制波可能在一个直流偏置下，如果直流偏置为0，则是过零点；如果直流偏置不为零，则为直流偏置值。由上述第八种可能的实现方式可知，采用单极性和双极性混合调制的方式，可以进一步提升功率变换电路的转换效率，以及进一步减小漏电流。
一种可能的实现方式，结合第二方面第八种可能的实现方式，在第九种可能的实现方式中，
所述第一预设角度范围包括(-α，β)，所述-α和β的值根据状态信息调整，所述状态信息包括所述正母线的电压、所述负母线的电压、所述交流侧电路的电压。
本申请第三方面提供功率变换电路，可以包括：
开关网络、控制电路、滤波电路、直流侧电路和交流侧电路；
所述开关网络与所述直流侧电路相连，所述开关网络与所述控制电路相连，所述开关网络与所述滤波电路相连，所述滤波电路与所述交流侧电路相连；
所述控制电路用于控制所述开关网络；
所述滤波电路包括第三功率电感和第四功率电感、共模电感、第一差模滤波电容、第一共模滤波电容和第二共模滤波电容；
所述共模电感包括第三绕组和第四绕组；
所述第三功率电感的第一端和所述第四功率电感的第一端都分别与所述开关网络相连，所述第三功率电感的第二端与所述第三绕组的第一端连接，所述第四功率电感的第二端与所述第四绕组的第一端连接；
所述第一差模滤波电容的两端分别连接到所述第三功率电感和所述第四功率电感的第二端；
所述第一共模滤波电容的第一端连接到所述第三绕组的第二端，所述第一共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接；
所述第二共模滤波电容的第一端连接到所述四绕组的第二端，所述第二共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接。
所述第一共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接中的低阻抗电路可以称为第一低阻抗电路，第二共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接中的低阻抗电路可以称为第二低阻抗电路。该第一低阻抗电路与第二低阻抗电路可以是同一低阻抗电路，也可以是不同的低阻抗电路。
由上述第三方面可知，第一差模滤波电容在滤波电路上连接在共模电感之前，可以有效避免高频电流分量流入到共模电感而造成的损耗，从而提高了提高功率变换电路的转换效率，另外第一共模滤波电容和第二共模滤波电容的第二端都连接到直流侧电路，可以提供共模电流低阻抗回路，从而有效减少功率变换电路对输出端口的共模漏电流。
本申请第三方面与第一方面的不同在于是通过第三功率电感和第四功率电感实现第一绕组和第二绕组的功能，实际上，也可以将第一绕组和第二绕组理解为是两个功率电感，第一方面和第三方面所提供的方案的本质还是相同的。
其余第三方面所涉及到的可能的实现方式都可以参阅第一方面的任一可能的实现方式 进行理解。
本申请第四方面提供一种混合调制的控制方法，该方法应用于功率变换电路，所述功率变换电路包括开关网络、控制电路、滤波电路、直流侧电路和交流侧电路；
所述开关网络与所述直流侧电路相连，所述开关网络与所述控制电路相连，所述开关网络与所述滤波电路相连，所述滤波电路与所述交流侧电路相连，所述开关网络包括第一换流桥臂和第二换流桥臂，所述第一换流桥臂包括第一开关器件和第二开关器件，所述第二换流桥臂包括第三开关器件和第四开关器件；
所述方法包括：
所述控制电路通过第一正弦调制波和第一载波控制所述第一换流桥臂的所述第一开关器件和所述第二开关器件；
所述控制电路通过第二正弦调制波和第二载波控制所述第二换流桥臂的所述第三开关器件和所述第四开关器件；
其中，所述第一载波和所述第二载波在第一预设角度范围采用双极性调制，在第二预设角度范围采用单极性调制，所述第一预设角度范围基于所述第一正弦调制波或所述第二正弦调制波的直流偏置处设定，所述第二预设角度范围为在所述第一正弦调制波或所述第二正弦调制波的正弦波周期内除所述第一预设角度范围的其他角度，所述双极性调制的开关频率高于单极性调制的开关频率。
由上述第四方面可知，采用单极性和双极性混合调制的方式，可以提升功率变换电路的转换效率，以及减小漏电流。
结合第四方面，在第一种可能的实现方式中，所述第一预设角度范围包括(-α，β)，所述-α和β的值根据状态信息调整，所述状态信息包括所述正母线的电压、所述负母线的电压、所述交流侧电路的电压。
本申请第五方面提供一种逆变器，包括上述第一方面、第一方面任一可能的实现方式，或者第二方面、第二方面任一可能的实现方式或第三方面、第三方面任一可能的实现方式所述的功率变换电路。
本申请第六方面提供一种光伏发电系统，可以包括：
光伏板、逆变器和交流网络；
所述光伏板与所述逆变器相连，所述逆变器与所述交流网络相连；
所述光伏板用于将所述光能转化为直流电；
所述逆变器包括上述第一方面、第一方面任一可能的实现方式，或者第二方面、第二方面任一可能的实现方式或第三方面、第三方面任一可能的实现方式所述的功率变换电路，用于将所述直流电转换为交流电；
所述交流网络用于传输所述交流电。
本申请的第七方面提供了一种计算机可读存储介质，所述计算机可读存储介质中存储有指令，当其在计算机上运行时，使得计算机执行上述第四方面所述的控制方法。
本申请实施例提供的方案，通过第一差模滤波电容在滤波电路上连接在共模电感之前，可以有效避免高频电流分量流入到共模电感而造成的损耗，从而提高了提高功率变换电路 的转换，另外第一共模滤波电容和第二共模滤波电容的第二端都连接到直流侧电路，可以提供共模电流低阻抗回路，从而有效减少功率变换电路对输出端口的共模漏电流。
附图说明
图1是本申请实施例所应用的一种场景示意图；
图2是本申请实施例所应用的另一种场景示意图；
图3是本申请实施例提供的一种功率变换电路的示意图；
图4是本申请实施例提供的一种功率变换电路的结构图；
图5是本申请实施例提供的另一种功率变换电路的结构图；
图6是本申请实施例提供的另一种功率变换电路的结构图；
图7是本申请实施例提供的另一种功率变换电路的结构图；
图8是本申请实施例提供的另一种功率变换电路的结构图；
图9是本申请实施例提供的另一种功率变换电路的结构图；
图10是本申请实施例提供的另一种功率变换电路的结构图；
图11是本申请实施例提供的一种基于单极性调制的电路波形图；
图12是本申请实施例提供的一种混合调制的控制方法波形示意图；
图13是本申请实施例提供的一种基于混合调制的电路波形图；
图14是本申请实施例提供的不同控制方式下的变流效率对比图；
图15是本申请实施例提供的一种逆变电压和共模电感电流的波形图。
具体实施方式
本申请实施例公开了一种功率变换电路，可以提高逆变器的转换效率，可以有效减少共模漏电流。本申请实施例还提供了相应的逆变器以及混合调制的控制方法。
为了使本技术领域的人员更好地理解本申请方案，下面将结合本申请实施例中的附图，对本申请实施例中的技术方案进行描述，显然，所描述的实施例仅仅是本申请一部分的实施例，而不是全部的实施例。基于本申请中的实施例，都应当属于本申请保护的范围。
本申请的说明书和权利要求书及上述附图中的术语“第一”、“第二”等是用于区别类似的对象，而不必用于描述特定的顺序或先后次序。应该理解这样使用的数据在适当情况下可以互换，以便这里描述的实施例能够以除了在这里图示或描述的内容以外的顺序实施。此外，术语“包括”和“具有”以及他们的任何变形，意图在于覆盖不排他的包含，例如，包含了一系列步骤或单元的过程、方法、系统、产品或设备不必限于清楚地列出的那些步骤或单元，而是可包括没有清楚地列出的或对于这些过程、方法、产品或设备固有的其它步骤或单元。
本申请提出一种功率变换电路、逆变器及混合调制的控制方法。该功率变换电路、逆变器及混合调制的控制方法可应用于如图1所示的场景架构中。
图1是本申请实施例所应用的一种场景示意图。
图1为不间断电源(uninterruptible power system，UPS)供电系统架构，正常情况下市电直接向负载供电，同时工频交流市电可以经过直流(direct current，DC)/交流(alternating current，AC)逆变器将交流电转换为直流电，并通过DC/DC逆变器对电池 充电；当市电发生故障时，电池经过DC/DC逆变器和DC/AC逆变器，将直流电转换为交流电，对负载进行供电，其中，本申请提供的功率变换电路及混合调制的控制方法可应用于AC/DC整流器或DC/AC逆变器的运行过程中。
在另一种可能的应用场景中，本申请提出的功率变换电路、逆变器及控制方法可应用于光伏并网发电系统中。
图2是本申请实施例所应用的另一种场景示意图。
如图2所示，光伏板输出的直流电经过DC/AC逆变器，将直流电转换为交流电，实现光伏板的并网发电，其中，本申请提供的功率变换电路及混合调制的控制方法可应用于DC/AC逆变器的运行过程中。
基于上述应用场景，下面以实施例的方式对本申请技术方案做进一步的说明。
图3是本申请实施例提供的一种功率变换电路的示意图。
如图3所示，该功率变换电路可以包括直流侧电路701、开关网络702、控制电路703、滤波电路704和交流侧电路705。
该开关网络702与该直流侧电路701相连，该开关网络702与该控制电路703相连，该开关网络702与该滤波电路704相连，该滤波电路704与该交流侧电路705相连。
该控制电路703用于控制该开关网络702按照预设的调制方案使得从直流侧电路701输入的直流电转换为交流电，该滤波电路704用于滤除开关网络702在调制过程中产生的高频纹波，并将处理后的交流电传输至交流侧电路705。
在一种可能的电路结构中，图3所示的功率变换电路可以采用如图4所示的结构图。
图4是本申请实施例提供的一种功率变换电路的结构图。
图4中所示的直流侧电路包括正母线、负母线和母线电容。
开关网络可以包括第一换流桥臂和第二换流桥臂，第一换流桥臂包括第一开关器件T1和第二开关器件T2，第二换流桥臂包括第三开关器件T3和第四开关器件T4。
可选地，第一开关器件T1、第二开关器件T2、第三开关器件T3和第四开关器件T4可以选用绝缘栅双极型晶体管(insulated gate bipolar transistor，IGBT)、氮化镓(gallium nitride，GaN)、金属氧化物半导体场效应晶体管(metal-oxide-semiconductor field effect transistor，MOSFET)或其他功率半导体器件。
滤波电路可以包括第一功率电感La1、共模电感Lcm、第一差模滤波电容Cdm、第一共模滤波电容Ccm1。一种可能的实现方式中，滤波电路还可以包括第二共模滤波电容Ccm2。
该第一功率电感La1包括第一绕组La11和第二绕组La12，可以理解的是，第一绕组La11和第二绕组La12的线圈数比例可以根据需求调节，在一种可能的场景中，可以采用相同的线圈数，当然具体的线圈数因实际场景而定，本申请实施例中对此不做限定。该共模电感Lcm包括第三绕组Lcm11和第四绕组Lcm12。
该第一绕组La11的第一端和该第二绕组La12的第一端都分别与该开关网络A点相连，该第一绕组La11的第二端与该第三绕组Lcm11的第一端连接，该第二绕组La12的第二端与该第四绕组Lcm12的第一端连接。
该第一差模滤波电容Cdm的第一端连接到第一绕组La11的第二端，该第一差模滤波电 容Cdm的第二端连接到该第二绕组La12的第二端。
该第一共模滤波电容Ccm1的第一端连接到该第三绕组Lcm11的第二端，该第一共模滤波电容Ccm1的第二端通过低阻抗电路与该直流侧电路连接。该实施例中，通过低阻抗电路可以理解为是第一共模滤波电容Ccm1的第二端直接与直流侧电路连接的导线，该场景下，低阻抗电路为零阻抗电路。
该第二共模滤波电容Ccm2的第一端连接到该四绕组Lcm12的第二端，该第二共模滤波电容Ccm2的第二端通过低阻抗电路与该直流侧电路连接。该实施例中，通过低阻抗电路可以理解为是第二共模滤波电容Ccm2的第二端直接与直流侧电路连接的导线，该场景下，低阻抗电路为零阻抗电路。
通过第一共模滤波电容Ccm1和第二共模滤波电容Ccm2将共模漏电流抽回直流侧电路，从而提供共模漏电流低阻抗回路，可显著降低逆变器的对地漏电流；而且图4所示电路结构中第一功率电感La1由两个绕组La11和La12组成，可提供共模阻抗，降低逆变器共模电压的影响，减小所需的共模电感。
需要说明的是，上述方案中第一功率电感La1也可以只是一个绕组，不需要分成第一绕组和第二绕组，这种情况下，所述第一功率电感的第一端与所述开关网络相连，所述第一功率电感的第二端与所述第三绕组的第一端连接，所述第四绕组的第一端与所述开关网络连接；所述第一差模滤波电容的第一端连接到所述第一功率电感的第二端，所述第一差模滤波电容的第二端连接到所述第三绕组的第一端。
可选地，第一功率电感La1也可以由两个独立的功率电感组成，可以参考图5进行理解。
图5是本申请实施例提供的另一种功率变换电路的结构图。
与图4相比，图5所示的实施例中，将图4中的第一功率电感La1替换为了第三功率电感La3和第四功率电感La4，可以理解的是，第三功率电感La3和第四功率电感La4的线圈数可以参照第一绕组La11和第二绕组La12进行选择，也可以根据需求进行调整，具体线圈数因实际场景而定。
可选地，为进一步降低逆变器高频纹波对交流端口的影响，也可以在滤波电路与交流侧电路之间加入第二功率电感La2，以提供阻抗，减小功率变换电路对交流端口的影响，可以参阅图6进行理解。
图6是本申请实施例提供的另一种功率变换电路的结构图。
如图6所示，第二功率电感La2包括第五绕组La21和第六绕组La22，第三绕组Lcm11的第二端与第五绕组La21的第一端连接，第四绕组Lcm12的第二端与第六绕组La22的第一端连接，第五绕组La21的第二端和第六绕组La21的第二端与交流侧电路连接。可以理解的是，第五绕组La21和第六绕组La22的线圈数比例可以根据需求调节，在一种可能的场景中，可以采用相同的线圈数。
基于上述电路可以发现，本申请实施例通过第一共模滤波电容Ccm1和第二共模滤波电容Ccm2将共模漏电流抽回直流侧电路，从而提供共模漏电流低阻抗回路，可显著降低逆变器的对地漏电流。然而，作为共模漏电流抽回目标的直流侧电路包括正母线和负母线，可 以理解的是，共模漏电流抽回目标可以包含以下多种情况，下面结合附图进行说明。
需要注意的是，改变共模漏电流抽回直流侧电路并不影响第一功率电感La1采用单独电感和在滤波电路与交流侧电路之间加入第二功率电感La2的方案，即基于下述方案，均可以与第一功率电感La1采用单独电感和在滤波电路与交流侧电路之间加入第二功率电感La2的方案进行结合或除去。下面以图6的电路为样本进行说明，具体的电路变化因实际场景而定，此处不做限制。
一、共模漏电流抽回负母线。
本实施例中，可参照上述图4-10的结构图，此处不再重复赘述。
二、共模漏电流抽回正母线。
本实施例中，可参照图7所示出的另一种功率变换电路的结构图。如图7所示，第一共模滤波电容Ccm1的第二端与正母线连接，第二共模滤波电容Ccm2的第二端与正母线连接，其余部分描述可参考图4-10中的相关内容进行理解，此处不再重复赘述。
三、共模漏电流抽回正母线和负母线。
本实施例中，可参照图8所示出的另一种功率变换电路的结构图。如图8所示，第一共模滤波电容Ccm1的第二端与正母线连接，第二共模滤波电容Ccm2的第二端与负母线连接，从而给共模电感Lcm的两个绕组提供完全等同的阻抗电路，共模电感不容易出现单边饱和现象。其余部分描述可参考图4-10中的相关内容进行理解，此处不再重复赘述。
四、共模漏电流抽回正母线和负母线的中点。
本实施例中，可参照图9所示出的另一种功率变换电路的结构图。如图9所示，该实施例中母线电容Cdc包括正母线电容Cdcp和负母线电容Cdcn。正母线电容Cdcp的第一端与正母线连接，正母线电容Cdcp的第二端与负母线电容Cdcn的第一端连接，负母线电容Cdcn的第二端与负母线连接，在正母线电容Cdcp与负母线电容Cdcn之间有母线中点M。第一共模滤波电容Ccm1的第二端和第二共模滤波电容Ccm2的第二端连接到母线中点M，从而可将输出共模电容Lcm的中点连接到母线中点M处。该滤波方式同样可提供共模漏电流的低阻抗电路，显著降低逆变器的输出共模漏电流。
另一方面，该滤波方式同样可给共模电感Lcm两个绕组提供完全等同的阻抗回路，共模电感Lcm不容易出现单边饱和现象。其余部分描述可参考图4-10中的相关内容进行理解，此处不再重复赘述。
可选地，在共模漏电流抽回直流侧电路的导线上可以接入一个低阻抗电阻R1作为低阻抗电路，如图10另一种功率变换电路的结构图中所示出的，在母线中点M与滤波电路之间接入一个低阻抗电阻R1，也就是低阻抗电路上有一个电阻R1，该低阻抗电阻R1可提供共模回路中的阻尼，有效抑制共模回路中的振荡，且低阻抗电阻R1可根据实测情况进行参数调整，以满足不同的电路状况。当然，该实施例中只是以加入一个低阻抗电阻R1为例进行说明，实际上，R1也可以替换为两个或多个串联的电阻。
可以理解的是，上述关于低阻抗电阻R1的接入也可以应用到上述一至四的母线连接点的方案中，具体接入方式参照图10的相关内容，此处不做赘述。
本申请实施例中的控制电路要基于控制策略对开关网络中的开关频率进行控制，将共 模漏电流抽回直流侧电路，从而提供共模漏电流低阻抗电路，可显著降低逆变器的对地漏电流。控制策略可以是基于单极性调制，也可以是基于单极性和双极性混合调制。
在一种基于单极性调制的可能场景中，基于图6所示的电路可以测得如图11所示的波形，图11是本申请实施例提供的一种基于单极性调制的电路波形图。从图11可以看出上述电路基于单极性调制的控制策略虽然可减小输出功率电感，但单极性调制在电压过零处导致极严重的共模电压。该共模电压一方面导致共模电感伏秒高，所需共模电感体积大，成本高；另一方面，该共模电压在共模电感上会导致极严重的高频损耗，显著降低单极性调制的优势。
为充分利用单极性调制的优势，同时减小所提电路中的共模电感体积，本申请实施例提出一种混合调制的控制方法，该控制方法通过控制电路703实现，具体的，该控制电路703通过第一正弦调制波和第一载波控制该第一换流桥臂的该第一开关器件T1和该第二开关器件T2；该控制电路通过第二正弦调制波和第二载波控制该第二换流桥臂的该第三开关器件T3和该第四开关器件T4。
本申请实施例中，可通过图12所示的单极性和双极性混合调制的方式控制开关网络的开关频率。图12是本申请实施例提供的一种混合调制的控制方法波形示意图。
图12所示出的第一载波和该第二载波在第一预设角度范围(-α，β)内采用双极性调制，在第二预设角度范围，也就是360度中除掉(-α，β)的角度范围内采用单极性调制。以一个360度的周期为例，也可以理解为在角度(0～β)，((π-α)～π)，(π～(π+β))，((2π-α)～2π)内，调制方式采用双极性调制；在角度(β～(π-α))和((π+β)～(2π-α))内采用单极性调制。该第一预设角度范围基于该第一正弦调制波或该第二正弦调制波的直流偏置处设定，该第二预设角度范围为在该第一正弦调制波或该第二正弦调制波的正弦波周期内除该第一预设角度范围的其他角度，该双极性调制的开关频率高于单极性调制的开关频率，且双极性调制的开关频率为单极性调制开关频率的两倍，在一些可能的场景中，可根据第一功率电感La1电流纹波情况设定双极性调制的频率为单极性调制开关频率的数倍，可能的范围在15～3倍之间。该调制方式可实现单极性调制和双极性调制的平滑过渡，滤波电感电流不会出现畸变，从而可抑制电流畸变导致的电路振荡和电磁兼容性(electro magnetic compatibility，EMC)问题。
应当注意的是，上述直流偏置处可以为任意值，此处以零点作为直流偏置处为结合实验测量场景说明所用，即根据不同的实验场景可以得到不同的直流偏置，此处不做限定；另外在直流偏置处所设定的第一正弦调制波或该第二正弦调制波可以是对称的，在数量上可以是一个，也可以是多个，具体数量因实际场景而定，此处不做限定。
可选地，在完成当前时刻的α和β的设定，并完成一个周期的控制切换后，可以通过调节载波的计数频率和计数器的计数变化实现单极性调制和双极性调制的切换。
可以理解的是，α和β的值根据状态信息调整，即α和β可以相同，也可以不同，该状态信息包括该正母线的电压、该负母线的电压、该交流侧电路的电压。
在一种可能的场景中，可以设置α为30°，β为30°，此时，可测得如图13所示的波形，如图13所示，是本申请实施例提供的一种基于混合调制的电路波形图，对比图11 所示结果，可发现抽回母线电流在交流电压过零处的幅值明显下降，无任何饱和现象问题，即同样的共模电感在单极性调制下无法满足逆变器正常工作要求，但在本申请所提控制方法下可保证逆变器的正常工作。通过该控制方法与本申请提供的电路结构相结合，可显著提升逆变器效率，降低无源器件体积，同时实现共模漏电流的有效控制。
通过上述控制方法，可以得到如图14所示，是本申请实施例提供的不同控制方式下的变流效率对比图，由于相比于双极性调制，单极性调制在工频电压过零处共模电压严重，在低负载下导致极严重的共模电感高频损耗，效率较低。随着输出功率的增加，单极性调制半导体开关损耗减半的优势得到体现，逆变器效率增加。图14中曲线显示，相比单极性调制和双极性调制，本申请实施例所提供的混合调制控制策略可充分利用单极性调制和双极性调制的优势，一方面降低半导体损耗，另一方面减小所需共模电感。
在一种可能的场景中，基于图6的电路采用上述控制方法可以得到如图15所示的数据图，如图15所示，是本申请实施例提供的一种逆变电压和共模电感电流的波形图，可明显的看出，本申请所提电路的共模电感绕组上无高频纹波电流，从而可减小共模电感上的高频损耗，进而提升逆变器性能；另一方面，本申请所提电路结构第一功率电感La1和第二功率电感La2可以由两个绕组组成，以提供共模阻抗，降低逆变器共模电压的影响，减小对于共模电感的需求。
本申请实施例提供的逆变器指的是包括上述所描述的功率变换电路的设备，可以参阅前述对功率变换电路的描述进行理解。
本领域普通技术人员可以理解上述实施例的各种电路操作的全部或部分步骤是可以通过程序来指令相关的硬件来完成，该程序可以存储于一计算机可读存储介质中，存储介质可以包括：U盘、移动硬盘、只读存储器(read-only memory，ROM)、随机存取存储器(random access memory，RAM)、磁碟或者光盘等各种可以存储程序代码的介质。
以上对本申请实施例所提供的信号放大电路及终端设备进行了详细介绍，本文中应用了具体个例对本申请的原理及实施方式进行了阐述，以上实施例的说明只是用于帮助理解本申请的方法及其核心思想；同时，对于本领域的一般技术人员，依据本申请的思想，在具体实施方式及应用范围上均会有改变之处，综上所述，本说明书内容不应理解为对本申请的限制。

Claims (15)

  1. 一种功率变换电路，其特征在于，包括：
    开关网络、控制电路、滤波电路、直流侧电路和交流侧电路；
    所述开关网络与所述直流侧电路相连，所述开关网络与所述控制电路相连，所述开关网络与所述滤波电路相连，所述滤波电路与所述交流侧电路相连；
    所述控制电路用于控制所述开关网络；
    所述滤波电路包括第一功率电感、共模电感、第一差模滤波电容、第一共模滤波电容和第二共模滤波电容；
    所述第一功率电感包括第一绕组和第二绕组，所述共模电感包括第三绕组和第四绕组；
    所述第一绕组的第一端和所述第二绕组的第一端都分别与所述开关网络相连，所述第一绕组的第二端与所述第三绕组的第一端连接，所述第二绕组的第二端与所述第四绕组的第一端连接；
    所述第一差模滤波电容的两端分别连接到所述第一绕组和所述第二绕组的第二端；
    所述第一共模滤波电容的第一端连接到所述第三绕组的第二端，所述第一共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接；
    所述第二共模滤波电容的第一端连接到所述四绕组的第二端，所述第二共模滤波电容Ccm2的第二端通过低阻抗电路与所述直流侧电路连接。
  2. 根据权利要求1所述的功率变换电路，其特征在于，所述低阻抗电路为零阻抗电路，或所述低阻抗电路中包括一个电阻或至少两个串联电阻。
  3. 根据权利要求1或2所述的功率变换电路，其特征在于，所述滤波电路还包括第二功率电感，所述第二功率电感包括第五绕组和第六绕组，所述第三绕组的第二端与所述第五绕组的第一端连接，所述第四绕组的第二端与所述第六绕组的第一端连接，所述第五绕组的第二端和所述第六绕组的第二端与所述交流侧电路连接。
  4. 根据权利要求1-3任一所述的功率变换电路，其特征在于，所述直流侧电路包括正母线、母线电容和负母线，所述母线电容的两端分别与所述正母线和所述负母线连接。
  5. 根据权利要求4所述的功率变换电路，其特征在于，
    所述第一共模滤波电容的第二端和所述第二共模滤波电容的第二端都分别连接到所述正母线。
  6. 根据权利要求4所述的功率变换电路，其特征在于，
    所述第一共模滤波电容的第二端和所述第二共模滤波电容的第二端都分别连接到所述负母线。
  7. 根据权利要求4所述的功率变换电路，其特征在于，
    所述第一共模滤波电容的第二端连接到所述正母线，所述第二共模滤波电容的第二端连接到所述负母线。
  8. 根据权利要求4所述的功率变换电路，其特征在于，所述母线电容包括正母线电容和负母线电容，所述正母线电容的第一端与所述正母线连接，所述正母线电容的第二端与所述负母线电容的第一端连接，所述负母线电容的第二端与所述负母线连接；
    所述第一共模滤波电容的第二端和所述第二共模滤波电容的第二端连接到所述正母线电容和所述负母线电容的中点。
  9. 根据权利要求1-8任一所述的功率变换电路，其特征在于，
    所述开关网络包括第一换流桥臂和第二换流桥臂，所述第一换流桥臂包括第一开关器件和第二开关器件，所述第二换流桥臂包括第三开关器件和第四开关器件；
    所述控制电路通过第一正弦调制波和第一载波控制所述第一换流桥臂的所述第一开关器件和所述第二开关器件；
    所述控制电路通过第二正弦调制波和第二载波控制所述第二换流桥臂的所述第三开关器件和所述第四开关器件；
    其中，所述第一载波和所述第二载波在第一预设角度范围采用双极性调制，在第二预设角度范围采用单极性调制，所述第一预设角度范围基于所述第一正弦调制波或所述第二正弦调制波的直流偏置处设定，所述第二预设角度范围为在所述第一正弦调制波或所述第二正弦调制波的正弦波周期内除所述第一预设角度范围的其他角度，所述双极性调制的开关频率高于单极性调制的开关频率。
  10. 根据权利要求9所述的功率变换电路，其特征在于，所述第一预设角度范围包括(-α，β)，所述-α和β的值根据状态信息调整，所述状态信息包括所述正母线的电压、所述负母线的电压、所述交流侧电路的电压。
  11. 一种功率变换电路，其特征在于，包括：
    开关网络、控制电路、滤波电路、直流侧电路和交流侧电路；
    所述开关网络与所述直流侧电路相连，所述开关网络与所述控制电路相连，所述开关网络与所述滤波电路相连，所述滤波电路与所述交流侧电路相连；
    所述控制电路用于控制所述开关网络；
    所述滤波电路包括第三功率电感和第四功率电感、共模电感、第一差模滤波电容Cdm、第一共模滤波电容和第二共模滤波电容；
    所述共模电感包括第三绕组和第四绕组；
    所述第三功率电感的第一端和所述第四功率电感的第一端都分别与所述开关网络相连，所述第三功率电感的第二端与所述第三绕组的第一端连接，所述第四功率电感的第二端与所述第四绕组的第一端连接；
    所述第一差模滤波电容的两端分别连接到所述第三功率电感La3和所述第四功率电感的第二端；
    所述第一共模滤波电容的第一端连接到所述第三绕组的第二端，所述第一共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接；
    所述第二共模滤波电容的第一端连接到所述四绕组的第二端，所述第二共模滤波电容的第二端通过低阻抗电路与所述直流侧电路连接。
  12. 一种混合调制的控制方法，其特征在于，所述方法应用于功率变换电路，所述功率变换电路包括开关网络、控制电路、滤波电路、直流侧电路和交流侧电路；
    所述开关网络与所述直流侧电路相连，所述开关网络与所述控制电路相连，所述开关 网络与所述滤波电路相连，所述滤波电路与所述交流侧电路相连，所述开关网络包括第一换流桥臂和第二换流桥臂，所述第一换流桥臂包括第一开关器件和第二开关器件，所述第二换流桥臂包括第三开关器件和第四开关器件；
    所述方法包括：
    所述控制电路通过第一正弦调制波和第一载波控制所述第一换流桥臂的所述第一开关器件和所述第二开关器件；
    所述控制电路通过第二正弦调制波和第二载波控制所述第二换流桥臂的所述第三开关器件和所述第四开关器件；
    其中，所述第一载波和所述第二载波在第一预设角度范围采用双极性调制，在第二预设角度范围采用单极性调制，所述第一预设角度范围基于所述第一正弦调制波或所述第二正弦调制波的直流偏置处设定，所述第二预设角度范围为在所述第一正弦调制波或所述第二正弦调制波的正弦波周期内除所述第一预设角度范围的其他角度，所述双极性调制的开关频率高于单极性调制的开关频率。
  13. 根据权利要求12所述的控制方法，其特征在于，所述第一预设角度范围包括(-α，β)，所述-α和β的值根据状态信息调整，所述状态信息包括所述正母线的电压、所述负母线的电压、所述交流侧电路的电压。
  14. 一种逆变器，其特征在于，包括权利要求1-11任一项所述的功率变换电路。
  15. 一种光伏发电系统，其特征在于，包括：
    光伏板、逆变器和交流网络；
    所述光伏板与所述逆变器相连，所述逆变器与所述交流网络相连；
    所述光伏板用于将所述光能转化为直流电；
    所述逆变器包括上述权利要求1-11任一项所述的功率变换电路，用于将所述直流电转换为交流电；
    所述交流网络用于传输所述交流电。
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