WO2020110315A1 - Motor driving device - Google Patents

Abstract

This motor driving device (100) is provided with an inverter (1), a controller (3), and a detector (4). The controller (3) is provided with a voltage control unit (50), a coordinate transformation unit (20), a pulsation extraction unit (30), and a phase synchronization calculation unit (40). The controller (3) controls the voltage which is output to a motor (2) by the inverter (1), and performs calculation for estimating a rotational position and a rotational frequency of the motor (2). The coordinate transformation unit (20) converts the motor currents obtained from the detector (4) to two-phase currents (20a, 20b) in a coordinate system at rest. The pulsation extraction unit (30) extracts pulsating currents (30a, 30b) from the two-phase currents (20a, 20b). The phase synchronization calculation unit (40) calculates an estimated pulsation phase (40a) and an estimated pulsation frequency (40b). When this calculation is performed, the voltage control unit (50) calculates and outputs a voltage command value with which any of the motor currents do not become zero.

Description

Motor drive

The present invention relates to a motor drive device that drives a synchronous motor having saliency without a sensor.

-A rotating magnetic field is generated when a multi-phase AC voltage is applied to the stator of a synchronous motor. Torque is generated in the synchronous motor due to the magnetic interaction between the rotating magnetic field and the rotor. The polyphase AC voltage is an AC voltage of three phases or four phases or more. When the synchronous motor rotates, the rotating magnetic field and the rotor need to be synchronized in phase and frequency. Therefore, in order to drive the synchronous motor, information on the rotational position or rotational frequency of the rotor is required. There is a method of using a position sensor or a speed sensor to obtain information on the rotational position or rotational frequency of the rotor. On the other hand, in order to reduce the number of parts and the number of wirings, the application of a drive system that does not use these sensors is expanding. A state and a driving method that do not include a sensor for acquiring information on the rotational position or the rotational frequency of the rotor are called “sensorless”.

When starting a synchronous motor without a sensor, it is necessary to start the switching control for the semiconductor elements of the inverter while synchronizing the phase and frequency of the output voltage with the rotation state of the rotor so that overcurrent does not occur. Therefore, in the sensorless drive system of the synchronous motor, it is usual to distinguish the case of starting the synchronous motor from the cases other than the case of starting. Specifically, the sensorless drive system of the synchronous motor is roughly composed of two algorithms called "steady state estimation" and "initial estimation".

▽ The algorithm of steady estimation is applied when the semiconductor element of the inverter that drives the motor is switching operation and the torque or rotation state of the motor is continuously controlled. On the other hand, the initial estimation algorithm is applied when starting the switching operation from the state where the semiconductor element of the inverter stops the switching operation. That is, the steady estimation is started using the information on the rotational position and the rotational frequency of the rotor obtained by the initial estimation.

As an example of the initial estimation, a method of estimating the rotation frequency of the rotor based on the cycle in which the current polarity of the specific phase reverses, and estimating the rotational position of the rotor based on the timing when the current polarity of the specific phase reverses, It is disclosed in Patent Document 1. For the sake of convenience, the inversion of the polarity of the phase current, that is, the sign of the phase current is called “zero cross”.

JP, 2004-336866, A

By the way, in an inverter whose output voltage is controlled by Pulse Width Modulation (PWM), the controller intends because of the ON voltage of the switching element that constitutes the inverter and the dead time when controlling the inverter. It is known that the actual output voltage deviates from the voltage command value. The voltage error, which is the difference between the voltage command value and the actual output voltage, becomes more significant as the current flowing through the switching element is smaller, that is, the motor current is smaller. The motor current is a current flowing through each phase of the motor, that is, a phase current of the motor.

In the initial estimation described in Patent Document 1, since the current of a specific phase is controlled so as to pulsate near zero, the voltage error mentioned above becomes remarkable. If the voltage error is large, the phase currents of the other phases become too small and the frequency estimation accuracy deteriorates. Alternatively, the phase currents of the other phases become excessively large, causing magnetic saturation, which deteriorates the estimation accuracy of the rotational position. The "other phase" here means a phase that is too small or too large with respect to a specific phase. As described above, when the voltage error is large, there arises a problem that the estimation accuracy of the frequency deteriorates or the estimation accuracy of the rotational position deteriorates.

Also, the signals of the voltage sensor and current sensor include noise that enters in the analog circuit. Therefore, the voltage sensor or the current sensor cannot always detect the accurate zero-cross timing.

Also, the number of times the current detection value can be sampled with a digital circuit is limited. In particular, during high-speed rotation of the motor, the pulsating cycle of the motor current becomes short, so that there are many timings at which current detection values cannot be obtained, and the zero-cross timing error becomes more prominent.

The present invention has been made in view of the above, and an object of the present invention is to provide a motor drive device capable of highly accurately estimating information on the rotational position or rotational frequency of a rotating synchronous motor.

In order to solve the problems described above and achieve the object, a motor drive device according to the present invention provides an inverter that drives a synchronous motor having saliency, a controller that controls the operating state of the inverter, and a phase current of the synchronous motor. A detector for detecting is provided. The controller is a voltage control unit that determines the output voltage of the inverter, a coordinate conversion unit that converts a phase current into a two-phase current in a stationary coordinate system, a pulsation extraction unit that extracts a pulsating current from the two-phase current, and the frequency of the pulsating current. And a phase synchronization calculator for estimating and calculating the phase. The voltage control unit outputs a voltage command value in which none of the phase currents becomes zero.

According to the motor drive device of the present invention, it is possible to highly accurately estimate the information on the rotational position or the rotational frequency of the rotating synchronous motor.

Configuration diagram of a motor drive device according to the first embodiment Circuit diagram showing the configuration of the inverter shown in FIG. Schematic diagram for explaining the voltage command value output by the voltage control unit according to the first embodiment. FIG. 3 is a diagram showing the relationship between the phase of the voltage command value vector and the magnitude of phase current in the first embodiment. Block diagram showing the configuration of a pulsation extraction unit according to the second embodiment FIG. 3 is a block diagram showing the configuration of the phase synchronization calculation unit according to the second embodiment. FIG. 3 is a block diagram showing the configuration of the phase synchronization calculation unit according to the third embodiment. FIG. 8 is a diagram for explaining the operation of switching the gain of the amplifier according to the third embodiment. Configuration diagram of a motor drive device according to a fourth embodiment Block diagram showing a configuration of a correction operation unit according to the fourth embodiment FIG. 4 is a block diagram showing an example of a hardware configuration that realizes an arithmetic function in the controller according to the fourth embodiment. The block diagram which shows another example of the hardware constitutions which implement|achieve the arithmetic function in the controller of Embodiment 4.

A motor drive device according to an embodiment of the present invention will be described in detail below with reference to the accompanying drawings. The present invention is not limited to the embodiments described below.

Embodiment 1.
FIG. 1 is a configuration diagram of a motor drive device according to the first embodiment. FIG. 2 is a circuit diagram showing a configuration of the inverter shown in FIG. 1, a motor drive device 100 includes a motor 2 having a rotor 2a, an inverter 1 for driving the motor 2, a controller 3 for controlling an operating state of the inverter 1, and a detector for detecting a phase current of the motor 2. 4 and.

The inverter 1 is supplied with DC power from the power source 110 and applies a voltage of variable amplitude and variable frequency to the motor 2. The inverter 1 adjusts the voltage applied to the motor 2 by pulse width modulation (PWM) control for a plurality of semiconductor elements (not shown in FIG. 1).

The motor 2 is a synchronous motor having saliency. An example of a salient-polarity synchronous motor is a synchronous reluctance motor (Synchronous Reluctance Motor: hereinafter referred to as “SynRM”). The SynRM rotor has a characteristic that the magnetic resistance in the radial direction changes according to the rotation angle with respect to the cylindrical axis. Such a characteristic is called "saliency". When a voltage is applied to the stator of the SynRM and a current flows through the stator, a magnetic field that radially intersects the circumference of the rotor is generated. At this time, torque is generated to rotate the rotor in the direction in which the magnetic flux increases, that is, the magnetic resistance in the magnetic path decreases. Thus, the torque generated due to the salient pole of the rotor is called reluctance torque.

FIG. 2 shows the circuit configuration when the inverter 1 is a three-phase inverter. In the inverter 1 shown in FIG. 2, the leg 10A in which the semiconductor element UP of the upper arm and the semiconductor element UN of the lower arm are connected in series, the semiconductor element VP of the upper arm and the semiconductor element VN of the lower arm are connected in series. The connected leg 10B and the leg 10C in which the upper arm semiconductor element WP and the lower arm semiconductor element WN are connected in series are provided. The leg 10A, the leg 10B, and the leg 10C are connected in parallel with each other.

A bus voltage is applied to the inverter 1 through the DC buses 15a and 15b. The inverter 1 drives the motor 2 by converting the DC power of the power source 110 supplied through the DC buses 15 a and 15 b into AC power and supplying the converted AC power to the motor 2.

FIG. 2 exemplifies a case where the semiconductor elements UP, UN, VP, VN, WP, WN are metal oxide semiconductor field effect transistors (Metal-Oxide-Semiconductor Field-Effect Transistor: MOSFET). The semiconductor element UP includes a transistor 10a and a diode 10b connected in antiparallel with the transistor 10a. The other semiconductor elements UN, VP, VN, WP, WN have the same configuration. The anti-parallel means that the anode side of the diode is connected to the first terminal corresponding to the source of the MOSFET and the cathode side of the diode is connected to the second terminal corresponding to the drain of the MOSFET.

The semiconductor elements UP, UN, VP, VN, WP, WN may be replaced by MOSFETs, for example, insulated gate bipolar transistors (Insulated Gate Bipolar Transistors: IGBT) may be used.

Also, although FIG. 2 shows a configuration including three legs in which the semiconductor element of the upper arm and the semiconductor element of the lower arm are connected in series, the configuration is not limited to this. The number of legs may be four or more. Further, in FIG. 2, one leg is composed of one upper and lower arm semiconductor element, but one leg may be composed of a plurality of pairs of upper and lower arm semiconductor elements.

When the transistor 10a of each of the semiconductor elements UP, UN, VP, VN, WP, WN is a MOSFET, at least one of the semiconductor elements UP, UN, VP, VN, WP, WN is made of silicon carbide or gallium nitride. It may be formed of a wide band gap semiconductor such as a system material or diamond.

Wide bandgap semiconductors generally have higher voltage resistance and heat resistance than silicon semiconductors. Therefore, if at least one of the semiconductor elements UP, UN, VP, VN, WP, WN is formed of a wide bandgap semiconductor MOSFET, it is possible to obtain the effects of withstand voltage and heat resistance.

A connection point 12 between the upper arm semiconductor element UP and the lower arm semiconductor element UN is connected to the first phase (for example, U phase) of the motor 2, and the upper arm semiconductor element VP and the lower arm semiconductor element VN are connected. The connection point 13 is connected to the second phase (for example, the V phase) of the motor 2, and the connection point 14 between the upper arm semiconductor element WP and the lower arm for the semiconductor element WN is the third phase (for example, the W phase) of the motor 2. )It is connected to the. In the inverter 1, the connection points 12, 13 and 14 form AC terminals.

Returning to FIG. 1, the description of the motor drive device 100 will be continued. The controller 3 includes a voltage control unit 50 that determines the output voltage of the inverter 1, a coordinate conversion unit 20 that converts a phase current into a two-phase current in a stationary coordinate system, and a pulsation extraction unit that extracts a pulsating current from the two-phase current. 30 and a phase synchronization calculation unit 40 that estimates and calculates the frequency and phase of the pulsating current.

The voltage control unit 50 calculates a voltage command value that is a command value of the voltage that the inverter 1 should output. The switching states of the semiconductor elements UP, UN, VP, VN, WP, WN of the inverter 1 are determined based on the voltage command value. Here, the voltage command value for setting the torque of the motor 2 to a desired value needs to be calculated based on the rotation position and the rotation frequency of the rotor 2a.

There is a method of using a position sensor or a speed sensor to obtain information on the rotation position or rotation frequency of the rotor 2a. However, these sensors are often installed coaxially with the motor, and if the installation space of the motor is restricted, the allowable axial length of the motor will be reduced. Therefore, the method using the position sensor or the speed sensor has a disadvantage that the motor output is limited as a result. Further, in the position sensor or the speed sensor, the signal line of the sensor needs to be wired to the hardware in which the controller 3 is mounted. Therefore, the method of using the position sensor or the speed sensor has a problem that the cost of parts increases and there is a risk of disconnection. For these reasons, the application of the sensorless drive system, which is a drive system that does not use the position sensor and the speed sensor, is expanding. The motor drive device 100 according to the first embodiment is also premised on driving the motor without a sensor.

When starting the motor 2 without a sensor, it is necessary to start the switching control of the inverter 1 after synchronizing the phase and frequency of the output voltage with the rotation state of the rotor 2a so that overcurrent does not occur. From this viewpoint, the sensorless drive system of the synchronous motor is divided into two algorithms, that is, steady estimation and initial estimation, as described above.

As described above, the steady estimation is an algorithm applied when the inverter 1 starts switching and the torque or rotation frequency of the motor 2 is continuously controlled.

One of the typical methods for steady estimation is to use the induced voltage of the motor 2 (also called "back electromotive force"). In this method, the induced voltage is calculated based on the mathematical model of the motor, and the position between the coordinate axis of the true dq coordinate system corresponding to the position of the rotor 2a and the coordinate axis of the estimated dq coordinate system where the voltage command value is calculated is calculated. Define the phase difference. Then, the estimated dq coordinate system is corrected so as to eliminate this phase difference, and as a result, the estimated values of the position and the rotation frequency of the rotor 2a are obtained. The dq coordinate system represents a rotating coordinate system when the motor 2 is vector-controlled, and is a widely known concept.

Another method of steady estimation is to apply a high frequency voltage to the motor 2 and use the current response at that time. In this method, when a high-frequency voltage is applied to the motor 2 having saliency, the current on the dq coordinates has an elliptical locus, which is a method of obtaining the estimated values of the rotational position and the rotational frequency of the rotor 2a. .. This method is often used under low speed operating conditions where the induced voltage is small.

Also, as described above, the initial estimation is an algorithm applied when switching is started from the state in which the inverter 1 has stopped switching. As described above, the steady estimation algorithm requires an initial value of one or both of the information on the rotational position and the rotational frequency of the rotor 2a when starting the calculation. If energization is started in a state where the difference between the initial value and the true value is large, inconveniences such as overcurrent may occur. Therefore, steady estimation is started using the information on the rotational position and the rotational frequency of the rotor 2a obtained by the initial estimation. In this way, the initial estimation algorithm is executed for a short time when the inverter 1 is started.

The operation of the voltage control unit 50 shown in FIG. 1 will be described below. In the first embodiment, a method of initial estimation will be described in detail. The method of steady estimation is not particularly limited.

When the voltage control unit 50 receives a command to start the inverter 1 from a higher-order control system (not shown), the voltage control unit 50 calculates a voltage command value in which none of the motor currents of the respective phases becomes zero. Furthermore, the voltage control unit 50 generates a voltage command value that is a DC voltage and has a voltage vector in the same direction or in the opposite direction to any phase of the motor 2. The reason for generating such a voltage command value will be described later.

FIG. 3 is a schematic diagram for explaining the voltage command value output by the voltage control unit 50 according to the first embodiment. Here, the motor 2 is a three-phase motor. Each phase of the three-phase motor is described as u phase, v phase and w phase. The u-phase, v-phase, and w-phase form a three-phase coordinate system. The uvw three-phase coordinate system is a stationary coordinate system. In FIG. 3, the voltage command value vector has the same direction as the u-phase as an example, but the present invention is not limited to this. The voltage command value vector may be in the same direction as the v phase or the w phase.

First, the average value of the u-phase current is represented by i _u0 , the average value of the v-phase current is represented by i _v0 , and the average value of the w-phase current is represented by i _w0 . Then, the inverter 1 outputs the voltage in accordance with the voltage command value shown in FIG. Then, the average value i _v0 of the v-phase current and the average value i _w0 of the w-phase current are opposite in _sign to the average value i _u0 of the u-phase current and have a magnitude of 1/2.

Moreover, the arrows of α and β in FIG. 3 are the coordinate axes when the voltage and current are converted into three phases and two phases. That is, α and β form a two-phase coordinate system. The αβ two-phase coordinate system is a stationary coordinate system like the uvw three-phase coordinate system. The conversion matrix from the uvw three-phase coordinate system to the αβ two-phase coordinate system is given by the following equation.

Although the transformation matrix differs from the above equation (1) depending on how the coordinate axes are defined, it is general that the α axis is defined so as to coincide with any of the uvw axes. When the αβ axis is defined as shown in FIG. 3 and the above equation (1), the voltage command value should be such that v _α is non-zero and v _β is zero. It should be noted that the dq axes shown in FIG. 3 are obtained by rotating the αβ two-phase coordinate system by the rotation angle θ of the rotor 2a.

By the way, when a DC voltage is applied to the rotating motor stator, pulsation occurs in the phase current due to the saliency. The nature of this pulsating current will be described in detail below.

First, the voltage equation of SynRM in the αβ coordinate system is expressed by the following equation.

In the above formula (2), v _α , v _β and i _α , i _β represent the two-phase converted voltage and current, respectively. Further, P represents a differential operator, and R _s represents winding resistance. Further, L _α , L _β and L _αβ are defined by the following equations.

In the above formula (3), L _α is the α-axis inductance, L _β is the β-axis inductance, and L _αβ is the αβ-axis mutual inductance. Further, θ is the rotation angle of the rotor 2a, L ₀ is the average inductance, L ₁ is the differential inductance, L _d is the d-axis inductance, and L _q is the q-axis inductance.

In the motor having the saliency, the d-axis inductance L _d and the q-axis inductance L _q are different, so that the differential inductance L ₁ becomes non-zero from the fifth equation of the above equation (3). Therefore, as shown in the first and second equations of the above equation (3), the α-axis inductance L _α and the β-axis inductance L _β change according to the rotation angle θ of the rotor 2a.

Further, the differential operator P in the equation (2) is applied to both the α-axis inductance L _α , the β-axis inductance L _β and the αβ-axis mutual inductance L _αβ , and the α-axis current i _α and the β-axis current i _β. To work. Therefore, when the term of the differential operator in the above equation (2) is expanded, the following equation is obtained.

In the above formula (4), ω represents the rotation frequency of the rotor 2a, and ω=Pθ. The energized state in FIG. 3 is equivalent to applying a clockwise rotating magnetic field to the stationary motor when viewed from above the rotor 2a. Thus, alpha-axis current i _alpha and beta pulsation phase axis current i _beta is towards the beta-axis current i _beta than alpha-axis current i _alpha is 90 ° advanced phase.

When viewed on the stator, the saliency of the inductance equally affects both of the αβ axes, so that the pulsation amplitude of the α-axis current i _α is equal to the pulsation amplitude of the β-axis current i _β . Based on these, the α-axis current i _α and the β-axis current i _β are divided into an average value component (i _α0 , i _β0 ) and a pulsating current component (i _α1 , i _β1 ) as shown in the following equation. Represented by.

In the above formula (5), φ represents an unknown phase angle and Δi represents the amplitude of the pulsating current. Here, the average values i _α0 and i _β0 of the two-phase current can be obtained by solving the equation by ignoring all the terms relating to the inductance component in the equation (2). Since the β-axis voltage v _β is zero, as described above, the α-axis current average value i _α0 and the β-axis current average value i _β0 are i _α0 =v _α /Rs, i _β0 =0. Become.

Also, by substituting the above equation (5) into the above equation (4), the following equation is obtained.

Further, by rearranging the formula in the first line of the above formula (6), the following formula is obtained.

However, in the above formula (7), δ in the formula is set as in the following formula.

From the condition that equation (7) is the same, the amplitude Δi of the pulsating current and the unknown phase angle φ are obtained as in the following equation and the following equations.

Here, in the above equation (9), the resistance component is sufficiently smaller than the reactance component. Therefore, the above equation (8) can be approximated to δ=π/2, and the above equation (10) can be approximated to φ=π/2. At this time, the α-axis pulsating current i _α1 and the β-axis pulsating current i _β1 can be transformed into the following equation.

The above equation (11) indicates that if the pulsating current is extracted from the two-phase current values on the stationary coordinate system and the phase of those current values is calculated, the rotational position of the rotor 2a can be estimated as a result. ..

Specifically, the controller 3 of the first embodiment shown in FIG. 1 performs the following operations. First, the coordinate conversion unit 20 converts the phase current acquired from the detector 4 into an α-axis current 20a and a β-axis current 20b, which are two-phase currents on the αβ two-phase coordinate system, and outputs them. As the conversion equation at this time, for example, the above equation (1) is used.

Next, the pulsation extraction unit 30 extracts the α-axis pulsation current 30a and the β-axis pulsation current 30b based on the α-axis current 20a and the β-axis current 20b and outputs them to the phase synchronization calculation unit 40. In the following description, the α-axis pulsating current 30a and the β-axis pulsating current 30b may be collectively referred to simply as “pulsating current”.

Then, the phase synchronization calculation unit 40 calculates and outputs the estimated pulsation phase 40a and the estimated pulsation frequency 40b based on the α-axis pulsation current 30a and the β-axis pulsation current 30b. The estimated pulsation phase 40a and the estimated pulsation frequency 40b calculated by the phase synchronization calculation unit 40 are converted into appropriate values, and the converted values are used in a steady estimation algorithm (not shown).

Next, the advantages of setting the voltage command value to a value that does not make any phase current zero, that is, making the u-phase current, the v-phase current, and the w-phase current all zero, will be described. In the following description, the u-phase current, v-phase current, and w-phase current may be collectively referred to as “three-phase current”.

First, not only the inverter 1 of the first embodiment, but also the PWM signal for controlling the semiconductor elements of the upper and lower arms of the inverter is provided with a pause period for giving an off command to both the semiconductor elements of the upper and lower arms. This rest period is called dead time. The pause period is provided to prevent the short circuit between the DC buses 15a and 15b from occurring.

Also, semiconductor devices have voltage drops due to the physical properties of semiconductor devices. If the semiconductor element is an IGBT, there is a collector-emitter voltage drop called a saturation voltage. If the semiconductor element is a MOSFET, there is a voltage drop due to the resistance between the drain and the source.

　To reduce the voltage error caused by the above factors, the voltage command value is often corrected. These corrections are called dead time correction, on-voltage correction, and the like.

Also, when controlling a semiconductor element, it is necessary to deal with the following characteristics unique to the semiconductor element.

(1) In the small current region where the current flowing through the semiconductor element is small, the switching transition time of the semiconductor element changes in a complicated manner.
(2) In the correction of the voltage command value, since the sign of the voltage correction amount is determined using the polarity of the current, chattering easily occurs. Chattering is an event in which the inversion of the voltage correction amount and the current polarity is repeated at an unexpectedly early cycle.
(3) When a dead zone is provided to prevent chattering, the effect of correction is reduced.

What can be said in common with the above items (1) to (3) is that the smaller the current flowing through the semiconductor element, the more difficult it becomes to correct the voltage command value. That is, the voltage error increases when the first phase current, which is one of the three phase currents, is small. When the voltage error becomes large, one of the phase currents other than the first phase current becomes smaller or larger. As a result, the average value i _α0 of the α-axis current and the average value i _β0 of the β-axis current become too _small or too large.

Here, as is clear from the equation (9), the amplitude Δi of the α-axis pulsating current i _α1 and the β-axis pulsating current i _β1 is proportional to the magnitude of the average value i _α0 of the α-axis current. If any of the three-phase currents is small, the average value i _α0 of the α-axis current becomes too small, and the estimation accuracy of the rotational position and the rotational frequency may deteriorate.

Further, although not modeled in the above formula (2), in SynRM, the magnetic saturation of the magnetic member progresses as the amount of energization increases. Further, in SynRM, the degree of magnetic saturation is significantly different depending on the direction of magnetizing the rotor. For this reason, when the average value i _α0 of the α-axis current and the average value i _β0 of the β-axis current become excessive and the degree of magnetic saturation becomes strong, the detected pulsating current includes harmonics, and the rotational position Also, the estimation accuracy of the rotation frequency deteriorates.

Based on the above points, the voltage control unit 50 according to the first embodiment calculates and outputs a voltage command value such that none of the three-phase currents becomes zero. As a result, it becomes easy to correct the voltage command value for setting the output voltage of the inverter 1 to a desired value. As a result, the electric current of the motor 2 can be controlled to an appropriate level, so that the estimation accuracy of the rotational position and the rotational frequency of the rotor 2a can be improved.

Note that when a voltage is applied to the rotating motor 2, the pulsating current due to the saliency can be observed regardless of the magnitude and phase of the voltage. However, it is desirable that the voltage command value output by the voltage control unit 50 be a DC voltage. The reason is as follows.

For example, assume that the voltage for initial estimation contains an AC component with a frequency f. Then, an AC component of frequency f is also generated in the phase current of the motor. That is, the pulsating component synchronized with the rotation frequency of the motor and the same frequency component as the voltage are mixed in the phase current. If a plurality of frequency components are mixed in the phase current, it becomes difficult to separate them and the accuracy of initial estimation deteriorates. Therefore, it is desirable that the voltage for initial estimation, that is, the voltage command value output by the voltage control unit 50 is a DC voltage.

Next, the advantage of making the direction of the voltage vector of the voltage command value the same as or opposite to the phase of any phase of the motor 2 will be described with reference to FIG. FIG. 4 is a diagram showing the relationship between the phase of the voltage command value vector and the magnitude of the phase current in the first embodiment. The “direction of the voltage vector” may be rephrased as the “phase of the voltage vector” and the “same or opposite direction” may be rephrased as the “same or opposite phase”. In this case, the same or opposite phase means, for example, when the phase is 60 [deg], the "same phase" is 60 [deg], and the "opposite phase" is obtained by adding 180 [deg], It means that it is 240 [deg].

The horizontal axis of FIG. 4 shows the phase of the voltage command value vector with respect to the α axis, and the vertical axis shows the amplitudes of various standardized phase currents. Specifically, the dotted line indicates the average value i _u0 of the u-phase current, the thin solid line indicates the average value i _v0 of the v-phase current, the alternate long and _short dash line indicates the average value i _w0 of the w-phase current, and the thin broken line indicates the average value of α-axis current. i _α0 and the thick broken line represent the average value i _β0 of the β-axis currents, respectively.

Further, the waveform of the thick solid line is a drawing of the waveform portion having the smallest absolute value in the average value (i _u0 , i _v0 , i _w0 ) of each phase current. Here, the phase having the smallest absolute value among the average values (i _u0 , i _v0 , i _w0 ) of the phase currents is defined as the “minimum phase”. The minimum phase current is defined as "minimum phase current".

In the energized state of FIG. 4, the phase of the voltage command vector corresponds to zero. At this time, the minimum phase is the v phase or the w phase, and the absolute value of the minimum phase current is “0.5”. It should be noted that this value of "0.5" is the maximum value that can be taken when the phase of the voltage command value vector is changed, which can be understood from the waveform of the thick solid line in FIG. 4, that is, the waveform of the minimum phase current. ..

As mentioned above, the smaller the phase current, the more difficult it becomes to correct the voltage command value to reduce the voltage error. In FIG. 3, the direction of the voltage command value vector is the same as that of the u-phase in order to “maximize the minimum phase current”. Further, according to the waveform of the minimum phase current in FIG. 4, points at intervals of 0 [deg] to 60 [deg] have the maximum values. That is, according to FIG. 4, it is understood that the voltage command value vector only needs to be oriented in the same direction or in the opposite direction to any one of the u phase, v phase, and w phase.

For example, when the phase of the voltage command value vector is 60 [deg], the voltage command value vector is in the opposite direction to the v phase. Further, for example, when the phase of the voltage command value vector is 120 [deg], the voltage command value vector is in the same direction as the w phase.

However, when the phase of the voltage command value vector is other than zero, the phase angle φ defined by the above equation (5) is different from the value shown by the above equation (10). Therefore, when the phase of the voltage command value vector is other than zero, appropriate correction is necessary in the processing of the phase synchronization calculation unit 40.

As described above, the voltage control unit according to the first embodiment calculates a DC voltage command value such that the phase of the voltage command value vector is in the same or opposite direction to any phase of the motor. Output. This facilitates the correction of the voltage command value for setting the output voltage of the inverter to a desired value. As a result, the motor current can be controlled to an appropriate level, and the estimation accuracy of the rotational position and rotational frequency of the rotor can be improved.

Embodiment 2.
In the second embodiment, detailed configurations and operations of the pulsation extraction unit 30 and the phase synchronization calculation unit 40 illustrated in FIG. 1 will be described.

FIG. 5 is a block diagram showing the configuration of the pulsation extraction unit 30 according to the second embodiment. As shown in FIG. 5, the pulsation extraction unit 30 according to the second embodiment has two high pass filters (HPF) 301 and 302 having the same characteristics. The α-axis current 20a and the β-axis current 20b, which are two-phase currents in the stationary coordinate system, are input to the high- pass filters 301 and 302, respectively.

The time required to remove the DC component contained in the α-axis current 20a and the β-axis current 20b depends on the cutoff frequencies of the high pass filters 301 and 302. The higher the cutoff frequency, the shorter the time required to remove the DC component, and the estimation calculation by the phase synchronization calculation unit 40 described later can be started quickly. However, the frequency of the pulsating current that must be extracted changes in conjunction with the rotation frequency of the rotor 2a. Therefore, if the cutoff frequency is too high, even the amplitude of the pulsating current may be attenuated, and the S/N ratio may decrease. Therefore, caution is required in the design.

Next, the phase synchronization calculation unit 40 will be described. FIG. 6 is a block diagram showing the configuration of the phase synchronization calculation unit 40 according to the second embodiment. The phase synchronization calculation unit 40 according to the second embodiment has a phase error calculation unit 401, an amplifier 402, and an integrator 403, as shown in FIG.

The α-axis pulsating current 30 a (i _α1 ), the β-axis pulsating current 30 b (i _β1 ), and the estimated pulsating phase 40 a (θ^ ₂ ) are input to the phase error calculation unit 401. The estimated pulsating phase 40a is the output of the integrator 403. The notation “θ^ ₂ ”is an alternative notation for the letter “θ” in “θ ₂ ”with a hat symbol “^” added above it. This alternative notation is used herein except for the mathematical formulas that are inserted in the image. The same applies to "ω^ ₂ " described later.

The phase error calculator 401 calculates the phase error 40f (Δθ ₂ ) according to the following equation.

Amplifier 402 outputs the estimated ripple frequency 40b amplifies the phase error _{40f (Δθ 2) (ω ^} 2). As the amplifier 402, it is preferable to use a PI controller that performs proportional integration (PI) as shown in FIG.

The integrator 403 integrates the estimated pulsation frequency 40b and outputs the integrated value as the estimated pulsation phase 40a. The estimated pulsation phase 40a is fed back to the phase error calculation unit 401.

In the above equation (12), when 2θ>θ^ ₂ , Δθ ₂ becomes positive, so the estimated pulsation frequency ω^ ₂ and the estimated pulsation phase θ^ ₂ are corrected in the increasing direction. Conversely, when the 2 [Theta] <theta ^ _2, since [Delta] [theta] ₂ is negative, the estimated pulsating frequency omega ^ ₂ and the estimated pulse phase theta ^ ₂ is modified to decrease direction. Finally, 2θ=θ^ ₂ , and the phase and frequency of the pulsating current are estimated. In this way, the phase-locked arithmetic unit 40 takes the form of a phase-locked loop (PLL).

By the way, when the phase of the voltage command value vector is set to zero as in the first embodiment, as shown in FIG. 4, the average value ( _iβ0 ) of the β-axis current 20b becomes zero. Therefore, at first glance, it may be considered that it is not necessary to use the high-pass filter 302 to extract the β-axis pulsating current 30b (i _β1 ) from the β-axis current 20b. However, if there is a difference in the presence or absence of the filtering process for extracting the pulsating current and the characteristics between the α axis and the β axis, the amplitude and the phase of the extracted pulsating current may be different between the α axis and the β axis. Becomes Therefore, two high pass filters 301 and 302 are required regardless of the phase of the voltage command value vector. Further, it is desirable that both have the same characteristics. In addition, the same characteristic here does not mean that the physical characteristics are completely the same, but means that the physical characteristics are designed and configured with the expectation that they have the same characteristics.

A method of calculating the pulsation frequency from the zero-cross interval and the pulsation phase from the zero-cross timing by using either the α-axis pulsating current 30a or the β-axis pulsating current 30b is also conceivable. However, since the signal from the detector 4 contains noise that enters the circuit of the detector 4, it is not always possible to detect an accurate zero-cross timing. Further, during high-speed rotation of the motor 2, the number of samplings per cycle in which the current pulsates decreases, so the error in the zero-cross timing becomes more significant. Further, since the pulsating current includes low-order harmonics due to the influence of the magnetic saturation of the motor 2 and the spatial harmonics, the relationship between the timing of zero crossing of the pulsating current and the position of the rotor 2a is more complicated. Become.

On the other hand, the phase synchronization calculation unit 40 according to the second embodiment is composed of a circuit corresponding to a PLL including a feedback path. When the circuit corresponding to the PLL is configured as shown in FIG. 6, since the integrator 403 is included in the path until the estimated pulsation phase 40a is obtained, it is unlikely to be affected by noise mixed in the signal path of the detector 4. Further, the estimated pulsation phase 40a is continuously calculated so as to follow the true pulsation phase. Therefore, even if the number of samplings per cycle in which the current pulsates is small, it is easy to correct the error caused by the discretization. Furthermore, even when disturbances such as magnetic saturation and spatial harmonics are mixed in the pulsating current, the estimated pulsating phase converges to the true phase on average.

As described above, the pulsation extraction unit 30 according to the embodiment has two high-pass filters that remove the DC component from the two-phase current in the stationary coordinate system and output the pulsation current. The two high pass filters have the same characteristics. Further, the phase synchronization calculation unit according to the second embodiment includes a phase error calculation unit that calculates a phase error based on the pulsating current and the estimated pulsation phase, an amplifier that amplifies the phase error and outputs the estimated pulsation frequency, And an integrator that integrates the estimated pulsation frequency and outputs it as an estimated pulsation phase. As a result, the influence of various disturbances is reduced, so that the rotational position and the rotational frequency of the rotor can be estimated with high accuracy.

Embodiment 3.
In the third embodiment, switching of operation modes when starting the inverter 1 will be described.

First, as described in the second embodiment, it takes time according to the cutoff frequency of the high-pass filter to remove the DC component from the two-phase current in the stationary coordinate system by the pulsation extraction unit 30.

Moreover, the estimated pulsation frequency does not converge while the DC component remains in the signal input to the phase synchronization calculation unit 40. If the amplifier 402 of the phase synchronization calculation unit 40 and the integrator 403 start calculation while the DC component remains, the estimated pulsation frequency 40b (ω^ ₂ ) and the estimated pulsation phase 40a (θ^). ₂ ) diverges or oscillates to inaccurate values. As a result, the time required for the initial estimation becomes long as a result. Therefore, it is desirable that the calculation of the amplifier 402 and the integrator 403 be started when a required time elapses after the inverter 1 starts energization.

Therefore, the phase synchronization calculation unit 41 according to the third embodiment has the configuration shown in FIG. FIG. 7 is a block diagram showing the configuration of the phase synchronization calculation unit 41 according to the third embodiment. Comparing FIG. 7 with FIG. 6, a gain switching signal 40c for controlling switching of the gain of the amplifier 402 is added to FIG.

In the phase synchronization calculation unit 41 of FIG. 7, the gain representing the amplification factor of the amplifier 402 is set to zero immediately after the inverter 1 starts energization, and after the first time has elapsed since the inverter 1 started energization, The gain of the amplifier 402 is switched to a value other than zero, that is, a value greater than zero. When the gain of the amplifier 402 is zero, the estimated pulsation frequency 40b remains zero no matter what phase error 40f is input to the amplifier 402. As a result, the input to the integrator 403 also becomes zero, and the estimated pulsation phase 40a also remains zero.

The time from the start of energization of the inverter 1 to the first switching of the gain of the amplifier 402 is determined based on the cutoff frequencies of the high pass filters 301 and 302 in the pulsation extraction unit 30. More specifically, when the cutoff frequencies of the high- pass filters 301 and 302 are high, the time required to remove the DC component may be relatively short. Therefore, a relatively short time after the inverter 1 starts energization. In the meantime, the operation of the amplifier 402 can be started. On the other hand, when the cutoff frequencies of the high- pass filters 301 and 302 are low, the time required to remove the DC component becomes relatively long, so that it is necessary to provide a relatively long grace period before starting the operation of the amplifier 402. There is.

By the way, the gain of the amplifier 402 must be relatively large until the estimated pulsation phase 40a converges to the first value close to the true value after the amplifier 402 starts the calculation. On the other hand, once the estimated pulsation phase 40a reaches the first value, a large gain is not required. The required gain of the amplifier 402 depends on whether or not the approximation of sin(2θ−θ^ ₂ )≈2θ−θ^ ₂ holds in the phase error definition equation shown in the above equation (12). .. That is, when the true value 2θ of the pulsating current and the estimated pulsating phase θ^ ₂ are close to each other, the gain required to make the estimated pulsating phase θ^ ₂ follow the true value 2θ is relatively large. Becomes smaller. On the contrary, when the deviation between the true value 2θ and the estimated pulsation phase θ^ ₂ is large, a relatively large gain is required to converge the estimated pulsation phase θ^ ₂ to the true value 2θ.

On the other hand, as described in the second embodiment, the α-axis pulsating current 30a and the β-axis pulsating current 30b have low-order harmonic components due to the spatial harmonics of the motor 2 and the magnetic saturation of the motor 2. include. Since such a harmonic is also amplified by the amplifier 402, the estimated pulsating phase θ^ ₂ becomes slightly oscillatory. If the estimated pulsation phase θ ₂ is oscillatory, the error from the true value may increase depending on the timing of holding the estimation result. Therefore, from the viewpoint of improving the accuracy of the initial estimation, it is desirable that the gain of the amplifier 402 be minimum.

Therefore, in the phase synchronization calculation unit 41 of FIG. 7, the characteristics are switched to the direction in which the gain of the amplifier 402 decreases after the second time has elapsed since the inverter 1 started energization. The second time period is longer than the first time period. However, the gain after switching is assumed to be larger than zero.

Alternatively, a method may be used in which the amplifier 402 holds a plurality of constants and one of the constants is selected based on the gain switching signal 40c. Further, the gain switching signal 40c itself may be a signal including a constant itself that determines the gain of the amplifier 402.

Next, the operation when the gain of the amplifier 402 is switched over time will be described with reference to FIG. FIG. 8 is a diagram for explaining the operation of switching the gain of the amplifier according to the third embodiment.

FIG. 8 shows various waveform examples when the gain of the amplifier 402 is switched. More specifically, in the first stage of FIG. 8, the u-phase current is shown by the alternate long and short dash line, the v-phase current is shown by the broken line, and the w-phase current is shown by the solid line. In the second row of FIG. 8, the α-axis current is shown by a solid line and the β-axis current is shown by a broken line. In the third row of FIG. 8, the α-axis pulsating current is shown by a solid line and the β-axis pulsating current is shown by a broken line. In the fourth row of FIG. 8, the estimated pulsation frequency is shown by a solid line and the true frequency is shown by a broken line. In the fifth row of FIG. 8, the estimated pulsation phase is shown by a solid line and the true phase is shown by a broken line.

In FIG. 8, first, at time t0, the gate start in which the inverter 1 starts outputting voltage is performed. When the inverter 1 starts the gate, the three-phase current starts to flow. At this time, as shown in the first-stage waveform, a pulsating current proportional to the magnitude of the DC component is superimposed. Further, the waveform of the second stage is obtained by converting the three-phase current of the first stage into the two-phase current of the stationary coordinate system. Furthermore, the two-phase current of the second stage is input to the pulsation extraction unit 30, and the pulsating current is extracted, and the result is the waveform of the third stage.

By referring to the waveform of the α-axis pulsating current in the third stage, it can be seen that the DC component is removed after a certain period of time between time t0 and time t1. At time t1, the gain of the amplifier 402 is switched from zero to a positive value, and the estimation calculation of the phase synchronization calculation unit 41 is started. The time from time t0 to time t1 is the time corresponding to the above-mentioned first time. At the time when the estimation calculation is started, the DC component of the two-phase current is sufficiently removed, so that the gain of the amplifier 402 is set relatively high. Therefore, the estimated pulsation frequency 40b and the estimated pulsation phase 40a rapidly converge to values close to the true value, as shown in the waveforms of the fourth and fifth stages.

Next, at time t2, the gain of the amplifier 402 is switched to a smaller value. As a result, the pulsation of the estimated pulsation frequency 40b and the estimated pulsation phase 40a is reduced. The time from time t0 to time t2 is the time corresponding to the above-mentioned second time. Finally, at time t3, the estimation result is held, and the steady estimation algorithm is started using the result.

As described above, the phase synchronization calculating unit according to the third embodiment switches the gain of the amplifier from zero to a value larger than zero after the first time has elapsed since the inverter started energizing. As a result, the time required for the initial estimation can be shortened. Further, the phase synchronization calculation unit according to the third embodiment switches the gain of the amplifier to a smaller direction after the second time has elapsed since the inverter started energizing. As a result, the degree to which the harmonics included in the pulsating current are amplified is suppressed, and the accuracy of the estimation result is improved.

Fourth Embodiment
Next, a synchronous motor drive device according to the fourth embodiment will be described. FIG. 9 is a configuration diagram of a motor drive device according to the fourth embodiment. Motor drive device 101 shown in FIG. 9 is obtained by adding correction calculation unit 60 to controller 3 in the configuration of motor drive device 100 according to the first embodiment shown in FIG. Further, the controller 3 is shown as a controller 3A by adding the correction calculation unit 60. It should be noted that other configurations are the same as or equivalent to those in FIG. 1, and the same or equivalent components are designated by the same reference numerals, and duplicate description will be omitted.

The output of the pulsation extraction unit 30 of the controller 3 is in a phase advanced state as compared with the pulsation current contained in the original two-phase current. The degree to which the phase advances depends on the cutoff frequency of the high-pass filter and the frequency of the pulsating current, that is, the rotation frequency of the rotor 2a. Therefore, the phase synchronization calculation unit 41 according to the third embodiment performs the estimation calculation on the signal with the advanced pulsation phase. Therefore, the rotational position of the rotor 2a obtained by converting the output of the phase synchronization calculation unit 41 includes an error. Hereinafter, in the fourth embodiment, a method for eliminating this error will be described in detail.

In FIG. 9, the correction calculator 60 calculates and outputs the estimated rotor phase 60a and the estimated rotor frequency 60b based on the estimated pulsation phase 40a and the estimated pulsation frequency 40b which are the outputs of the phase synchronization calculation unit 41.

Next, the detailed configuration of the correction calculation unit 60 will be described with reference to FIG. FIG. 10 is a block diagram showing the configuration of the correction calculation unit according to the fourth embodiment. As shown in FIG. 10, the correction calculation unit 60 according to the fourth embodiment has a low pass filter (LPF) 601, a lookup table 602, a subtractor 603, and a conversion unit 604.

The low-pass filter 601 has a high-frequency cutoff characteristic, smoothes the estimated pulsation frequency 40b, and outputs a smoothed pulsation frequency 60d. The look-up table 602 outputs the phase correction amount 60e based on the smoothed pulsation frequency 60d. The subtractor 603 subtracts the phase correction amount 60e from the estimated pulsation phase 40a, and outputs the subtraction result as the corrected pulsation phase 60c. The conversion unit 604 converts the corrected pulsation phase 60c with a constant to obtain an estimated rotor phase 60a, and converts the smoothed pulsation frequency 60d with a constant to obtain an estimated rotor frequency 60b.

The estimated pulsation frequency 40b is pulsating due to the influence of magnetic saturation of the motor 2 and spatial harmonics. The third embodiment has described the method of reducing this pulsation by switching the gain of the amplifier 402 in the direction of decreasing with the passage of time. However, the pulsation cannot be completely eliminated by switching the gain. Therefore, in order to improve the accuracy further, the estimated pulsation frequency 40b is smoothed by using the low-pass filter 601.

The type of low-pass filter may be a filter whose transfer function is in a temporary delay form, or may be a calculation for averaging an input signal over a set time. Note that the higher the high-frequency cutoff performance of the low-pass filter, the stronger the pulsation of the estimated pulsation frequency 40b can be removed, but it should be noted that the settling time of the smoothed pulsation frequency 60d becomes longer. In other words, the characteristics of the low-pass filter 601 must be determined so that the smoothed pulsation frequency 60d is settled within the target time of the initial estimation. Here, the “target time of initial estimation” is the time from the start of energization of the inverter 1 to the start of steady estimation.

Further, the lookup table 602 is determined according to the phase characteristics of the high pass filters 301 and 302 in the pulsation extraction unit 30. The lookup table 602 holds data indicating how much the phase of the signal that has passed through the pulsation extraction unit 30 changes according to the frequency of the signal. In the subtractor 603, the phase correction amount 60e is subtracted from the estimated pulsation phase 40a. As a result, the corrected pulsation phase 60c exactly matches the pulsation phase of the two-phase current before being processed by the high- pass filters 301 and 302.

Finally, the purpose of the conversion unit 604 will be explained. The corrected pulsation phase 60c and the smoothed pulsation frequency 60d are the phase and frequency of the pulsation component superimposed on the two-phase current. As shown in the above equation (11), the two-phase current pulsates at a frequency twice the rotation angle θ of the rotor 2a. Therefore, in order to obtain information on the rotational position and the rotational frequency of the rotor 2a, the phase and frequency of the pulsating current may be multiplied by 0.5, respectively.

Also, in the fourth embodiment, the units of the estimated rotor phase 60a and the estimated rotor frequency 60b are not particularly limited. Further, in the conversion unit 604, conversion of a mechanical angle and an electrical angle, conversion of a power method and a radian method, etc. may be performed at the same time depending on the configuration of an algorithm for steady estimation (not shown).

As described above, the controller according to the fourth embodiment includes the low pass filter that smoothes the estimated pulsation frequency. Thereby, the harmonics included in the estimated pulsation frequency are further reduced, and the estimation accuracy is improved. Further, the controller according to the fourth embodiment includes a look-up table that refers to the phase correction amount based on the output of the low-pass filter, and refers to the look-up table to correct the estimated pulsation phase with the phase correction amount. This look-up table is determined based on the frequency-phase characteristics of the high-pass filter that constitutes the pulsation extraction unit. As a result, the phase advance of the signal in the pulsation extraction unit is corrected, and the accurate pulsation current phase is obtained. As a result, accurate information on the rotational position of the rotor can be obtained.

In the fourth embodiment, the example in which the method for smoothing the estimated pulsation frequency and correcting the estimated pulsation phase using the lookup table is applied to the third embodiment has been described, but the present invention is not limited to this. The same correction may be applied to the second embodiment, and the same effect can be obtained.

Further, in the first to fourth embodiments, the description has been made assuming that the motor 2 is SynRM, but another type of motor may be used. For example, the motor 2 may be an embedded permanent magnet synchronous motor (Interior Permanent Magnet Synchronous Motor: IPMSM). As described above, in the first to fourth embodiments, when the DC voltage is applied to the rotating motor 2, the fact that the pulsation of double frequency is generated in the motor current due to the saliency is utilized to utilize the rotor 2a. The information of the rotation position and the rotation frequency of is estimated. Therefore, if the IPMSM is designed to obtain not only the magnet torque but also the reluctance torque, the initial estimation algorithm according to the first to fourth embodiments can be applied.

Embodiment 5.
Next, a hardware configuration for realizing the arithmetic function in the controller 3A of the fourth embodiment will be described with reference to the drawings of FIGS. 11 and 12. FIG. 11 is a block diagram showing an example of a hardware configuration that realizes the arithmetic function in the controller according to the fourth embodiment. FIG. 12 is a block diagram showing another example of the hardware configuration that realizes the arithmetic function in the controller of the fourth embodiment.

When a part or all of the arithmetic functions in the controller 3A according to the fourth embodiment are realized by software, as shown in FIG. 11, a processor 90 for performing arithmetic and a program read by the processor 90 are stored. It can be configured to include a memory 91 and an interface 92 for inputting and outputting signals.

The processor 90 may be a computing unit such as a computing device, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor). In addition, the memory 91 includes a nonvolatile or volatile semiconductor memory such as a RAM (Random Access Memory), a ROM (Read Only Memory), a flash memory, an EPROM (Erasable Programmable ROM), and an EEPROM (registered trademark) (Electrically EPROM). Examples include magnetic disks, flexible disks, optical disks, compact disks, mini disks, and DVDs (Digital Versatile Discs).

The memory 91 stores a program that executes all or some of the arithmetic functions of the controller 3A. The processor 90 exchanges necessary information via the interface 92, and the processor 90 executes the program stored in the memory 91 to estimate the PWM control of the inverter 1 and the rotational position and rotational frequency of the motor 2. An initial estimate can be made.

The processor 90 and the memory 91 shown in FIG. 11 may be replaced with a processing circuit 93 as shown in FIG. The processing circuit 93 corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof.

As described above, in the fifth embodiment, the hardware configuration for realizing the arithmetic function of the controller 3A of the fourth embodiment has been described, but the present invention is not limited to this. It goes without saying that the controller 3 of the first to third embodiments can also be realized with the same hardware configuration.

Further, the configurations shown in the above embodiments are examples of the content of the present invention, and can be combined with another known technique, and the configurations are within the scope not departing from the gist of the present invention. It is also possible to omit or change a part of.

1 inverter, 2 motor, 2a rotor, 3,3A controller, 4 detector, 10A, 10B, 10C leg, 10a transistor, 10b diode, 12, 13, 14 connection point, 20 coordinate conversion unit, 20a α-axis current, 20b β-axis current, 30 pulsation extraction unit, 301, 302 high-pass filter, 30a α-axis pulsation current, 30b β-axis pulsation current, 40, 41 phase synchronization calculation unit, 401 phase error calculation unit, 402 amplifier, 403 integrator, 40a Estimated pulsation phase, 40b estimated pulsation frequency, 40c gain switching signal, 40f phase error, 50 voltage control unit, 60 correction calculation unit, 601 low pass filter, 602 lookup table, 603 subtractor, 604 conversion unit, 60a estimated rotor phase, 60b estimated rotor frequency, 60c post-correction pulsation phase, 60d smoothing pulsation frequency, 60e phase correction amount, 90 processor, 91 memory, 92 interface, 93 processing circuit, 100, 101 motor drive device, 110 power source.

Claims (10)

1. An inverter that drives a synchronous motor having saliency;
   A controller for controlling the operating state of the inverter,
   A detector for detecting the phase current of the synchronous motor,
   Equipped with
   The controller is
   A voltage control unit that determines the output voltage of the inverter,
   A coordinate conversion unit for converting the phase current into a two-phase current in a stationary coordinate system,
   A pulsation extraction unit that extracts a pulsation current from the two-phase current,
   A phase synchronization calculation unit for estimating and calculating the frequency and phase of the pulsating current;
   Equipped with
   The voltage control unit outputs a voltage command value in which none of the phase currents becomes zero,
   A motor drive device characterized by the above.
2. The voltage command value is a DC voltage,
   The motor drive device according to claim 1, wherein:
3. The direction of the voltage vector of the voltage command value is the same as or opposite to any phase of the synchronous motor,
   The motor drive device according to claim 2, wherein:
4. The pulsation extraction unit has two high-pass filters for removing the DC component of the two-phase current,
   The two high-pass filters have the same characteristics,
   The motor drive device according to claim 1, wherein the motor drive device is a motor drive device.
5. The phase synchronization calculation unit,
   A phase error calculation unit that calculates a phase error based on the pulsating current and the estimated pulsating phase,
   An amplifier that outputs the estimated pulsation frequency by amplifying the phase error,
   An integrator that integrates the estimated pulsation frequency and outputs the estimated pulsation phase,
   The motor drive device according to claim 4, further comprising:
6. Switching the gain of the amplifier from zero to a value greater than zero after a first time has elapsed since the inverter started to energize;
   The motor drive device according to claim 5, wherein:
7. The gain of the amplifier is changed to a smaller value after a second time has elapsed from the start of energization of the inverter,
   The motor drive device according to claim 5, wherein the motor drive device is a motor drive device.
8. The controller is
   A low-pass filter that smoothes the estimated pulsation frequency,
   A lookup table that refers to a phase correction amount based on the output of the low-pass filter,
   Equipped with
   An estimated rotor phase is obtained by correcting the estimated pulsating phase with the phase correction amount, and an estimated rotor frequency is obtained by multiplying the output of the low-pass filter by a constant number.
   The motor drive device according to claim 5, wherein the motor drive device is a motor drive device.
9. The look-up table is determined based on the frequency-to-phase characteristic of the high pass filter,
   The motor drive device according to claim 8, wherein:
10. The voltage controller outputs a voltage command value that is executed at the time of starting the inverter and does not become zero in any of the plurality of phase currents during an initial estimation period in which the rotational position and the frequency of the synchronous motor are estimated. The motor drive device according to claim 1, wherein the motor drive device is a motor drive device.

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