JP2011250630A - Control device of induction motor, and control method of the same - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a control device of an induction motor realizing a control system to reduce vibration of a motor by removing direct current components of estimated magnetic flux causing the vibration, and a control method of the same.SOLUTION: A control device 10 of an induction motor includes: a three phase/two phase coordinate converter 108 to convert a current supplied to the induction motor 106 to an α phase current and a β phase current on a stationary coordinate; a dq/two phase converter 120 to convert a magnetic flux voltage command of a d axis component on a rotational coordinate and a torque phase current command of a q axis component to an α phase voltage command and an β phase voltage command on the stationary coordinate; a magnetic flux estimator 11 to compute an α phase magnetic flux estimated value and a β phase magnetic flux estimated value by inputting the α phase current and the β phase current and the α phase voltage command and the β phase voltage command; an off-set removal processing part 12 to remove an off-set from the α phase magnetic flux estimated value and the β phase magnetic flux estimated value; and a speed estimator 112 to estimate the speed of the induction motor 106 by inputting the α phase magnetic flux estimated value and the β phase magnetic flux estimated value from the off-set removal processing part 12.

Description

  The present invention relates to an induction motor control device and control method, and more particularly to a control device and control method for an induction motor that is vector-controlled without a speed sensor.

  Conventionally, an induction motor control device includes an inverter circuit that supplies a three-phase drive current to an induction motor, and a PWM circuit that supplies a sinusoidal PWM signal to the inverter circuit. Various induction motor control devices that output a control signal to a PWM circuit based on a magnetic flux command signal have been proposed. (For example, see Patent Document 1)

  FIG. 8 is a block diagram showing a configuration of a conventional control device for an induction motor. An induction motor control device 100 (hereinafter, control device 100) shown in FIG. 8 is a control device that is connected to a three-phase AC power source 101 and an induction motor (IM) 106 and controls the induction motor 106, and is a diode rectifier circuit 102. , Smoothing circuit 103, inverter circuit 104, PWM gate signal generator 105, current sensor 107, three-phase / two-phase coordinate converter 108, two-phase / dq coordinate converter 109, magnetic flux estimator 110, high-pass filter (HPF) 111 , Speed estimator 112, low-pass filter (LPF) 113, slip calculator 114, integrator 115, magnetic flux PI controller 116, magnetic flux current PI controller 117, speed PI controller 118, torque current PI controller 119, dq / A two-phase coordinate converter 120 and a two-phase / three-phase coordinate converter 121 are provided.

The control device 100 drives the induction motor 106 using the inverter circuit 104 with the magnetic flux command value φ _d ^* and the speed command value ω _m ^* as targets. The function of the inverter circuit 104 is realized by a power switching element. Each function described below is configured by a CPU (Central Processing Unit), a ROM (Read Only Memory), a RAM (Random Access Memory), and the like. This is realized by a computer and a program stored in a ROM, a RAM, or the like.

First, the control device 100 calculates a deviation Δφ _d between the magnetic flux estimated value φ _d ^ estimated by the magnetic flux estimator 110 and the magnetic flux command value φ _d ^* by the deviation calculating unit 122, and the obtained deviation Δφ _d is calculated as follows. Input to the magnetic flux PI controller 116. The magnetic flux PI controller 116 generates a magnetic flux current command value I _d ^* based on the input deviation Δφ _d . Then, the magnetic flux current command value _I ^{d *} and the deviation [Delta] _d between the magnetic flux current _{i d} of d-axis which is output from the 2-phase / dq coordinate converter 109 calculates the deviation calculation unit 123, the resulting deviation [Delta] I _d Is input to the flux current PI controller 117. Flux current PI controller 117 generates a d-axis voltage command _V ^{d *} on the basis of the input deviation [Delta] I _d.

On the other hand, the control device 100 calculates a deviation Δω _m between the estimated speed value ω _m ^ ′ estimated by the speed estimator 112 and obtained through the low-pass filter 113 and the speed command value ω _m ^* by the deviation calculating unit 124. The obtained deviation Δω _m is input to the speed PI controller 118. The speed PI controller 118 generates a torque current command value I _q ^* based on the input deviation Δω _m . Next, the deviation ΔI _q between the torque current command value I _q ^* and the q-axis torque current i _q output from the two-phase / dq coordinate converter 109 is calculated by the deviation calculating unit 125, and the obtained deviation ΔI _q Is input to the torque current PI controller 119. The torque current PI controller 119 generates a q-axis voltage command V _q ^* based on the input deviation ΔI _q .

The d-axis voltage command V _d ^* output from the magnetic flux current PI controller 117 and the q-axis voltage command V _q ^* output from the torque current PI controller 119 are input to the dq / 2-phase coordinate converter 120. . The dq / 2-phase coordinate converter 120 generates a two-phase voltage command V for the α-axis and β-axis based on the d-axis voltage command V _d ^* , the q-axis voltage command V _q ^*, and the phase angle θ output from the integrator 115. _α ^* and _Vβ ^* are output. The two-phase voltage commands V _α ^* and V _β ^* are input to the two-phase / 3-phase coordinate converter 121. The two-phase / three-phase coordinate converter 121 uses the U-phase, V-phase, and W-phase three-phase voltage commands V _u ^* , V _v ^* , V based on the input two-phase voltage commands V _α ^* , V _β ^*. Generate _w ^* . The generated three-phase voltage commands V _u ^* , V _v ^* , and V _w ^* are input to a PWM (Pulse Width Moduration) gate signal generator 105. The PWM gate signal generator 105 controls the output voltage of the inverter 104 according to the three-phase voltage commands V _u ^* , V _v ^* , and V _w ^* by PWM control.

In the block diagram of FIG. 8, control the apparatus 100 is connected to the induction motor 106 three-phase, by detecting the current i _w and the U-phase current i _u and W-phase current sensor 107 is an induction motor 106 Yes. The current sensor 107 detects currents i _w and _iu of two phases (W phase, U phase) out of three phases (U phase, V phase, W phase). sum current i _v of from zero other one phase (V-phase) is determined uniquely.

The currents i _w and i _u detected by the current sensor 107 are input to the three-phase / two-phase coordinate converter 108. The three-phase / two-phase coordinate converter 108 performs three-phase / two-phase coordinate conversion based on the currents i _w and i _u to generate _α- phase and β-axis two-phase currents i _α and i _β. / Dq output to the coordinate converter 109. The two-phase / dq coordinate converter 109 outputs a d-axis magnetic flux current _id and a q-axis torque current i _q based on the two-phase currents i _α and i _β and the phase angle θ output from the integrator 115. To do.

On the other hand, the magnetic flux estimator 110 includes the two-phase voltage commands V _α ^* and V _β ^* output from the dq / 2-phase coordinate converter 120 and the two-phase current i output from the three-phase / 2-phase coordinate converter 108. Based on _α and i _β , α-axis and β-axis estimated magnetic flux values φ _α ^ and φ _β ^ are generated and subjected to high-pass filter processing by the high-pass filter 111 to obtain α-axis and β-axis estimated magnetic flux values. φ _α ^ ′ and φ _β ^ ′ are output to the speed estimator 112. The speed estimator 112 generates a speed estimated value ω _m ^ based on the α-axis and β-axis magnetic flux estimated values φ _α ^ ′ and φ _β ^ ′ obtained by performing the high-pass filter process, and uses the speed estimated value ω _m ^ Outputs the estimated speed value ω _m ^ ′ obtained by performing the low-pass filter processing.

The slip calculator 114 slides the induction motor 106 based on the magnetic flux current command value I _d ^* output from the magnetic flux PI controller 116 and the torque current command value I _q ^* output from the speed PI controller 118. The speed estimated value ω _s is calculated, and the computing unit 126 inputs the speed estimated value ω _m ^ ′ subjected to the low-pass filter processing and the slip speed estimated value ω _s, and sets ω _e = ω _m ^ ′ + ω _s . The power source angular frequency (inverter output angular frequency) ω _e is output. This ω _e is converted into a phase angle θ by the integrator 115 and input to the two-phase / dq coordinate converter 109 and the dq / 2-phase coordinate converter 120.

As described above, conventionally, when the induction motor (motor) 106 is controlled without a speed sensor, speed sensorless vector control for individually controlling magnetic flux and torque is used. The torque is estimated by using the motor parameters and the two-phase currents i _α and i _β obtained from the motor parameters and the two-phase voltage commands V _α ^* and V _β ^* and the currents i _w and i _{u of the} induction motor 106. By calculating the estimated speed value ω _m ^ based on the estimated magnetic flux values φ _α ^ and φ _β ^, the speed signal (ω _m ^ ') of the induction motor 106 is fed back and the speed command value ω _m ^* and The torque current command value I _q ^* is generated by performing PI (proportional integration) control of the deviation Δω _m of. On the other hand, the magnetic flux generates a flux current command value I _d ^* by PI control of a deviation Δφ _d between the flux command value φ _d ^* and the estimated flux value φ _d ^ estimated by the flux estimator 110. The current command values I _q ^* , I _d ^* and the currents i _q , i _d on the rotation coordinates (q axis, d axis) obtained by coordinate conversion of the currents i _w , i _u of the induction motor 106. deviation [Delta] I _q, the [Delta] I _d by PI control conversion, * the voltage command value _{V ^q,} generates a _V ^{d *,} voltage command value finally three-phase alternating current _V ^u _*, ^V v _*, the _V ^{w *} Then, a PWM gate signal is generated to control switching of the inverter circuit 104 to drive the induction motor 106.

When the induction motor (motor) 106 is vector-controlled without a speed sensor, in order to estimate the motor speed (speed estimated value ω _m ^), the motor magnetic flux (magnetic flux estimated values φ _α ^, φ _β ^) It is calculated from the voltage command values V _α ^* and V _β ^* and the current detection values i _α and i _β . If an offset occurs in the estimated magnetic flux values φ _α ^ and φ _β ^, a vibration component appears in the speed estimated value ω _m ^, and if speed control is performed as it is, a pulsating component is generated in the torque of the motor and the motor vibrates. In the conventional system as a countermeasure, in addition to the technique of reducing the speed control gain and avoiding the vibration region by the command frequency jump, as shown in FIG. 8, a high-pass filter 111 for removing the offset component of the estimated magnetic flux value and the speed estimation The motor vibration was reduced by adding a low-pass filter 113 that removes the vibration component of the value ω _m ^.

JP 2000-31499 A

However, in the control apparatus for a conventional induction motor, in order to perform low-pass filtering the estimated speed value omega _{m ^,} it is necessary to set a lower cut-off frequency to remove the vibration component of the velocity estimation value omega _{m ^,} There is a problem that the response of the motor becomes slow. In addition, the high-pass filter processing of the estimated flux values φ _α ^ and φ _β ^ is digitally processed, and it is difficult to completely remove only the DC offset, which is a problem when controlling the motor at a low frequency. There is a problem that occurs.

  In view of the above problems, the object of the present invention is to remove the DC component of the estimated magnetic flux that causes vibration by a method that does not use a high-pass filter, and always use a sinusoidal estimated magnetic flux that has no offset. Another object of the present invention is to provide a control device or control method for an induction motor that realizes a control method for reducing vibration of the induction motor by performing speed estimation calculation.

In order to achieve the above object, an induction motor control apparatus or control method according to the present invention is configured as follows.
A control device for a first induction motor (corresponding to claim 1) is a control device for an induction motor that is vector-controlled without a speed sensor. The current supplied to the induction motor is expressed by an α-phase current and a β-phase current on stationary coordinates. A three-phase / two-phase coordinate converter that converts to a d-axis component magnetic flux voltage command and a q-axis component torque voltage command on rotating coordinates are converted into an α-phase voltage command and a β-phase voltage command on stationary coordinates. a dq / 2 phase coordinate converter, a magnetic flux estimator for calculating an α phase magnetic flux estimated value and a β phase magnetic flux estimated value by inputting an α phase current and a β phase current, an α phase voltage command and a β phase voltage command, and The offset removal processing unit for removing the offset from the α-phase magnetic flux estimation value and the β-phase magnetic flux estimation value, and the post-offset removal α-phase magnetic flux estimation value and the post-offset removal β-phase magnetic flux estimation value from the offset removal processing unit are input. To estimate the speed of the induction motor And a speed estimator.
In the above configuration, the second induction motor control device (corresponding to claim 2) is preferably configured such that the offset removal processing unit has a positive peak and a negative peak of the α-phase magnetic flux estimation value input from the magnetic flux estimator. A peak detection unit that detects a peak, a positive peak and a negative peak of a β-phase magnetic flux estimation value, an offset calculation unit that calculates an offset from the positive peak and the negative peak, and an offset that stores the calculated offset A storage unit and an offset removal calculation unit that performs an operation for removing the offset stored in the offset storage unit from the α-phase magnetic flux estimation value and the β-phase magnetic flux estimation value input from the magnetic flux estimator, and outputs the calculation result. To do.
In the third induction motor control device (corresponding to claim 3), in the above configuration, preferably, the offset obtained by the offset removal processing unit is fed back to the magnetic flux estimator, and the magnetic flux estimator calculates the offset. The α-phase magnetic flux estimated value and the β-phase magnetic flux estimated value are used to calculate.
A first induction motor control method (corresponding to claim 4) is a control method for an induction motor that is vector-controlled without a speed sensor, and the current supplied to the induction motor is expressed by an α-phase current and a β-phase current on stationary coordinates. A three-phase / 2-phase coordinate conversion step for converting to a d-axis component magnetic flux voltage command and a q-axis component torque voltage command on rotation coordinates are converted into an α-phase voltage command and a β-phase voltage command on stationary coordinates. a dq / 2-phase coordinate conversion step, a flux estimation step for calculating an α-phase flux estimated value and a β-phase flux estimated value by inputting an α-phase current and a β-phase current, an α-phase voltage command and a β-phase voltage command, and , The offset removal processing step for removing the offset from the α-phase magnetic flux estimated value and the β-phase magnetic flux estimated value, and the post-offset-removed α-phase magnetic flux estimated value and the β-phase magnetic flux estimated after the offset removal processing obtained in the offset removing processing step. And a speed estimation step of estimating a speed of the induction motor by inputting a value.
In the control method of the second induction motor (corresponding to claim 5), in the above method, preferably, the offset removal processing step includes a positive peak and a negative peak of the α-phase magnetic flux estimated value obtained in the magnetic flux estimating step. Peak detection step for detecting a peak, a positive peak and a negative peak of a β-phase magnetic flux estimated value, an offset calculation step for calculating an offset from the positive peak and the negative peak, and an offset for storing the calculated offset A storage step and an offset removal calculation step of performing an operation for removing the offset stored in the offset storage step from the α-phase magnetic flux estimation value and the β-phase magnetic flux estimation value obtained in the magnetic flux estimation step and outputting them. To do.
In the third induction motor control method (corresponding to claim 6), in the above method, preferably, the offset obtained in the offset removal processing step is used in the magnetic flux estimation step, and the magnetic flux estimation step uses the offset. And calculating an α-phase magnetic flux estimated value and a β-phase magnetic flux estimated value.

According to the present invention, in a control device for an induction motor that is vector-controlled without a speed sensor, a three-phase / two-phase coordinate converter that converts a current supplied to the induction motor into an α-phase current and a β-phase current on stationary coordinates. A dq / 2-phase coordinate converter for converting the d-axis component magnetic flux voltage command and the q-axis component torque voltage command on the rotation coordinate into an α-phase voltage command and a β-phase voltage command on the stationary coordinate; A magnetic flux estimator that calculates an α-phase flux estimated value and a β-phase flux estimated value by inputting a current and a β-phase current, an α-phase voltage command and a β-phase voltage command, and an α-phase flux estimated value and a β-phase flux estimate An offset removal processing unit that removes an offset from the value, and a speed estimator that estimates the speed of the induction motor by inputting the estimated α-phase magnetic flux estimated value after the offset removal processing and the estimated β-phase magnetic flux estimated value after the offset removal processing from the offset removing processing unit To prepare for The DC component of the estimated magnetic flux that causes vibration is removed by a method that does not use a high-pass filter, and the speed estimation calculation is always performed using a sine wave-shaped estimated magnetic flux without an offset. It is possible to provide a control device for an induction motor that realizes a reduction control method.
Further, according to the present invention, in a control method for an induction motor that is vector-controlled without a speed sensor, a three-phase / 2-phase coordinate that converts a current supplied to the induction motor into an α-phase current and a β-phase current on a stationary coordinate. A conversion step, and a dq / 2-phase coordinate conversion step for converting a d-axis component magnetic flux voltage command and a q-axis component torque voltage command on rotational coordinates into an α-phase voltage command and a β-phase voltage command on stationary coordinates, A flux estimation step for calculating an α-phase flux estimated value and a β-phase flux estimated value by inputting an α-phase current and a β-phase current, an α-phase voltage command and a β-phase voltage command, and an α-phase flux estimated value and a β-phase The offset removal processing step for removing the offset from the magnetic flux estimation value, and the post-offset removal α-phase magnetic flux estimation value and the post-offset removal β-phase magnetic flux estimation value obtained in the offset removal processing step are input to input the speed of the induction motor. A speed estimation step for estimating the degree of vibration, so that the DC component of the estimated magnetic flux causing the vibration is removed by a method that does not use a high-pass filter, and a sine wave-like estimated magnetic flux without an offset is always used. Since the speed estimation calculation is performed, it is possible to provide a control method for an induction motor that realizes a control method for reducing vibration of the induction motor.

It is a block diagram which shows the structure of the control apparatus of the induction motor which concerns on this embodiment of this invention. It is a block diagram of the offset removal process part in the control apparatus of the induction motor which concerns on this embodiment of this invention. It is a block diagram of the magnetic flux estimator in the control apparatus of the induction motor which concerns on this embodiment of this invention. It is an estimated speed characteristic diagram when there is no offset in the estimated magnetic flux. It is an estimated speed characteristic diagram when there is an offset in the estimated magnetic flux. It is explanatory drawing of the method of removing an offset from an estimated magnetic flux. It is a flowchart explaining the method of removing an offset from an estimated magnetic flux. It is a block diagram which shows the structure of the control apparatus of the conventional induction motor.

  DESCRIPTION OF EMBODIMENTS Preferred embodiments (examples) of the present invention will be described below with reference to the accompanying drawings.

  FIG. 1 is a block diagram showing a configuration of a control device for an induction motor according to this embodiment of the present invention. The basic block configuration has the same components as those of the conventional control device 100 shown in FIG. 8, but the high-pass filter 111 for performing vibration suppression control performs the offset removal processing that is a feature of the present invention. Therefore, the offset removal processing unit 12 is changed. The offset amount obtained by the offset removal processing unit 12 is fed back to the magnetic flux estimator 11. In addition, the same code | symbol is attached | subjected to the component same as the conventional control apparatus 100 shown in FIG. 8, and description is abbreviate | omitted.

  An induction motor control device 10 (hereinafter, control device 10) shown in FIG. 1 is a control device that is connected to a three-phase AC power source 101 and an induction motor (IM) 106 and controls the induction motor 106, and is a diode rectifier circuit 102. , Smoothing circuit 103, inverter circuit 104, PWM gate signal generator 105, current sensor 107, three-phase / two-phase coordinate converter 108, two-phase / dq coordinate converter 109, magnetic flux estimator 11, offset removal processing unit 12, Speed estimator 112, low-pass filter 13, slip calculator 114, integrator 115, magnetic flux PI controller 116, magnetic flux current PI controller 117, speed PI controller 118, torque current PI controller 119, dq / 2 phase coordinate transformation 120 and a two-phase / three-phase coordinate converter 121.

In the above configuration, the three-phase / 2-phase coordinate converter 108, the current _i u supplied to the induction motor _106, i w, a _{i v,} alpha phase currents on the stationary coordinates αβ two orthogonal axes i _alpha and β performing a 3-phase / 2-phase coordinate conversion step of converting the phase current i _beta. The dq / 2-phase coordinate transformer 120 converts the d-axis component magnetic flux voltage command V _d ^* and the q-axis component torque voltage command V _q ^* on the dq orthogonal two-axis rotational coordinates on the αβ orthogonal two-axis stationary coordinates. The dq / 2-phase coordinate conversion step of converting into the α-phase voltage command V _α ^* and β-phase voltage command V _β ^* is executed.

The magnetic flux estimator 11 inputs an α-phase current i _α and a β-phase current i _β and an α-phase voltage command V _α ^* and a β-phase voltage command V _β ^* and inputs an α-phase flux estimated value φ _α ^ and a β phase. The magnetic flux estimation step for calculating the magnetic flux estimation value φ _β ^ is executed.

The offset removal processing unit 12 removes the offset from the α-phase magnetic flux estimated value φ _α ^ and the β-phase magnetic flux estimated value φ _β ^, and after the offset removal, the α-phase magnetic flux estimated value φ _α ^ 'and the offset-removed β-phase magnetic flux. An offset removal processing step for outputting the estimated value φ _β ^ ′ is executed.

The speed estimator 112 receives the post-offset removal α-phase magnetic flux estimated value φ _α ^ ′ and the post-offset removal β-phase magnetic flux estimated value φ _β ^ ′ from the offset removal processing unit 12 and inputs the speed ω of the induction motor 106. A speed estimation step is performed to estimate _m ^.

In the control device 10, the offsets φ _αoffset ^ and φ _βoffset ^ obtained by the offset removal processing unit 12 are fed back to the magnetic flux estimator 11, and the magnetic flux estimator 11 uses the offsets φ _αoffset ^ and φ _βoffset ^ to The phase magnetic flux estimated value φ _α ^ and the β phase magnetic flux estimated value φ _β ^ are calculated.

  As shown in FIG. 8, the conventional induction motor control apparatus 100 is provided with a high-pass filter 111 after the magnetic flux estimator 110, and the offset generated in the magnetic flux estimator 110 is removed by the high-pass filter 111. On the other hand, in the present embodiment, the offset generated in the magnetic flux estimator 11 as described above is removed by the offset removal processing unit 12. The offset obtained by the offset removal processing unit 12 is fed back to the magnetic flux estimator 11, and the offset is also removed by the magnetic flux estimator 11.

FIG. 2 is a block diagram of the offset removal processing unit 12. The offset removal processing unit 12 includes a peak detection unit 20, an offset calculation unit 21, an offset storage unit 22, and an offset removal calculation unit 23 that execute an offset removal processing step. The peak detection unit 20 performs a peak detection step of detecting a peak of time-varying magnetic flux estimation values φ _α ^ and φ _β ^ input from the magnetic flux estimator 11 and a peak of an opposite sign next to those peaks. Each peak value and the peak value of the next opposite sign are output to the offset calculation unit 21. The offset calculation unit 21 performs an offset calculation step of calculating the respective offset amounts φ _αoffset and φ _βoffset from each peak value output from the peak detection unit 20 and the peak value of the next reverse sign, and the offset storage unit 22 Output to. The offset storage unit 22 performs an offset storage step of storing each offset amount φ _αoffset and φ _βoffset output from the offset calculation unit 21 until the next offset amount is input, and outputs the offset amount to the offset removal calculation unit 23. . If the next offset amount is input, the stored offset amount is updated. The offset removal calculation unit 23 performs calculation to remove the offset amounts φ _αoffset and φ _βoffset input from the offset storage unit 22 from the estimated magnetic flux values φ _α ^ and φ _β ^ input from the magnetic flux estimator 11. An offset removal calculation step is performed, and the calculated magnetic flux estimation values φ _α ^ ′ and φ _β ^ ′ are output to the speed estimator 112. Further, the offset storage unit 22 feeds back and outputs the stored offset amounts φ _αoffset and φ _βoffset to the magnetic flux estimator 11.

FIG. 3 is a block configuration diagram of the magnetic flux estimator 11. The magnetic flux estimator 11 calculates the magnetic flux estimation calculation values φ _α0 ^ and φ _β0 ^ by calculating with formulas (7), (8) and formula (7) to be described later by substituting the suffix α with d. and parts 30, offset phi _Arufaoffset fed back from the offset storage unit 22 of the offset removing unit 12, with phi _Betaoffset, flux estimation calculated value _φ α0 ^, φ β0 ^ from the offset amount _φ αoffset, φ βoffset respectively An offset subtracting unit 31 is provided that outputs the estimated magnetic flux values φ _α ^ and φ _β ^ by subtraction. Here, φ _α ^ and φ _β ^ on the left side of equations (7) and (8) are replaced with φ _α0 ^ and φ _β0 ^.

  Next, the reason why the vibration of the motor in the speed sensorless vector control can be suppressed by configuring the control device 10 as shown in FIGS. 1 to 3 will be described below with reference to FIG.

The relationship between the phase θ of the rotating magnetic field and the secondary magnetic fluxes φ _α and φ _β expressed on the stationary coordinates is expressed by equation (1).

θ = tan ⁻¹ (φ _β / φ _α ) (1)

  The power supply angular frequency ωe is calculated by equation (2) obtained by time differentiation of equation (1).

ωe = dθ / dt = (( dφ β / dt) · φ α - (dφ α / dt) · φ β) / (φ α 2 + φ β 2) ········ (2)

  Further, the slip ωs can be calculated by equation (3).

ωs = Rr · (Lm / Lr ) · (φ α i β -φ β i α) / (φ α 2 + φ β 2)
(3)

  The motor angular velocity (electrical angle conversion) ωme is calculated by equation (4).

  ωme = ωe−ωs (4)

  At this time, since the magnetic flux on the stationary coordinate cannot be directly measured, it is calculated from the voltage and current on the stationary coordinate according to the equations (5) and (6).

φ _α = ∫ (V _α −R _s i _α ) dt−l _s i _α (5)
φ _β = ∫ (V _β −R _s i _β ) dt−l _s i _β (6)

The variables in the equations (1) to (6) show the following values.
V _α : α phase voltage, V _β : β phase voltage, i _α : α phase current, i _β : β phase current, φ _α : α phase magnetic flux, φ _β : β phase magnetic flux, Rs: primary resistance, Rr: two Secondary resistance, ls: primary leakage inductance, L _m : mutual inductance, L _r : secondary inductance

In the conventional magnetic flux estimator 110, α-phase voltage command V _α ^* , β-phase voltage command V _β ^* , α-phase current i _α , and β-phase current i _β are used to calculate α using equations (5) and (6). Phase and β-phase estimated magnetic flux values φ _α ^ and φ _β ^ are calculated according to the equations (7) and (8).

φ _α ^ = ∫ (V _α ^* −R _s i _α ) dt−l _s i _α (7)
φ _β ^ = ∫ (V _β ^* −R _s i _β ) dt−l _s i _β (8)

Here, since the equations (7) and (8) are integral operations, the α phase voltage command V _α ^* , β phase voltage command V _β ^* , α phase current i _α , and β phase current i _β are offset. When this occurs, the offset amount is also integrated at the same time, and an offset also appears in the α-phase and β-phase magnetic flux estimated values φ _α ^ and φ _β ^ as outputs. Since the voltage uses the α-phase and β-phase voltage command values V _α ^* and V _β ^* , the offset amount is almost zero, but since the current uses the detected values i _α and i _β , the current sensor If even a small amount of offset 107 remains, a large amount of offset appears in the α-phase and β-phase magnetic flux estimated values φ _α ^ and φ _β ^ when a certain amount of time has passed.

When the α-phase and β-phase magnetic flux estimated values φ _α ^ and φ _β ^ including the offset amount are used as described above, the estimated value ωe ^ of the power supply angular frequency ωe is the estimated value of the phase θ of the rotating magnetic field θ ^ Then, the calculation is performed from the equation (9) using the equation (2).

ωe ^ = dθ ^ / dt = ((dφ β ^ / dt) · φ α ^ - (dφ α ^ / dt) · φ β ^) / (φ α ^ 2 + φ β ^ 2) ······ (9)

  The estimated value ωs ^ of the slip ωs is calculated from the equation (10) using the equation (3).

ωs ^ = Rr · (Lm / Lr) · (φ α ^ i β -φ β ^ iα) / (φ α ^ 2 + φ β ^ 2)
(10)

  The estimated value ωme ^ of the motor angular velocity (electrical angle conversion) ωme is calculated from the equation (11) using the equation (4).

  ωme ^ = ωe ^ -ωs ^ (11)

4, the magnetic flux estimation value phi _alpha ^, (9) when there is no offset in the phi _beta ^ equation (11) estimates .omega.e ^ and the motor angular velocity (electrical angle equivalent) of the calculated power supply angular frequency .omega.e from equation It is a figure which shows the time change of estimated value (omega) me ^ of omegame, estimated speed (omega) mr, and rotation speed Nm. If there is no offset in the estimated magnetic flux value from FIG. 4, the vibration is caused by the time variation of the estimated value ωe ^ of the power supply angular frequency ωe, the estimated value ωme ^ of the motor angular velocity (electrical angle conversion) ωme, the estimated speed ωmr, and the rotational speed Nm. can not see.

5, the magnetic flux estimation value phi _alpha ^, (9) when the offset occurs in the phi _beta ^ equation (11) estimates .omega.e ^ and the motor angular velocity (electrical angle Conversion calculated power supply angular frequency .omega.e from equation ) It is a figure showing the time change of the estimated value ωme ^ of ωme, the estimated speed ωmr, and the rotation speed Nm. When an offset occurs in the estimated magnetic flux value from FIG. 5, the estimated value ωe ^ of the power source angular frequency ωe, the estimated value ωme ^ of the motor angular velocity (electrical angle conversion) ωme, the estimated speed ωmr, and the rotational speed Nm vibrate with time. Ingredients appear.

  Therefore, when speed control is performed in a state where the estimated speed ωmr includes a vibration component, the control system operates to compensate for vibration at a speed that does not actually exist, torque pulsation occurs, and the motor is vibrated. Result.

Therefore, the offset removal processing unit 12 in the control device 10 according to the present embodiment is provided with a peak detection unit 20 and an offset calculation unit 21, and the offsets of the estimated magnetic flux values φ _α ^ and φ _β ^ are set as magnetic fluxes as shown in FIG. A method of obtaining from the positive and negative peak values of the estimated magnetic flux values φ _α ^ and φ _β ^ output from the estimator 11 was adopted. Here, as an example, the estimated magnetic flux values φ _α ^ and φ _β ^ are represented by φ. When an offset as shown in FIG. 6A occurs in the sine waveform of the estimated magnetic flux φ, the sine waveform is divided into sections n to n + 3 every 90 degrees of phase, and the positive peak value (absolute value) φ at that time _{maxn, φ maxn + 1, φ} maxn + 2 and negative peak values (absolute value) phi _minn, previously obtained the phi _{minn + 1.} The offset amounts φ _offsetn , φ _{offsetn + 1} , φ _{offsetn +2} , and φ _{offsetn +3} vary from positive peak values (absolute values) to negative peak values (absolute values) shown in the following equations (12) to (15) for each section. It can be determined by a value that is 1/2 of the subtracted difference.

Section n: _φoffsetn = ( _φmaxn − _{φminn −1} ) / 2 (12)
Interval _{n + 1: φ offsetn + 1} = (φ maxn -φ minn) / 2 (13)
Interval _{n + 2: φ offsetn + 2} = (φ maxn + 1 -φ minn) / 2 (14)
Interval n + 3: _{φoffsetn + 3} = ( _{φmaxn + 2−φminn + 1} ) / 2 (15)

Specifically, as shown in the flowcharts of FIGS. 2, 6, and 7, the peak detector 20 sets the flag to zero when φ is increasing, and sets the flag to 1 when φ is decreasing. And φ at the moment when the flag changes from zero to 1 is detected as a positive peak value (step S11). Further, φ at the moment when the flag changes from 1 to zero is detected as a negative peak value (step S12). In the offset calculation unit 21, the offset amount φ _offsetn according to the equation (16) based on the continuous peak value (absolute value) φ _maxn output from the peak detection unit 20 and the peak value (absolute value) φ _{minn −1} of the opposite sign. Is calculated (step S13). Here, n represents an integer.

_φoffsetn = ( _φmaxn − _{φminn −1} ) / 2 (16)

The offset storage unit 22 stores the offset amount φ _offsetn calculated by the offset calculation unit 21 (step S14), and outputs it to the offset removal calculation unit 23. In step S15, it is determined whether the next peak has been detected. If the next peak is not detected in step S15, the offset removal calculation unit 23 subtracts the offset amount _φoffsetn from the estimated magnetic flux φ ^ output from the magnetic flux estimator 11 as shown in equation (17). The estimated magnetic flux φ ^ ′ after the offset removal process is output (step S16).

φ ^ '= φ ^ -φ _offsetn (17)

In step S17, the offset amount _φoffsetn is fed _back to the magnetic flux estimator 11. In step S18, it is determined whether or not the drive is stopped. If the drive is not stopped, step S15 is executed. If the drive is stopped, the end.

  FIG. 6B is a time change of the estimated magnetic flux value output from the offset removal calculating unit 23.

  If the next peak is detected by the peak detection unit 20 in step S15, the peak value is input to the offset calculation unit 21, and the next offset is calculated and output to the offset storage unit 22. Thereby, the offset storage unit 22 updates and stores the next offset, and outputs it to the offset removal calculation unit 23.

In the case of this method, the offset is detected with a delay of 90 degrees of the phase of the sine wave, but there is a merit that the offset can be detected regardless of a low-frequency sine waveform. Further, the offset detected by this processing is fed back to the magnetic flux estimator 11 as shown in FIG. 2, and the magnetic flux estimation calculated values φ _α0 ^ and φ _β0 ^ of the equations (7) and (8) are integrated, By subtracting the offset amounts φ _αoffset and φ _βoffset from the value by the offset subtracting unit 31, the offset is reduced at the output stage of the magnetic flux estimator 11, and the _remaining offset in the magnetic flux estimator 11 is eliminated by the offset removal processing unit 12. It becomes possible to do. Thereby, since the offset removal is performed not only in the offset removal calculation unit 23 but also in the offset subtraction unit 31, the offset removal effect can be further improved. As a result, the vibration component included in the estimated speed is reduced, and the motor vibration can be reduced even if the frequency band and speed control gain that have been vibrated up to now are increased.

  Since this method only removes the offset of the estimated magnetic flux value, the motor vibration reduction effect can be obtained with any speed sensorless drive regardless of the estimated speed algorithm. Further, since the processing of the offset removal processing unit 12 in FIG. 2 can be used as a method for simply removing the offset from the periodic waveform, a high-pass filter processing (or 0.25 Hz, for example) at an extremely low frequency (or, for example) It can also be applied as an offset removal process.

  As described above, by combining a method capable of removing the offset of the DC component caused by the estimation calculation of the secondary magnetic flux regardless of the frequency, and further by subtracting the offset amount by the integration calculation of the magnetic flux estimation value, It becomes possible to suppress the vibration of the motor.

  In the present embodiment, the offset removal of the magnetic flux estimation value is performed by both the offset removal calculation unit 23 of the offset removal processing unit 12 and the offset subtraction unit 31 of the magnetic flux estimator 11. The offset removal of the magnetic flux estimation value can be performed only by the offset removal calculation unit 23 of the offset removal processing unit 12 or can be performed only by the offset subtraction unit 31 of the magnetic flux estimator 11.

  The configurations, shapes, sizes, and arrangement relationships described in the above embodiments are merely schematically shown to the extent that the present invention can be understood and implemented, and the numerical values and the compositions (materials) of the respective components Is just an example. Therefore, the present invention is not limited to the described embodiments, and can be modified in various forms without departing from the scope of the technical idea shown in the claims.

  The control apparatus and control method for an induction motor according to the present invention are used as an apparatus and a method for controlling driving of an induction motor.

DESCRIPTION OF SYMBOLS 10 Control apparatus of induction motor 11 Magnetic flux estimator 12 Offset removal process part 13 Low pass filter 20 Peak detection part 21 Offset calculation part 22 Offset memory | storage part 23 Offset removal calculation part 101 Three-phase alternating current power supply 102 Diode rectifier circuit 103 Smoothing circuit 104 Inverter circuit DESCRIPTION OF SYMBOLS 105 PWM gate signal generator 106 Induction motor 107 Current sensor 108 3 phase / 2 phase coordinate converter 109 2 phase / dq coordinate converter 112 Speed estimator 114 Slip calculator 115 Integrator 116 Magnetic flux PI controller 117 Magnetic flux current PI control 118 Speed PI controller 119 Torque current PI controller 120 dq / 2 phase coordinate converter 121 2 phase / 3 phase coordinate converter

Claims (6)

In a control device for an induction motor that is vector-controlled without a speed sensor,
A three-phase / two-phase coordinate converter for converting the current supplied to the induction motor into an α-phase current and a β-phase current on a stationary coordinate;
A dq / 2-phase coordinate converter for converting a d-axis component magnetic flux voltage command and a q-axis component torque voltage command on rotational coordinates into an α-phase voltage command and a β-phase voltage command on the stationary coordinate;
A magnetic flux estimator that inputs the α-phase current and β-phase current and the α-phase voltage command and β-phase voltage command to calculate an α-phase flux estimated value and a β-phase flux estimated value;
An offset removal processing unit for removing an offset from the α-phase magnetic flux estimated value and the β-phase magnetic flux estimated value;
A speed estimator that estimates the speed of the induction motor by inputting the post-offset removal α-phase magnetic flux estimated value and the post-offset removal β-phase magnetic flux estimated value from the offset removal processing unit;
An induction motor control device comprising:   The offset removal processing unit detects a positive peak and a negative peak of the α-phase magnetic flux estimated value input from the magnetic flux estimator and a positive peak and a negative peak of the β-phase magnetic flux estimated value. A detection unit; an offset calculation unit that calculates an offset from the positive peak and the negative peak; an offset storage unit that stores the calculated offset; the α-phase magnetic flux estimation value input from the magnetic flux estimator; The induction motor control device according to claim 1, further comprising an offset removal calculation unit that performs calculation for removing the offset stored in the offset storage unit from a β-phase magnetic flux estimated value and outputs the calculation result.   The offset obtained by the offset removal processing unit is fed back to the magnetic flux estimator, and the magnetic flux estimator calculates an α-phase magnetic flux estimated value and a β-phase magnetic flux estimated value using the offset. The control apparatus for an induction motor according to claim 1 or 2. In a control method of an induction motor that is vector-controlled without a speed sensor,
A three-phase / 2-phase coordinate conversion step for converting the current supplied to the induction motor into an α-phase current and a β-phase current on a stationary coordinate;
A dq / 2-phase coordinate conversion step for converting a d-axis component magnetic flux voltage command and a q-axis component torque voltage command on the rotation coordinate into an α-phase voltage command and a β-phase voltage command on the stationary coordinate;
A magnetic flux estimation step for inputting the α-phase current and β-phase current and the α-phase voltage command and β-phase voltage command to calculate an α-phase magnetic flux estimated value and a β-phase magnetic flux estimated value;
An offset removal processing step of removing an offset from the α-phase magnetic flux estimated value and the β-phase magnetic flux estimated value;
A speed estimation step of estimating the speed of the induction motor by inputting the post-offset removal α phase magnetic flux estimated value and the offset removal post processing β phase magnetic flux estimated value obtained in the offset removal processing step;
An induction motor control method comprising:   The offset removal processing step detects a positive peak and a negative peak of the α-phase magnetic flux estimated value obtained in the magnetic flux estimation step, and a peak for detecting a positive peak and a negative peak of the β-phase magnetic flux estimated value. A detecting step; an offset calculating step for calculating an offset from the positive peak and the negative peak; an offset storing step for storing the calculated offset; the α-phase magnetic flux estimated value obtained in the magnetic flux estimating step; 5. The induction motor control method according to claim 4, further comprising: an offset removal calculation step for performing calculation for removing the offset stored in the offset storage step from a β-phase magnetic flux estimated value and outputting the calculation result.   The offset obtained in the offset removal processing step is used in the magnetic flux estimation step, and the magnetic flux estimation step calculates an α-phase magnetic flux estimated value and a β-phase magnetic flux estimated value using the offset. The method for controlling an induction motor according to claim 4 or 5.

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