DE112018008176T5 - MOTOR DRIVING DEVICE - Google Patents

Abstract

A motor drive device (100) includes an inverter (1), a controller (3) and a detector (4). The controller (3) contains a voltage control unit (50), a coordinate transformation unit (20), a pulsation extraction unit (30) and a phase synchronization calculation unit (40). The controller (3) controls the voltage output from the converter (1) to the motor (2) and performs a calculation that estimates a rotational position and a rotational frequency of the motor (2). The coordinate transformation unit (20) transforms the motor currents detected by the detector (4) into a two-phase current (20a, 20b) in a stationary coordinate system. The pulsation extraction unit (30) extracts a pulsating current (30a, 30b) from the two-phase current (20a, 20b), and the phase synchronization calculating unit (40) calculates an estimated pulsation phase (40a) and an estimated pulsation frequency (40b). During this calculation, the voltage control unit (50) calculates a nominal voltage value at which none of the motor currents is equal to zero, and outputs it.

Description

TECHNICAL AREA

The present invention relates to a motor drive device which drives a synchronous motor with salience without a sensor.

STATE OF THE ART

A rotating magnetic field is generated when a multiphase alternating voltage is applied to the stator of a synchronous motor. The synchronous motor generates torque through magnetic interaction between the rotating magnetic field and a rotor. The polyphase alternating voltage is an alternating voltage with three phases or four or more phases. When the synchronous motor rotates, the phase and frequency between the rotating magnetic field and the rotor must be synchronized. In order to drive or control the synchronous motor, information about the rotational position or the rotational frequency of the rotor is required. There is a method of using a position sensor or a speed sensor to get the information about the rotational position or rotational frequency of the rotor. On the other hand, in order to reduce the number of parts and the number of wirings, the application of a drive system that does not use such sensors is also spreading. The state and the drive system without a sensor to record the information about the rotational position or rotational frequency of the rotor are called "sensorless".

When starting a synchronous motor without a sensor, it is necessary to start a switching control for semiconductor elements of the inverter while synchronizing the phase and frequency of the output voltage with the rotating state of the rotor so that an overcurrent or the like does not occur. Therefore, the sensorless drive system of the synchronous motor normally distinguishes the case in which the synchronous motor is started from the other case in which the synchronous motor is not started. Specifically, the sensorless drive system of the synchronous motor is roughly divided into two algorithms called “steady state estimation” and “initial estimation”.

The “steady state estimate” algorithm is used when the semiconductor elements of the inverter driving the motor are switching and the torque or the rotational state of the motor is continuously controlled. On the other hand, the initial estimation algorithm is applied when the semiconductor elements of the inverter start the switching operation from a state in which the switching operation is stopped. That is, the estimation of the steady state is started using information on the rotational position and the rotational frequency of the rotor obtained by the initial estimation.

As an example of the initial estimation, Patent Document 1 describes a method of estimating the rotational frequency of the rotor based on a period with which the current polarity of a certain phase is reversed and estimating the rotational position of the rotor based on the timing at which the current polarity is reversed a certain phase is reversed. It should be noted that, for the sake of simplicity, the reversal of the polarity of the phase current, i.e. the sign of the phase current, is referred to as the "zero crossing".

STATE OF THE ART

Patent Document 1: Japanese Patent Application Laid-Open JP 2004-336 866 A

BRIEF DESCRIPTION OF THE INVENTION

Problems to be Solved by the Invention

In a converter whose output voltage is controlled by pulse width modulation (PWM), it is known that the actual output voltage can deviate from a voltage setpoint intended by a control due to the influence of the switch-on voltage of a switching element contained in the converter, the dead time in the control of the converter and the like . A voltage error, ie a difference between the nominal voltage value and the actual output voltage, becomes more noticeable the smaller the current flowing through the switching element, ie the smaller the motor current. It's closed Note that the motor current is a current that flows through each phase of the motor, that is, a phase current of the motor.

In the initial estimation described in Patent Document 1, the current of each phase is controlled to pulsate near zero, so that the above-mentioned voltage error is significant. When the voltage error is large, the phase current of another phase becomes too small and the accuracy of estimating the frequency is lowered. Alternatively, the phase current of another phase becomes too large, causing magnetic saturation and reducing the accuracy of the estimation of the rotational position. It should be noted that “other phase” here means a phase whose phase current is too small or too large in relation to the specific phase. As described above, a large voltage error causes a problem that the accuracy of the estimation of the frequency or the rotational position is reduced.

In addition, the signals from a voltage sensor and a current sensor contain noise that enters an analog circuit. Therefore, the voltage sensor or the current sensor cannot always recognize the exact time of the zero crossing.

In addition, the number of times that a detected value of the current can be sampled by a digital circuit is limited. In particular, during high-speed rotation of the motor, the motor current pulsates with a short period, so that the detected value of the current is often not detected and the error of the zero-crossing time becomes noticeable.

The present invention has been made in view of the above problems, and an object of the present invention is to provide a motor drive device which can estimate information on a rotational position or rotational frequency of a synchronous motor that is rotating with high accuracy.

Means of solving the problems

In order to solve the above problem, the motor drive device according to the present invention includes an inverter that drives a synchronous motor with salience, a controller that controls the operating state of the inverter, and a detector that detects phase currents of the synchronous motor. The controller includes a voltage control unit that determines an output voltage of the converter, a coordinate transformation unit that transforms the phase currents into a two-phase current in a stationary coordinate system, a pulsation extraction unit that extracts a pulsating current from the two-phase current, and a phase synchronization calculation unit that the Estimates and calculates the frequency and phase of the pulsating current. The voltage control unit outputs a voltage setpoint at which none of the phase currents is equal to zero.

Effect of the invention

The motor drive device according to the present invention can estimate the information on the rotational position or the rotational frequency of the rotating synchronous motor with high accuracy.

Figure list

• 1 Fig. 13 is a block diagram showing a configuration of a motor driving device according to a first embodiment;
• 2 is a circuit diagram showing a configuration of an in 1 shown converter shows;
• 3 Fig. 13 is a schematic diagram for explaining a voltage target value output from a voltage control unit according to the first embodiment;
• 4th Fig. 13 is a diagram showing a relationship between a phase of a voltage command value vector and the magnitude of a phase current in the first embodiment;
• 5 Fig. 13 is a block diagram illustrating a configuration of a pulsation extraction unit according to a second embodiment;
• 6th Fig. 13 is a block diagram illustrating a configuration of a phase synchronization calculating unit according to the second embodiment;
• 7th Fig. 13 is a block diagram illustrating a configuration of a phase synchronization calculating unit according to a third embodiment;
• 8th Fig. 13 is a diagram for explaining an operation for switching the gain of an amplifier according to the third embodiment;
• 9 Fig. 13 is a block diagram showing a configuration of a motor driving device according to a fourth embodiment;
• 10 Fig. 13 is a block diagram showing a configuration of a correction calculation unit according to the fourth embodiment;
• 11 FIG. 13 is a block diagram illustrating an example of a hardware configuration that implements arithmetic functions of a controller of the fourth embodiment, and FIG
• 12th Fig. 13 is a block diagram showing another example of a hardware configuration that implements the arithmetic functions of the controller of the fourth embodiment.

DESCRIPTION OF THE EMBODIMENTS

Motor driving devices according to embodiments of the present invention will now be described in detail with reference to the drawings. It should be noted that the present invention is not limited to the following embodiments.

Embodiment 1

1 Fig. 13 is a block diagram of a configuration of a motor driving device according to a first embodiment. 2 is a circuit diagram showing the configuration of the in 1 shown converter shows. In 1 includes a motor drive device 100 an engine 2 with a rotor 2a , a converter 1 who is the engine 2 drives, a controller 3 that indicate an operating state of the converter 1 controls, and a detector 4th , the one phase current of the motor 2 recorded.

The converter 1 receives direct current from an energy source 110 and applies a voltage of variable amplitude and variable frequency to the motor 2 at. The converter 1 puts the to the engine 2 applied voltage by applying pulse width modulation (PWM) control to a variety of in 1 performs semiconductor elements, not shown.

The motor 2 is a synchronous motor with salience. An example of a synchronous motor with salience is a synchronous reluctance motor (hereinafter referred to as "SynRM"). The rotor of the SynRM has the property that the magnetic resistance changes in the radial direction depending on an angle of rotation to a cylindrical axis or axis of rotation. Such a trait is known as "salience". When a current flows through a stator of the SynRM by applying a voltage to it, a magnetic field is generated that traverses the circumference of the rotor in a radial direction. At this time, a torque is generated to rotate the rotor in a direction in which the magnetic flux increases, that is, in a direction in which the magnetic resistance of a magnetic path decreases. The torque generated in this way due to the salience of the rotor is referred to as the reluctance torque.

2 shows a circuit arrangement when the inverter 1 is a three-phase converter. The in 2 inverter shown 1 includes a branch 10A , in which a semiconductor element UP of an upper branch and a semiconductor element UN of a lower branch are connected in series, a branch 10B , in which an upper arm semiconductor element VP and a lower arm semiconductor element VN are connected in series, and one arm 10C , in which a semiconductor element WP of an upper branch and a semiconductor element WN of a lower branch are connected in series. The branches 10A , 10B and 10C are connected in parallel to each other.

A bus voltage is applied to the converter via the DC busbars 15a and 15b 1 created. The converter 1 drives the engine 2 by taking the DC power supplied through the DC bus bars 15a and 15b to the power source 110 is converted to AC power and the AC power obtained by the conversion to the motor 2 supplies.

2 shows, as an example, a case where the semiconductor elements UP, UN, VP, VN, WP and WN are metal-oxide-semiconductor field effect transistors (MOSFETs). The semiconductor element UP contains a transistor 10a and a diode 10b that are anti-parallel to the transistor 10a is switched. The other semiconductor elements UN, VP, VN, WP and WN each have a similar configuration. Antiparallel means that one anode side the diode is connected to a first terminal corresponding to a source of the MOSFET, and a cathode side of the diode is connected to a second terminal corresponding to a drain of the MOSFET.

It should be noted that the semiconductor elements UP, UN, VP, VN, WP and WN can be, for example, insulated gate bipolar transistors (IGBTs) instead of the MOSFETs.

In addition, shows 2 the three-branch configuration in which the semiconductor element of the upper branch and the semiconductor element of the lower branch are connected in series. However, the invention is not limited to this configuration. The number of branches can be four or more. In addition, in 2 a branch may have a pair of the semiconductor elements of the upper and lower branches, but a branch may have a plurality of pairs of the semiconductor elements of the upper and lower branches.

Even if the transistor 10a each of the semiconductor elements UP, UN, VP, VN, WP and WN is a MOSFET, at least one of the semiconductor elements UP, UN, VP, VN, WP and WN may be formed from a wide band gap semiconductor such as silicon carbide, gallium nitride or diamond .

Wide band gap semiconductors generally have higher breakdown voltage characteristics and heat resistance than silicon semiconductors. Therefore, when the MOSFET made of the wide band gap semiconductor is used for at least one of the semiconductor elements UP, UN, VP, VN, WP and WN, improved effects of the breakdown voltage characteristic and the heat resistance can be obtained.

A connection point 12th between the semiconductor element UP of the upper branch and the semiconductor element UN of the lower branch is with a first phase (for example a U-phase) of the motor 2 connected, a connection point 13th between the semiconductor element VP of the upper arm and the semiconductor element VN of the lower arm is with a second phase (for example a V-phase) of the motor 2 connected and a connection point 14th between the semiconductor element WP of the upper arm and the semiconductor element WN of the lower arm is with a third phase (e.g. a W phase) of the motor 2 tied together. In the converter 1 form the connection points 12th , 13th and 14th an alternating current connection each.

Coming back to 1 becomes the description of the motor driving device 100 continued. The control 3 includes a voltage control unit 50 that is the output voltage of the converter 1 determined a coordinate transformation unit 20th , which transforms phase currents into a two-phase current in a stationary coordinate system, a pulsation extraction unit 30th that extracts a pulsating current from the two-phase current, and a phase synchronization calculating unit 40 that estimates and calculates the frequency and phase of the pulsating current.

The voltage control unit 50 calculates a voltage setpoint that is a setpoint from the inverter 1 voltage to be output. The switching states of the semiconductor elements UP, UN, VP, VN, WP and WN of the converter are determined on the basis of the voltage setpoint 1 determined. The voltage setpoint must be used to set the torque of the motor 2 to a desired value based on the rotational position and the rotational frequency of the rotor 2a be calculated.

There is a method of using a position sensor or a speed sensor to get information about the rotational position or rotational frequency of the rotor 2a to obtain. However, since such sensors are often installed coaxially with the motor, the permissible wavelength of the motor is reduced if the installation space of the motor is limited. Therefore, the method of using the position sensor or the speed sensor has the disadvantage that the motor output is limited in the end. In addition, the position sensor or the speed sensor require the signal line of the sensor to be wired to the hardware on which the controller is connected 3 is mounted. Therefore, the use of the position sensor or the speed sensor has a problem that the cost of the components increases and there is a fear of disconnection. For these reasons, the application of the sensorless drive system, that is, a drive system that uses neither the position sensor nor the speed sensor, is spreading. The motor driver device 100 According to the first embodiment, it is also used for the sensorless drive of the motor.

When the engine 2 is started without a sensor, it is necessary to have a switching control of the converter 1 start at which the phase and frequency of the output voltage match the rotating state of the rotor 2a are synchronized so that an overcurrent or the like does not occur. From this point of view, will divided the sensorless drive system of the synchronous motor into two algorithms, ie a steady state estimate and an initial estimate as described above.

As described above, steady state estimation is the algorithm that is applied when the inverter is running 1 starts to switch and the torque or the rotational frequency of the motor 2 is controlled continuously.

A typical method of steady-state estimation is a method using an induced voltage (also called "back electromotive voltage") of the motor 2 used. This method calculates the induced voltage based on a mathematical model of the motor and defines a phase difference between a coordinate axis of a true dq coordinate system that is the position of the rotor 2a and a coordinate axis of an estimated dq coordinate system in which the voltage target value is calculated. Then, the estimated dq coordinate system is corrected to eliminate the phase difference, and as a result, estimated values of the position and the rotational frequency of the rotor are obtained 2a . It should be noted that the dq coordinate system is a rotating coordinate system when the motor 2 is vector driven and that it is a well known concept.

Another method for estimating the steady state is a method in which a high frequency voltage is applied to the motor 2 is applied and a current response is used at that time. This method obtains estimated values of the rotational position and the rotational frequency of the rotor 2a by taking advantage of the fact that the current on the dq coordinates has an elliptical profile when the high frequency voltage is applied to the motor 2 is applied with salience. This method is often used in low speed operating conditions where the induced voltage is small.

As described above, the initial guess is the algorithm that is applied when the inverter is running 1 starts shifting from a state in which shifting is stopped. As described above, the steady state estimation algorithm requires an initial value of either or both pieces of information about the rotational position and the rotational frequency of the rotor 2a when the calculation starts. If the calculation is started while the difference between the initial value and the true value is large, problems such as overcurrent will arise. For this reason, the steady state estimate is made using the information on the rotational position and the rotational frequency of the rotor 2a obtained by the initial estimate. So the initial estimation algorithm is only executed for a short time when the inverter is 1 is started.

In the 1 Functioning of the voltage control unit shown 50 will now be described. In the first embodiment, a method of the initial estimation is described in detail. It should be noted that the steady state estimation method is not limited to any particular method.

When the voltage control unit 50 a command to start the inverter 1 from a higher-level controller (not shown), the voltage control unit calculates 50 a voltage setpoint so that no motor current of the phases is equal to zero. Furthermore, the voltage control unit generates 50 the voltage setpoint, which is a DC voltage and a voltage vector in the same direction as or in the opposite direction to one of the phases of the motor 2 Has. A reason for generating such a voltage command will be described later.

3 Fig. 13 is a diagram for explaining that of the voltage control unit 50 output voltage target value according to the first embodiment. Here it is assumed that the engine 2 is a three-phase motor. The phases of the three-phase motor are called u-phase, v-phase and w-phase. The u-phase, v-phase and w-phase form a three-phase coordinate system. The uvw three-phase coordinate system is a stationary coordinate system. It should be noted that although a voltage setpoint vector lies in the same direction as the u-phase, as in 3 but the direction is not limited thereto. The voltage setpoint vector can be in the same direction as the v-phase or the w-phase.

First, an average value of a u-phase _{current is represented} by “i u0”, an average value of a v-phase _{current is represented by “i v0} ”, and an average value of a w-phase _{current is} represented by “i w0”. The converter 1 then gives a voltage corresponding to that in 3 voltage setpoint shown. As a result, the mean value of the V-phase current and the mean value of the W-phase current have the opposite sign and half the size of the mean value of the U-phase current.

In addition, the arrows “α” and “β” point in 3 Coordinate axes when the voltage and current are subjected to a three-phase to two-phase transformation. This means that “α” and “β” form a two-phase coordinate system. The αβ two-phase coordinate system is a stationary coordinate system like the uvw three-phase coordinate system. A transformation matrix from the uvw three-phase coordinate system to the αβ two-phase coordinate system is given by the following formula. $$\left(\begin{array}{c}\mathrm{\alpha }\\ \mathrm{\beta }\end{array}\right)=\sqrt{2\text{/}3}\left(\begin{array}{lll}1\hfill & -1\text{/}2\hfill & -1\text{/}2\hfill \\ 0\hfill & -\sqrt{3}\text{/}2\hfill & \sqrt{3}\text{/}2\hfill \end{array}\right)\left(\begin{array}{c}\text{u}\\ \text{v}\\ \text{w}\end{array}\right)$$

It should be noted that the transformation matrix differs from that represented by the above formula (1) depending on how the coordinate axes are defined, but it is common to define the coordinate axes so that the α-axis coincides with one of the following -, v- and w-axes coincide. When the α and β axes are as in 3 and the above formula (1) are defined, the voltage setpoint only needs to be set such that “v _α ” is not equal to zero and “v _β ” is equal to zero. It should be noted that the in 3 Axes d and q shown are those obtained by a rotating coordinate transformation of the αβ two-phase coordinate system by an angle of rotation θ of the rotor 2a can be obtained.

If a direct voltage is now applied to the stator of the rotating motor, the phase current pulsates due to the salience. The properties of the pulsating current are described in detail below.

First, a voltage equation of the SynRM in the αβ coordinate system is represented by the following formula. $$\left(\begin{array}{c}{\text{v}}_{\mathrm{\alpha }}\\ {\text{v}}_{\mathrm{\beta }}\end{array}\right)=\left(\begin{array}{cc}{\text{R.}}_{\text{s}}+{\text{PL}}_{\mathrm{\alpha }}& {\text{PL}}_{\mathrm{\alpha \beta }}\\ {\text{PL}}_{\mathrm{\alpha \beta }}& {\text{R.}}_{\text{s}}+{\text{PL}}_{\mathrm{\beta }}\end{array}\right)\left(\begin{array}{c}{\text{i}}_{\mathrm{\alpha }}\\ {\text{i}}_{\mathrm{\beta }}\end{array}\right)$$

In the above formula (2), “v _α ” and “v _β ” represent the voltages obtained by two-phase transformation, and “i _α ” and “i _β ” represent the currents obtained by two-phase transformation. In addition, “P” stands for a differential operator and “R _s ” for a coil resistance. Furthermore, “L _α ”, “L _β ” and “L _αβ ” are defined by the following formula. $$\begin{array}{l}{\text{L.}}_{\mathrm{\alpha }}={\text{L.}}_0+{\text{L.}}_1\text{<figure-callout id="2" label="cos" filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png" state="{{state}}">cos</figure-callout>}2\mathrm{\theta }\hfill \\ {\text{L.}}_{\mathrm{\beta }}={\text{L.}}_0-{\text{L.}}_1\text{<figure-callout id="2" label="cos" filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png" state="{{state}}">cos</figure-callout>}2\mathrm{\theta }\hfill \\ {\text{L.}}_{\mathrm{\alpha \beta }}={\text{L.}}_1\text{<figure-callout id="2" label="sin" filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png" state="{{state}}">sin</figure-callout>}2\mathrm{\theta }\hfill \\ {\text{L.}}_0=\left({\text{L.}}_{\text{d}}+{\text{L.}}_{\text{q}}\right)\text{/}2\hfill \\ {\text{L.}}_1=\left({\text{L.}}_{\text{d}}-{\text{L.}}_{\text{q}}\right)\text{/}2\hfill \end{array}$$

In the above formula (3), α-axis inductance stands for β-axis inductance, and “L _αβ ” stands for αβ-axis mutual inductance. Furthermore, “θ” stands for an angle of rotation of the rotor 2a , "L ₀ " for an average inductance, "L ₁ " for a differential inductance, "L _d " for a d-axis inductance and "L _q " for a q-axis inductance.

In the motor with salience, the d-axis inductance L _d and the q-axis inductance L _{q are} different, so that the differential inductance L ₁ according to the fifth equation of the above formula (3) is not equal to zero. Therefore, as shown by the first and second equations of the above formula (3), the α-axis inductance L _α and the β-axis inductance L _{β change} according to the rotation angle θ of the rotor 2a .

In addition, the differential operator P in the above formula (2) acts on the total α-axis inductance L _α , the β-axis inductance L _β and the αβ-axis mutual inductance L _αβ and the α-axis current i _α and the β-axis current i _β . Thus, by expanding the terms of the differential operator in the above formula (2), the following formula is obtained. $$\left(\begin{array}{c}{\text{v}}_{\mathrm{\alpha }}\\ {\text{v}}_{\mathrm{\beta }}\end{array}\right)=\left\{{\text{R.}}_{\text{s}}+2\mathrm{\omega }{\text{L.}}_1\left(\begin{array}{cc}-\text{<figure-callout id="2" label="sin" filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png" state="{{state}}">sin</figure-callout>}2\mathrm{<figure-callout\; id="2"\; label="\theta \; cos"\; filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png"\; state="\{\{state\}\}">\theta </figure-callout>}& \text{cos}2\mathrm{<figure-callout\; id="2"\; label="\theta \; cos"\; filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png"\; state="\{\{state\}\}">\theta </figure-callout>}& \\ \text{cos}2\mathrm{<figure-callout\; id="2"\; label="\theta \; sin"\; filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png"\; state="\{\{state\}\}">\theta </figure-callout>}& \text{sin}2\mathrm{\theta }\end{array}\right)\right\}\left(\begin{array}{c}{\text{i}}_{\mathrm{\alpha }}\\ {\text{i}}_{\mathrm{\beta }}\end{array}\right)+\left\{{\text{L.}}_0+{\text{L.}}_1\left(\begin{array}{cc}\text{<figure-callout id="2" label="cos" filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png" state="{{state}}">cos</figure-callout>}2\mathrm{<figure-callout\; id="2"\; label="\theta \; sin"\; filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png"\; state="\{\{state\}\}">\theta </figure-callout>}& \text{sin}2\mathrm{<figure-callout\; id="2"\; label="\theta \; sin"\; filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png"\; state="\{\{state\}\}">\theta </figure-callout>}& \\ \text{sin}2\mathrm{\theta }& -\text{<figure-callout id="2" label="cos" filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png" state="{{state}}">cos</figure-callout>}2\mathrm{\theta }\end{array}\right)\right\}\left(\begin{array}{c}{\text{pi}}_{\mathrm{\alpha }}\\ {\text{pi}}_{\mathrm{\beta }}\end{array}\right)$$

In the above formula (4), “ω” represents the frequency of rotation of the rotor 2a and ω = Pθ. The excited state in 3 is equivalent to applying a clockwise rotating magnetic field to the motor, which extends away from the rotor 2a from the point of view of being at rest. Therefore, the pulsation phase of the α-axis current i _α and the β-axis current i _{β is} such that the α-axis current i _{α leads} the β-axis current i _β by 90 degrees.

In addition, the salience of the inductance, seen from the stator, acts equally on the α- and β-axes, so that the amplitude of the pulsation of the α-axis current i _α and the amplitude of the pulsation of the β-axis current i _{β are the} same. On this basis, the α-axis current i _α and the β-axis current i _{β are divided} into an average _{component (i α0} , i _β0 ) and a pulsating current component (i _α1 , i _β1 ) and then represented by the following formula . $$\begin{array}{c}{\text{i}}_{\mathrm{\alpha }}:={\text{i}}_{\mathrm{\alpha }0}+{\text{i}}_{\mathrm{\alpha }1}={\text{i}}_{\mathrm{\alpha }0}-\mathrm{\Delta }\text{i sin}\left(2\mathrm{\theta }+\mathrm{\varphi }\right)\\ {\text{i}}_{\mathrm{\beta }}:={\text{i}}_{\mathrm{\beta }0}+{\text{i}}_{\mathrm{\beta }1}={\text{i}}_{\mathrm{\beta }0}-\mathrm{\Delta }\text{i cos}\left(2\mathrm{\theta }+\mathrm{\varphi }\right)\end{array}$$

In the above formula (5), “φ” stands for an unknown phase angle and “Δi” for the amplitude of the pulsating current. Here, the average _values i α0 and i _{β0 of} the two-phase current are obtained by solving the equation obtained by ignoring all the terms related to the inductance component in the above formula (2). Since the β-axis voltage v _β is zero as described above, the mean value i _{α0 of} the α-axis current and the mean value i _{β0 of} the β-axis current are i _α0 = v _α / R _s and i _β0 = 0, respectively.

In addition, the following formula is obtained by substituting the above formula (5) into the above formula (4). $$0=2\mathrm{\omega }{\text{L.}}_1{\text{i}}_{\mathrm{\alpha }0}\left(\begin{array}{c}-\text{<figure-callout id="2" label="sin" filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png" state="{{state}}">sin</figure-callout>}2\mathrm{<figure-callout\; id="2"\; label="\theta \; cos"\; filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png"\; state="\{\{state\}\}">\theta </figure-callout>}& \\ \text{cos}2\mathrm{\theta }\end{array}\right)+{\text{R.}}_{\text{s}}\mathrm{\Delta }\text{i}\left(\begin{array}{c}-\text{sin}\left(2\mathrm{\theta }+\mathrm{\varphi }\right)\\ \text{cos}\left(2\mathrm{\theta }+\mathrm{\varphi }\right)\end{array}\right)-2\mathrm{\omega }{\text{L.}}_0\mathrm{\Delta }\text{i}\left(\begin{array}{c}\text{cos}\left(2\mathrm{\theta }+\mathrm{\varphi }\right)\\ \text{sin}\left(2\mathrm{\theta }+\mathrm{\varphi }\right)\end{array}\right)$$

The following formula can also be obtained by rearranging the formula in the first line of formula (6) above. $$0=2\mathrm{\omega }{\text{L.}}_1{\text{i}}_{\mathrm{\alpha }0}\text{sin}\left(2\mathrm{\theta }\right)+\mathrm{\Delta }\text{i}\sqrt{{\text{R.}}_{\text{s}}^2+{\left(2\mathrm{\omega }{\text{L.}}_0\right)}^2}\text{sin}\left(2\mathrm{\theta }+\mathrm{\varphi }+\mathrm{\delta }\right)$$

Note that in the above formula (7), “δ” is set as in the following formula. $$\mathrm{\delta }={\text{tan}}^{-1}\left(2\mathrm{\omega }{\text{L.}}_0{\text{/ R}}_{\text{s}}\right)$$

$$\mathrm{\varphi }=\mathrm{\pi }-\mathrm{\delta }$$

Based on the condition that the above formula (7) is an identity, the amplitude Δi of the pulsating current and the unknown phase angle φ are obtained as in the following formulas. $$\mathrm{\Delta }\text{i}={\text{i}}_{\mathrm{\alpha }0}\times \frac{2\mathrm{\omega }{\text{L.}}_1}{\sqrt{{\text{R.}}_{\text{s}}^2+{\left(2\mathrm{\omega }{\text{L.}}_0\right)}^2}}$$

$$\mathrm{\varphi }=\mathrm{\pi }-\mathrm{\delta }$$

Here, in the above formula (9), the resistance component is sufficiently smaller than the reactance component. Therefore, the above formula (8) can be approximated as δ = π / 2, and the above formula (10) can be approximated as φ = π / 2. At this time, the α-axis pulsating current i _α1 and the β-axis pulsating current i _β1 can be modified as in the following formula. $$\begin{array}{c}{\text{i}}_{\mathrm{\alpha }1}=-\mathrm{\Delta }\text{icos}\left(2\mathrm{\theta }\right)\\ {\text{i}}_{\mathrm{\beta }1}=-\mathrm{\Delta }\text{isin}\left(2\mathrm{\theta }\right)\end{array}$$

The above formula (11) shows that by extracting the pulsating currents from the values of the two-phase current in the stationary coordinate system and calculating the phases of these current values as a result, the rotational position of the rotor 2a can be appreciated.

Specifically, the control leads 3 the in 1 illustrated first embodiment performs the following operations. First, the coordinate transformation unit transforms 20th those from the detector 4th detected phase currents into an α-axis current 20a and a β-axis current 20b representing the two-phase current on the αβ-two-phase coordinate system, and outputs the α-axis current and the β-axis current. A transformation formula used at this point in time is e.g. B. the above formula (1).

Next, the pulsation extraction unit extracts 30th an α-axis pulsating current 30a and a β-axis pulsating current 30b on the basis of the α-axis current 20a and the β-axis current 20b, and outputs the α-axis pulsating current and the β-axis pulsating current to the phase synchronization calculating unit 40 the end. It should be noted that in the following description, the α-axis pulsating current 30a and the β-axis pulsating current 30b in some cases collectively referred to as "pulsating current".

Then the phase synchronization calculating unit calculates 40 an estimated pulsation phase 40a and an estimated pulsation frequency 40b based on the α-axis pulsating current 30a and the β-axis pulsating current 30b and outputs the estimated pulsation phase and the estimated pulsation frequency. The one from the phase synchronization calculation unit 40 calculated estimated pulsation phase 40a and the estimated pulsation frequency 40b are converted into corresponding values, and these values obtained by the conversion are used in the algorithm for estimating the steady state (not shown).

Next, an advantage of setting the voltage command value at a value at which none of the phase currents is zero, that is, a value at which none of the u-phase current, the v-phase current and the w-phase current is zero will be described. Note that in the following description, the u-phase current, the v-phase current, and the w-phase current are collectively referred to as “three-phase current” in some cases.

First of all, a PWM signal has to control the semiconductor elements of the upper and lower branches of a converter, not limited to the converter 1 of the first embodiment, a pause period in which an OFF command is given to all semiconductor elements of the upper and lower branches. This pause time is called the dead time. The pause time is set to prevent a short circuit between the DC busbars 15a and 15b.

The semiconductor element also experiences a voltage drop due to the physical properties of the semiconductor element. When the semiconductor element is an IGBT, there is a collector-emitter voltage drop called the saturation voltage. If the semiconductor element is a MOSFET, there is a voltage drop due to the resistance between the drain and source.

To reduce a voltage error caused by the above factors, the voltage setpoint is often corrected. These corrections are referred to as dead time correction, turn-on voltage correction, and the like.

• (1) In einem kleinen Strombereich, in dem der durch das Halbleiterelement fließende Strom klein ist, ändert sich die Schaltübergangszeit des Halbleiterelements auf komplizierte Weise.
• (2) Bei der Korrektur des Spannungssollwerts wird das Vorzeichen eines Spannungskorrekturbetrags anhand der Polarität des Stroms bestimmt, so dass wahrscheinlich Chattering auftritt. Es ist zu beachten, dass Chattering ein Ereignis ist, bei dem sich die Umkehrung des Spannungskorrekturbetrags und der Polarität des Stroms in einem unerwartet schnellen Zyklus wiederholt.
• (3) Wenn eine Totzone eingestellt wird, um Chattering zu verhindern, wird die Wirkung der Korrektur reduziert.

In addition, in controlling the semiconductor element, it is necessary to consider the following properties inherent in the semiconductor element.

• (1) In a small current range in which the current flowing through the semiconductor element is small, the switching transition time of the semiconductor element changes in a complicated manner.
• (2) In correcting the voltage target value, the sign of a voltage correction amount is determined based on the polarity of the current, so that chattering is likely to occur. It should be noted that chattering is an event in which the inversion of the voltage correction amount and the polarity of the current repeats in an unexpectedly fast cycle.
• (3) If a dead zone is set to prevent chattering, the effect of the correction will be reduced.

The above items (1) to (3) have in common that the smaller the current flowing through the semiconductor element, the more difficult it becomes to correct the voltage command value. That is, the voltage error increases when a first phase current, which is a phase current below the three-phase current, is small. When the voltage error increases, the phase current of one of the other phase currents other than the first phase current is increased or decreased. As a result, the mean value i _{α0 of} the α-axis current and the mean value i _{β0 of} the β-axis current become too small or too large.

Here, as can be seen from the above formula (9), the amplitude Δi of the α-axis pulsating current i _α1 and the β-axis pulsating current i _{β1 is} proportional to the size of the mean value i _{α0 of} the α-axis current. If one of the three-phase currents is small, the mean value i _{α0 of} the α-axis current may be too small and the accuracy of the estimation of the rotational position and the rotational frequency may be reduced.

Although not modeled by the above formula (2), in SynRM, the magnetic saturation of a magnetic element progresses as the excitation increases. In addition, in the SynRM, the ease of magnetic saturation differs significantly depending on the direction in which the rotor is magnetized. Therefore, when the mean value i _{α0 of} the α-axis current and the mean value i _{β0 of} the β-axis current become too large to cause the degree of magnetic saturation to increase, the detected pulsating current contains harmonics, so the accuracy of the Estimation of the rotational position and the rotational frequency is reduced.

Based on the above points, the voltage control unit calculates 50 According to the first embodiment, the voltage setpoint at which no phase current of the three-phase current is equal to zero, and outputs this. This makes it easier to correct the voltage setpoint for setting the output voltage of the converter 1 to a desired value. This allows the current of the motor 2 can be controlled to an appropriate amount, and thus the accuracy of the estimation of the rotational position and the rotational frequency of the rotor 2a increase.

It should be noted that when a voltage is applied to the rotating motor 2 is applied, the pulsating current can be observed regardless of the size and phase of the voltage due to the salience. However, it is desirable that that from the voltage control unit 50 output voltage setpoint is a DC voltage. The reason for this is as follows.

For example, it is assumed that the voltage for the initial estimate contains an AC component with a frequency “f”. Then an alternating current component with the frequency of "f" is also generated in the phase current of the motor. That is, the phase current contains a pulsating component that is synchronized with the rotational frequency of the motor, and the frequency component has the same frequency as the voltage. When a plurality of frequency components are mixed in the phase current, it is difficult to separate them, so that the accuracy of the initial estimation is lowered. It is therefore desirable that the voltage for the initial estimation, that is, that from the voltage control unit 50 output voltage setpoint is a direct current voltage.

Next, referring to FIG 4th an advantage of setting the direction of the voltage vector of the voltage setpoint so that it is equal to or opposite to any of the phases of the motor 2 is. 4th Fig. 13 is a diagram showing a relationship between the phase of the voltage command value vector and the magnitude of the phase current in the first embodiment. It should be noted that the “direction of the voltage vector” can be reformulated as “phase of the voltage vector” and the “same or opposite direction” as “same or opposite phase”. In this case, the same or opposite phase means that when the phase is e.g. B. is 60 [deg], the "same phase" is 60 [deg] and the "opposite phase" is 240 [deg], which is obtained by adding 180 [deg] to it.

The horizontal axis of 4th _{represents the phase of the voltage setpoint vector} with respect to the α-axis, and the vertical axis represents the amplitudes of the various phase currents in normalized form The average value i _{v0 of} the v-phase current, the dash-dotted line represents the Represents the average value i _{w0 of} the w-phase current, the thin broken line represents the average value i _{α0 of} the α-axis current, and the thick dashed line represents the average value i _{β0 of} the β-axis current.

In addition, the waveform of the thick solid line is obtained by drawing a waveform _portion having the smallest absolute value among the mean values (i u0, i _v0 , i _w0 ) of the phase currents. Here, the phase with the smallest absolute value among the average values (i _u0 , i _v0 , i _{w0) of the phase currents is} defined as the “minimum phase”. The current of the minimum phase is also defined as the “minimum phase current”.

When energized by 4th the phase of the voltage setpoint vector corresponds to the value zero. At this time, the minimum phase is the v-phase or the w-phase, and the absolute value of the minimum phase current is “0.5”. Note that from the waveform of the thick solid line in 4th , ie the waveform of the minimum phase current, it can be seen that the value of “0.5” is the maximum possible value when the phase of the voltage setpoint vector is changed.

As described above, the smaller the phase current, the more difficult it is to correct the voltage command value to reduce the voltage error. In 3 the direction of the voltage setpoint vector is set to match that of the u-phase in order to "maximize the minimum phase current". In addition, according to the waveform of the minimum phase current in 4th the maximum value in 60 [degree] increments from 0 [degree]. That is, according to 4th it can be seen that the voltage setpoint vector only needs to be aligned in the same direction as the u-phase, the v-phase and the w-phase or in the opposite direction to these.

For example, if the phase of the voltage setpoint vector 60 [Grd], the voltage setpoint vector is oriented in the opposite direction to the v-phase. For example, when the phase of the voltage command value vector is 120 [deg], the voltage command value vector points in the same direction as the w-phase.

However, when the phase of the voltage command value vector is not zero, the phase angle φ defined by the above formula (5) is different from the value shown by the above formula (10). Therefore, if the phase of the voltage command value vector is not equal to zero, there is a corresponding correction in the processing by the phase synchronization calculation unit 40 necessary.

As described above, the voltage control unit according to the first embodiment calculates and outputs the DC voltage setpoint value in such a way that the phase of the voltage setpoint value vector is in the same direction or in the opposite direction to any phase of the motor. This makes it easier to correct the voltage setpoint in order to set the output voltage of the converter to a desired value. As a result, the motor current can be controlled to an appropriate amount, and thus the accuracy of the estimation of the rotational position and the rotational frequency of the rotor can be increased.

Embodiment 2

In a second embodiment, detailed configurations and modes of operation of the in 1 illustrated pulsation extraction unit 30th and the phase synchronization calculating unit 40 described.

5 Fig. 13 is a block diagram showing a configuration of the pulsation extraction unit 30th according to the second embodiment. As in 5 includes the pulsation extraction unit 30th according to the second embodiment, two high-pass filters (HPF) 301 and 302 with the same properties. The α-axis current 20a and the β-axis current 20b, which represent the two-phase current in the stationary coordinate system, become the high-pass filters 301 respectively. 302 fed.

The time required for removing the DC component contained in the α-axis current 20a and the β-axis current 20b depends on the cutoff frequency of the high-pass filter 301 and 302 away. The higher the cutoff frequency, the shorter the time it takes to remove the DC component and the later-described estimation calculation by the phase synchronization calculation unit 40 can be started quickly. However, the frequency of the pulsating current to be extracted changes with the rotational frequency of the rotor 2a . So if the cutoff frequency is too high, even the amplitude of the pulsating current, which may reduce the S / N ratio, so care must be taken in the design.

Next, the phase synchronization calculating unit 40 described. 6th Fig. 13 is a block diagram showing a configuration of the phase synchronization calculating unit 40 according to the second embodiment. As in 6th shown, includes the phase synchronization calculation unit 40 according to the second embodiment, a phase error calculation unit 401 , an amplifier 402 and an integrator 403 .

The phase error calculation unit 401 receives the α-axis pulsating current 30a (i _α1 ), the β-axis pulsating current 30b (i _β1 ), and the estimated pulsation phase 40a (θ ^ ₂ ). The estimated pulsation phase 40a is the output of the integrator 403 . The notation “θ ^ ₂ ” is an alternative notation to a notation in which a cone symbol “^” is placed on the tip of the character “θ” in “θ ₂ ”. In the present specification, the alternative notation is used with the exception of the mathematical formulas inserted as images. The same applies to “ω ^ ₂ ”, which will be described later.

The phase error calculation unit 401 calculates a phase error 40f (Δθ ₂ ) according to the following formula. $$\mathrm{\Delta }{\mathrm{\theta }}_2:={\text{i}}_{\mathrm{\alpha }1}\text{sin}{\widehat{\theta }}_2-{\text{<figure-callout id="1" label="i β" filenames="DE112018008176T5\_0013.png,DE112018008176T5\_0014.png" state="{{state}}">i</figure-callout>}}_{\mathrm{\beta }}\text{cos}{\widehat{\theta }}_2=\mathrm{\Delta }\text{i sin}\left(2\mathrm{\theta }-{\widehat{\theta }}_2\right)\cong \mathrm{\Delta }\text{i}\left(2\mathrm{\theta }-{\widehat{\theta }}_2\right)\cong \mathrm{\Delta }\text{i}\left(2\mathrm{\theta }-{\widehat{\theta }}_2\right)$$

The amplifier 402 amplifies the phase error 40f (Δθ ₂ ) and gives the estimated pulsation frequency 40b (ω ^ ₂ ). As in 6th indicated is the amplifier 402 preferably a proportional-integral (PI) controller that performs proportional-integral control.

The integrator 403 integrates the estimated pulsation frequency 40b and gives the integrated value as the estimated pulsation phase 40a the end. The estimated pulsation phase 40a is applied to the phase error calculation unit 401 returned.

In the above formula (12), when 2θ> θ ^ ₂ , is positive, so the estimated pulsation frequency ω ^ ₂ and the estimated pulsation _{phase θ ^ 2 are} corrected to be increased. On the contrary, when 2θ <θ ^ ₂ is negative, so the estimated pulsation frequency ω ^ ₂ and the estimated pulsation _{phase θ ^ 2} are corrected to be decreased. Finally, 2θ is estimated to be equal to θ ^ ₂ and thus the phase and frequency of the pulsating current is estimated. The phase synchronization calculation unit 40 thus has the form of a phase locked loop (PLL).

_{Now, when the phase of the voltage command} value vector is set to zero as in the first embodiment, the mean value (i β0) of the β-axis current 20b is zero as in FIG 4th shown. Therefore, at first glance, it seems unnecessary to use the high-pass filter 302 to use the β-axis pulsating current 30b (i _β1 ) from the β-axis stream 20b. However, when there is a difference between the α-axis and the β-axis in the presence / absence and the characteristics of the filtering for the pulsating current extraction, the amplitude and the phase of the pulsating current to be extracted differ between the α-axis and the β-axis. Hence the two high pass filters 301 and 302 required regardless of the phase of the voltage setpoint vector. It is also desirable that both have the same characteristics. It should be noted that the same properties in this case do not mean that the physical properties are completely the same, but that they are designed and configured with the expectation of having the same properties.

A method is also conceivable in which either the α-axis pulsating current 30a or the β-axis pulsating current 30b is used to calculate the pulsation frequency from the zero crossing interval and the pulsation phase from the zero crossing time. As the signal from the detector 4th however, it contains noise that is in the circuitry of the detector 4th penetrates, the zero crossing time cannot always be precisely determined. It also decreases during high speed rotation of the engine 2 the number of samples per cycle with which the current pulses from, so that an error in the zero crossing time becomes more noticeable. Because the pulsating current is also due to the influence of magnetic saturation and spatial harmonics of the motor 2 Contains low order harmonics, the relationship between the zero crossing time of the pulsating current and the position of the rotor 2a more complicated.

On the other hand, includes the phase synchronization calculating unit 40 according to the second embodiment, a circuit corresponding to a PLL with a feedback path. When the circuit corresponding to the PLL is as shown in 6th is configured is the integrator 403 included in the path before the estimated pulsation phase 40a is obtained so that it is not easily affected by the noise that is in the signal path of the detector 4th entry. In addition, the estimated pulsation phase 40a continuously calculated so that it follows the true pulsation phase. As a result, it is easily possible to correct an error caused by discretization even with a small number of samples per cycle with which the current pulsates. In addition, the estimated pulsation phase converges on average with the true phase even if disturbances such as magnetic saturation and spatial harmonics are included in the pulsating current.

As described above, contains the pulsation extraction unit 30th According to the embodiment, the two high-pass filters, which remove the direct current component from the two-phase current in the stationary coordinate system and output the pulsating currents. The two high-pass filters have the same properties. In addition, the phase synchronization calculating unit according to the second embodiment includes the phase error calculating unit that calculates the phase error from the pulsating current and the estimated pulsation phase, the amplifier that amplifies the phase error and outputs the estimated pulsation frequency, and the integrator that outputs the estimated pulsation frequency integrated and outputs as an estimated pulsation phase. This makes it less susceptible to various disturbances, so that the rotational position and the rotational frequency of the rotor can be estimated with a high degree of accuracy.

Embodiment 3

In a third embodiment, the operating mode is switched when the converter is started 1 described.

First of all, the pulsation extraction unit is required 30th , as described in the second embodiment, a time corresponding to the cutoff frequency of the high-pass filter in order to remove the direct current component from the two-phase current in the stationary coordinate system.

In addition, the estimated pulsation frequency does not converge while the direct current component at the signal input to the phase synchronization calculation unit 40 remains. When the amplifier 402 and the integrator 403 the phase synchronization calculation unit 40 start the calculation while the DC component remains, the estimated pulsation frequency diverges 40b (ω ^ ₂ ) and the estimated pulsation phase 40a (θ ^ ₂ ) or oscillate to imprecise values. As a result, the time required for the initial estimation becomes long. It is therefore desirable that the amplifier 402 and the integrator 403 start the calculation at a point in time when a required time after the inverter is switched on 1 has passed.

A phase synchronization calculation unit 41 according to the third embodiment is as in FIG 7th shown constructed. 7th Fig. 13 is a block diagram showing the configuration of the phase synchronization calculating unit 41 according to the third embodiment. When comparing 7th with 6th is in 7th a gain switching signal 40c added which is switching the gain of the amplifier 402 controls.

In the phase synchronization calculation unit 41 from 7th becomes the gain, which is the gain of the amplifier 402 represents immediately after the drive starts to be energized 1 set to zero and after an initial time has elapsed since the converter started to be energized 1 becomes the gain of the amplifier 402 set to a value not equal to zero, ie a value greater than zero. When the gain of the amplifier 402 is set to zero, the estimated pulsation rate remains 40b equal to zero, regardless of the phase error 40f into the amplifier 402 is entered. As a result, there is also the input of the integrator 403 equal to zero, and the estimated pulsation phase 40a also remains zero.

The time from the start of the drive being energized 1 until the amplifier gain is switched for the first time 402 is based on the cutoff frequency of the high-pass filter 301 and 302 in the pulsation extraction unit 30th certainly. More specifically, when the cutoff frequency is the high-pass filter 301 and 302 is high, it takes a relatively short time to remove the DC component, so the amplifier 402 the calculation within a relatively short time after the drive starts to be energized 1 can begin. In contrast, when the cutoff frequency becomes the high pass filter 301 and 302 is low, one takes a relatively long time to remove the DC component, so that a relatively long waiting time must be set before the amplifier 402 the calculation begins.

The gain of the amplifier 402 must be relatively large from the time the amplifier is installed 402 the calculation begins until the estimated pulsation phase 40a converges to a first value that is close to the actual value. On the other hand, the required gain is not so great once the estimated pulsation phase 40a reached the first value. The required gain of the amplifier 402 depends on whether or not the approximation of sin (2θ-θ ^ ₂ ) ≈2θ-θ ^ ₂ in the defining equation of the phase error represented by the above formula (12) holds. That is, when 2θ as the true value of the phase of the pulsating current and the estimated pulsation phase θ ^ _{2 are} approximately close to each other, a relatively small gain is required for the estimated pulsation _{phase θ ^ 2} to follow the true value 2θ. On the contrary, when the difference between the true value 2θ and the estimated pulsation phase θ ^ _{2 is} large, a relatively large gain is required to make the estimated pulsation phase θ ^ ₂ approach the true value 2θ.

On the other hand, the α-axis contain pulsating current 30a and the β-axis pulsating current 30b as described in the second embodiment, due to the influence of the spatial harmonics of the motor 2 and the magnetic saturation of the motor 2 the harmonic components of the lower order. Because such harmonics are also caused by the amplifier 402 are amplified, the estimated pulsation phase θ ^ _{2 is} slightly oscillating. When the estimated pulsation phase θ ^ _{2 is} oscillatory, an error from the actual value may be large depending on the timing of holding an estimation result. It is therefore desirable to increase the gain of the amplifier 402 is minimized with a view to improving the accuracy of the initial estimate.

Therefore, in the phase synchronization calculation unit 41 from 7th the characteristic switched so that the gain of the amplifier 402 after a second time has elapsed since the converter began to be energized 1 is reduced. The second time is longer than the first time. However, it is assumed that the gain after switching is greater than zero.

A method can also be used in which a plurality of constants in the amplifier 402 and one of the constants based on the gain switching signal 40c is selected. Alternatively, the gain switching signal 40c itself be a signal that contains the constant itself that is the gain of the amplifier 402 certainly.

Next is the operation when the gain of the amplifier 402 is switched according to the elapsed time, referring to FIG 8th described. 8th Fig. 13 is a diagram for explaining the operation in switching the gain of the amplifier according to the third embodiment.

8th illustrates various examples of waveforms when increasing the gain of the amplifier 402 is switched. In particular, illustrates a first level of FIG 8th the u-phase current by a dot-dash line, the v-phase current by a dashed line, and the w-phase current by a solid line. A second level of 8th represents the α-axis current by a solid line and the β-axis current by a broken line. A third plane of FIG 8th illustrates the α-axis pulsating current by a solid line and the β-axis pulsating current by a broken line. A fourth level of 8th shows the estimated pulsation frequency by a solid line and the true frequency by a dashed line. A fifth level of 8th illustrates the estimated pulsation phase with a solid line and the true phase with a dashed line.

In 8th the gate start is first carried out at time t0, at which the converter 1 begins with the output of the voltage. As soon as the gate start by the inverter 1 is done, the three-phase current begins to flow. At this point in time, as shown by the curves in the first level, the pulsating currents, which are proportional to the magnitude of the direct current component, are superimposed. In addition, the three-phase current in the first level is converted into the two-phase current in the stationary coordinate system, which is represented by the curve shapes in the second level. Furthermore, the two-phase flow is fed into the pulsation extraction unit in the second level 30th is input, and a result of the extraction of the pulsating currents is represented by the waveforms in the third level.

Looking at the waveform of the α-axis pulsating current in the third plane, it can be seen that the direct current component is removed after a certain time has elapsed between time t0 and time t1. At time t1 the gain of the amplifier becomes 402 from zero is switched to a positive value and the estimation calculation by the phase synchronization calculation unit 41 is started. The time from time t0 to time t1 is the time that corresponds to the first time described above. At the time of starting the estimation calculation, the direct current component of the two-phase current is sufficiently removed so that the gain of the amplifier 402 is set relatively high. Therefore, the estimated pulsation frequency converge 40b and the estimated pulsation phase 40a quickly to values that are close to the true values, as shown by the waveforms in the fourth and fifth levels.

Next, at time t2, the gain of the amplifier becomes 402 switched so that it is decreased. As a result, the estimated pulsation frequency 40b and the estimated pulsation phase 40a be reduced. The time from time t0 to time t2 is the time that corresponds to the second time described above.

Finally, at time t3, the estimation result is recorded, and the algorithm for estimating the steady state is started using the result.

As described above, the phase synchronization calculating unit according to the third embodiment switches the gain of the amplifier from zero to the value greater than zero after the first time has elapsed since the converter was energized. This can reduce the time required for the initial estimation. In addition, according to the third embodiment, the phase synchronization calculating unit switches the gain of the amplifier so that it is decreased after the second time that has elapsed since the inverter was energized. This reduces the degree of amplification of the harmonics contained in the pulsating current and improves the accuracy of the estimation result.

Embodiment 4

Next, a motor driving device for a synchronous motor according to a fourth embodiment will be described. 9 Fig. 13 is a block diagram showing a configuration of a motor driving device according to the fourth embodiment. One in 9 illustrated motor drive device 101 results from adding a correction calculation unit 60 for controlling 3 in the configuration of the motor driver device 100 according to the in 1 illustrated first embodiment. Due to the addition of the correction calculation unit 60 is the controller 3 also as a control 3A shown. Note that the other configurations are identical or equivalent to those of 1 are and therefore with the same reference numerals as in 1 are designated, whereby a redundant description is dispensed with.

The output of the pulsation extraction unit 30th the control 3 has the phase that is advanced compared to the pulsating current contained in the original two-phase current. The degree of advance of the phase depends on the cutoff frequency of the high-pass filter and the frequency of the pulsating current, ie the rotational frequency of the rotor 2a . Therefore, the phase synchronization calculation unit performs 41 according to the third embodiment, performs the estimation calculation on the signal whose pulsation phase is advanced. This creates an error in the rotational position of the rotor 2a included by converting the output of the phase synchronization calculation unit 41 is obtained. In the fourth embodiment, a method for eliminating the error will be described in detail below.

In 9 calculates and gives the correction calculation unit 60 an estimated rotor phase 60a and an estimated rotor frequency 60b based on the estimated pulsation phase 40a and the estimated pulsation frequency 40b as the output of the phase synchronization calculation unit 41 the end.

Next, a detailed configuration of the correction calculation unit will be presented 60 with reference to 10 described. 10 Fig. 13 is a block diagram showing a configuration of the correction calculation unit according to the fourth embodiment. As in 10 shown includes the correction calculation unit 60 according to the fourth embodiment, a low-pass filter (LPF) 601 , a look-up table 602 , a subtracter 603 and a conversion unit 604 .

The low pass filter 601 has a high-frequency blocking characteristic and smooths the estimated pulsation frequency 40b for outputting a smoothed pulsation frequency 60d . The look-up table 602 gives a phase correction amount 60e based on the smoothed pulsation frequency 60d the end. The subtracter 603 subtracts the phase correction amount 60e from the estimated pulsation phase 40a , and gives a result of the subtraction as the corrected pulsation phase 60c the end. The conversion unit 604 calculates the corrected Pulsation phase 60c with a constant order to get the estimated rotor phase 60a and calculates the smoothed pulsation frequency 60d with a constant around to get the estimated rotor frequency 60b to obtain.

The estimated pulsation rate 40b pulsates due to the influence of the magnetic saturation and the spatial harmonics of the motor 2 . In the third embodiment, the method is described in which the pulsation is controlled by switching the gain of the amplifier 402 is reduced according to the elapsed time. Switching the gain, however, cannot completely eliminate the pulsation. Therefore, in order to further improve the accuracy, the estimated pulsation frequency 40b by using the low pass filter 601 smoothed.

The low pass filter may be of the type such as a filter in which the transfer function is a temporary delay, or it may perform an operation of averaging the input signal over a certain time. It should be noted that the pulsation is the estimated pulsation frequency 40b the more it can be removed, the higher the limit power of the low-pass filter for high frequencies, but the settling time for the smoothed pulsation frequency 60d gets longer. In other words: the characteristics of the low-pass filter 601 must be determined so that the smoothed pulsation frequency 60d is controlled within a target time for the initial estimate. The "target time for the initial estimation" is the time from when the converter began to be energized 1 until the beginning of the steady state estimation.

Furthermore, the look-up table 602 according to the phase characteristic of the high-pass filter 301 and 302 in the pulsation extraction unit 30th determined. The look-up table 602 contains data indicating how much the phase of the signal sent by the pulsation extraction unit has changed 30th has passed through, depending on the frequency of the signal changes. In the subtracter 603 becomes the phase correction amount 60e from the estimated pulsation phase 40a subtracted. As a result, the corrected pulsation phase is correct 60c exactly matched the pulsation phase of the two-phase current before being passed by the high-pass filters 301 and 302 is processed.

Finally, there is another purpose of the conversion unit 604 described. The corrected pulsation phase 60c and the smoothed pulsation frequency 60d are the phase and frequency of the pulsation component superimposed on the two-phase current. As shown in the above formula (11), the two-phase current pulsates at a frequency twice as large as the rotation angle θ of the rotor 2a . To get information about the rotational position and the rotational frequency of the rotor 2a To obtain, therefore, the phase and the frequency of the pulsating current only need to be multiplied by 0.5 each.

In addition, in the fourth embodiment, the units are the estimated rotor phase 60a and the estimated rotor frequency 60b not particularly limited. In addition, the conversion unit 604 perform a conversion between a mechanical angle and an electrical angle, a conversion between a degree and a radian measure, and the like at the same time depending on the configuration of the steady-state estimation algorithm (not shown).

As described above, according to the fourth embodiment, the controller includes a low-pass filter that smooths the estimated pulsation frequency. This further reduces the harmonics contained in the estimated pulsation frequency and improves the estimation accuracy. The controller according to the fourth embodiment further includes the look-up table for referencing the phase correction amount based on the output of the low-pass filter and corrects the estimated pulsation phase with the phase correction amount with reference to the look-up table. The look-up table is determined on the basis of the frequency-phase characteristic of the high-pass filter contained in the pulsation extraction unit. This corrects the phase shift of the signal in the pulsation extraction unit and precisely determines the phase of the pulsating current. Therefore, accurate information on the rotational position of the rotor can be obtained.

Note that, in the fourth embodiment, an example will be described in which the method of smoothing the estimated pulsation frequency and correcting the estimated pulsation phase using the look-up table is applied to the third embodiment, but the present invention is not applicable to this limited. A similar correction can also be applied to the second embodiment, whereby a similar effect can be obtained.

The description of the first to fourth embodiments assumes that the engine 2 is a SynRM, but another type of motor can be used. For example, the engine can 2 be an internal permanent magnet synchronous motor (IPMSM). As described above, in the first through fourth Embodiment the information about the rotational position and the rotational frequency of the rotor 2a estimated using the fact that the motor current when the DC voltage is applied to the rotating motor 2 because of the salience it pulsates at twice the frequency. Therefore, an IPMSM designed to be able to obtain not only the magnetic torque but also the reluctance torque can apply the initial estimation algorithm according to the first to fourth embodiments.

Embodiment 5

Next, a hardware configuration is made to implement the arithmetic functions of the controller 3A of the fourth embodiment with reference to FIG 11 and 12th described. 11 Fig. 13 is a block diagram showing an example of the hardware configuration that implements the arithmetic functions of the controller of the fourth embodiment. 12th Fig. 13 is a block diagram showing another example of the hardware configuration that implements the arithmetic functions of the controller of the fourth embodiment.

When some or all of the arithmetic functions of the controller 3A of the fourth embodiment can be implemented by software as shown in FIG 11 shown, the configuration can have a processor 90 that performs arithmetic operations, a memory 91 by the processor 90 saves programs to be read, and an interface 92 that input and output signals.

The processor 90 can be an arithmetic means such as an arithmetic unit, a microprocessor, a microcomputer, a central processing unit (CPU) or a digital signal processor (DSP). The memory 91 For example, it can be a non-volatile or volatile semiconductor memory such as a random access memory (RAM), a read-only memory (ROM), a flash memory, an erasable programmable ROM (EPROM) or an electrical EPROM (EEPROM (registered trademark)), a magnetic disk , a flexible disc, an optical disc, a compact disc, a mini-disc, or a digital versatile disc (DVD).

The memory 91 stores programs to perform all or some of the controller's arithmetic functions 3A . The processor 90 sends and receives necessary information via the interface 92 and performs that in memory 91 saved programs to the PWM control of the inverter 1 and the initial estimate showing the rotational position and the rotational frequency of the motor 2 appreciate being able to perform.

In addition, the in 11 illustrated processor 90 and the memory 91 by a processing circuit 93 as in 12th be replaced. The processing circuit 93 corresponds to a single circuit, a complex circuit, an application-specific integrated circuit (ASIC), a field-programmable gate array (FPGA) or a combination thereof.

As described above, in a fifth embodiment, the hardware configuration for implementing the arithmetic functions is the controller 3A the fourth embodiment has been described, but the present invention is not limited thereto. Of course, the control can also 3 of the first to third embodiments can be implemented with a similar hardware configuration.

The configurations shown in the aforementioned embodiments merely represent examples of the content of the present invention and can therefore be combined with another known technique or partially omitted and / or modified without departing from the scope of the present invention.

List of reference symbols

1

Converter

2

engine

2a

rotor

3, 3A

steering

4th

detector

10A, 10B, 10C

poor

10a

transistor

10b

diode

12, 13, 14

Connection point

20th

Coordinate transformation unit

20a

α-axis current

20b

β-axis current

30th

Pulsation extraction unit

301, 302

High pass filter

30a

α-axis pulsating current

30b

β-axis pulsating current

40, 41

Phase synchronization calculation unit

401

Phase error calculation unit

402

amplifier

403

Integrator

40a

estimated pulsation phase

40b

estimated pulsation frequency

40c

Gain switching signal

40f

Phase error

50

Voltage control unit

60

Correction calculation unit

601

Low pass filter

602

Lookup table

603

Subtracter

604

Conversion unit

60a

estimated rotor phase

60b

estimated rotor frequency

60c

corrected pulsation phase

60d

smoothed pulsation frequency

60e

Phase correction amount

90

processor

91

Storage

92

interface

93

Processing circuit

100, 101

Motor driving device

110

Energy source

QUOTES INCLUDED IN THE DESCRIPTION

This list of the documents listed by the applicant was generated automatically and is included solely for the better information of the reader. The list is not part of the German patent or utility model application. The DPMA assumes no liability for any errors or omissions.

Patent literature cited

• JP 2004336866 A [0006]

Claims (10)

A motor driving device comprising: a converter for driving a synchronous motor with salience; a controller to control the operating state of the converter; and a detector for detecting phase currents of the synchronous motor, the controller comprising: a voltage control unit for determining an output voltage of the converter; a coordinate transformation unit for transforming the phase currents into a two-phase current in a stationary coordinate system; a pulsation extraction unit for extracting a pulsating stream from the two-phase stream; and a phase synchronization calculating unit for estimating and calculating a frequency and a phase of the pulsating current, the voltage control unit outputting a voltage target value at which none of the phase currents is equal to zero. Motor driver device according to Claim 1 , where the voltage setpoint is a DC voltage. Motor driver device according to Claim 2 , wherein the direction of the voltage vector of the voltage setpoint is equal to or opposite to a phase of the synchronous motor. Motor driving device according to one of the Claims 1 until 3 wherein the pulsation extraction unit includes two high-pass filters for removing a direct current component of the two-phase current, and the two high-pass filters have the same characteristic. Motor driver device according to Claim 4 wherein the phase synchronization calculating unit comprises: a phase error calculating unit for calculating a phase error on the basis of the pulsating current and an estimated pulsation phase; an amplifier for amplifying the phase error and outputting an estimated pulsation frequency; and an integrator to integrate the estimated pulsation frequency and output a result of the integration as an estimated pulsation phase. Motor driver device according to Claim 5 , the gain of the amplifier being switched from zero to a value greater than zero after a first time has elapsed since the converter was energized. Motor driver device according to Claim 5 or 6th wherein the gain of the amplifier is changed so that it is decreased after a second time that has elapsed since the converter was started to be energized. Motor driving device according to one of the Claims 5 until 7th wherein the controller comprises: a low pass filter for smoothing the estimated pulsation frequency; and a look-up table for referencing a phase correction amount based on the output of the low-pass filter, wherein the estimated pulsation phase is corrected by the phase correction amount to obtain an estimated rotor phase, and the output of the low-pass filter is multiplied by a constant to become an estimated rotor frequency obtain. Motor driver device according to Claim 8 , wherein the look-up table is determined on the basis of a frequency-phase characteristic of the high-pass filter. Motor driving device according to one of the Claims 1 until 9 , wherein the voltage control unit outputs the voltage setpoint at which none of the plurality of phase currents is equal to zero, during a period of the initial estimation which is carried out when the converter is started and which estimates the rotational position and a frequency of the synchronous motor.

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