WO2019163103A1

WO2019163103A1 - Vacuum cleaner

Info

Publication number: WO2019163103A1
Application number: PCT/JP2018/006768
Authority: WO
Inventors: 裕次 ▲高▼山; 和徳畠山; 酒井　顕
Original assignee: 三菱電機株式会社
Priority date: 2018-02-23
Filing date: 2018-02-23
Publication date: 2019-08-29
Also published as: JP6925497B2; JPWO2019163103A1

Abstract

A vacuum cleaner (1) includes a single phase motor (12), a motor drive device (2) that drives the single phase motor (12), and a suction port body (63) that sucks in dust. The motor drive device (2) includes: a single phase inverter (11) that applies AC power to the single phase motor (12); and a processor (31) that calculates the opening percentage of the gap formed between a suction port (69) of the suction port body (63) and a cleaning surface on the basis of a position sensor (21) that outputs a position sensor signal which indicates the position of the magnetic pole of a rotor of the single phase motor (12) and of changes in the rotational speed of the single phase motor (12) calculated on the basis of the position sensor signal. When the opening percentage is within a first range, the amplitude value of the AC power is a first value, and when the opening percentage is within a second range, the amplitude value of the AC power is a second value that is different from the first value.

Description

Electric vacuum cleaner

The present invention relates to a vacuum cleaner equipped with a single-phase motor driven by a motor driving device.

A single-phase motor has the following advantages compared to a three-phase motor having three phases.
(1) While it is necessary to use a three-phase inverter for a three-phase motor, a single-phase inverter having a simpler configuration than a three-phase inverter may be used for a single-phase motor.
(2) A three-phase inverter using a full-bridge inverter requires six switching elements, whereas a single-phase motor can be configured with four switching elements even if a full-bridge inverter is used.
(3) Due to the features of (1) and (2) above, a device using a single-phase motor can be made smaller than a device using a three-phase motor.

Due to the above features, single-phase motors are often applied from the viewpoint of miniaturization in cordless vacuum cleaners equipped with batteries.

The vacuum cleaner includes, as basic components, an electric blower that generates a suction force, a suction port for sucking dust, and a dust collection chamber for storing sucked dust. In addition to these, the electric vacuum cleaner disclosed in Patent Literature 1 includes an optical floor detection sensor that detects the type of floor to be cleaned. Examples of floor types include carpets, tatami mats, and flooring. The floor surface detection sensor of Patent Document 1 includes a wheel for detecting a floor surface, a holding frame for the wheel, two photo switches, a light shielding unit for shielding an optical axis of one of the two photo switches, And a light shielding lever for shielding the optical axis of the other photoswitch.

The vacuum cleaner of Patent Document 1 performs control to automatically switch the operation mode of the vacuum cleaner according to the detection information on the floor surface. The detection information of the floor surface in Patent Document 1 is information regarding whether the floor surface is a carpet, whether the floor surface is a plank floor or a tatami floor, or whether the suction port is away from the floor surface. is there.

Japanese Patent Laid-Open No. 2-52620

Due to the diversification of lifestyles in recent years, the function of detecting the type of floor surface to be cleaned has been recognized as an essential component. However, as shown in Patent Document 1, the method using an optical floor detection sensor uses a large number of parts and has a complicated structure. As a result, the conventional vacuum cleaner has a problem that design and manufacturing costs increase, and reliability decreases due to an increase in the number of parts. For this reason, it is desired to realize a function of detecting a floor type without using an optical floor detection sensor.

The present invention has been made in view of the above, and an object of the present invention is to obtain a vacuum cleaner that can detect a floor type without using an optical floor detection sensor.

In order to solve the above-described problems and achieve the object, a vacuum cleaner according to the present invention includes a single-phase motor, a motor driving device that drives the single-phase motor, and a suction tool that sucks dust. The motor driving device includes a single-phase inverter that applies an AC voltage to the single-phase motor, a position sensor that outputs a position sensor signal indicating a rotor magnetic pole position of the single-phase motor, and a rotational speed of the single-phase motor based on the position sensor signal. A calculation unit is provided that calculates and calculates an opening ratio of a gap generated between the suction port of the suction tool and the cleaning surface based on the calculated change in the rotation speed. When the aperture ratio is within the first range, the amplitude value of the AC voltage is the first value, and when the aperture ratio is within the second range, the amplitude value of the AC voltage is a second value different from the first value. The value of

The vacuum cleaner according to the present invention has an effect that the floor type can be detected without using an optical floor detection sensor.

The basic block diagram of the vacuum cleaner provided with the motor drive device in the embodiment The block diagram which shows the structure of the motor drive system containing the motor drive device in embodiment Circuit diagram of single-phase inverter shown in FIG. The block diagram which shows the carrier comparison part implement | achieved by the processor shown in FIG. 2, and the carrier production | generation part shown in FIG. The block diagram which shows in detail the structure of the carrier comparison part and carrier generation part which are shown in FIG. FIG. 5 is a time chart showing waveform examples of a positive voltage command, a negative voltage command, a pulse width modulation (PWM) signal, and an inverter output voltage. The figure which shows the change of the inverter output voltage according to the modulation factor The block diagram which shows the function structure for calculating the advance angle phase input into the carrier comparison part shown in FIG.4 and FIG.5 The figure which shows an example of the calculation method of the advance angle phase in embodiment A position sensor signal output from the position sensor shown in FIG. 2, a rotor mechanical angle that is an angle from the reference position of the rotor shown in FIG. 2, a reference phase that is a phase obtained by converting the rotor mechanical angle into an electrical angle, The figure which shows the relationship with the voltage command shown in FIG. The figure which shows the time change of the voltage amplitude command in embodiment The figure which uses for description of the definition of the aperture ratio in embodiment The figure which serves for description of the control method of the voltage command in the embodiment The figure which uses for description of the calculation method of the aperture ratio in embodiment The figure which serves for description of the control method of the advance angle phase in the embodiment FIG. 3 is a schematic cross-sectional view showing a schematic structure of a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor) that can be used as a switching element shown in FIG. The 1st figure which shows the path | route of the motor current by the polarity of the inverter output voltage output from the single phase inverter shown in FIG. 2nd figure which shows the path | route of the motor current by the polarity of the inverter output voltage output from the single phase inverter shown in FIG. 3rd figure which shows the path | route of the motor current by the polarity of the inverter output voltage output from the single phase inverter shown in FIG. The figure for demonstrating the modulation control in the motor drive device which concerns on embodiment

Hereinafter, a vacuum cleaner according to an embodiment of the present invention will be described in detail with reference to the drawings. The present invention is not limited to the following embodiments.

Embodiment.
FIG. 1 is a basic configuration diagram of a vacuum cleaner according to an embodiment. Moreover, FIG. 2 is a block diagram which shows a function structure when the vacuum cleaner which concerns on embodiment is seen as a motor drive system. As shown in FIG. 1, the vacuum cleaner 1 includes a suction port body 63, an extension pipe 62, and a cleaner body 6. The suction port body 63 is for sucking together dust and dust (hereinafter simply referred to as “dust”) on the surface to be cleaned such as the floor surface together with air. A suction port 69 that opens downward is formed on the lower surface of the suction port body 63. The suction port body 63 sucks dust together with air from the suction port 69. The extension pipe 62 is connected between the suction port body 63 and the cleaner body 6 and is made of a straight member having a hollow cylindrical shape.

The vacuum cleaner body 6 is for separating dust from the air taken in and discharging the air from which the dust has been removed. The cleaner body 6 accommodates the motor driving device 2, the electric blower 64, and the dust collecting chamber 65. The motor driving device 2 is a device for driving the electric blower 64, and the electric blower 64 has a single-phase motor (not shown) and blades directly connected to the single-phase motor for generating high-speed airflow. Air containing dust is sucked from the suction port 69 by the high-speed airflow generated by driving the electric blower 64. The dust collection chamber 65 is for separating dust from air containing dust and temporarily storing the separated dust.

Further, the cleaner body 6 is provided with an operation unit 66 and a battery 67. The operation unit 66 is for the user of the vacuum cleaner 1 to hold and operate, and the operation unit 66 is provided with an operation switch (not shown) for the user to operate the vacuum cleaner 1. ing. The battery 67 is a DC power source that supplies DC power to the motor driving device 2.

As shown in FIG. 2, the motor driving device 2 is connected to a battery 67 as a power source and the single-phase motor 12 provided in the electric blower 64 shown in FIG. 1. A voltage sensor 20 is provided between the battery 67 and the motor drive device 2. A current sensor 22 is provided between the motor driving device 2 and the single-phase motor 12. An example of the single phase motor 12 is a brushless motor. The motor drive device 2 drives the single-phase motor 12 by supplying AC power to the single-phase motor 12. The voltage sensor 20 is a sensor that detects a DC voltage V _dc applied from the battery 67 to the motor driving device 2. The position sensor 21 is a sensor that detects a rotor magnetic pole position that is a magnetic pole position of the rotor 12 a built in the single-phase motor 12. The current sensor 22 is a sensor that detects a motor current that is a current flowing through the single-phase motor 12.

The vacuum cleaner 1 may be configured to have a plurality of operation modes. One of the plurality of operation modes is a “normal operation mode”, and the other one of the plurality of operation modes is a “save operation mode”. The “normal operation mode” is an operation mode in which operation is performed at the motor rotation speed during normal operation. The “save operation mode” is an operation mode in which power consumption is suppressed by setting the motor rotation speed lower than the motor rotation speed during normal operation.

The switching between the “normal operation mode” and the “save operation mode” may be performed by a changeover switch (not shown) or may be performed by control of the motor drive device 2 according to the remaining battery level.

In the present embodiment, the voltage sensor 20 detects the DC voltage V _dc , but the detection target is not limited to the DC voltage V _dc . The detection target may be an inverter output voltage that is an output voltage of the motor drive device 2. “Inverter output voltage” has the same meaning as “motor applied voltage” described later.

Further, the position sensor 21 may be any sensor that can detect the rotor magnetic pole position. A position detection element such as a Hall IC and a Hall element, or a circuit that detects the rotor magnetic pole position from the motor induced voltage may be used. The motor induced voltage is a voltage induced in a winding (not shown) in the stator 12b of the single phase motor 12.

Next, the internal configuration of the motor drive device 2 will be described. As shown in FIG. 2, the motor drive device 2 includes a single-phase inverter 11, a control unit 25, an analog / digital converter 30, and a drive signal generation unit 32.

The single-phase inverter 11 is connected to the single-phase motor 12 and applies an AC voltage to the single-phase motor 12. The analog-digital converter 30 converts analog data, which is the DC voltage V _dc detected by the voltage sensor 20, into digital data. The analog / digital converter 30 converts the analog data of the motor current detected by the current sensor 22 into digital data.

The control unit 25 generates pulse width modulation (PWM) signals Q1, Q2, Q3, and Q4, which are signals for controlling the motor current into a sine wave. The drive signal generation unit 32 generates a drive signal for driving the switching elements in the single-phase inverter 11 based on the PWM signals Q1, Q2, Q3, and Q4 output from the control unit 25.

The control unit 25 generates PWM signals Q1, Q2, Q3, and Q4 based on the DC voltage converted by the analog-digital converter 30 and the position sensor signal that is the rotational position detection signal output from the position sensor 21. To do. The position sensor signal is a binary digital signal that changes in accordance with the direction of the magnetic flux generated in the rotor 12a.

Next, the internal configuration of the control unit 25 will be described. The control unit 25 includes a processor 31, a carrier generation unit 33, and a memory 34.

The processor 31 is a processing unit that performs various calculations related to PWM control and advance angle control. A function of the carrier comparison unit 38 and a function of the advance phase calculation unit 44 described later are realized by the processor 31. The processor 31 may be called a CPU (Central Processing Unit), a microprocessor, a microcomputer, or a DSP (Digital Signal Processor).

The memory 34 stores a program read by the processor 31. The memory 34 is used as a work area when the processor 31 performs arithmetic processing. The memory 34 is generally a nonvolatile or volatile semiconductor memory such as a RAM (Random Access Memory), a flash memory, an EPROM (Erasable Programmable ROM), or an EEPROM (registered trademark) (Electrically EPROM). Details of the configuration of the carrier generation unit 33 will be described later.

The drive signal generation unit 32 converts the PWM signals Q1, Q2, Q3, and Q4 output from the processor 31 into drive signals for driving the single-phase inverter 11, and outputs the drive signals to the single-phase inverter 11.

When the single-phase motor 12 is a brushless motor, a plurality of permanent magnets (not shown) are arranged in the circumferential direction on the rotor 12a of the single-phase motor 12. The plurality of permanent magnets are arranged so that the magnetization direction is alternately reversed in the circumferential direction, and form a plurality of magnetic poles of the rotor 12a. A winding (not shown) is wound around the stator 12 b of the single-phase motor 12. Hereinafter, the winding of the stator 12b is referred to as “stator winding”. The alternating current flowing through the stator winding corresponds to the “motor current” described above. In the present embodiment, it is assumed that the number of magnetic poles of the rotor 12a is four, but the number of magnetic poles of the rotor 12a may be other than four.

FIG. 3 is a circuit configuration diagram of the single-phase inverter shown in FIG. The single-phase inverter 11 has a plurality of switching

elements

51, 52, 53, and 54 that are bridge-connected. Each of the two switching

elements

51 and 53 located on the high potential side is referred to as an upper arm switching element. Each of the two switching

elements

52 and 54 located on the low potential side is referred to as a lower arm switching element. The connection end 11a between the switching element 51 and the switching element 52 and the connection end 11b between the switching element 53 and the switching element 54 constitute an AC end in the bridge circuit. A single-phase motor 12 is connected to the connection ends 11a and 11b.

Each of the plurality of switching

elements

51, 52, 53, 54 is a MOSFET which is a metal oxide semiconductor field effect transistor. MOSFET is an example of FET (Field-Effect Transistor).

In the switching element 51, a body diode 51a connected in parallel between the drain and source of the switching element 51 is formed. In the switching element 52, a body diode 52a connected in parallel between the drain and source of the switching element 52 is formed. In the switching element 53, a body diode 53a connected in parallel between the drain and source of the switching element 53 is formed. The switching element 54 is formed with a body diode 54 a connected in parallel between the drain and source of the switching element 54. Each of the plurality of

body diodes

51a, 52a, 53a, 54a is a parasitic diode formed inside the MOSFET, and is used as a freewheeling diode.

At least one of the plurality of switching

elements

51, 52, 53, and 54 is not limited to a MOSFET formed of a silicon-based material, but is formed of a wide band gap semiconductor such as silicon carbide, a gallium nitride-based material, or diamond. A MOSFET may be used.

Generally, wide band gap semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using a wide band gap semiconductor for at least one of the plurality of switching

elements

51, 52, 53, and 54, the withstand voltage and allowable current density of the switching element are increased, and the semiconductor module incorporating the switching element Can be miniaturized. In addition, since the wide band gap semiconductor has high heat resistance, it is possible to reduce the size of the heat radiating portion for radiating the heat generated in the semiconductor module. Furthermore, the wide band gap semiconductor can simplify the heat dissipation structure that dissipates heat generated in the semiconductor module.

FIG. 4 is a block diagram showing a carrier comparison unit realized by the processor shown in FIG. 2 and a carrier generation unit shown in FIG. FIG. 5 is a block diagram showing in detail the configuration of the carrier comparison unit and the carrier generation unit shown in FIG. As described above, the function of generating the PWM signals Q1, Q2, Q3, and Q4 can be realized by the carrier generation unit 33 and the carrier comparison unit 38 illustrated in FIG.

In FIG. 4, the carrier comparison unit 38 receives an advance angle phase θ _v subjected to advance angle control and a reference phase θ _e used when generating a voltage command V _m described later. The reference phase θ _e is a phase obtained by converting the rotor mechanical angle θ _m that is an angle from the reference position of the rotor 12a into an electrical angle. Here, “advance angle phase” represents “advance angle”, which is the “advance angle” of the voltage command, in terms of phase. The “advance angle” referred to here is a phase difference between the motor applied voltage applied to the stator winding by the single-phase inverter 11 and the motor induced voltage induced in the stator winding. When the motor applied voltage is ahead of the motor induced voltage, the “lead angle” takes a positive value. The motor induced voltage is a signal synchronized with the output signal of the position sensor 21. For this reason, the “lead angle” is also a phase difference between the position sensor signal and the motor applied voltage. The calculation of the advanced angle phase theta _v, it is necessary to information about the voltage applied to the motor, it may be used motor current instead of the voltage applied to the motor. In other words, the phase difference between the position sensor signal and the motor current may be the “advance angle”.

In addition, the carrier comparison unit 38 includes the advance value θ _v and the reference phase θ _e , the carrier generated by the carrier generation unit 33, the DC voltage V _dc, and the amplitude value of the voltage command V _m. A voltage amplitude command V * is input. The carrier comparison unit 38 generates PWM signals Q1, Q2, Q3, and Q4 based on the carrier, the advance angle phase θ _v , the reference phase θ _e , the DC voltage V _dc, and the voltage amplitude command V *.

As illustrated in FIG. 5, a carrier frequency f _C [Hz] that is a carrier frequency is set in the carrier generation unit 33. At the tip of the arrow of the carrier frequency f _C , as an example of the carrier waveform, a waveform of a triangular wave carrier that goes up and down between “0” and “1” is shown. Note that the PWM control of the single-phase inverter 11, and the synchronous PWM control, there are asynchronous PWM control, when the asynchronous PWM control, the control for synchronizing the carrier to advance the phase theta _v is not necessary.

As shown in FIG. 5, the carrier comparison unit 38 includes an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38d, an addition unit 38e, a multiplication unit 38f, a comparison unit 38g, a comparison unit 38h, and an output inversion unit. 38i and an output inverting unit 38j.

The absolute value calculation unit 38a calculates the absolute value | V * | of the voltage amplitude command V *. In the dividing unit 38b, the absolute value | V * | is divided by the DC voltage V _dc detected by the voltage sensor 20. The battery voltage, which is the output voltage of the battery 67, varies as the current continues to flow. By dividing the absolute value | V * | by the DC voltage _Vdc , the motor applied voltage does not decrease due to a decrease in the battery voltage. In addition, the modulation rate can be increased.

In the multiplication unit 38c, a sine value of “θ _e + θ _v ” obtained by adding the advance phase θ _v to the reference phase θ _e is calculated. The calculated sine value of “θ _e + θ _v ” is multiplied by the output of the division unit 38b. In the multiplication unit 38d, the voltage command V _m that is the output of the multiplication unit 38c is multiplied by ½. In the adder 38e, ½ is added to the output of the multiplier 38d. The multiplication unit 38f multiplies the output of the addition unit 38e by “−1”. The output of the addition unit 38e is input to the comparison unit 38g as a positive voltage command V _m1 for driving the two switching

elements

51, 53 of the upper arm among the plurality of switching

elements

51, 52, 53, 54. , the output of the multiplication unit 38f is input to the comparison unit 38h as a voltage command _{V m2} of the negative side for driving the two switching

elements

52, 54 of the lower arm.

The comparison unit 38g compares the positive-side voltage command V _m1 with the carrier amplitude. The output of the comparison unit 38g becomes the PWM signal Q1 to the switching element 51, and the output of the output inversion unit 38i obtained by inverting the output of the comparison unit 38g becomes the PWM signal Q2 to the switching element 52. Similarly, the comparison unit 38h compares the negative-side voltage command V _m2 with the carrier amplitude. The output of the comparison unit 38h is a PWM signal Q3 to the switching element 53, and the output of the output inversion unit 38j obtained by inverting the output of the comparison unit 38h is the PWM signal Q4 to the switching element 54. The switching element 51 and the switching element 52 are not simultaneously turned on by the output inverting part 38i, and the switching element 53 and the switching element 54 are not simultaneously turned on by the output inverting part 38j.

FIG. 6 is a time chart showing waveform examples of the positive voltage command, the negative voltage command, the PWM signal, and the inverter output voltage shown in FIG. In FIG. 6, in order from the top, the waveform of the positive voltage command V _m1 output from the adder 38e, the waveform of the negative voltage command V _m2 output from the multiplier 38f, and the PWM signals Q1, Q2, and The waveforms of Q3 and Q4 and the waveform of the inverter output voltage are shown. By using the voltage commands V _m1 and V _m2 , PWM signals Q1, Q2, Q3, and Q4 are generated. The motor drive device 2 shown in FIG. 2 controls the plurality of switching

elements

51, 52, 53, and 54 in the single-phase inverter 11 by using the PWM signals Q1, Q2, Q3, and Q4, so that FIG. An inverter output voltage as shown, that is, a PWM-controlled voltage pulse is applied to the single-phase motor 12.

The modulation method used when generating the PWM signals Q1, Q2, Q3, and Q4 includes bipolar modulation that outputs a voltage pulse that changes at a positive or negative potential, and a voltage that changes at three potentials every half cycle of the power supply. There is known a pulse, that is, unipolar modulation that outputs a voltage pulse that changes to a positive potential, a negative potential, and a zero potential. The waveform shown in FIG. 6 is based on unipolar modulation. In the motor drive device 2 of the present embodiment, any modulation method may be used. In applications where it is necessary to control the motor current waveform to a sine wave, it is preferable to employ unipolar modulation with a lower harmonic content than bipolar modulation.

FIG. 7 is a diagram illustrating a change in the inverter output voltage in accordance with the modulation rate. The upper part of FIG. 7 shows the voltage command V _m when the modulation factor is 1.0, the carrier, and the inverter output voltage. The middle part of FIG. 7 shows the voltage command V _m , the carrier, and the inverter output voltage when the modulation factor = 1.2. The lower part of FIG. 7 shows the voltage command V _m , the carrier, and the inverter output voltage when the modulation rate is 2.0.

As described with reference to FIG. 5, the voltage command V _m1 on the positive side is compared with the carrier amplitude in the comparison unit 38g, and the voltage command V _m2 on the negative side is compared with the carrier amplitude in the comparison unit 38h. When the voltage commands V _m1 and V _m2 are larger than the carrier amplitude, the switching element of the single-phase inverter 11 is turned on. When the voltage commands V _m1 and V _m2 are smaller than the carrier amplitude, the switching element of the single-phase inverter 11 is turned off. By these operations, an inverter output voltage subjected to PWM control as shown in FIG. 6 is applied to the single-phase motor 12.

There are various definitions of the modulation rate. In this specification, the ratio between the voltage amplitude command V * and the carrier amplitude, that is, “voltage amplitude command V * / carrier amplitude” is defined as the modulation rate. To do. The upper part of FIG. 7 shows a waveform when the modulation rate is 1.0, but the same waveform is obtained when the modulation rate is less than 1.0. When the modulation factor is less than 1.0, an inverter output voltage is generated according to the carrier frequency, and therefore, the inverter output voltage also outputs a voltage pulse according to the carrier frequency.

On the other hand, when the modulation rate exceeds 1.0, the waveforms are as shown in the middle and lower parts of FIG. When the modulation rate exceeds 1.0, it is called “overmodulation”, and the region where the modulation rate exceeds 1.0 is called “overmodulation region”. In the overmodulation region, since the voltage command V _m exceeds the carrier amplitude, there is a section in which an inverter drive signal cannot be generated according to the carrier frequency. In this section, since the inverter output voltage is fixed to a positive power supply voltage or a negative power supply voltage, the inverter output voltage can obtain a larger output voltage than when the modulation factor is 1.0.

Next, we will explain the problems and countermeasures when using a battery as a power source.

A battery is a structure that exhibits the nature of internal impedance. For this reason, the battery output voltage varies greatly according to the current output from the battery. Specifically, it is known that when a current of 20 [A] flows in a battery of 20 [V], the battery output voltage decreases to approximately 17 [V]. Further, in the above-described region where the modulation factor is 1.0 or more, it is known that the output voltage cannot be accurately obtained with respect to the voltage command V _m because the number of output voltage pulses decreases. It has been. Furthermore, since the battery current becomes a pulsating current due to the influence of switching by the single-phase inverter 11, it is known that the voltage output from the battery also pulsates. In order to solve these problems, if the advance angle is controlled so as to be sequentially changed without being a constant value, the variation in the voltage applied from the battery to the single-phase inverter 11 and the voltage output from the single-phase inverter 11 Both variations can be suppressed.

Next, the advance angle control in the present embodiment will be described. FIG. 8 is a block diagram illustrating a functional configuration for calculating an advance phase input to the carrier comparison unit illustrated in FIGS. 4 and 5. The function for calculating the advance angle phase θ _v can be realized by a rotation speed calculation unit 42 and an advance angle phase calculation unit 44 as shown in FIG.

The rotation speed calculation unit 42 calculates the rotation speed ω of the single-phase motor 12 based on the position sensor signal detected by the position sensor 21. In addition, the rotation speed calculation unit 42 calculates a reference phase θ _{e obtained} by converting a rotor mechanical angle θ _m that is an angle from the reference position of the rotor 12a into an electrical angle. The advance phase calculation unit 44 calculates the advance phase θ _v based on the rotation speed ω and the reference phase θ _e calculated by the rotation speed calculation unit 42 and the position sensor signal.

FIG. 9 is a diagram illustrating an example of a method of calculating the advance phase according to the embodiment. The horizontal axis of FIG. 9 shows the rotational speed N, the vertical axis of FIG. 9 shows an advanced phase theta _v. Note that when the rotation speed is expressed in units of [rpm], the notation “N” is used. As shown in FIG. 9, the advance phase θ _v can be determined using a function in which the advance phase θ _v increases as the rotational speed N increases. Further, in FIG. 9, the advance angle adjustment width Δθdel indicates a variation range of the attachment position of the position sensor 21. In the example of FIG. 9, and determines the advance angle phase theta _v by first-order linear function, but is not limited to first-order linear function. If either advanced angle phase theta _v according to an increase of the rotational speed N is equal or greater relationship may be used functions other than first-order linear function.

10 shows a position sensor signal output from the position sensor shown in FIG. 2, a rotor mechanical angle that is an angle from the reference position of the rotor 12a shown in FIG. 2, and a phase obtained by converting the rotor mechanical angle into an electrical angle. FIG. 5 is a diagram showing a relationship between a certain reference phase and a voltage command shown in FIG. 4. The bottom portion of Figure 10, the rotor mechanical angle theta _m when the rotor 12a is rotated in the clockwise direction is 0 °, 45 °, 90 ° , the single-phase motor 12 is 135 ° and 180 ° are shown . The rotor 12a of the single phase motor 12 is provided with four magnets. Four teeth 12b1 are provided on the outer periphery of the rotor 12a. If the rotor 12a is rotated clockwise, the position sensor signals corresponding to the rotor mechanical angle theta _m is detected. The control unit 25 calculates a reference phase θ _e converted into an electrical angle in accordance with the detected position sensor signal.

The voltage command V _m shown as “Example 1” in the middle part of FIG. 10 is a voltage command in the case of the advance angle phase θ _v = 0. When the advance angle phase θ _v = 0, a sinusoidal voltage command V _m having the same phase as the reference phase θ _e is output. The amplitude of the voltage command V _m at this time is determined based on the voltage amplitude command V * as described above.

The voltage command V _m shown as “Example 2” in the middle part of FIG. 10 is a voltage command in the case of the advance angle phase θ _v = π / 4. When the advance angle phase θ _v = π / 4, a sine-wave voltage command V _m advanced by π / 4, which is a component of the advance angle phase θ _v , from the reference phase θ _e is output.

Next, how to give the voltage amplitude command V * will be described. FIG. 11 is a diagram illustrating a time change of the voltage amplitude command V * in the embodiment. In the present embodiment, the voltage amplitude command V * is an operation mode that changes stepwise according to time, as shown. More specifically, first, a predetermined first voltage V ₁ set in advance is applied at the time of startup. In startup period, the first voltage V ₁ is maintained. In the acceleration section, the voltage amplitude command V * is increased so as to obtain a preset acceleration rate. After the acceleration, when shifting to the steady operation section, the increase of the voltage amplitude command V * is stopped. In the steady operation section, the voltage amplitude command V * at the time of acceleration stop is maintained. Thus, in the steady operation period, the second voltage V ₂ is applied a larger constant than the first voltage V _1. In other words, in the present embodiment, the voltage amplitude command V * is controlled to be constant in the start-up section and the steady operation section. Incidentally, in the start-up interval, the time τ1 providing a first voltages V ₁ may be set to any time in consideration of the stabilization time of the control system.

Next, the effect of the voltage amplitude command V * being constant will be described. The following effects can be obtained by maintaining the voltage amplitude command V * constant in the steady operation section.

(1) Even when the load suddenly changes, a constant voltage command can be output based on the phase detected from the position sensor signal.
(2) Since the voltage amplitude is not affected even when the rotation speed fluctuates, the output voltage can be kept stable.

In addition, in the case of the vacuum cleaner 1, the “load” referred to here means a closed state of the suction port body 63. The closed state of the suction port body 63 is related to a gap generated between the suction port body 63 and the floor surface which is a cleaning surface. This gap can be defined by the concept of “aperture ratio”. Details of the aperture ratio will be described later.

By the way, in the constant rotation speed control performed in a general vacuum cleaner, an overcurrent may flow through the motor. The reason why the overcurrent flows is that the current fluctuates abruptly in order to keep the rotation speed constant when the load fluctuates. More specifically, if the rotational speed constant control is performed at the time of transition from “light load”, that is, “load torque is small” to “heavy load”, that is, “load torque is large”, it is the same. This is because the motor output torque must be increased to maintain the rotation speed, and the amount of change in the motor current increases.

On the other hand, in the present embodiment, as described above, in the steady operation section, the voltage amplitude command V * is controlled to be constant. Here, when the voltage amplitude command V * is constant, the voltage amplitude command V * does not change when the load becomes heavy. Therefore, the rotation speed of the motor decreases as the load torque increases. Even if the rotational speed of the motor is reduced, a sudden change in motor current and an overcurrent can be prevented, so that a sudden change in the rotational speed of the motor can be suppressed. Thereby, the vacuum cleaner which rotates stably is realizable.

In the case of a vacuum cleaner, the load torque increases with an increase in the rotational speed of the blade, which is the load of the single-phase motor, and also increases with an increase in the diameter of the air passage. The diameter of the air passage means the width of the suction port of the suction port body when an electric vacuum cleaner is taken as an example. When the suction port is wide, for example, when nothing is in contact with the suction port, a force for sucking wind is required, so that the load torque when the blades are rotating at the same rotational speed is increased. On the other hand, when the suction port is narrow, for example, when the suction port is in contact with some object and blocked, the load when the blades are rotating at the same rotation speed is not necessary because the force to suck in the wind is not necessary. Torque is reduced.

Next, the aperture ratio will be described. FIG. 12 is a diagram for explaining the definition of the aperture ratio in the embodiment. Here, the “opening ratio” in the embodiment is a percentage that represents the degree of a gap formed between the suction port 69 of the suction port body 63 and the cleaning surface shown in FIG. FIG. 12 schematically shows the suction port body 63 of the electric vacuum cleaner 1 shown in FIG. 1, the extension pipe 62 connected to the suction port body 63, and the air passage 72 in the extension pipe 62. Yes. Solid arrows in the figure indicate the flow of air. In addition, in FIG. 12, the suction inlet of the suction inlet body 63 is not shown with the code | symbol.

As shown on the left side of FIG. 12, a state where the suction port of the suction port body 63 is closed and sealed by the floor surface 76 is defined as an opening ratio = 0%. The sealed state is a state where air does not flow. On the other hand, as shown on the right side of FIG. 12, the case where the suction port of the suction port body 63 is not in contact with the floor surface 76 is defined as the opening ratio = 100%. In the center of FIG. 12, the middle of the two, that is, the state of the aperture ratio = 50% is shown.

開口 The aperture ratio varies depending on the floor type. In addition to the floor surface type, the opening ratio changes depending on the suction mode such as when the suction port is clogged with dust or when the suction port sucks the curtain. Therefore, in this specification, the range of the aperture ratio corresponding to the floor type or the suction mode is defined as follows.

-Opening ratio when suction material is clogged in the suction port body 63: 0 to 10%
・ Opening ratio when floor 76 is carpet: 10-50%
・ Opening ratio when floor 76 is flooring: 50-80%
-Opening ratio in a state where the suction port of the suction port body 63 is separated from the floor surface 76: 80 to 100%

Note that the opening ratio according to the floor type or suction mode is affected by the structure of the suction port body of the vacuum cleaner or the degree of dust clogging at the suction port. For this reason, the numerical value shown here is an example to the last.

As described above, in the present embodiment, control is performed with the voltage amplitude command V * kept constant during steady operation. When this control is performed, the rotational speed of the single-phase motor 12 changes according to load fluctuations. In other words, the load fluctuation can be indirectly observed by observing the rotational speed of the single-phase motor 12. Here, the load fluctuation appears as a result of the fluctuation of the aperture ratio. The variation in the aperture ratio is a variation in floor type or a variation in suction mode. Therefore, by observing the motor rotation speed, it is possible to recognize a change in floor type or a change in suction mode. In other words, the control according to the present embodiment in which the voltage amplitude command V * is constant makes it possible to incorporate control according to the floor type and suction mode.

FIG. 13 is a diagram for explaining a voltage command control method according to the embodiment. The horizontal axis of FIG. 13 represents the rotation speed N, the vertical axis of FIG. 13 illustrates an advanced phase theta _v and the voltage amplitude command V *.

In the example of FIG. 13, the control area corresponding to the rotation speed and the aperture ratio is divided into four areas from area 1 to area 4. Region 1 assumes that the opening ratio is 80% to 100%, that is, the state where the suction port 63 is not touching the floor surface. Region 2 assumes that the aperture ratio is 50% to 80%, that is, the floor is flooring. Region 3 assumes that the opening ratio is 10% to 50%, that is, the floor is a carpet. The region 4 assumes that the opening ratio is 0% to 10%, that is, the suction port body 63 is in a sealed state, or the suction port body 63 is clogged with suction.

As the rotational speed N increases, the advance angle phase θ _v changes as shown by a thick broken line in FIG. A voltage amplitude command V * is set according to the rotation speed N.

The transition from the region 1 to the region 2 is performed when the aperture ratio falls below 80% that is the threshold value between the regions. When the opening ratio is less than 80%, it is considered that the suction port body 63 of the vacuum cleaner 1 has moved from the state where it does not touch the floor surface 76 to the floor surface 76 which is a flooring. In this case, the rotation speed N is further increased by changing the voltage amplitude command V * from the first command value Vc1 to the second command value Vc2. The second command value Vc2 is larger than the first command value Vc1. At this time, as described above, increases in accordance with an increase of the rotational speed N advanced phase theta _v slope value K. Note that the transition from the region 2 to the region 1 is performed in the same manner. The transition from the region 2 to the region 1 is performed when the aperture ratio exceeds 80% which is the threshold value between the regions.

The transition from the region 2 to the region 3 is performed when the aperture ratio falls below 50% which is the threshold value between the regions. The case where the opening ratio falls below 50% is a case where the suction port body 63 of the vacuum cleaner 1 is considered to have moved from the flooring to the carpet. Carpets need more suction than flooring. For this reason, the rotational speed N is further increased by setting the voltage amplitude command V * to a larger value. Specifically, the voltage amplitude command V * is changed from the second command value Vc2 to the third command value Vc3. The third command value Vc3 is larger than the second command value Vc2. Further, as the rotational speed N increases, the advance angle phase θ _v is also increased by the inclination value K. The suction force can be further increased by changing the voltage amplitude command V * from the second command value Vc2 to the third command value Vc3. The transition from the region 3 to the region 2 is performed in the same manner. The transition from the region 3 to the region 2 is performed when the aperture ratio exceeds 50% which is the threshold value between the regions.

The transition from the region 3 to the region 4 is performed when the aperture ratio falls below 10% which is the threshold value between the regions. When the opening ratio is less than 10%, it is assumed that the suction port body 63 is clogged with suction material during cleaning of the carpet. When the suctioned object is clogged, the value of the voltage amplitude command V * is decreased in order to decrease the rotational speed N. Specifically, the voltage amplitude command V * is changed from the third command value Vc3 to the fourth command value Vc4. The fourth command value Vc4 is smaller than the first command value Vc1. Moreover, to reduce the rotational speed N, to reduce the slope of the advanced angle phase theta _v value "-K". By changing the voltage amplitude command V * to the fourth command value Vc4, an increase in the rotational speed N can be suppressed and useless heat generation in the single-phase motor 12 can be suppressed. The transition from the region 4 to the region 3 is performed in the same manner. The transition from the region 4 to the region 3 is performed when the aperture ratio exceeds 10% which is a threshold value between the regions.

It should be noted that the example shown in FIG. 13, that is, the example of changing the voltage amplitude command V * according to the aperture ratio is an example, and is not limited to this. The magnitude relationship between the voltage command values may be changed according to the application. Moreover, when there are a plurality of operation modes of the product, the first to fourth command values described above may have a plurality of values for each operation mode.

In the above description, the transition between the region 1 and the region 2, the transition between the region 2 and the region 3, and the transition between the region 3 and the region 4 are described. Transition between, transition between region 1 and region 4, and transition between region 2 and region 4 can also occur. The transition from the region 1 to the region 3 is performed when the threshold value is 50%, which is the upper limit value of the aperture ratio range in the region 3, and the aperture ratio falls below the threshold value. The transition from the region 3 to the region 1 is performed when the threshold value is 80%, which is the lower limit value of the aperture ratio range in the region 1, and the aperture ratio exceeds the threshold value. The transition from the region 1 to the region 4 is performed when the upper limit value of the aperture ratio range in the region 4 is 10% as a threshold value and the aperture ratio falls below the threshold value. The transition from the region 4 to the region 1 is performed when the threshold value is 80%, which is the lower limit value of the aperture ratio range in the region 1, and the aperture ratio exceeds the threshold value. The transition from the region 2 to the region 4 is performed when the upper limit value of the aperture ratio range in the region 4 is 10% as a threshold value and the aperture ratio falls below the threshold value. The transition from the region 4 to the region 2 is performed when the threshold value is 50% that is the lower limit value of the aperture ratio range in the region 2 and the aperture ratio exceeds the threshold value.

Next, a method for calculating the aperture ratio will be described. FIG. 14 is a diagram for explaining the calculation method of the aperture ratio in the embodiment.

FIG. 14 shows three curves composed of “Vc1”, “Vc2” and “Vc3” shown in FIG. 13 with respect to the voltage amplitude command V * which changes according to the aperture ratio. The range of the aperture ratio is as described above. “No floor” means that the suction port 63 is separated from the floor 76.

In FIG. 14, when the rotational speed is less than N1, the voltage amplitude command V * = Vc1, and the operating point moves on the thick solid line L1. When the rotational speed N1 is reached, the voltage amplitude command V * = Vc2, and the operating point moves on the thick solid line L2. At this time, the rotational speed N reaches N2. At a rotational speed of N2 or more and less than N3, the operating point moves on the thick solid line L2. When the rotation speed N3 is reached, the voltage amplitude command V * = Vc3, and the operating point moves on the thick solid line L3. At this time, the rotational speed N reaches N4. At a rotational speed of N4 or higher, the operating point moves on the thick solid line L3.

As described above, when the rotation speed N is changed, the relationship of the voltage amplitude command V * to the aperture ratio changes. For this reason, the relationship between the rotational speed N and the voltage amplitude command V * is grasped in advance and held in a table (not shown) in the control unit 25. In the example shown in FIG. 14, the voltage amplitude command V * = Vc1 when the rotational speed is an aperture ratio smaller than N1 = 100% to 80%. Similarly, when the rotational speed is N1 to N3, the aperture ratio is 80% to 50%, and the voltage amplitude command V * = Vc2. When the rotation speed is N3 or higher, the aperture ratio is 50% to 10%, and the voltage amplitude command V * = Vc3. In addition, when the rotational speed suddenly rises and falls, the rotational speed is larger than the rotational speed after the sudden rise, and may be smaller than the rotational speed after the sudden drop. The aperture ratio does not correspond to 14 thick solid lines. Based on the rotation speed ω calculated by the rotation speed calculation unit 42 shown in FIG. 8, by referring to a table in which the relationship between the aperture ratio, the rotation speed N, and the voltage amplitude command V * is maintained, the aperture ratio can be calculated. The corresponding voltage amplitude command V * can be calculated.

In the example of FIG. 13, the gradient of increasing the advance phase theta _v is a constant, depending on the range of the aperture ratio, in the range of the aperture ratio has output a voltage amplitude command V * is a constant value However, the present invention is not limited to this. Voltage amplitude command V * is a constant, may be changed inclination increases the advanced angle phase theta _v. Hereinafter, this method will be described with reference to FIG. That is, FIG. 15 is a diagram for explaining the control method of the advance phase in the embodiment.

In FIG. 15, the transition from region 1 to region 2 is performed when the aperture ratio falls below 80%, which is the threshold value between the regions. At this time, no voltage amplitude command V * is changed, the slope of increasing the advance phase theta _v is changed from the first inclination value K1 to the second inclination value K2. Both the first slope value K1 and the second slope value K2 are positive. The second slope value K2 is larger than the first slope value K1. Note that the transition from the region 2 to the region 1 is performed in the same manner. The transition from the region 2 to the region 1 is performed when the aperture ratio exceeds 80% which is the threshold value between the regions. At this time, the slope of increasing the advance phase theta _v is changed from the second inclination value K2 to the first inclination value K1.

The transition from the region 2 to the region 3 is performed when the aperture ratio falls below 50% which is the threshold value between the regions. In this case, no voltage amplitude command V * is changed, the slope of increasing the advance phase theta _v is changed from the second inclination value K2 in a third inclination value K3. Both the second slope value K2 and the third slope value K3 are positive. The third inclination value K3 is larger than the second inclination value K2. The transition from the region 3 to the region 2 is performed in the same manner. The transition from the region 3 to the region 2 is performed when the aperture ratio exceeds 50% which is the threshold value between the regions. At this time, the slope of increasing the advance phase theta _v is changed from the third slope value K3 to the second inclination value K2.

The transition from the region 3 to the region 4 is performed when the aperture ratio falls below 10% which is the threshold value between the regions. In this case, no voltage amplitude command V * is changed, the slope of increasing the advance phase theta _v is changed from the third slope value K3 to a fourth inclination value K4. The fourth slope value K4 is negative. The transition from the region 4 to the region 3 is performed in the same manner. The transition from the region 4 to the region 3 is performed when the aperture ratio exceeds 10% which is a threshold value between the regions. At this time, the slope of increasing the advance phase theta _v is changed from the fourth inclination value K4 in the third inclination value K3.

Note that modification of example, i.e. the slope value of the advanced angle phase theta _v corresponding to the aperture ratio shown in FIG. 15 is an example, not limited to this. Magnitude relationship between the gradient values of the advanced angle phase theta _v in each region may be changed depending on the application. Moreover, when there are a plurality of operation modes of the product, the first to fourth inclination values described above may have a plurality of values for each operation mode.

Next, the effect of the advance angle control will be described. First, if to increase the advance phase theta _v according to an increase of the rotational speed N, it is possible to widen the rotational speed range. The advance phase theta _v when 0, the rotational speed N is saturated where the voltage applied to the motor and the motor induced voltage is balanced. To further increase the rotational speed N advances the advanced angle phase theta _v, suppresses motor induced voltage by weakening the magnetic flux to be generated in the stator 12b by the armature reaction, increasing the rotational speed N. Therefore, by selecting the advance angle phase θ _{v according} to the rotation speed N, a wider rotation speed region can be obtained.

Next, the effect of providing the advance angle adjustment width Δθdel for the advance angle control will be described. First, by providing the advance angle adjustment width Δθdel, even if a position shift of the position sensor 21 occurs at the time of manufacture or a characteristic shift of sensitivity inherent to the position sensor 21 occurs, Deviation or characteristic deviation can be corrected. Thereby, the stable specific rotational speed N can be obtained. Further, since the correction of the positional deviation or the characteristic deviation can be omitted from the manufacturing process, the generation of the cost for correcting or adjusting the positional deviation or the characteristic deviation can be suppressed.

As described above, according to the advance angle control according to the present embodiment, when the opening ratio of the suction port body 63 is in the first range, the voltage amplitude command V * is kept constant and the rotation speed N is increased. increasing the advance phase theta _v is a lead angle of the voltage command in accordance with the. This control enables stable driving in a wide rotational speed range. Further, by providing the advance angle adjustment width Δθdel, the influence on the rotational speed N can be suppressed even when the position sensor 21 is displaced.

Further, according to the advance angle control according to the present embodiment, when the opening ratio of the suction port body 63 changes from the first range to the second range, the voltage amplitude command V * corresponds to the first range. The first command value is changed to a second command value corresponding to the second range. Thereby, the control according to the fluctuation | variation of the floor surface kind or the fluctuation | variation of a suction aspect is attained, and the preferable suction | attraction force according to an aperture ratio can be obtained. Note that changing the voltage amplitude command V * from the first command value corresponding to the first range to the second command value corresponding to the second range means that the single-phase inverter 11 to the single-phase motor 12 This is equivalent to changing the applied voltage from the first amplitude value corresponding to the first range to the second amplitude value corresponding to the second range.

Incidentally, if the advanced angle phase theta _v is set by the function according to the rotational speed N, it is possible to change the rotation speed by changing the slope of the straight line or curve indicated by function. That is, when the opening ratio of the suction port body 63 is in the first range, the slope of the straight line or curve representing the advanced angle phase theta _v is set to a first inclination value, the opening ratio of the suction port body 63 of the second for range, it sets the slope of the straight line or curve representing the advanced angle phase theta _v to the second gradient value. Thus, it is possible to calculate the advance phase theta _v quickly. Incidentally, the function representing the advanced angle phase theta _v be a curve, it is possible to use the gradient of the tangent at the control point of the curve.

Next, the loss reduction technique in the present embodiment will be described with reference to the drawings of FIGS. FIG. 16 is a schematic cross-sectional view showing a schematic structure of a MOSFET. FIG. 16 illustrates an n-type MOSFET. FIG. 17 is a first diagram illustrating a motor current path according to the polarity of the inverter output voltage output from the single-phase inverter illustrated in FIG. 3. FIG. 18 is a second diagram illustrating a motor current path according to the polarity of the inverter output voltage. FIG. 19 is a third diagram illustrating a motor current path according to the polarity of the inverter output voltage.

In the case of an n-type MOSFET, a p-type semiconductor substrate 80 is used as shown in FIG. A source electrode 82, a drain electrode 83, and a gate electrode 84 are formed on the semiconductor substrate 80 having the p-type region 81. At a portion in contact with the source electrode 82 and the drain electrode 83, an n-type region 85 is formed by ion implantation of a high concentration impurity. In the p-type semiconductor substrate 80, an oxide insulating film 86 is formed between a portion where the n-type region 85 is not formed and the gate electrode 84. That is, the oxide insulating film 86 is interposed between the gate electrode 84 and the p-type region 81 in the semiconductor substrate 80.

When a positive voltage is applied to the gate electrode 84, electrons are attracted to the interface between the p-type region 81 and the oxide insulating film 86 in the semiconductor substrate 80, and are negatively charged. Where the electrons gather, the electron density is higher than that of the holes and becomes n-type. This n-type portion becomes a current path and is called a channel. The example of FIG. 16 is an example where the n-type channel 87 is formed. In the case of a p-type MOSFET, a p-type channel is formed.

When the polarity of the inverter output voltage is positive, the current flows into the single-phase motor 12 through the channel of the switching element 51, which is the upper arm of the first phase, as shown by the thick solid line (a) in FIG. It flows out of the single-phase motor 12 through the channel of the switching element 54 which is a two-phase lower arm. When the polarity of the inverter output voltage is negative, current flows into the single-phase motor 12 through the channel of the switching element 53, which is the upper arm of the second phase, as shown by the thick broken line (b) in FIG. And flows out of the single-phase motor 12 through the channel of the switching element 52 which is the lower arm of the first phase.

Next, the current path when the inverter output voltage is zero, that is, when zero voltage is output from the single-phase inverter 11, will be described. When the inverter output voltage becomes zero after the positive inverter output voltage is generated, no current flows from the power source side, as shown by a thick solid line (c) in FIG. 18, and the single-phase inverter 11 and the single-phase motor 12 It becomes the recirculation | reflux mode in which an electric current passes between. At this time, since the direction of the current that flows immediately before the single-phase motor 12 does not change, the current that flows from the single-phase motor 12 flows between the channel of the switching element 54 that is the lower arm of the second phase and the first phase. It returns to the single-phase motor 12 through the body diode 52a of the switching element 52 which is the lower arm. In addition, when the inverter output voltage becomes zero after the negative inverter output voltage is generated, the direction of the current that has flowed immediately before is reversed, so that as shown by the thick broken line (d) in FIG. The direction of the current is reversed. More specifically, the current flowing out of the single-phase motor 12 passes through the body diode 51a of the switching element 51 that is the upper arm of the first phase and the channel of the switching element 53 that is the upper arm of the second phase. Return to the single-phase motor 12.

As described above, in the reflux mode in which the current flows back between the single-phase motor 12 and the single-phase inverter 11, a current flows through the body diode in one of the first phase and the second phase. Generally, it is known that the conduction loss is smaller when a current is passed through a MOSFET channel than when a current is passed in the forward direction of a diode. Therefore, in the present embodiment, in the return mode in which the return current flows, the MOSFET on the side having the body diode is controlled to be turned on in order to reduce the current flowing through the body diode.

In the reflux mode, the switching element 52 is controlled to be turned on at the timing when the reflux current shown by the thick solid line (c) in FIG. 18 flows. If controlled in this way, as indicated by a thick solid line (e) in FIG. 19, most of the return current flows through the channel side of the switching element 52 having a small resistance value. Thereby, the semiconductor loss in the switching element 52 is reduced. Further, the switching element 51 is controlled to be turned on at the timing when the return current indicated by the thick broken line (d) in FIG. 18 flows. If controlled in this way, as indicated by a thick broken line (f) in FIG. 19, most of the reflux current flows through the channel side of the switching element 51 having a small resistance value. Thereby, the semiconductor loss in the switching element 51 is reduced.

As described above, the loss of the switching element can be reduced by turning on the MOSFET on the side having the body diode at the timing when the reflux current flows through the body diode. For this reason, the structure of the MOSFET is made a surface mount type so that heat can be radiated on the substrate, and if necessary, part or all of the switching element is formed of a wide band gap semiconductor, so that the MOSFET generates heat only on the substrate. The structure which suppresses is realized. Note that if heat can be radiated only by the substrate, a heat sink is unnecessary, which contributes to the miniaturization of the inverter and can lead to the miniaturization of the product.

In addition to the heat dissipation method described above, a further heat dissipation effect can be obtained by installing the substrate in the air path. By installing the substrate in the air path, the semiconductor element on the substrate can be radiated by the wind generated by the electric blower, so that the heat generation of the semiconductor element can be significantly suppressed.

The electric vacuum cleaner 60 is a product whose motor rotation speed varies from 0 [rpm] to 100,000 or more [rpm]. Thus, when the single-phase motor 12 drives a product that rotates at high speed, the control method according to the above-described embodiment is suitable. And constant voltage amplitude command V *, by changing the advanced angle phase theta _v in accordance with the rotational speed, it is possible while expanding the rotational speed drive range from a low speed to a high speed rotation region, corresponding to the sudden load change. In addition, since the motor current can be controlled to a sine wave by PWM control, high-efficiency driving can be achieved, so that the operation time can be extended.

FIG. 20 is a diagram for explaining the modulation control in the motor drive device of the present embodiment. On the left side of the figure, the relationship between the rotational speed and the modulation rate is shown. Also, on the right side of the figure, a waveform of the inverter output voltage when the modulation rate is 1 or less and a waveform of the inverter output voltage when the modulation rate exceeds 1 are shown. In general, the load torque of the rotating body increases as the number of rotations increases. For this reason, it is necessary to increase the motor output torque as the rotational speed increases. In general, the motor output torque increases in proportion to the motor current, and the inverter output voltage needs to be increased to increase the motor current. Therefore, the number of revolutions can be increased by increasing the modulation rate and increasing the inverter output voltage.

Next, the rotational speed control in the present embodiment will be described. In the following description, an electric blower is assumed as a load, and an operation range of the electric blower is divided as follows.
(A) Low speed range (low speed range): 0 [rpm] to 80,000 [rpm]
(B) High speed rotation range (high rotation speed range): 80,000 [rpm] or more

In addition, the area | region between said (A) and said (B) is a gray zone, and depending on a use, it may be contained in a low-speed rotation area, and may be included in a high-speed rotation area.

First, the control in the low speed rotation region will be described. In the low speed rotation range, PWM control is performed with a modulation rate of 1 or less. Note that, by setting the modulation rate to 1 or less, the motor current can be controlled to a sine wave, and the efficiency of the motor can be increased. When operating at the same carrier frequency in the low-speed rotation region and the high-speed rotation region, the carrier frequency becomes a carrier frequency that matches the high-speed rotation region, and therefore the PWM pulse tends to increase more than necessary in the low-speed rotation region. . For this reason, in the low-speed rotation region, a method of reducing the carrier frequency and reducing the switching loss may be used. Further, by changing the carrier frequency in synchronization with the rotation speed, control may be performed so that the pulse number does not change according to the rotation speed.

Next, control in the high speed rotation range will be described. In the high-speed rotation range, the modulation rate is set to a value larger than 1. By increasing the modulation factor above 1, the increase in switching loss can be suppressed by increasing the inverter output voltage and reducing the number of switching operations performed by the switching elements in the inverter. Here, when the modulation rate exceeds 1, the motor output voltage increases, but since the number of switching times decreases, there is a concern about current distortion. However, during high speed rotation, the reactance component of the motor increases and di / dt, which is a change component of the motor current, decreases. Therefore, current distortion is smaller than in the low speed rotation range, and the influence on waveform distortion is small. Become. Therefore, in the high-speed rotation region, the modulation rate is set to a value larger than 1 and control is performed to reduce the number of switching pulses. By this control, an increase in switching loss can be suppressed and higher efficiency can be achieved.

In addition, as described above, the boundary between the low-speed rotation range and the high-speed rotation range is ambiguous. For this reason, the control unit 25 is set with a first rotation speed that determines the boundary between the low-speed rotation region and the high-speed rotation region, and the control unit 25 is configured when the rotation speed of the motor or the load is equal to or lower than the first rotation speed. The modulation rate is set to 1 or less, and when the rotational speed of the motor or load exceeds the first rotational speed, the modulation rate may be set to exceed 1.

The configuration described in the above embodiment shows an example of the contents of the present invention, and can be combined with another known technique, and can be combined with other configurations without departing from the gist of the present invention. It is also possible to omit or change the part.

1 electric vacuum cleaner, 2 motor drive device, 6 vacuum cleaner body, 11 single-phase inverter, 11a, 11b connection end, 12 single-phase motor, 12a rotor, 12b stator, 12b1 tooth, 20 voltage sensor, 21 position sensor, 22 current Sensor, 25 control unit, 30 analog-digital converter, 31 processor, 32 drive signal generation unit, 33 carrier generation unit, 34 memory, 38 carrier comparison unit, 38a absolute value calculation unit, 38b division unit, 38c, 38d, 38f multiplication Unit, 38e addition unit, 38g, 38h comparison unit, 38i, 38j output inversion unit, 42 rotation speed calculation unit, 44 advance phase calculation unit, 51, 52, 53, 54 switching element, 51a, 52a, 53a, 54a body Diode, 62 extension tube, 63 suction Mouth, 64 electric blower, 65 dust collection chamber, 66 operation unit, 67 battery, 69 suction port, 72 air passage, 76 floor, 80 semiconductor substrate, 81 p-type region, 82 source electrode, 83 drain electrode, 84 gate Electrode, 85 n-type region, 86 oxide insulating film.

Claims

A vacuum cleaner comprising a single-phase motor, a motor driving device for driving the single-phase motor, and a suction tool for sucking dust,
The motor driving device is
A single-phase inverter that applies an AC voltage to the single-phase motor;
A position sensor that outputs a position sensor signal indicating a rotor magnetic pole position of the single-phase motor;
Based on the position sensor signal, the rotational speed of the single-phase motor is calculated, and based on the calculated change in the rotational speed, the aperture ratio of the gap generated between the suction port of the suction tool and the cleaning surface is calculated. A calculating unit to
With
When the aperture ratio is within the first range, the amplitude value of the AC voltage is a first value, and when the aperture ratio is within the second range, the amplitude value of the AC voltage is the first value. A vacuum cleaner with a second value different from
A vacuum cleaner comprising a single-phase motor, a motor driving device for driving the single-phase motor, and a suction tool for sucking dust,
The motor driving device is
A single-phase inverter that applies an AC voltage to the single-phase motor;
A position sensor that outputs a position sensor signal indicating a rotor magnetic pole position of the single-phase motor;
A current sensor for detecting a motor current flowing in the single-phase motor;
Based on the position sensor signal, the rotational speed of the single-phase motor is calculated, and based on the calculated change in the rotational speed, the aperture ratio of the gap generated between the suction port of the suction tool and the cleaning surface is calculated. A calculating unit to
With
Controlling an advance angle which is a phase difference between the motor current and the position sensor signal;
The advance angle has a slope that increases as the rotational speed of the single-phase motor increases.
When the aperture ratio is in the first range, the slope of the advance angle is a first slope value, and when the aperture ratio is in the second range, the slope of the advance angle is the first slope value. A vacuum cleaner with a different second slope value.
A vacuum cleaner comprising a single-phase motor, a motor driving device for driving the single-phase motor, and a suction tool for sucking dust,
The motor driving device includes a single-phase inverter that applies an AC voltage to the single-phase motor,
When the opening ratio of the gap generated between the suction port of the suction tool and the cleaning surface is within the first range, the rotational speed of the single-phase motor is the first rotational speed and the amplitude value of the AC voltage Is the first value,
When the aperture ratio is in the second range, the rotation speed of the single-phase motor is a second rotation speed different from the first rotation speed, and the amplitude value of the AC voltage is the first value. Is a different second value vacuum cleaner.
A vacuum cleaner comprising a single-phase motor, a motor driving device for driving the single-phase motor, and a suction tool for sucking dust,
The motor driving device is
A single-phase inverter that applies an AC voltage to the single-phase motor;
A position sensor that outputs a position sensor signal indicating a rotor magnetic pole position of the single-phase motor;
With
Controlling the advance angle which is the phase difference between the voltage applied to the motor and the position sensor signal;
The advance angle has a slope that increases as the rotational speed of the single-phase motor increases.
When the opening ratio of the gap formed between the suction port of the suction tool and the cleaning surface is within the first range, the rotation speed of the single-phase motor is the first rotation speed and the inclination of the advance angle is A first slope value,
When the aperture ratio is in the second range, the rotation speed of the single-phase motor is a second rotation speed different from the first rotation speed, and the inclination of the advance angle is the first inclination value. Is a vacuum cleaner with a different second slope value.
A plurality of ranges of the aperture ratio, the upper limit value or the lower limit value of the range of the aperture ratio as a threshold value, the amplitude value of the AC voltage or the advance angle each time the aperture ratio exceeds or falls below the threshold value The vacuum cleaner of Claim 2 or 4 which changes an inclination value.
The electric vacuum cleaner according to claim 2, 4 or 5, wherein the AC voltage has a plurality of amplitude values or inclination values of the advance angle for each operation mode.
The single-phase inverter includes a plurality of switching elements,
The vacuum cleaner according to any one of claims 1 to 6, wherein at least one of the plurality of switching elements is formed of a wide band gap semiconductor.
The electric vacuum cleaner according to claim 7, wherein the wide band gap semiconductor is silicon carbide, gallium nitride, or diamond.