WO2018098975A1 - 一种脉冲宽度调制方法、脉冲宽度调制系统及控制器 - Google Patents

一种脉冲宽度调制方法、脉冲宽度调制系统及控制器 Download PDF

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WO2018098975A1
WO2018098975A1 PCT/CN2017/081600 CN2017081600W WO2018098975A1 WO 2018098975 A1 WO2018098975 A1 WO 2018098975A1 CN 2017081600 W CN2017081600 W CN 2017081600W WO 2018098975 A1 WO2018098975 A1 WO 2018098975A1
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modulation
modulated wave
phase
max
wave
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PCT/CN2017/081600
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English (en)
French (fr)
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刘方诚
辛凯
郭海滨
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华为技术有限公司
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Priority to EP17875630.0A priority Critical patent/EP3537589B1/en
Publication of WO2018098975A1 publication Critical patent/WO2018098975A1/zh
Priority to US16/426,171 priority patent/US10630163B2/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/501—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode sinusoidal output voltages being obtained by the combination of several pulse-voltages having different amplitude and width
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53873—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with digital control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53875—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
    • H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/085—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation wherein the PWM mode is adapted on the running conditions of the motor, e.g. the switching frequency
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/50—Reduction of harmonics
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H02M1/123—Suppression of common mode voltage or current

Definitions

  • the present application relates to the field of circuit technologies, and in particular, to a pulse width modulation method, a pulse width modulation system, and a controller.
  • converters as core energy control devices have become one of the key factors for clean energy applications.
  • three-phase converters are one of the most widely used converters for connecting three-phase AC power systems and DC power systems and enabling energy transfer between the two systems. According to the difference of energy flow direction, it is divided into two working states: rectification and inverter.
  • the transmission of energy from the DC system to the AC system is called inverter, and the transmission from the AC system to the DC system is called rectification.
  • Conversion efficiency and power quality are two key technical indicators of three-phase converters, and the modulation method directly affects the on-off state of the switching device, so it is one of the key factors affecting its conversion efficiency and power quality.
  • the pulse width modulation method of the commonly used three-phase converter is Pulse Width Modulation (PWM), which controls the width of the driving pulse of each device in the switching network.
  • PWM Pulse Width Modulation
  • the most straightforward implementation is to compare the carrier with the modulated wave and compare the results to control the on/off state of the switching device.
  • PWM can be divided into Continuous Pulse Width Modulation (CPWM) and Discontinuous Pulse Width Modulation (DPWM).
  • CPWM Continuous Pulse Width Modulation
  • DPWM Discontinuous Pulse Width Modulation
  • DPWM Discontinuous Pulse Width Modulation
  • DPWM Discontinuous Pulse Width Modulation
  • DPWM Discontinuous Pulse Width Modulation
  • DPWM Discontinuous Pulse Width Modulation
  • the harmonic distortion rate is generally higher than that of CPWM.
  • the high harmonic content of the injection is more likely to cause system resonance, and the lower the modulation degree of the three-phase converter, the harmonic of DPWM injection.
  • the common mode injection method of the DPWM is improved to reduce the common mode content of the injection, thereby reducing the total
  • the harmonic content generated by the mold injection has become the focus of research.
  • the two existing technologies are as follows: 1. Limiting the rising rate of the common mode voltage to reduce the rapid change due to the common mode voltage.
  • the generated harmonics are equivalent to extending the common mode voltage change time and reducing the switching state of the three-phase converter; second, taking the DC bus voltage as a reference object, measuring the DC bus voltage to calculate the switching tube clamping time and Corresponding conduction angle and increasing amplitude range, calculating the common mode modulation voltage of the three phases according to the DC bus voltage, the clamp tube clamping time and the phase information, and calculating according to the fundamental frequency sinusoidal modulation voltage and the common mode modulation voltage of the three phases The corresponding final modulation voltage is output.
  • the second method is not applicable to the scenario where the three-phase converter adopts DPWM, and the calculation process is complicated, and the calculation takes too long, which is not applicable in the case where the switching frequency of the three-phase converter is high.
  • the application provides a pulse width modulation method, a pulse width modulation system and a controller.
  • the rate of change of the common mode component of the three-phase converter changes with the modulation degree of the converter, and the three-phase current is improved. Stability and harmonic characteristics of the device, and flexible adaptive adjustment.
  • a first aspect of the present invention provides a pulse width modulation method for a three-phase converter, comprising:
  • the preset modulation parameter includes a preset maximum modulation degree, a preset minimum modulation degree, and a preset minimum modulation degree of the three-phase current converter, The minimum rate of common mode change between the preset positive and negative clamp states, the difference between the maximum rate and the minimum rate of the common mode change between the preset positive and negative clamp states, and the modulation curvature parameter;
  • the three-phase initial modulated wave is superimposed with the common mode modulated wave to obtain a three-phase output modulated wave.
  • the energy transfer between the three-phase AC power system and the DC power system is generally realized by a three-phase converter.
  • the modulation mode of the three-phase converter is DPWM, and the DPWM is realized when the converter modulation is relatively high.
  • the common mode component of the injection is small, and the output waveform is close to a sine wave, which can meet the requirements of reducing the loss and satisfying the distortion of the output current. At this time, the rate of change of the injected common mode component should be less limited, when the converter is adjusted.
  • the common mode component of DPWM injection is gradually larger. At this time, the output waveform distortion is severe, and the rate of change of the injected common mode component should be reduced.
  • the rate of change of the common mode component is determined by the rate constant of the common mode component. Determining, the common mode component change rate adjustment coefficient is calculated by the preset modulation parameter and the converter modulation degree, and the preset modulation parameter includes a preset maximum modulation degree of the three-phase converter, a preset minimum modulation degree, The minimum rate of common mode change between the preset positive and negative clamp states, the difference between the maximum rate and the minimum rate of the common mode change between the preset positive and negative clamp states, and the modulation curvature parameter.
  • the modulated wave set corresponding to the three-phase initial modulation wave is calculated, from the modulated wave set.
  • the modulated wave with the smallest absolute value is selected as the common mode modulated wave.
  • the three-phase initial modulated wave is superimposed with the common mode modulated wave to obtain a three-phase output modulated wave, which is compared with the prior art method 1 because of the common mode component.
  • the rate of change is determined by the modulation degree of the converter, and changes with the modulation degree of the converter.
  • the calculating a common mode component change rate adjustment coefficient according to the preset modulation coefficient and the converter modulation degree including:
  • a preset modulation factor is required, wherein the maximum modulation degree M max and the minimum modulation degree M min are the maximum modulation degrees allowed by the three-phase converter (generally M max does not exceed 1.15) and the minimum modulation degree (generally M min is less than 1), which is determined by the converter product design and application scenarios.
  • the maximum modulation degree M max and the minimum modulation degree M min are equivalent to two fixed values;
  • the common mode between the positive and negative clamp states the minimum amount of change in the rate of K b is a scene in which a three-phase converter of the decision, the user may be freely set; common modulus clamped state between the positive and negative maximum rate of change and the difference between K a K b
  • K a is equal to 1-K b
  • the value of the modulation curvature parameter N is preset, N is In the case of 1, the common mode component change rate adjustment coefficient and the converter modulation degree are in a linear relationship, and when N is greater than zero and not equal to 1, the common mode component change rate adjustment coefficient and the converter modulation degree are between For the curve relationship.
  • the component can be calculated by the formula change rate of the common mode adjustment factor K value, the value of K is by M It is decided to achieve the goal of flexibly adjusting the rate of change of the common mode component by the converter modulation degree M.
  • the preset modulation wave maximum limit value, the modulation wave minimum limit value, and the three-phase initial modulation wave sum is calculated to obtain a modulated wave set corresponding to the three-phase initial modulated wave, and the modulated wave having the smallest absolute value is selected from the modulated wave set as a common mode modulated wave, including:
  • An absolute value is obtained for each of the modulated wave variables in the common mode modulated wave set, and a modulated wave variable having the smallest absolute value is selected as the common mode modulated wave.
  • the modulation wave maximum limit value v max and the modulation degree minimum limit value v min are preset. Since each corresponding switch in the three-phase converter has three clamp states for each switch, then according to v max , v min, the common mode component change rate modifier initial coefficient K and three-phase modulation wave in a first initial phase modulation wave v a, v a is calculated to obtain a set of modulated wave corresponding to ⁇ v max -v a, K * (v max /2+v min /2-v a ),v min -v a ⁇ ,v max -v a is the modulation wave variable in the positive clamp state corresponding to v a , and v min -v a is the negative corresponding to v a
  • K*(v max /2+v min /2-v a ) is the modulation wave variable in the switching state corresponding to v a
  • the second phase initial modulation wave v b is calculated in turn.
  • the three-phase initial modulated wave and the common mode modulation Waves are superimposed on the waveform to obtain a three-phase output modulated wave, including:
  • a waveform in which the initial modulated waves of each of the three-phase initial modulated waves are respectively in one-to-one correspondence with the common mode modulated waves is superimposed to obtain a three-phase output modulated wave.
  • the three-phase converter After obtaining the common mode modulated wave, the three-phase converter needs to superimpose the waveforms of the initial modulated waves of each of the common mode modulated waves and the common mode modulated waves, thereby obtaining a three-phase output modulated wave.
  • obtaining the initial modulation wave and the converter modulation degree includes:
  • the three-phase converter is detected to obtain three-phase grid voltage, three-phase current and preset carrier amplitude, and the three-phase grid voltage is sent to the phase-locked loop to obtain the phase of the rotating coordinate system, and the three-phase current is performed according to the phase of the rotating coordinate system.
  • Coordinate transformation obtain the rotating coordinate system current, obtain the preset current reference value of the three-phase current coordinate transformation, and calculate the difference between the preset current reference value and the rotating coordinate system current to obtain the current difference, and the current difference through the PI regulator
  • the value is processed to obtain the adjustment component of the rotating coordinate system, and the inverse coordinate transformation is performed on the adjustment component of the rotating coordinate system to obtain a three-phase initial modulated wave, and a three-phase initial is obtained.
  • the amplitude of the modulated wave is modulated, and the converter modulation is obtained according to the ratio of the amplitude of the modulated wave to the amplitude of the preset carrier.
  • a second aspect of the present invention provides a pulse width modulation system for a three-phase converter, comprising:
  • An acquisition module configured to acquire a three-phase initial modulation wave and a converter modulation degree
  • a calculation module configured to calculate a common mode component change rate adjustment coefficient according to the preset modulation parameter and the converter modulation degree, where the preset modulation parameter includes a preset maximum modulation degree of the three-phase converter, and a preset The minimum modulation degree, the minimum rate of common mode change between the preset positive and negative clamp states, the difference between the maximum rate and the minimum rate of the common mode change between the preset positive and negative clamp states, and the modulation curvature parameter;
  • a calculation module configured to calculate the three-phase initial modulation according to a preset modulation wave maximum limit value, a modulation wave minimum limit value, the three-phase initial modulation wave, and the common mode component change rate adjustment coefficient a set of modulated waves corresponding to the wave, and selecting a modulated wave having the smallest absolute value as the common mode modulated wave from the modulated wave set;
  • the modulation module is further configured to superimpose the three-phase initial modulated wave and the common mode modulated wave to obtain a three-phase output modulated wave.
  • the energy transfer between the three-phase AC power system and the DC power system is generally realized by a three-phase converter.
  • the modulation mode of the three-phase converter is DPWM, and the DPWM is realized when the converter modulation is relatively high.
  • the common mode component of the injection is small, and the output waveform is close to a sine wave, which can meet the requirements of reducing the loss and satisfying the distortion of the output current. At this time, the rate of change of the injected common mode component should be less limited, when the converter is adjusted.
  • the common mode component of DPWM injection is gradually larger. At this time, the output waveform distortion is severe, and the rate of change of the injected common mode component should be reduced.
  • the rate of change of the common mode component is determined by the rate constant of the common mode component. Determining, the common mode component change rate adjustment coefficient is calculated by the preset modulation parameter and the converter modulation degree, and the preset modulation parameter includes a preset maximum modulation degree of the three-phase converter, a preset minimum modulation degree, The minimum rate of common mode change between the preset positive and negative clamp states, the difference between the maximum rate and the minimum rate of the common mode change between the preset positive and negative clamp states, and the modulation curvature parameter.
  • the modulated wave set corresponding to the three-phase initial modulation wave is calculated, from the modulated wave set.
  • the modulated wave with the smallest absolute value is selected as the common mode modulated wave.
  • the three-phase initial modulated wave is superimposed with the common mode modulated wave to obtain a three-phase output modulated wave, which is compared with the prior art method 1 because of the common mode component.
  • the rate of change is determined by the modulation degree of the converter, and changes with the modulation degree of the converter.
  • the acquiring module is further configured to acquire a preset maximum modulation degree M max , a preset minimum modulation degree M min , and a minimum modulus change rate K b between the preset positive and negative clamp states preset state clamped between positive and negative common modulus change rate of the maximum rate and the minimum difference between the curvature K a and modulation parameters N, N is greater than 0;
  • the calculation module is further configured to bring the M max , M min , K b , K a , N and the converter modulation degree M into a formula
  • the common mode component change rate adjustment coefficient K is calculated.
  • a preset modulation factor is required, wherein the maximum modulation degree M max and the minimum modulation degree M min are the maximum modulation degrees allowed by the three-phase converter (generally M max does not exceed 1.15) and the minimum modulation degree (generally M min is less than 1), which is determined by the converter product design and application scenarios.
  • the maximum modulation degree M max and the minimum modulation degree M min are equivalent to two fixed values; common mode between positive and negative clamp states the minimum amount of change in the rate of K b is a scene in which a three-phase converter of the decision, the user may be freely set; common modulus clamped state between the positive and negative maximum rate of change and the difference between K a K b Assuming that the maximum rate of common mode change between the positive and negative clamp states allowed by the three-phase converter is 1, then K a is equal to 1-K b ; the value of the modulation curvature parameter N is preset, N is In the case of 1, the common mode component change rate adjustment coefficient and the converter modulation degree are in a linear relationship, and when N is greater than zero and not equal to 1, the common mode component change rate adjustment coefficient and the converter modulation degree are between For the curve relationship.
  • the component can be calculated by the formula change rate of the common mode adjustment factor K value, the value of K is by M It is decided to achieve the goal of flexibly adjusting the rate of change of the common mode component by the converter modulation degree M.
  • the acquiring module is further configured to acquire a preset modulation wave maximum limit value v max and a modulation degree minimum limit value v min ;
  • the calculating module is further configured to calculate, according to the v max , the v min , the common mode component change rate adjustment coefficient K, and the first phase initial modulated wave v a of the three-phase initial modulated wave
  • the calculating module is further configured to calculate, according to the v max , the v min , the common mode component change rate adjustment coefficient K, and the second phase initial modulated wave v b of the three-phase initial modulated wave, a set of modulated waves corresponding to v b ⁇ v max -v b , K*(v max /2+v min /2-v b ), v min -v b ⁇ , wherein v max -v b is v b corresponds to a modulated wave variable in a positive clamp state, the v min -v b being a modulated wave variable in a negative clamp state corresponding to the v b , the K*(v max /2+v min /2-v b ) is a modulated wave variable in a switching state corresponding to the v b ;
  • the calculating module is further configured to calculate, according to the v max , the v min , the common mode component change rate adjustment coefficient K, and the second phase initial modulated wave v c of the three-phase initial modulated wave a set of modulated waves corresponding to v c ⁇ v max -v c , K*(v max /2+v min /2-v c ), v min -v c ⁇ , wherein v max -v c is v c corresponds to a modulated wave variable in a positive clamp state, the v min -v c being a modulated wave variable in a negative clamp state corresponding to the v c , the K*(v max /2+v min /2-v c ) is a modulation wave variable in a switching state corresponding to the v c ;
  • the calculation module is further configured to obtain, according to the modulation wave set corresponding to the v a , the v b and the v c, a modulated wave set corresponding to the three-phase initial modulated wave ⁇ v max -v a , v Max -v b , v max -v c , K*(v max /2+v min /2-v a ), K*(v max /2+v min /2-v b ), K*(v max /2+v min /2-v c ),v min -v a, v min -v b ,v min -v c ⁇ ;
  • the calculation module is further configured to take an absolute value for each modulated wave variable in the common mode modulated wave set, and select a modulated wave variable with the smallest absolute value as the common mode modulated wave.
  • the modulation wave maximum limit value v max and the modulation degree minimum limit value v min are preset. Since each corresponding switch in the three-phase converter has three clamp states for each switch, then according to v max , v min, the common mode component change rate modifier initial coefficient K and three-phase modulation wave in a first initial phase modulation wave v a, v a is calculated to obtain a set of modulated wave corresponding to ⁇ v max -v a, K * (v max /2+v min /2-v a ),v min -v a ⁇ ,v max -v a is the modulation wave variable in the positive clamp state corresponding to v a , and v min -v a is the negative corresponding to v a
  • K*(v max /2+v min /2-v a ) is the modulation wave variable in the switching state corresponding to v a
  • the second phase initial modulation wave v b is calculated in turn.
  • the modulation module is specifically configured to superimpose waveforms of the initial modulated waves of each of the three-phase initial modulated waves and the common mode modulated waves, to obtain a three-phase output modulated wave.
  • the three-phase converter After obtaining the common mode modulated wave, the three-phase converter needs to superimpose the waveforms of the initial modulated waves of each of the common mode modulated waves and the common mode modulated waves, thereby obtaining a three-phase output modulated wave.
  • the acquiring module is configured to acquire a three-phase grid voltage, a three-phase current, and a preset carrier amplitude
  • the calculation module is further configured to perform phase lock processing on the three-phase grid voltage to obtain a phase of a rotating coordinate system
  • the calculation module is further configured to perform coordinate transformation on the three-phase current according to the phase of the rotating coordinate system to obtain a rotating coordinate system current;
  • the acquiring module is further configured to acquire a preset current reference value of the three-phase current coordinate transformation, and calculate a difference between the preset current reference value and the rotating coordinate system current to obtain a current difference value;
  • the calculation module is further configured to process the current difference value by a proportional-integral PI regulator to obtain a rotating coordinate system adjustment component;
  • the calculation module is further configured to perform inverse coordinate transformation on the rotating coordinate system adjustment component to obtain a three-phase initial modulation wave;
  • the calculation module is further configured to obtain a modulation wave amplitude of the three-phase initial modulated wave, and obtain a converter modulation degree according to a ratio of the modulation wave amplitude to the preset carrier amplitude.
  • the acquisition module detects the three-phase converter to obtain the three-phase grid voltage, the three-phase current and the preset carrier amplitude, and the calculation module sends the three-phase grid voltage to the phase-locked loop to obtain the phase of the rotating coordinate system, according to the phase of the rotating coordinate system,
  • the three-phase current is coordinate-transformed to obtain the rotating coordinate system current, obtain the preset current reference value of the three-phase current coordinate transformation, and calculate the difference between the preset current reference value and the rotating coordinate system current to obtain the current difference, which is adjusted by PI.
  • the current difference is processed to obtain the rotating coordinate system adjustment component, and the rotating coordinate system adjustment component is inverse coordinate transformed to obtain a three-phase initial modulated wave, and the modulated wave amplitude of the three-phase initial modulated wave is obtained, and according to the modulation amplitude
  • the ratio of the value to the preset carrier amplitude gives the converter modulation.
  • a third aspect of the present invention provides a controller for a three-phase converter, including:
  • the processor, the memory and the signal interface are connected to each other, and the running instruction of the processor is stored in the memory.
  • the processor, the memory and the signal interface are connected to each other, and the running instruction of the processor is stored in the memory.
  • the signal interface is configured to acquire a three-phase initial modulation wave and a converter modulation degree
  • the processor is configured to calculate a common mode component change rate adjustment coefficient according to the preset modulation parameter and the converter modulation degree, where the preset modulation parameter includes a preset maximum modulation degree of the three-phase current converter , preset minimum modulation, The minimum rate of common mode change between the preset positive and negative clamp states, the difference between the maximum rate and the minimum rate of the common mode change between the preset positive and negative clamp states, and the modulation curvature parameter;
  • the processor is further configured to calculate the three-phase according to a preset modulation wave maximum limit value, a modulation wave minimum limit value, the three-phase initial modulation wave, and the common mode component change rate adjustment coefficient. a modulated wave set corresponding to the initial modulated wave, and a modulated wave having the smallest absolute value is selected from the modulated wave set as a common mode modulated wave;
  • the processor is further configured to superimpose the three-phase initial modulated wave and the common mode modulated wave to obtain a three-phase output modulated wave.
  • the common mode component change rate adjustment coefficient is calculated by the processor according to a preset modulation parameter and a converter modulation degree
  • the preset modulation parameter includes a preset maximum modulation degree of the three-phase converter, a preset minimum modulation degree, The minimum rate of common mode change between the preset positive and negative clamp states, the difference between the maximum rate and the minimum rate of the common mode change between the preset positive and negative clamp states, and the modulation curvature parameter, according to the preset modulation wave maximum limit
  • the amplitude, the minimum amplitude of the modulated wave, the three-phase initial modulated wave and the common mode component change rate adjustment coefficient are calculated, and the modulated wave set corresponding to the three-phase initial modulated wave is calculated, and the modulated wave with the smallest absolute value is selected from the modulated wave set.
  • the common mode modulates the wave.
  • the three-phase initial modulated wave is superimposed with the common mode modulated wave to obtain a three-phase output modulated wave, which is compared with the prior art method 1 because the rate of change of the common mode component is adjusted by the converter.
  • the system determines that as the converter modulation degree changes, there is no limiting link, so the influence of improper value of the limiting value can be avoided, and the stability of the three-phase converter is improved.
  • Wave characteristics, compared with the prior art two, rate of change of the common mode component is determined by the inverter modulation, adaptive achieved.
  • FIG. 1 is a system structural diagram of a three-phase converter provided by the present application.
  • FIG. 3 is a graph showing a relationship between a modulation degree of a converter and a harmonic distortion rate provided by the present application
  • FIG. 6 is a schematic flow chart of one embodiment of a pulse width modulation method provided by the present application.
  • FIG. 7 is a schematic structural diagram of a three-level grid-connected photovoltaic inverter provided by the present application.
  • FIG. 8 is a schematic diagram of a control algorithm for a three-level grid-connected photovoltaic inverter provided by the present application.
  • FIG. 9 is a relationship diagram of a modulation curvature parameter, a converter modulation degree, and a common mode component change rate adjustment coefficient provided by the present application;
  • FIG. 12 is a schematic structural diagram of an embodiment of a pulse width modulation system provided by the present application.
  • FIG. 13 is a schematic structural diagram of an embodiment of a controller provided by the present application.
  • FIG. 14 is a schematic structural diagram of a two-level rectifier and a current converter provided by the present application.
  • 15 is a schematic structural diagram of a three-level rectifier and a converter provided by the present application.
  • 16 is a schematic structural diagram of a five-level rectifier and a converter provided by the present application.
  • FIG. 17 is a schematic structural diagram of a cascaded multilevel rectifier and a converter provided by the present application.
  • the application provides a pulse width modulation method, a pulse width modulation system and a controller.
  • the rate of change of the common mode component of the three-phase converter changes with the modulation degree of the converter, and the three-phase current is improved. Stability and harmonic characteristics of the device, and flexible adaptive adjustment.
  • the invention is applied to a converter, in particular to the most widely used three-phase converter, which is used to connect a three-phase AC power system and a DC power system and realize energy transfer between the two systems. According to the difference of energy flow direction, it is divided into two working states: rectification and inverter.
  • the transmission of energy from the DC system to the AC system is called inverter, and the transmission from the AC system to the DC system is called rectification. Therefore, in most application scenarios, both rectification and inverter can be implemented by the same system.
  • the system structure of a typical three-phase converter is shown in Figure 1, including DC system, switching network, controller, filter and The AC system, the filter is used for filtering the alternating current during the rectification process, the switch network includes a switching device, and the controller controls the conduction or disconnection action of each switching device in the switch network by the modulation mode, thereby turning the AC system on and off. Energy transfer between the DC system and the DC system.
  • the pulse width modulation method adopted by the three-phase converter is PWM, that is, the width of the driving pulse of each device in the switching network is controlled.
  • PWM can be divided into CPWM and DPWM.
  • CPWM means that there is always one switching action per bridge arm in each switching cycle.
  • the common method is Sinusoidal Pulse Width Modulation (SPWM);
  • SPWM Sinusoidal Pulse Width Modulation
  • DPWM refers to the converter in a certain switching period.
  • a phase bridge arm is clamped to a positive DC bus or a negative DC bus. During this clamp time, the phase switching device is normally open or normally off, and there is no switching action.
  • Common DPWM modulation includes DPWM0, DPWM1, DPWM2, DPWM3, DPWMMAX, DPWMMIN and GDPWM (General DPWM).
  • the modulated wave of the DPWM can be realized by superimposing the equivalent common mode component on the modulated wave of the CPWM. For example, comparing a DPWM modulation waveform in a power frequency cycle (for example, 50 Hz) with a SPWM (a type of CPWM) modulation waveform, as shown in FIG. 2, the difference between the DPWM modulation wave and the power frequency sine wave is in FIG. The waveform of the common mode signal shown. Since the modulated wave of DPWM can be equivalent to the sum of SPWM modulated wave and common mode signal, the output characteristic of DPWM is affected by the SPWM output characteristic and the common mode signal output characteristic.
  • FIG. 3 is a graph showing the relationship between the modulation degree M of the converter and the harmonic distortion rate HDF in a common PWM modulation mode.
  • the harmonic distortion rate of DPWM modulation mode is generally higher than that of CPWM, but when the modulation degree is relatively high, the harmonic distortion rates of DPWM and CPWM are relatively close, and the DPWM modulation is injected.
  • Common mode signal components are less, while modulation When the degree is low, the common mode signal injected to achieve the clamp is large. At this time, the harmonic distortion rate of DPWM is much larger than that of CPWM. Therefore, it is necessary to improve the common mode injection of DPWM when the modulation degree is low, and reduce the injection common mode content, thereby reducing the harmonic content generated by the common mode injection.
  • the common mode signal for DPWM injection is small, the output waveform is close to a sine wave, and the output current distortion is small.
  • conventional DPWM modulation should be used, and the rate of change of the injected common mode component is less limited; when the converter modulation is relatively small
  • the common mode signal for DPWM injection is large.
  • the output waveform distortion is serious, and the rate of change of the injected common mode component should be reduced and reduced.
  • the high-frequency component of the common-mode component reduces the impact of the dramatic change of the common-mode component on the converter and the distortion of the current.
  • each bridge arm can generate positive clamp, negative clamp and switch three states, and the three bridge arms correspond to a total of nine states. 9 kinds of modulated waves.
  • the three-phase converter can only achieve one clamp state at any time, otherwise the converter will be out of control. Therefore, there may be three clamp states in the three-phase converter: positive clamp state and negative clamp state. And switch status.
  • the efficiency improvement of the three-phase converter is much smaller than that of the positive clamp and the negative clamp, and the switching state is a transition state of the positive clamp state and the negative clamp state, and the component of the common mode voltage in the switched state is directly Affects the common superimposed component of the final overlay.
  • the invention provides a pulse width modulation method for solving the influence of a common mode component on a three-phase converter in a switching state, as follows:
  • an embodiment of the present invention provides a pulse width modulation method, including:
  • the three-phase initial modulated wave is an initial modulated wave of each phase of the three-phase AC control loop
  • the control loop includes one of a voltage control loop, a current control loop, and a power control loop.
  • the converter modulation degree can be obtained by calculating the ratio of the peak value of the AC voltage phase of the converter and the DC voltage, or by calculating the peak value and the carrier peak value of the three-phase initial modulated wave, and the three-phase converter is adjusted.
  • the modulated wave is used to control the voltage and current of the converter, so the modulated wave is dynamically changing.
  • the modulated wave is periodically repeated.
  • the peak value of the modulated wave can also be regarded as a fixed value, and the converter modulation degree can be accurately obtained.
  • the preset modulation parameters include a preset maximum modulation degree of the three-phase converter, a preset minimum modulation degree, a minimum rate of common mode change between the preset positive and negative clamp states, and a preset positive and negative clamp.
  • the difference between the maximum rate and the minimum rate of the common mode change between states and the modulation curvature parameter, wherein the preset maximum modulation degree and the preset minimum modulation degree are determined by the three-phase converter product design and application scenarios.
  • the difference between the minimum rate of common mode change between the positive and negative clamp states and the change of the common mode between the positive and negative clamp states is determined by the scene in which the three-phase converter is located. It can also be set freely by the user, and the modulation curvature parameter is also preset.
  • the values of the parameters in the preset modulation parameters are known before the implementation of the scheme, and are calculated according to the preset modulation parameters and the converter modulation degree.
  • the common mode component change rate adjustment coefficient is obtained, and the common mode component change rate adjustment coefficient is used for the rate of change of the common mode component. The larger the value, the faster the rate of change of the common mode component.
  • the maximum amplitude limit value of the modulation wave and the minimum limit value of the modulation degree are preset. Since each corresponding switch in the three-phase current converter has three clamp states for each switch, the three-phase Each of the modulated waves in the initial modulated wave corresponds to a modulated wave of three clamp states, and the modulated wave set corresponding to the three-phase initial modulated wave is calculated to include nine modulated waves, from the modulated wave set. Among the nine modulated waves, the modulated wave having the smallest absolute value is selected as the common mode modulated wave.
  • the three-phase initial modulated wave and the common mode modulated wave are superimposed on the waveform to obtain a corresponding three-phase output modulated wave, and the waveform distortion of the three-phase output modulated wave can be observed, the converter modulation degree and the common mode
  • the rate of change of the component is proportional.
  • the change speed of the common mode component is determined by the common mode component change rate adjustment coefficient
  • the common mode component change rate adjustment coefficient is calculated by the preset modulation parameter and the converter modulation degree.
  • the modulation parameters include the preset maximum modulation degree of the three-phase converter, the preset minimum modulation degree, the minimum rate of common mode change between the preset positive and negative clamp states, and the common mode between the preset positive and negative clamp states.
  • the difference between the maximum rate and the minimum rate of change and the modulation curvature parameter are calculated according to the preset modulation wave maximum limit value, the modulation wave minimum limit value, the three-phase initial modulation wave, and the common mode component change rate adjustment coefficient.
  • the modulated wave set corresponding to the initial modulated wave is selected from the modulated wave set as the common mode modulated wave, and finally, the three-phase initial modulated wave is superimposed with the common mode modulated wave to obtain a three-phase output modulated wave.
  • the rate of change of the common mode component is determined by the degree of modulation of the converter, as the degree of modulation of the converter changes, there is no limiting link. The influence of improper value of the limiting value can be avoided, and the stability and harmonic characteristics of the three-phase converter are improved.
  • the rate of change of the common mode component is determined by the modulation degree of the converter, Adaptive adjustment.
  • the three-phase converter can be divided into an inverter state and a rectifier state.
  • the solution of the present invention can be applied to an inverter or a rectifier.
  • the inverter structure is The two-level structure, the DC bus is connected to the photovoltaic panel, and the AC port is connected to the three-phase AC grid through the L filter.
  • the positive direction of the current is defined as the direction of the outflow inverter bridge arm port.
  • the algorithm diagram of the inverter is shown.
  • obtaining a three-phase initial modulation wave and a converter modulation degree including:
  • the three-phase current is coordinate-converted to obtain a rotating coordinate system current
  • the current difference is processed by the PI regulator to obtain a rotating coordinate system adjustment component
  • the amplitude of the modulated wave of the three-phase initial modulated wave is obtained, and the converter modulation is obtained according to the ratio of the amplitude of the modulated wave to the amplitude of the preset carrier.
  • the three-phase grid voltages v ga , v gb and v gc , and the three-phase currents i a , i b and i c , and the converter modulation degree are obtained.
  • the three-phase grid voltages v ga , v gb and v gc are sent to a phase-locked loop (PLL) to obtain a phase ⁇ of the rotating coordinate system, and then the three-phase currents i a , i b and i c are coordinate-transformed to obtain Its rotation coordinate system currents i d and i q , the coordinate transformation is calculated as Park Transformation, and its calculation equation is:
  • the calculated i d and i q are compared with their reference values i dref and i qref , and the difference values are respectively sent to the regulators G d and G q to obtain the d-axis and q-axis rotating coordinate system adjustment components v d and v q , where G d and G q are Proportional Integral (PI) regulators.
  • PI Proportional Integral
  • the inverse coordinate transformation is performed on the adjustment components v d and v q to obtain their equivalent values v a , v b and v c in the stationary coordinate system.
  • the calculation method is Inverse Park Transformation, and the calculation equation is:
  • the inverter modulation degree is obtained, and the inverter modulation degree is the converter modulation degree.
  • the above embodiment obtains a three-phase initial modulated wave according to the coordinate transformation of the three-phase grid voltage and the three-phase current, but only one way of obtaining the initial modulated wave. In practical applications, there may be other ways, specifically not Make a limit.
  • the common mode component change rate adjustment coefficient is calculated according to the preset modulation coefficient and the converter modulation degree, including:
  • the formula for calculating the common mode component change rate adjustment coefficient K is:
  • the maximum modulation degree M max and the minimum modulation degree M min are the maximum modulation degrees allowed by the three-phase converter (generally M max does not exceed 1.15) and the minimum modulation degree (generally M min is less than 1), which is determined by the converter The product design and application scenarios are determined. Therefore, the maximum modulation degree M max and the minimum modulation degree M min are equivalent to two fixed values; the minimum rate K b of the common mode change between positive and negative clamp states is changed from three phases.
  • a user may be freely set; common modulus change between positive and negative clamp state maximum difference rate K b K a, assuming a three-phase converter will permit
  • K a is equal to 1-K b ;
  • the value of modulation curvature parameter N is preset, and when N is 1, the preset common mode component is adjusted.
  • the formula is a straight line formula. When N is greater than zero and not equal to 1, the preset common mode component adjustment formula is a curve equation.
  • the obtaining module M, M max, M min values, K b, K a and N a calculation module able to change the common mode component calculated by the equation rate adjustment value of the coefficient K, and M max, M
  • the values of min , K b , K a and N are all preset and adjustable, so the value of K is determined by M, and the goal of flexibly adjusting the rate of change of the common mode component by the converter modulation degree M is realized.
  • the specific modulation curvature parameter N needs to be changed according to the degree of change of the desired adaptive adjustment, for example, FIG.
  • the three-phase initial modulation is calculated according to a preset modulation wave maximum limit value, a modulation wave minimum limit value, a three-phase initial modulation wave, and a common mode component change rate adjustment coefficient.
  • the modulated wave set corresponding to the wave, and the modulated wave having the smallest absolute value is selected from the modulated wave set as the common mode modulated wave, including:
  • An absolute value is obtained for each of the modulated wave variables in the common mode modulated wave set, and a modulated wave variable having the smallest absolute value is selected as the common mode modulated wave.
  • the maximum amplitude value and the minimum amplitude value of the modulated wave need to be preset, the preset modulation wave limit value is obtained, and the three-phase initial modulation wave, the common mode component change rate adjustment coefficient, and the modulation wave limit value are obtained.
  • the preset common mode modulated wave variable set includes the initial modulated wave of each phase.
  • the number of modulation wave variables is nine, which is the smallest to affect the stability of the system. The smallest one of the nine modulation wave variables is selected.
  • the calculation process is:
  • the three-phase initial modulated wave is superimposed with the common mode modulated wave to obtain a three-phase output modulated wave, including:
  • a waveform in which the initial modulated waves of each of the three phase initial modulated waves are respectively in one-to-one correspondence with the common mode modulated waves is superimposed to obtain a three-phase output modulated wave.
  • the common mode modulated wave v Z and the three-phase initial modulated waves v a , v b and v c are respectively added to obtain new three-phase output modulated waves v a — mod , v b — mod and v c — mod .
  • FIG. 10(a) and FIG. 10(b) are converters.
  • FIG. 10(c) and FIG. 10(d) are the DPWM modulation waveform and the injected common mode spectrum of the present invention. It can be seen from the comparison of Fig. 10 that when the modulation degree M is high, the scheme and the conventional DPWM The method modulation waveform and spectral characteristics are very close. At this time, the high-frequency component of the injected common-mode signal is small, and there is no need to limit the common-mode voltage source, which can maximize the clamping effect of the converter and improve the efficiency.
  • FIG. 11(c) and FIG. 11(d) are diagrams showing the DPWM modulation waveform and the injected common mode spectrum of the present invention.
  • the modulation degree decreases, the common mode component of the conventional DPWM injection increases.
  • the modulation waveform of this scheme and the conventional DPWM method are quite different.
  • the common mode component injected by this scheme is relatively smooth, and the spectral component is much smaller than the conventional DPWM scheme.
  • the present invention provides a pulse width modulation system applied to a three-phase converter, including:
  • the obtaining module 1201 is configured to acquire a three-phase initial modulation wave and a converter modulation degree
  • the calculation module 1202 is configured to calculate a common mode component change rate adjustment coefficient according to the preset modulation parameter and the converter modulation degree, where the preset modulation parameter includes a preset maximum modulation degree of the three-phase converter, a preset minimum modulation degree, The minimum rate of common mode change between the preset positive and negative clamp states, the difference between the maximum rate and the minimum rate of the common mode change between the preset positive and negative clamp states, and the modulation curvature parameter;
  • the calculating module 1202 is further configured to calculate a modulated wave corresponding to the three-phase initial modulated wave according to a preset modulation wave maximum limit value, a modulation wave minimum limit amplitude value, a three-phase initial modulation wave, and a common mode component change rate adjustment coefficient. a set, selecting a modulated wave having the smallest absolute value from the modulated wave set as a common mode modulated wave;
  • the modulation module 1203 is further configured to superimpose the three-phase initial modulated wave and the common mode modulated wave to obtain a three-phase output modulated wave.
  • the rate of change of the common mode component is determined by the common mode component change rate adjustment coefficient, and the common mode component change rate adjustment coefficient is calculated by the calculation module 1202 according to the converter modulation degree acquired by the acquisition module 1201. And the calculation module 1202 obtains a common mode modulated wave according to the three-phase initial modulation wave and the common mode component change rate adjustment coefficient, and the modulation module 1203 obtains the three-phase output modulated wave according to the three-phase initial modulation wave and the common mode modulated wave, and the existing Compared with the first technique, since the rate of change of the common mode component is determined by the modulation degree of the converter, and the modulation degree of the converter changes, there is no limiting link, so that the influence of improper value of the limiting value can be avoided, and the effect is improved. Compared with the prior art 2, the change rate of the common mode component is determined by the converter modulation degree, and adaptive adjustment is realized.
  • the obtaining module 1201 is further configured to acquire a preset maximum modulation degree M max of the three-phase converter, a preset minimum modulation degree M min , a common mode variable minimum rate K b between the preset positive and negative clamp states, and a preset common modulus clamping state change between positive and negative maximum rate and the minimum rate and the difference between K a modulation curvature parameters N, N is greater than 0;
  • the calculation module 1202 is further configured to bring the M max , M min , K b , K a , N and the converter modulation degree M into the formula
  • the common mode component change rate adjustment coefficient K is calculated.
  • a preset modulation coefficient is required, wherein the maximum modulation degree M max and the minimum modulation degree M min are maximum modulation degrees allowed by the three-phase converter (generally M max does not exceed 1.15) and minimum modulation degree ( Generally, M min is less than 1), which is determined by the design and application scenarios of the converter. Therefore, the maximum modulation degree M max and the minimum modulation degree M min are equivalent to two fixed values;
  • the minimum rate of change of modulus K b is determined by the scene in which the three-phase converter is located, or it can be set freely by the user; the difference between the maximum rate of change of the common mode between the positive and negative clamp states and K b a .
  • K a is equal to 1-K b ;
  • the value of the modulation curvature parameter N is preset, N
  • the preset common mode component adjustment formula is a straight line formula.
  • the preset common mode component adjustment formula is a curve equation.
  • the calculation module 1202 can calculate the value of the common mode component change rate adjustment coefficient K by the formula, and M max
  • M min , K b , K a and N are all preset and adjustable, then the value of K is determined by M, and the modulation rate M of the converter is flexibly adjusted by the converter modulation degree M. aims.
  • the obtaining module 1201 is further configured to obtain a preset modulation wave maximum limit value v max and a modulation degree minimum limit value v min ;
  • the calculation module 1202 is further configured to calculate a modulation wave set corresponding to v a according to v max , v min , a common mode component change rate adjustment coefficient K, and a first phase initial modulated wave v a of the three-phase initial modulated wave.
  • the calculation module 1202 is further configured to calculate a modulation wave set corresponding to v b according to v max , v min , a common mode component change rate adjustment coefficient K, and a second phase initial modulated wave v b of the three-phase initial modulated wave.
  • Max -v b K*(v max /2+v min /2-v b ),v min -v b ⁇ ,v max -v b is the modulation wave variable in the positive clamp state corresponding to v b
  • v Min -v b is the modulation wave variable in the negative clamp state corresponding to v b
  • K*(v max /2+v min /2-v b ) is the modulation wave variable in the switching state corresponding to v b ;
  • the calculation module 1202 is further configured to calculate a modulation wave set corresponding to v c according to v max , v min , a common mode component change rate adjustment coefficient K, and a second phase initial modulation wave v c of the three-phase initial modulated wave.
  • Max -v c K*(v max /2+v min /2-v c ), v min -v c ⁇ , v max -v c is the modulation wave variable in the positive clamp state corresponding to v c , v Min -v c is the modulation wave variable in the negative clamp state corresponding to v c , and K*(v max /2+v min /2-v c ) is the modulation wave variable in the switching state corresponding to v c ;
  • the calculating module 1202 is further configured to obtain, according to the modulated wave set corresponding to v a , v b , and v c , a modulated wave set corresponding to the three-phase initial modulated wave ⁇ v max -v a , v max -v b , v max -v c , K*(v max /2+v min /2-v a ), K*(v max /2+v min /2-v b ), K*(v max /2+v min /2-v c ), v min -v a, v min -v b ,v min -v c ⁇ ;
  • the calculation module 1202 is further configured to take an absolute value for each modulated wave variable in the common mode modulated wave set, and select a modulated wave variable having the smallest absolute value as the common mode modulated wave.
  • the modulation wave maximum amplitude value and the minimum amplitude value are preset, and the obtaining module 1201 obtains the preset modulation wave maximum amplitude value and the modulation wave minimum amplitude value, because each corresponding one of the three-phase converters
  • the switch has three clamp states for each switch, and the calculation module 1202 calculates according to v max , v min , the common mode component change rate adjustment coefficient K, and the first phase initial modulation wave v a of the three-phase initial modulated wave.
  • v a corresponding modulation wave set ⁇ v max -v a , K*(v max /2+v min /2-v a ), v min -v a ⁇ , v max -v a is the positive clamp corresponding to v a
  • the modulation wave variable in the bit state, v min -v a is the modulation wave variable in the negative clamp state corresponding to v a
  • K*(v max /2+v min /2-v a ) is the switching of v a
  • the modulated wave variable in the state is sequentially calculated to obtain the modulated wave set corresponding to the second phase initial modulated wave v b ⁇ v max -v b , K*(v max /2+v min /2-v b ), v min -v b ⁇
  • the modulated wave set corresponding to the third phase initial modulated wave v b is calculated ⁇ v max -v b , K*(v max /2+v min
  • the modulation module 1203 is specifically configured to superimpose waveforms of the initial modulation waves of each of the three-phase initial modulated waves and the common mode modulated waves, to obtain a three-phase output modulated wave.
  • the obtaining module 1201 is configured to acquire a three-phase grid voltage, a three-phase current, and a preset carrier amplitude
  • the calculation module 1202 is further configured to perform phase-locking processing on the three-phase grid voltage to obtain a phase of the rotating coordinate system;
  • the calculation module 1202 is further configured to perform coordinate transformation on the three-phase current according to the phase of the rotating coordinate system to obtain a rotating coordinate system current;
  • the obtaining module 1201 is further configured to acquire a preset current reference value of the three-phase current coordinate transformation, and calculate a difference between the preset current reference value and the rotating coordinate system current to obtain a current difference value;
  • the calculating module 1202 is further configured to process the current difference value by using a proportional-integral PI regulator to obtain a rotating coordinate system adjustment component;
  • the calculation module 1202 is further configured to perform inverse coordinate transformation on the rotating coordinate system adjustment component to obtain a three-phase initial modulation wave;
  • the calculating module 1202 is further configured to obtain a modulated wave amplitude value of the three-phase initial modulated wave, and obtain a converter modulation degree according to a ratio of the modulated wave amplitude value and the preset carrier amplitude.
  • the acquisition module 1201 detects the three-phase converter to obtain the three-phase grid voltage, the three-phase current, and the preset carrier amplitude, and the calculation module 1202 sends the three-phase grid voltage to the phase-locked loop to obtain the phase of the rotating coordinate system.
  • phase of the rotating coordinate system coordinate transformation of the three-phase current to obtain a rotating coordinate system current, obtain a preset current reference value of the three-phase current coordinate transformation, and calculate a difference between the preset current reference value and the rotating coordinate system current,
  • the current difference is obtained, the current difference is processed by the PI regulator, the adjustment component of the rotating coordinate system is obtained, and the inverse coordinate transformation is performed on the adjustment component of the rotating coordinate system to obtain a three-phase initial modulated wave, and a modulated wave of the three-phase initial modulated wave is obtained.
  • the amplitude is obtained, and the converter modulation is obtained according to the ratio of the amplitude of the modulation wave to the amplitude of the preset carrier.
  • the above embodiment has explained the pulse width modulation method and the pulse width modulation system in detail.
  • the physical device of the pulse width modulation system will be described below.
  • the physical device is the controller shown in FIG.
  • the present invention provides a controller, including:
  • the processor 1301, the memory 1302, and the signal interface 1303 are connected to each other, and the memory 1302 stores the running instruction of the processor 1301.
  • a signal interface 1303, configured to acquire a three-phase initial modulation wave and a converter modulation degree
  • the processor 1301 is configured to calculate a common mode component change rate adjustment coefficient according to a preset modulation parameter and a converter modulation degree, where the preset modulation parameter includes a preset maximum modulation degree of the three-phase converter, a preset minimum modulation degree, The minimum rate of common mode change between the preset positive and negative clamp states, the difference between the maximum rate and the minimum rate of the common mode change between the preset positive and negative clamp states, and the modulation curvature parameter;
  • the processor 1301 is further configured to calculate a modulated wave corresponding to the three-phase initial modulated wave according to a preset modulation wave maximum limit value, a modulation wave minimum limit value, a three-phase initial modulation wave, and a common mode component change rate adjustment coefficient. Set, from modulation The modulated wave with the smallest absolute value is selected as the common mode modulated wave in the wave set;
  • the processor 1301 is further configured to superimpose a three-phase initial modulated wave and a common mode modulated wave to obtain a three-phase output modulated wave.
  • the common mode component change rate adjustment coefficient is calculated by the processor 1301 according to the preset modulation parameter and the converter modulation degree acquired by the signal interface 1303, and the preset modulation parameter includes a three-phase converter.
  • the preset modulation parameter includes a three-phase converter.
  • the processor 1301 calculates a modulated wave corresponding to the three-phase initial modulated wave according to a preset modulation wave maximum limit value, a modulated wave minimum limit value, a three-phase initial modulation wave, and a common mode component change rate adjustment coefficient.
  • the set selects the modulated wave with the smallest absolute value from the modulated wave set as the common mode modulated wave, and finally, superimposes the three-phase initial modulated wave and the common mode modulated wave to obtain a three-phase output modulated wave, which is compared with the prior art method.
  • the rate of change of the common mode component is determined by the converter modulation degree, as the converter modulation degree changes, there is no limiter link, so the limit value can be avoided. Effects caused by improper, enhance the stability of the three-phase harmony Potter converter, as compared with the prior art two, rate of change of the common mode component is determined by the inverter modulation, adaptive achieved.
  • the controller shown in Figure 13 also includes one or more storage media 1304 (e.g., one or one of the Shanghai quantity storage devices) that store application 1305 or data 1306.
  • the memory 1302 and the storage medium 1304 may be short-term storage or persistent storage.
  • the program stored on storage medium 1304 can include one or more modules (not shown), each of which can include a series of instruction operations in the server.
  • the processor 1301 can be configured to communicate with the storage medium 1304 to perform a series of instruction operations in the storage medium 1304 on the server.
  • the controller may also include one or more operating systems 1307, such as Windows ServerTM, Mac OS XTM, UnixTM, LinuxTM or FreeBSDTM, and the like.
  • operating systems 1307 such as Windows ServerTM, Mac OS XTM, UnixTM, LinuxTM or FreeBSDTM, and the like.
  • the photovoltaic inverter application system shown in FIG. 7 as exemplified in the present invention can be equivalent to the two-level rectifier and the inverter structure as shown in FIG. 14 , in addition, for the three-level rectifier and converter structure shown in FIG. 15, it can also be the five-level rectifier and converter structure shown in FIG. 16, and can also be the cascaded multi-level rectifier and the variable shown in FIG. Streamer structure.
  • the size of the sequence numbers of the above processes does not mean the order of execution, and the order of execution of each process should be determined by its function and internal logic, and should not be taken to the embodiments of the present invention.
  • the implementation process constitutes any limitation.

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Abstract

一种脉冲宽度调制方法、脉冲宽度调制系统及控制器，其随着变流器调制度的变化而改变三相变流器的共模分量的变化速率，从而提升三相变流器的稳定性和谐波特性，实现灵活地自适应调节。脉冲宽度调制方法包括：获取三相初始调制波及变流器调制度（601）；根据预设调制参数和变流器调制度计算得到共模分量变化速率调节系数；根据预设的调制波最大限幅值、调制波最小限幅值、三相初始调制波和共模分量变化速率调节系数，计算得到三相初始调制波对应的调制波集合，从调制波集合中选取绝对值最小的调制波作为共模调制波（603）；将三相初始调制波与共模调制波进行波形叠加，得到三相输出调制波。

Description

一种脉冲宽度调制方法、脉冲宽度调制系统及控制器
本申请要求于2016年11月30日提交中国专利局、申请号为201611080630.X、发明名称为“一种脉冲宽度调制方法、脉冲宽度调制系统及控制器”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本申请涉及电路技术领域，尤其涉及一种脉冲宽度调制方法、脉冲宽度调制系统及控制器。
背景技术
随着经济社会的发展，能源危机逐步凸显以及全球环境的逐渐恶化，发展和使用清洁替代能源已成为的能源行业的重要目标。伴随新能源发电、储能以及新能源汽车产业的不断发展，作为核心能源控制装置的变流器成为清洁能源应用的关键因素之一。在众多种类的变流器中，三相变流器是应用最为广泛的变流器之一，用于连接三相交流电力系统以及直流电力系统并实现两个系统之间能量传递。根据能量流向的不同又区分为整流和逆变两种工作状况，其中能量从直流系统传递到交流系统被称为逆变，而从交流系统传递到直流系统称为整流。转换效率和电能质量是三相变流器的两个关键技术指标，而调制方式直接影响着开关器件的通断状态，因此是影响其转换效率和电能质量的关键因素之一。
常用的三相变流器的脉宽调制手段为脉冲宽度调制(Pulse Width Modulation，PWM)，即对开关网络中各器件的驱动脉冲的宽度进行控制。最直接的实现形式为将载波与调制波进行比较，以比较结果来控制开关器件的通断状态。PWM又可以分为连续脉冲宽度调制(Continuous Pulse Width Modulation，CPWM)和不连续脉冲宽度调制(Discontinuous Pulse Width Modulation，DPWM)，DPWM和CPWM相比，DPWM的开关次数更少，因此开关损耗较小，由此带来的益处即是能够提高变流器的能量转换效率。但是，采用的调制方式为DPWM时，谐波畸变率普遍高于CPWM的方式，注入谐波含量高，更容易引起系统谐振，并且三相变流器调制度越低时，DPWM注入的谐波含量越高，而三相变流器调制度较高时，DPWM和CPWM注入水平接近。因此在三相变流器调制度较低的同时，又要兼顾能量转换效率，就需要采用DPWM，此时，对DPWM的共模注入方法进行改进，降低其注入共模含量，从而降低由共模注入产生的谐波含量就成为了研究的重点。
为了降低三相变流器采用DPWM时，由于共模注入产生的谐波含量，现有的两种技术分别为：一、对共模电压的上升率进行限制，以减少由于共模电压迅速变化而产生的谐波，等效延长共模电压变化时间并减少三相变流器的切换状态的时间；二、以直流母线电压为参考对象，量测直流母线电压计算出开关管钳位时间及对应的导通角度，及增加限幅度环节，根据直流母线电压、开关管钳位时间及相位信息计算出三相对应的共模调制电压，根据三相的基频正弦调制电压与共模调制电压计算出对应的最终调制电压。
但是，方法一中如果限幅值取值太小，则无法明显减少三相变流器在DPWM下的切换状态的时间，效率提升的优势减小，若限幅值取值太大，则难以抑制高频谐波，实施起来不够灵活；根据实际测验，在三相变流器中决定共模调制电压大小的并不是直流母线电压， 因此方法二不适用于三相变流器采用DPWM的场景中，而且计算过程比较复杂，计算耗时过长，在三相变流器这种开关频率较高的场合中不适用。
发明内容
本申请提供了一种脉冲宽度调制方法、脉冲宽度调制系统及控制器，用于三相变流器的共模分量的变化速率随着变流器调制度的变化而改变，提升三相变流器的稳定性和谐波特性，并且实现灵活地自适应调节。
本发明第一方面提供一种脉冲宽度调制方法，应用于三相变流器，包括：
获取三相初始调制波及变流器调制度；
根据预设调制参数和所述变流器调制度计算得到共模分量变化速率调节系数，所述预设调制参数包括所述三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数；
根据预设的调制波最大限幅值、调制波最小限幅值、所述三相初始调制波和所述共模分量变化速率调节系数，计算得到所述三相初始调制波对应的调制波集合，从所述调制波集合中选取绝对值最小的调制波作为共模调制波；
将所述三相初始调制波与所述共模调制波进行波形叠加，得到三相输出调制波。
三相交流电力系统和直流电力系统之间的能量传递，一般是通过三相变流器实现的，三相变流器的调制方式为DPWM情况下，变流器调制度比较高时，实现DPWM注入的共模分量较小，此时输出波形接近正弦波，既可满足降低损耗亦可满足输出电流畸变小的要求，此时应较少限制注入共模分量的变化速率，当变流器调制度逐渐降低时，实现DPWM注入的共模分量逐渐较大，此时输出波形畸变严重，应降低注入共模分量的变化速率，而共模分量的变化速度是由共模分量变化速率调节系数所决定的，共模分量变化速率调节系数是由预设调制参数和变流器调制度计算得来的，预设调制参数包括三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数，根据预设的调制波最大限幅值、调制波最小限幅值、三相初始调制波和共模分量变化速率调节系数，计算得到三相初始调制波对应的调制波集合，从调制波集合中选取绝对值最小的调制波作为共模调制波，最后，将三相初始调制波与共模调制波进行波形叠加，得到三相输出调制波，与现有技术的方法一相比，因为共模分量的变化速率由变流器调制度决定，随着变流器调制度变化而改变，没有限幅环节，因此可以避免限幅值取值不当造成的影响，提升了三相变流器的稳定性和谐波特性，与现有技术二相比，共模分量的变化速率由变流器调制度决定，实现了自适应调节。
结合本发明第一方面，本发明第一方面第一实施方式中，所述根据预设调制系数和所述变流器调制度，计算得到共模分量变化速率调节系数，包括：
获取所述三相变流器的预置最大调制度Mmax、预置最小调制度Mmin、预置正负钳位状态之间共模量变化最小速率Kb、预置正负钳位状态之间共模量变化最大速率与最小速率的 差值Ka及调制曲率参数N，N大于0；
将所述Mmax、所述Mmin、所述Kb、所述Ka、所述N及所述变流器调制度M带入公式
Figure PCTCN2017081600-appb-000001
计算得到共模分量变化速率调节系数K。
在方案实施之前，需要预设调制系数，其中，最大调制度Mmax和最小调制度Mmin是三相变流器所容许的最大调制度(一般Mmax不超过1.15)和最小调制度(一般Mmin小于1)，是由变流器产品设计和应用场景所决定的，因此，最大调制度Mmax和最小调制度Mmin相当于两个固定的数值；正负钳位状态之间共模量变化最小速率Kb是由三相变流器所处的场景所决定的，也可以是用户自由设置的；正负钳位状态之间共模量变化最大速率与Kb的差值Ka，假设三相变流器所能允许的正负钳位状态之间共模量变化最大速率为1，那么Ka等于1-Kb；调制曲率参数N的取值是预先设置的，N为1的情况下，共模分量变化速率调节系数和变流器调制度之间为直线关系，N大于零且不等于1的情况下，共模分量变化速率调节系数和变流器调制度之间为曲线关系。在已知M、Mmax、Mmin、Kb、Ka及N的值的情况下，通过公式就能计算得出共模分量变化速率调节系数K的值，那么K的值是由M所决定，实现了通过变流器调制度M灵活地调节共模分量变化速率的目标。
结合本发明第一方面第一实施方式，本发明第一方面第二实施方式中，所述根据预设的调制波最大限幅值、调制波最小限幅值、所述三相初始调制波和所述共模分量变化速率调节系数，计算得到所述三相初始调制波对应的调制波集合，从所述调制波集合中选取绝对值最小的调制波作为共模调制波，包括：
获取预设的调制波最大限幅值vmax和调制度最小限幅值vmin；
根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第一相初始调制波va，计算得到所述va对应的调制波集合{vmax-va，K*(vmax/2+vmin/2-va)，vmin-va}，所述vmax-va为所述va对应的正钳位状态下的调制波变量，所述vmin-va为所述va对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-va)为所述va对应的切换状态下的调制波变量；
根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第二相初始调制波vb，计算得到所述vb对应的调制波集合{vmax-vb，K*(vmax/2+vmin/2-vb)，vmin-vb}，所述vmax-vb为所述vb对应的正钳位状态下的调制波变量，所述vmin-vb为所述vb对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-vb)为所述vb对应的切换状态下的调制波变量；
根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第二相初始调制波vc，计算得到所述vc对应的调制波集合{vmax-vc，K*(vmax/2+vmin/2-vc)，vmin-vc}，所述vmax-vc为所述vc对应的正钳位状态下的调制波变量，所述vmin-vc为所述vc对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-vc)为所述vc对应的切换状态下的调制波变量；
根据所述va、所述vb和所述vc对应的调制波集合，得到所述三相初始调制波对应的调制波集合{vmax-va，vmax-vb，vmax-vc，K*(vmax/2+vmin/2-va)，K*(vmax/2+vmin/2-vb)， K*(vmax/2+vmin/2-vc)，vmin-va，vmin-vb，vmin-vc}；
对所述共模调制波集合中每一个调制波变量取绝对值，选择绝对值最小的调制波变量作为共模调制波。
调制波最大限幅值vmax和调制度最小限幅值vmin是预先设置的，由于三相变流器中每一相对应一个开关，每一个开关存在三种钳位状态，那么根据vmax、vmin、共模分量变化速率调节系数K及三相初始调制波中的第一相初始调制波va，计算得到va对应的调制波集合{vmax-va，K*(vmax/2+vmin/2-va)，vmin-va}，vmax-va为va对应的正钳位状态下的调制波变量，vmin-va为va对应的负钳位状态下的调制波变量，K*(vmax/2+vmin/2-va)为va对应的切换状态下的调制波变量，依次对计算得到第二相初始调制波vb对应的调制波集合{vmax-vb，K*(vmax/2+vmin/2-vb)，vmin-vb}，计算得到第三相初始调制波vb对应的调制波集合{vmax-vb，K*(vmax/2+vmin/2-vb)，vmin-vb}，将三个相的初始调制波的调制波集合合并为三相初始调制波对应的调制波集合{vmax-va，vmax-vb，vmax-vc，K*(vmax/2+vmin/2-va)，K*(vmax/2+vmin/2-vb)，K*(vmax/2+vmin/2-vc)，vmin-va，vmin-vb，vmin-vc}，对共模调制波集合中每一个调制波变量取绝对值，选择绝对值最小的调制波变量作为共模调制波。
结合本发明第一方面、第一方面第一实施方式或第一方面第二实施方式，本发明第一方面第三实施方式中，所述将所述三相初始调制波与所述共模调制波进行波形叠加，得到三相输出调制波，包括：
将所述三相初始调制波中的每一相初始调制波分别与所述共模调制波进行一一对应的波形叠加，得到三相输出调制波。
在得到共模调制波之后，三相变流器需要通过将共模调制波中每一相初始调制波分别与共模调制波进行一一对应的波形叠加，从而得到三相输出调制波。
结合本发明第一方面，本发明第一方面第四实施方式中，获取初始调制波及变流器调制度，包括：
获取三相电网电压、三相电流及预设载波幅值；
对所述三相电网电压进行锁相处理，得到旋转坐标系相位；
根据所述旋转坐标系相位，对所述三相电流进行坐标变换，得到旋转坐标系电流；
获取所述三相电流坐标变换的预设电流参考值，并计算所述预设电流参考值与所述旋转坐标系电流的差值，得到电流差值；
通过PI调节器对所述电流差值进行处理，得到旋转坐标系调节分量；
对所述旋转坐标系调节分量进行反坐标变换，得到三相初始调制波；
获得所述三相初始调制波的调制波幅值，并根据所述调制波幅值和所述预置载波幅值的比值，得到变流器调制度。
检测三相变流器得到三相电网电压、三相电流及预设载波幅值，把三相电网电压送入锁相环，得到旋转坐标系相位，根据旋转坐标系相位，对三相电流进行坐标变换，得到旋转坐标系电流，获取三相电流坐标变换的预设电流参考值，并计算预设电流参考值与旋转坐标系电流的差值，得到电流差值，通过PI调节器对电流差值进行处理，得到旋转坐标系调节分量，对旋转坐标系调节分量进行反坐标变换，得到三相初始调制波，获得三相初始 调制波的调制波幅值，并根据调制波幅值和预置载波幅值的比值，得到变流器调制度。
本发明第二方面提供一种脉冲宽度调制系统，应用于三相变流器，包括：
获取模块，用于获取三相初始调制波及变流器调制度；
计算模块，用于根据预设调制参数和所述变流器调制度计算得到共模分量变化速率调节系数，所述预设调制参数包括所述三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数；
计算模块，还用于根据预设的调制波最大限幅值、调制波最小限幅值、所述三相初始调制波和所述共模分量变化速率调节系数，计算得到所述三相初始调制波对应的调制波集合，从所述调制波集合中选取绝对值最小的调制波作为共模调制波；
调制模块，还用于将所述三相初始调制波与所述共模调制波进行波形叠加，得到三相输出调制波。
三相交流电力系统和直流电力系统之间的能量传递，一般是通过三相变流器实现的，三相变流器的调制方式为DPWM情况下，变流器调制度比较高时，实现DPWM注入的共模分量较小，此时输出波形接近正弦波，既可满足降低损耗亦可满足输出电流畸变小的要求，此时应较少限制注入共模分量的变化速率，当变流器调制度逐渐降低时，实现DPWM注入的共模分量逐渐较大，此时输出波形畸变严重，应降低注入共模分量的变化速率，而共模分量的变化速度是由共模分量变化速率调节系数所决定的，共模分量变化速率调节系数是由预设调制参数和变流器调制度计算得来的，预设调制参数包括三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数，根据预设的调制波最大限幅值、调制波最小限幅值、三相初始调制波和共模分量变化速率调节系数，计算得到三相初始调制波对应的调制波集合，从调制波集合中选取绝对值最小的调制波作为共模调制波，最后，将三相初始调制波与共模调制波进行波形叠加，得到三相输出调制波，与现有技术的方法一相比，因为共模分量的变化速率由变流器调制度决定，随着变流器调制度变化而改变，没有限幅环节，因此可以避免限幅值取值不当造成的影响，提升了三相变流器的稳定性和谐波特性，与现有技术二相比，共模分量的变化速率由变流器调制度决定，实现了自适应调节。
结合本发明第二方面，本发明第二方面第一实施方式中，
所述获取模块，还用于获取所述三相变流器的预置最大调制度Mmax、预置最小调制度Mmin、预置正负钳位状态之间共模量变化最小速率Kb、预置正负钳位状态之间共模量变化最大速率与最小速率的差值Ka及调制曲率参数N，N大于0；
所述计算模块，还用于将所述Mmax、Mmin、Kb、Ka、N及所述变流器调制度M带入公式
Figure PCTCN2017081600-appb-000002
计算得到共模分量变化速率调节系数K。
在方案实施之前，需要预设调制系数，其中，最大调制度Mmax和最小调制度Mmin是三相变流器所容许的最大调制度(一般Mmax不超过1.15)和最小调制度(一般Mmin小于1)， 是由变流器产品设计和应用场景所决定的，因此，最大调制度Mmax和最小调制度Mmin相当于两个固定的数值；正负钳位状态之间共模量变化最小速率Kb是由三相变流器所处的场景所决定的，也可以是用户自由设置的；正负钳位状态之间共模量变化最大速率与Kb的差值Ka，假设三相变流器所能允许的正负钳位状态之间共模量变化最大速率为1，那么Ka等于1-Kb；调制曲率参数N的取值是预先设置的，N为1的情况下，共模分量变化速率调节系数和变流器调制度之间为直线关系，N大于零且不等于1的情况下，共模分量变化速率调节系数和变流器调制度之间为曲线关系。在已知M、Mmax、Mmin、Kb、Ka及N的值的情况下，通过公式就能计算得出共模分量变化速率调节系数K的值，那么K的值是由M所决定，实现了通过变流器调制度M灵活地调节共模分量变化速率的目标。
结合本发明第二方面第一实施方式，本发明第二方面第二实施方式中，
所述获取模块，还用于获取预设的调制波最大限幅值vmax和调制度最小限幅值vmin；
所述计算模块，还用于根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第一相初始调制波va，计算得到所述va对应的调制波集合{vmax-va，K*(vmax/2+vmin/2-va)，vmin-va}，所述vmax-va为所述va对应的正钳位状态下的调制波变量，所述vmin-va为所述va对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-va)为所述va对应的切换状态下的调制波变量；
所述计算模块，还用于根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第二相初始调制波vb，计算得到所述vb对应的调制波集合{vmax-vb，K*(vmax/2+vmin/2-vb)，vmin-vb}，所述vmax-vb为所述vb对应的正钳位状态下的调制波变量，所述vmin-vb为所述vb对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-vb)为所述vb对应的切换状态下的调制波变量；
所述计算模块，还用于根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第二相初始调制波vc，计算得到所述vc对应的调制波集合{vmax-vc，K*(vmax/2+vmin/2-vc)，vmin-vc}，所述vmax-vc为所述vc对应的正钳位状态下的调制波变量，所述vmin-vc为所述vc对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-vc)为所述vc对应的切换状态下的调制波变量；
所述计算模块，还用于根据所述va、所述vb和所述vc对应的调制波集合，得到所述三相初始调制波对应的调制波集合{vmax-va，vmax-vb，vmax-vc，K*(vmax/2+vmin/2-va)，K*(vmax/2+vmin/2-vb)，K*(vmax/2+vmin/2-vc)，vmin-va，vmin-vb，vmin-vc}；
所述计算模块，还用于对所述共模调制波集合中每一个调制波变量取绝对值，选择绝对值最小的调制波变量作为共模调制波。
调制波最大限幅值vmax和调制度最小限幅值vmin是预先设置的，由于三相变流器中每一相对应一个开关，每一个开关存在三种钳位状态，那么根据vmax、vmin、共模分量变化速率调节系数K及三相初始调制波中的第一相初始调制波va，计算得到va对应的调制波集合{vmax-va，K*(vmax/2+vmin/2-va)，vmin-va}，vmax-va为va对应的正钳位状态下的调制波变量，vmin-va为va对应的负钳位状态下的调制波变量，K*(vmax/2+vmin/2-va)为va对应的切换状态下的调制波变量，依次对计算得到第二相初始调制波vb对应的调制波集合{vmax-vb， K*(vmax/2+vmin/2-vb)，vmin-vb}，计算得到第三相初始调制波vb对应的调制波集合{vmax-vb，K*(vmax/2+vmin/2-vb)，vmin-vb}，将三个相的初始调制波的调制波集合合并为三相初始调制波对应的调制波集合{vmax-va，vmax-vb，vmax-vc，K*(vmax/2+vmin/2-va)，K*(vmax/2+vmin/2-vb)，K*(vmax/2+vmin/2-vc)，vmin-va，vmin-vb，vmin-vc}，对共模调制波集合中每一个调制波变量取绝对值，选择绝对值最小的调制波变量作为共模调制波。
结合本发明第二方面、第二方面第一实施方式或第二方面第二实施方式，本发明第二方面第三实施方式中，
所述调制模块，具体用于将所述三相初始调制波中的每一相初始调制波分别与所述共模调制波进行一一对应的波形叠加，得到三相输出调制波。
在得到共模调制波之后，三相变流器需要通过将共模调制波中每一相初始调制波分别与共模调制波进行一一对应的波形叠加，从而得到三相输出调制波。
结合本发明第二方面，本发明第二方面第四实施方式中，
所述获取模块，用于获取三相电网电压、三相电流及预设载波幅值；
所述计算模块，还用于对所述三相电网电压进行锁相处理，得到旋转坐标系相位；
所述计算模块，还用于根据所述旋转坐标系相位，对所述三相电流进行坐标变换，得到旋转坐标系电流；
所述获取模块，还用于获取所述三相电流坐标变换的预设电流参考值，并计算所述预设电流参考值与所述旋转坐标系电流的差值，得到电流差值；
所述计算模块，还用于通过比例积分PI调节器对所述电流差值进行处理，得到旋转坐标系调节分量；
所述计算模块，还用于对所述旋转坐标系调节分量进行反坐标变换，得到三相初始调制波；
所述计算模块，还用于获得所述三相初始调制波的调制波幅值，并根据所述调制波幅值和所述预置载波幅值的比值，得到变流器调制度。
获取模块检测三相变流器得到三相电网电压、三相电流及预设载波幅值，计算模块把三相电网电压送入锁相环，得到旋转坐标系相位，根据旋转坐标系相位，对三相电流进行坐标变换，得到旋转坐标系电流，获取三相电流坐标变换的预设电流参考值，并计算预设电流参考值与旋转坐标系电流的差值，得到电流差值，通过PI调节器对电流差值进行处理，得到旋转坐标系调节分量，对旋转坐标系调节分量进行反坐标变换，得到三相初始调制波，获得三相初始调制波的调制波幅值，并根据调制波幅值和预置载波幅值的比值，得到变流器调制度。
本发明第三方面提供一种控制器，应用于三相变流器，包括：
处理器、存储器及信号接口之间互相连接，存储器中存储有处理器的运行指令，
处理器、存储器及信号接口之间互相连接，存储器中存储有处理器的运行指令，
所述信号接口，用于获取三相初始调制波及变流器调制度；
所述处理器，用于根据预设调制参数和所述变流器调制度计算得到共模分量变化速率调节系数，所述预设调制参数包括所述三相变流器的预置最大调制度、预置最小调制度、 预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数；
所述处理器，还用于根据预设的调制波最大限幅值、调制波最小限幅值、所述三相初始调制波和所述共模分量变化速率调节系数，计算得到所述三相初始调制波对应的调制波集合，从所述调制波集合中选取绝对值最小的调制波作为共模调制波；
所述处理器，还用于将所述三相初始调制波与所述共模调制波进行波形叠加，得到三相输出调制波。
共模分量变化速率调节系数是由处理器根据预设调制参数和变流器调制度计算得来的，预设调制参数包括三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数，根据预设的调制波最大限幅值、调制波最小限幅值、三相初始调制波和共模分量变化速率调节系数，计算得到三相初始调制波对应的调制波集合，从调制波集合中选取绝对值最小的调制波作为共模调制波，最后，将三相初始调制波与共模调制波进行波形叠加，得到三相输出调制波，与现有技术的方法一相比，因为共模分量的变化速率由变流器调制度决定，随着变流器调制度变化而改变，没有限幅环节，因此可以避免限幅值取值不当造成的影响，提升了三相变流器的稳定性和谐波特性，与现有技术二相比，共模分量的变化速率由变流器调制度决定，实现了自适应调节。
附图说明
为了更清楚地说明本申请实施例技术方案，下面将对实施例和现有技术描述中所需要使用的附图作简单地介绍。
图1为本申请提供的三相变流器的系统结构图；
图2为本申请提供的共模信号的一个波形示意图；
图3为本申请提供的变流器调制度和谐波畸变率的曲线关系图；
图4为本申请提供的变流器调制度M＝1.1时的DPWM调制波的波形示意图；
图5为本申请提供的变流器调制度M＝0.88时的DPWM调制波的波形示意图；
图6为本申请提供的脉冲宽度调制方法的一个实施例的流程示意图；
图7为本申请提供的三电平并网光伏逆变器的结构示意图；
图8为本申请提供的三电平并网光伏逆变器控制算法示意图；
图9为本申请提供的调制曲率参数、变流器调制度与共模分量变化速率调节系数的关系图；
图10为本申请提供的变流器调制度M＝1.1时本发明DPWM与常规DPWM效果对比图；
图11为本申请提供的变流器调制度M＝0.8时本发明DPWM与常规DPWM效果对比图；
图12为本申请提供的脉冲宽度调制系统的一个实施例的结构示意图；
图13为本申请提供的控制器的一个实施例结的构示意图；
图14为本申请提供的两电平整流器和变流器的结构示意图；
图15为本申请提供的三电平整流器和变流器的结构示意图；
图16为本申请提供的五电平整流器和变流器的结构示意图；
图17为本申请提供的级联多电平整流器和变流器的结构示意图。
具体实施方式
本申请提供了一种脉冲宽度调制方法、脉冲宽度调制系统及控制器，用于三相变流器的共模分量的变化速率随着变流器调制度的变化而改变，提升三相变流器的稳定性和谐波特性，并且实现灵活地自适应调节。
下面将结合本申请中的附图，对本申请中的技术方案进行清楚、完整地描述，显然，所描述的实施例仅仅是本发明一部分实施例，而不是全部的实施例。
首先简单介绍本发明应用的系统构架或场景。
本发明应用于变流器，尤其是目前应用最广泛的三相变流器，三相变流器用于连接三相交流电力系统以及直流电力系统并实现两个系统之间能量传递。根据能量流向的不同又区分为整流和逆变两种工作状况，其中能量从直流系统传递到交流系统被称为逆变，而从交流系统传递到直流系统称为整流。因此，大部分应用场景下，整流和逆变均可采用同样的系统来实现，典型的三相变流器的系统结构如图1所示，包括直流系统、开关网络、控制器、滤波器及交流系统，滤波器用于在整流的过程中，进行交流电的过滤，开关网络包括开关器件，控制器通过调制方式控制开关网络中的各开关器件的导通或断开动作，从而开启和关闭交流系统和直流系统之间的能量传递。
三相变流器采用的脉宽调制手段为PWM，即对开关网络中各器件的驱动脉冲的宽度进行控制。最直接的实现形式为将载波与调制波进行比较，以比较结果来控制开关器件的通断状态。PWM又可以分为CPWM和DPWM。CPWM是指在每个开关周期内每相桥臂总有一个开关动作，常见的方式有正弦脉宽调制(Sinusoidal Pulse Width Modulation，SPWM)；DPWM是指在一定的开关时期内，变流器的某相桥臂被钳位在正直流母线或者负直流母线，在该钳位时间内，该相开关器件常通或常断，没有开关动作。常见的DPWM调制包括DPWM0，DPWM1，DPWM2，DPWM3，DPWMMAX，DPWMMIN和GDPWM(General DPWM)等方式。
在具体实现上，DPWM的调制波可以通过在CPWM的调制波上叠加等效共模分量来实现。例如，将一个工频周期(例如50Hz)内的DPWM调制波形与SPWM(CPWM的一种)调制波形进行对比，如图2所示，DPWM调制波和工频正弦波之差即是图2中所示共模信号的波形。由于DPWM的调制波可以等效于SPWM调制波与共模信号之和，因此DPWM的输出特性受到SPWM输出特性与共模信号输出特性的共同影响。注入的共模信号会影响变流器电能质量，甚至会产生谐振，从而对系统稳定性产生影响。图3为常见PWM调制方式下，变流器调制度M与谐波畸变率HDF的关系曲线图。如图3可以看出，DPWM的调制方式的谐波畸变率普遍高于CPWM的方式，但是在调制度比较高的时候，DPWM和CPWM的谐波畸变率较为接近，此时DPWM调制中注入的共模信号分量较少，而在调制 度较低时，为了实现钳位而注入的共模信号量较大，此时DPWM的谐波畸变率要比CPWM的大很多。因此，需要对调制度低时的DPWM的共模注入进行改进，降低其注入共模含量，从而降低由共模注入产生的谐波含量。
根据现有研究可知，当变流器调制度比较大时，实现DPWM注入的共模信号较小，如图4所示为变流器调制度M＝1.1时DPWM调制波的示意图。此时实现DPWM注入的共模信号较小，输出波形接近正弦波，输出电流畸变小，此时应采用常规DPWM调制，较少限制注入共模分量的变化速率；当变流器调制度比较小时，实现DPWM注入的共模信号较大,如图5所示为变流器调制度M＝0.88时DPWM调制波的示意图，此时输出波形畸变严重，应降低注入共模分量的变化速率，减少共模分量的高频分量，降低共模分量剧烈变化对变流器带来的冲击以及电流的畸变。
在目前的DPWM方案中，由于三相变流器有3个桥臂，每个桥臂又能够产生正钳位、负钳位和切换3种状态，这3个桥臂总共9个状态分别对应9种调制波。而三相变流器在任一时刻仅能实现一种钳位状态，否则变流器就失控，因此，三相变流器可能存在的钳位状态有三种：正钳位状态、负钳位状态和切换状态。根据分析，切换对于三相变流器效率提升远小于正钳位和负钳位，而且切换状态又是正钳位状态和负钳位状态的一个过渡状态，切换状态下的共模电压的分量直接影响到最终叠加的共模分量。
本发明提出一种脉冲宽度调制方法用于解决切换状态下共模分量对三相变流器所造成的影响，具体如下：
请参阅图6，本发明实施例提供一种脉冲宽度调制方法，包括：
601、获取三相初始调制波及变流器调制度；
本实施例中，三相初始调制波为三相交流器控制环路的每一相的初始调制波，该控制环路包括电压控制环路、电流控制环路以及功率控制环路中的一种或多种组合，变流器调制度可以通过计算变流器交流电压相峰值和直流电压的比值得到，或者通过计算三相初始调制波的峰值和载波峰值来获得，三相变流器通过调整调制波来对变流器的电压和电流进行控制，因此调制波是动态变化的。当变流器进入稳定运行状态，调制波是周期性重复的，此时调制波的峰值也可以认为是一个固定值，那么变流器调制度是可以准确的获得的。
602、根据预设调制参数和变流器调制度计算得到共模分量变化速率调节系数；
本实施例中，预设调制参数包括三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数，其中，预置最大调制度和预置最小调制度是由三相变流器产品设计和应用场景所决定的，预置正负钳位状态之间共模量变化最小速率及预置正负钳位状态之间共模量变化最大速率与最小速率的差值是由三相变流器所处的场景所决定的，也可以是用户自由设置的，调制曲率参数也是预先设置的，因此，在方案实施之前预设调制参数中各参数的值均是已知的，根据预设调制参数和变流器调制度计算得到共模分量变化速率调节系数，共模分量变化速率调节系数是用于共模分量的变化速率的，取值越大，表示共模分量的变化速率越快。
603、根据预设的调制波最大限幅值、调制波最小限幅值、三相初始调制波和共模分量变化速率调节系数，计算得到三相初始调制波对应的调制波集合，从调制波集合中选取绝对值最小的调制波作为共模调制波；
本实施例中，调制波最大限幅值和调制度最小限幅值是预先设置的，由于三相变流器中每一相对应一个开关，每一个开关存在三种钳位状态，那么三相初始调制波中的每一相调制波均对应的可以得到三种钳位状态的调制波，由此计算得到三相初始调制波对应的调制波集合包含九个调制波，从调制波集合中的九个调制波中选取绝对值最小的调制波作为共模调制波。
604、将三相初始调制波与共模调制波进行波形叠加，得到三相输出调制波。
本实施例中，将三相初始调制波和共模调制波进行波形叠加，得到对应的三相输出调制波，从三相输出调制波的波形畸变情况可以观察到，变流器调制度与共模分量的变化速率是成正比的。
本发明实施例中，共模分量的变化速度是由共模分量变化速率调节系数所决定的，共模分量变化速率调节系数是由预设调制参数和变流器调制度计算得来的，预设调制参数包括三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数，根据预设的调制波最大限幅值、调制波最小限幅值、三相初始调制波和共模分量变化速率调节系数，计算得到三相初始调制波对应的调制波集合，从调制波集合中选取绝对值最小的调制波作为共模调制波，最后，将三相初始调制波与共模调制波进行波形叠加，得到三相输出调制波，与现有技术的方法一相比，因为共模分量的变化速率由变流器调制度决定，随着变流器调制度变化而改变，没有限幅环节，因此可以避免限幅值取值不当造成的影响，提升了三相变流器的稳定性和谐波特性，与现有技术二相比，共模分量的变化速率由变流器调制度决定，实现了自适应调节。
三相变流器可以分为逆变器状态和整流器状态，本发明的方案可以应用于逆变器或整流器中，以图7所示的光伏逆变器应用系统为例，逆变器结构为两电平结构，直流母线与光伏电池板相连，交流端口通过L滤波器与三相交流电网相连。在控制系统中定义电流正方向为流出逆变器桥臂端口方向，如图8所示为逆变器的算法图。
可选的，本发明的一些实施例中，获取三相初始调制波及变流器调制度，包括：
获取三相电网电压、三相电流及预设载波幅值；
对三相电网电压进行锁相处理，得到旋转坐标系相位；
根据旋转坐标系相位，对三相电流进行坐标变换，得到旋转坐标系电流；
获取三相电流坐标变换的预设电流参考值，并计算预设电流参考值与旋转坐标系电流的差值，得到电流差值；
通过PI调节器对电流差值进行处理，得到旋转坐标系调节分量；
对旋转坐标系调节分量进行反坐标变换，得到三相初始调制波；
获得三相初始调制波的调制波幅值，并根据调制波幅值和预置载波幅值的比值，得到变流器调制度。
本发明实施例中，结合图8所示的光伏逆变器，获取三相电网电压vga、vgb和vgc，及三相电流ia、ib和ic，及变流器调制度M，首先将三相电网电压vga、vgb和vgc送入锁相环(PLL)，得到旋转坐标系相位θ，再将三相电流ia、ib和ic进行坐标变换，得到其在旋转坐标系电流id和iq，坐标变换的计算方法为帕克变换(Park Transformation)，其计算方程为：
Figure PCTCN2017081600-appb-000003
将计算得到的id和iq与其参考值idref和iqref进行比较，将其差值分别送入调节器Gd和Gq，得到d轴和q轴旋转坐标系调节分量vd和vq，其中Gd和Gq均为比例积分(Proportional Integral，PI)调节器。
对调节分量vd和vq进行反坐标变换，得到其在静止坐标系中的等效值va、vb和vc，计算方法为逆帕克变换(Inverse Park Transformation)，其计算方程为：
Figure PCTCN2017081600-appb-000004
根据三相初始调制波va、vb和vc的调制波幅值，以及预置载波幅值的比值，得到逆变器调制度，逆变器调制度即变流器调制度。
需要说明的是，上述实施例根据三相电网电压和三相电流的坐标变换得到三相初始调制波，只是获取初始调制波的一种方式，在实际应用中，还可以有其他方式，具体不做限定。
可选的，本发明的一些实施例中，根据预设调制系数和变流器调制度，计算得到共模分量变化速率调节系数，包括：
获取三相变流器的预置最大调制度Mmax、预置最小调制度Mmin、预置正负钳位状态之间共模量变化最小速率Kb、预置正负钳位状态之间共模量变化最大速率与最小速率的差值Ka及调制曲率参数N，N大于0；
将Mmax、Mmin、Kb、Ka、N及变流器调制度M带入公式
Figure PCTCN2017081600-appb-000005
计算得到共模分量变化速率调节系数K。
本发明实施例中，根据变流器调制度M及预设调制系数，计算得到共模分量变化速率调节系数K的公式为：
Figure PCTCN2017081600-appb-000006
其中，最大调制度Mmax和最小调制度Mmin是三相变流器所容许的最大调制度(一般Mmax不超过1.15)和最小调制度(一般Mmin小于1)，是由变流器产品设计和应用场景所决定的，因此，最大调制度Mmax和最小调制度Mmin相当于两个固定的数值；正负钳位状态之间共模量变化最小速率Kb是由三相变流器所处的场景所决定的，也可以是用户自由设置的；正负钳位状态之间共模量变化最大速率与Kb的差值Ka，假设三相变流器所能允许的正负钳位状态之间共模量变化最大速率为1，那么Ka等于1-Kb；调制曲率参数N的取值是预先设置的，N为1的情况下，预设共模分量调节公式为直线公式，N大于零且不等于1的情况下，预设共模分量调节公式是曲线方程。在获取模块获取M、Mmax、Mmin、Kb、Ka及N的值的情况下，计算模块通过公式就能计算得出共模分量变化速率调节系数K的值，并且Mmax、Mmin、Kb、Ka及N的值都是预先设置并且可调节的，那么K的值是由M所决定，实现了通过变流器调制度M灵活地调节共模分量变化速率的目标。
假设Mmax为1.15，Mmin为0.8，N为1，Ka和Kb均为0.5，则可以根据M计算出共模分量变化速率调节系数K，其数学关系为：
Figure PCTCN2017081600-appb-000007
得到的变流器调制度M与共模分量变化速率调节系数K的关系图，如图9中N＝1的直线，具体调制曲率参数N需要根据期望的自适应调节变化程度来改变，例如图9中，当N＝0.01时，可以看到大部分范围内K值均接近或等于1，这样切换状态的范围就比较大，而当K值不等于1时，所处的过渡状态覆盖范围比较小。这个需要结合三相变流器硬件系统结构以及参数来共同设定。类似，若N＝99时，可以看到过渡状态(K不等于1)占据了大部分空间，而切换状态(K＝1)仅在最大调制度时方才实现，因此，预设调制系数可以实现减少三相变流器切换状态的时间，进一步提高三相变流器的效率。
可选的，本发明的一些实施例中，根据预设的调制波最大限幅值、调制波最小限幅值、三相初始调制波和共模分量变化速率调节系数，计算得到三相初始调制波对应的调制波集合，从调制波集合中选取绝对值最小的调制波作为共模调制波，包括：
获取预设的调制波最大限幅值vmax和调制度最小限幅值vmin；
根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第一相初始调制波va，计算得到所述va对应的调制波集合{vmax-va，K*(vmax/2+vmin/2-va)，vmin-va}，所述vmax-va为所述va对应的正钳位状态下的调制波变量，所述vmin-va为所述va对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-va)为所述va对应的切换状态下的调制波变量；
根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第二相初始调制波vb，计算得到所述vb对应的调制波集合{vmax-vb，K*(vmax/2+vmin/2-vb)，vmin-vb}，所述vmax-vb为所述vb对应的正钳位状态下的调制波变量，所述vmin-vb为所述vb对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-vb)为所述vb对应的切换状态下的调制波变量；
根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波 中的第二相初始调制波vc，计算得到所述vc对应的调制波集合{vmax-vc，K*(vmax/2+vmin/2-vc)，vmin-vc}，所述vmax-vc为所述vc对应的正钳位状态下的调制波变量，所述vmin-vc为所述vc对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-vc)为所述vc对应的切换状态下的调制波变量；
根据所述va、所述vb和所述vc对应的调制波集合，得到所述三相初始调制波对应的调制波集合{vmax-va，vmax-vb，vmax-vc，K*(vmax/2+vmin/2-va)，K*(vmax/2+vmin/2-vb)，K*(vmax/2+vmin/2-vc)，vmin-va，vmin-vb，vmin-vc}；
对所述共模调制波集合中每一个调制波变量取绝对值，选择绝对值最小的调制波变量作为共模调制波。
本发明实施例中，需要预先设置调制波的最大幅度值和最小幅度值，获取预设的调制波限幅值，将三相初始调制波、共模分量变化速率调节系数及调制波限幅值代入预置共模调制波变量集，由于三相变流器中每一相对应一个开关，每一个开关存在三种钳位状态，那么预置共模调制波变量集中包括每一相初始调制波对应的正钳位状态、负钳位状态和切换状态的调制波变量，所以调制波变量的数量为九个，为最小的影响系统的稳定性，选取九个调制波变量中绝对值最小的一个作为共模调制波。假设调制波限幅值为vmax＝1，vmin＝-1，将三相初始调制波va、vb和vc，共模分量变化速率调节系数K及调制波限幅值vmax＝1，vmin＝-1，代入预置共模调制波变量集{1-va，1-vb，1-vc，K*(1/2+-1/2-va)，K*(1/2+-1/2-vb)，K*(1/2+-1/2-vc)，-1-va，-1-vb，-1-vc}，其中包括九个变量，选择九个变量中绝对值最小的变量，作为共模调制波vZ，计算过程为：
1、当1-max{va,vb,vc}的绝对值小于-1-min{va,vb,vc}的绝对值，K*min{va,vb,vc}的绝对值以及K*max{va,vb,vc}的绝对值时，vZ＝1-max{va,vb,vc}；
2、当K*min{va,vb,vc}的绝对值小于1-max{va,vb,vc}的绝对值，-1-min{va,vb,vc}的绝对值以及K*max{va,vb,vc}的绝对值时，vZ＝-K*min{va,vb,vc}；
3、当K*max{va,vb,vc}的绝对值小于1-max{va,vb,vc}的绝对值，-1-min{va,vb,vc}的绝对值以及K*min{va,vb,vc}的绝对值时，vZ＝-K*max{va,vb,vc}；
4、当-1-min{va,vb,vc}的绝对值小于1-max{va,vb,vc}的绝对值，K*min{va,vb,vc}的绝对值以及K*max{va,vb,vc}的绝对值时，vZ＝-1-min{va,vb,vc}。
可选的，本发明的一些实施例中，将三相初始调制波与共模调制波进行波形叠加，得到三相输出调制波，包括：
将三相初始调制波中的每一相初始调制波分别与共模调制波进行一一对应的波形叠加，得到三相输出调制波。
本发明实施例中，共模调制波vZ与三相初始调制波va、vb和vc分别相加，得到新的三相输出调制波va_mod、vb_mod和vc_mod。
将本发明技术方案的实现效果与常规DPWM方案效果进行对比，其中变流器调制度M＝1.1时的对比结果如图10所示，图10(a)和图10(b)为变流器调制度M＝1.1时的常规DPWM调制波形及注入共模频谱图，图10(c)和图10(d)为本发明的DPWM调制波形及注入共模频谱图。经过图10的对比可以看出，在调制度M较高时，本方案和常规DPWM 方法调制波形以及频谱特性非常接近。此时，注入共模信号的高频分量较小，无需对共模电压源进行限制，可以使得变流器开关钳位效果最大化，提升效率。
变流器调制度M＝0.8时的对比结果如图11所示，图11(a)和图11(b)为变流器调制度M＝0.8时的常规DPWM调制波形及注入共模频谱图，图11(c)和图11(d)为本发明的DPWM调制波形及注入共模频谱图，经过图11的对比可以看出，随着调制度降低，常规DPWM注入的共模分量增大。而在调制度较低时，本方案和常规DPWM方法调制波形差距较大，本方案注入的共模分量较为平滑，频谱分量也远小于常规DPWM方案。
以上实施例描述的是脉冲宽度调制方法，下面通过实施例对脉冲宽度调制系统进行说明。
请参阅图12，本发明提供一种脉冲宽度调制系统，应用于三相变流器，包括：
获取模块1201，用于获取三相初始调制波及变流器调制度；
计算模块1202，用于根据预设调制参数和变流器调制度计算得到共模分量变化速率调节系数，预设调制参数包括三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数；
计算模块1202，还用于根据预设的调制波最大限幅值、调制波最小限幅值、三相初始调制波和共模分量变化速率调节系数，计算得到三相初始调制波对应的调制波集合，从调制波集合中选取绝对值最小的调制波作为共模调制波；
调制模块1203，还用于将三相初始调制波与共模调制波进行波形叠加，得到三相输出调制波。
本发明实施例中，共模分量的变化速度是由共模分量变化速率调节系数所决定的，共模分量变化速率调节系数是由计算模块1202根据获取模块1201获取到的变流器调制度计算得来的，并且计算模块1202根据三相初始调制波及共模分量变化速率调节系数得到共模调制波，调制模块1203根据三相初始调制波及共模调制波得到三相输出调制波，与现有技术一相比，因为共模分量的变化速率由变流器调制度决定，随着变流器调制度变化而改变，没有限幅环节，因此可以避免限幅值取值不当造成的影响，提升了三相变流器的稳定性和谐波特性，与现有技术二相比，共模分量的变化速率由变流器调制度决定，实现了自适应调节。
可选的，本发明的一些实施例中，
获取模块1201，还用于获取三相变流器的预置最大调制度Mmax、预置最小调制度Mmin、预置正负钳位状态之间共模量变化最小速率Kb、预置正负钳位状态之间共模量变化最大速率与最小速率的差值Ka及调制曲率参数N，N大于0；
计算模块1202，还用于将Mmax、Mmin、Kb、Ka、N及变流器调制度M带入公式
Figure PCTCN2017081600-appb-000008
计算得到共模分量变化速率调节系数K。
本发明实施例中，需要预设调制系数，其中，最大调制度Mmax和最小调制度Mmin是三相变流器所容许的最大调制度(一般Mmax不超过1.15)和最小调制度(一般Mmin小于1)，是由变流器产品设计和应用场景所决定的，因此，最大调制度Mmax和最小调制度Mmin相 当于两个固定的数值；正负钳位状态之间共模量变化最小速率Kb是由三相变流器所处的场景所决定的，也可以是用户自由设置的；正负钳位状态之间共模量变化最大速率与Kb的差值Ka，假设三相变流器所能允许的正负钳位状态之间共模量变化最大速率为1，那么Ka等于1-Kb；调制曲率参数N的取值是预先设置的，N为1的情况下，预设共模分量调节公式为直线公式，N大于零且不等于1的情况下，预设共模分量调节公式是曲线方程。在获取模块1201获取M、Mmax、Mmin、Kb、Ka及N的值的情况下，计算模块1202通过公式就能计算得出共模分量变化速率调节系数K的值，并且Mmax、Mmin、Kb、Ka及N的值都是预先设置并且可调节的，那么K的值是由M所决定，实现了通过变流器调制度M灵活地调节共模分量变化速率的目标。
可选的，本发明的一些实施例中，
获取模块1201，还用于获取预设的调制波最大限幅值vmax和调制度最小限幅值vmin；
计算模块1202，还用于根据vmax、vmin、共模分量变化速率调节系数K及三相初始调制波中的第一相初始调制波va，计算得到va对应的调制波集合{vmax-va，K*(vmax/2+vmin/2-va)，vmin-va}，vmax-va为va对应的正钳位状态下的调制波变量，vmin-va为va对应的负钳位状态下的调制波变量，K*(vmax/2+vmin/2-va)为va对应的切换状态下的调制波变量；
计算模块1202，还用于根据vmax、vmin、共模分量变化速率调节系数K及三相初始调制波中的第二相初始调制波vb，计算得到vb对应的调制波集合{vmax-vb，K*(vmax/2+vmin/2-vb)，vmin-vb}，vmax-vb为vb对应的正钳位状态下的调制波变量，vmin-vb为vb对应的负钳位状态下的调制波变量，K*(vmax/2+vmin/2-vb)为vb对应的切换状态下的调制波变量；
计算模块1202，还用于根据vmax、vmin、共模分量变化速率调节系数K及三相初始调制波中的第二相初始调制波vc，计算得到vc对应的调制波集合{vmax-vc，K*(vmax/2+vmin/2-vc)，vmin-vc}，vmax-vc为vc对应的正钳位状态下的调制波变量，vmin-vc为vc对应的负钳位状态下的调制波变量，K*(vmax/2+vmin/2-vc)为vc对应的切换状态下的调制波变量；
计算模块1202，还用于根据va、vb和vc对应的调制波集合，得到三相初始调制波对应的调制波集合{vmax-va，vmax-vb，vmax-vc，K*(vmax/2+vmin/2-va)，K*(vmax/2+vmin/2-vb)，K*(vmax/2+vmin/2-vc)，vmin-va，vmin-vb，vmin-vc}；
计算模块1202，还用于对共模调制波集合中每一个调制波变量取绝对值，选择绝对值最小的调制波变量作为共模调制波。
本发明实施例中，预先设置了调制波最大幅度值和最小幅度值，获取模块1201获取预设的调制波最大幅度值和调制波最小幅度值，由于三相变流器中每一相对应一个开关，每一个开关存在三种钳位状态，那么计算模块1202根据vmax、vmin、共模分量变化速率调节系数K及三相初始调制波中的第一相初始调制波va，计算得到va对应的调制波集合{vmax-va，K*(vmax/2+vmin/2-va)，vmin-va}，vmax-va为va对应的正钳位状态下的调制波变量，vmin-va为va对应的负钳位状态下的调制波变量，K*(vmax/2+vmin/2-va)为va对应的切换状态下的调制波变量，依次对计算得到第二相初始调制波vb对应的调制波集合{vmax-vb，K*(vmax/2+vmin/2-vb)，vmin-vb}，计算得到第三相初始调制波vb对应的调制波集合{vmax-vb，K*(vmax/2+vmin/2-vb)，vmin-vb}，计算模块1202将三个相的初始调制波的调制波集合合并为三相初始调制波对应的 调制波集合{vmax-va，vmax-vb，vmax-vc，K*(vmax/2+vmin/2-va)，K*(vmax/2+vmin/2-vb)，K*(vmax/2+vmin/2-vc)，vmin-va，vmin-vb，vmin-vc}，对共模调制波集合中每一个调制波变量取绝对值，选择绝对值最小的调制波变量作为共模调制波。
可选的，本发明的一些实施例中，
调制模块1203，具体用于将三相初始调制波中的每一相初始调制波分别与共模调制波进行一一对应的波形叠加，得到三相输出调制波。
可选的，本发明的一些实施例中，
获取模块1201，用于获取三相电网电压、三相电流及预设载波幅值；
计算模块1202，还用于对三相电网电压进行锁相处理，得到旋转坐标系相位；
计算模块1202，还用于根据旋转坐标系相位，对三相电流进行坐标变换，得到旋转坐标系电流；
获取模块1201，还用于获取三相电流坐标变换的预设电流参考值，并计算预设电流参考值与旋转坐标系电流的差值，得到电流差值；
计算模块1202，还用于通过比例积分PI调节器对电流差值进行处理，得到旋转坐标系调节分量；
计算模块1202，还用于对旋转坐标系调节分量进行反坐标变换，得到三相初始调制波；
计算模块1202，还用于获得三相初始调制波的调制波幅值，并根据调制波幅值和预置载波幅值的比值，得到变流器调制度。
本发明实施例中，获取模块1201检测三相变流器得到三相电网电压、三相电流及预设载波幅值，计算模块1202把三相电网电压送入锁相环，得到旋转坐标系相位，根据旋转坐标系相位，对三相电流进行坐标变换，得到旋转坐标系电流，获取三相电流坐标变换的预设电流参考值，并计算预设电流参考值与旋转坐标系电流的差值，得到电流差值，通过PI调节器对电流差值进行处理，得到旋转坐标系调节分量，对旋转坐标系调节分量进行反坐标变换，得到三相初始调制波，获得三相初始调制波的调制波幅值，并根据调制波幅值和预置载波幅值的比值，得到变流器调制度。
以上实施例对脉冲宽度调制方法和脉冲宽度调制系统进行了详细说明，下面对脉冲宽度调制系统的实体装置进行说明，实体装置为图13中所示的控制器，具体如下：
请参阅图13，本发明提供一种控制器，包括：
处理器1301、存储器1302及信号接口1303之间互相连接，存储器1302中存储有处理器1301的运行指令，
信号接口1303，用于获取三相初始调制波及变流器调制度；
处理器1301，用于根据预设调制参数和变流器调制度计算得到共模分量变化速率调节系数，预设调制参数包括三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数；
处理器1301，还用于根据预设的调制波最大限幅值、调制波最小限幅值、三相初始调制波和共模分量变化速率调节系数，计算得到三相初始调制波对应的调制波集合，从调制 波集合中选取绝对值最小的调制波作为共模调制波；
处理器1301，还用于将三相初始调制波与共模调制波进行波形叠加，得到三相输出调制波。
本发明实施例中，共模分量变化速率调节系数是由处理器1301根据预设调制参数和信号接口1303获取的变流器调制度计算得来的，预设调制参数包括三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数，处理器1301根据预设的调制波最大限幅值、调制波最小限幅值、三相初始调制波和共模分量变化速率调节系数，计算得到三相初始调制波对应的调制波集合，从调制波集合中选取绝对值最小的调制波作为共模调制波，最后，将三相初始调制波与共模调制波进行波形叠加，得到三相输出调制波，与现有技术的方法一相比，因为共模分量的变化速率由变流器调制度决定，随着变流器调制度变化而改变，没有限幅环节，因此可以避免限幅值取值不当造成的影响，提升了三相变流器的稳定性和谐波特性，与现有技术二相比，共模分量的变化速率由变流器调制度决定，实现了自适应调节。
图13所示的控制器还包括一个或一个以上存储应用程序1305或数据1306的存储介质1304(例如一个或一个以上海量存储设备)。其中，存储器1302和存储介质1304可以是短暂存储或持久存储。存储在存储介质1304的程序可以包括一个或一个以上模块(图示没标出)，每个模块可以包括对服务器中的一系列指令操作。更进一步地，处理器1301可以设置为与存储介质1304通信，在服务器上执行存储介质1304中的一系列指令操作。
控制器还可以包括一个或一个以上操作系统1307，例如Windows ServerTM，Mac OS XTM，UnixTM,LinuxTM或FreeBSDTM等等。
需要说明的是，本发明中所举的如图7所示的光伏逆变器应用系统可以等效为如图14所示的两电平整流器和逆变器结构，除此之外，还可以为图15所示的三电平整流器和变流器结构，还可以为图16所示的五电平整流器和变流器结构，还可以为图17所示的级联多电平整流器和变流器结构。
应理解，在本发明的各种实施例中，上述各过程的序号的大小并不意味着执行顺序的先后，各过程的执行顺序应以其功能和内在逻辑确定，而不应对本发明实施例的实施过程构成任何限定。
以上所述，以上实施例仅用以说明本发明的技术方案，而非对其限制；尽管参照前述实施例对本申请进行了详细的说明，本领域的普通技术人员应当理解：其依然可以对前述各实施例所记载的技术方案进行修改，或者对其中部分技术特征进行等同替换；而这些修改或者替换，并不使相应技术方案的本质脱离本申请各实施例技术方案的范围。

Claims (11)

  1. 一种脉冲宽度调制方法，应用于三相变流器，其特征在于，所述方法包括：
    获取三相初始调制波及变流器调制度；
    根据预设调制参数和所述变流器调制度计算得到共模分量变化速率调节系数，所述预设调制参数包括所述三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数；
    根据预设的调制波最大限幅值、调制波最小限幅值、所述三相初始调制波和所述共模分量变化速率调节系数，计算得到所述三相初始调制波对应的调制波集合，从所述调制波集合中选取绝对值最小的调制波作为共模调制波；
    将所述三相初始调制波与所述共模调制波进行波形叠加，得到三相输出调制波。
  2. 根据权利要求1所述的脉冲宽度调制方法，其特征在于，所述根据预设调制参数和所述变流器调制度计算得到共模分量变化速率调节系数，包括：
    获取所述三相变流器的预置最大调制度Mmax、预置最小调制度Mmin、预置正负钳位状态之间共模量变化最小速率Kb、预置正负钳位状态之间共模量变化最大速率与最小速率的差值Ka及调制曲率参数N，N大于0；
    将所述Mmax、所述Mmin、所述Kb、所述Ka、所述N及所述变流器调制度M带入公式
    Figure PCTCN2017081600-appb-100001
    计算得到共模分量变化速率调节系数K。
  3. 根据权利要求2所述的脉冲宽度调制方法，其特征在于，所述根据预设的调制波最大限幅值、调制波最小限幅值、所述三相初始调制波和所述共模分量变化速率调节系数，计算得到所述三相初始调制波对应的调制波集合，从所述调制波集合中选取绝对值最小的调制波作为共模调制波，包括：
    获取预设的调制波最大限幅值vmax和调制度最小限幅值vmin；
    根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第一相初始调制波va，计算得到所述va对应的调制波集合{vmax-va，K*(vmax/2+vmin/2-va)，vmin-va}，所述vmax-va为所述va对应的正钳位状态下的调制波变量，所述vmin-va为所述va对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-va)为所述va对应的切换状态下的调制波变量；
    根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第二相初始调制波vb，计算得到所述vb对应的调制波集合{vmax-vb，K*(vmax/2+vmin/2-vb)，vmin-vb}，所述vmax-vb为所述vb对应的正钳位状态下的调制波变量，所述vmin-vb为所述vb对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-vb)为所述vb对应的切换状态下的调制波变量；
    根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第二相初始调制波vc，计算得到所述vc对应的调制波集合{vmax-vc，K*(vmax/2+vmin/2-vc)，vmin-vc}，所述vmax-vc为所述vc对应的正钳位状态下的调制波变量，所述vmin-vc为所述vc 对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-vc)为所述vc对应的切换状态下的调制波变量；
    根据所述va、所述vb和所述vc对应的调制波集合，得到所述三相初始调制波对应的调制波集合{vmax-va，vmax-vb，vmax-vc，K*(vmax/2+vmin/2-va)，K*(vmax/2+vmin/2-vb)，K*(vmax/2+vmin/2-vc)，vmin-va，vmin-vb，vmin-vc}；
    对所述共模调制波集合中每一个调制波变量取绝对值，选择绝对值最小的调制波变量作为共模调制波。
  4. 根据权利要求1至3中任一项所述的脉冲宽度调制方法，其特征在于，所述将所述三相初始调制波与所述共模调制波进行波形叠加，得到三相输出调制波，包括：
    将所述三相初始调制波中的每一相初始调制波分别与所述共模调制波进行一一对应的波形叠加，得到三相输出调制波。
  5. 根据权利要求4所述的脉冲宽度调制方法，其特征在于，所述获取三相初始调制波及变流器调制度，包括：
    获取三相电网电压、三相电流及预设载波幅值；
    对所述三相电网电压进行锁相处理，得到旋转坐标系相位；
    根据所述旋转坐标系相位，对所述三相电流进行坐标变换，得到旋转坐标系电流；
    获取所述三相电流坐标变换的预设电流参考值，并计算所述预设电流参考值与所述旋转坐标系电流的差值，得到电流差值；
    通过比例积分PI调节器对所述电流差值进行处理，得到旋转坐标系调节分量；
    对所述旋转坐标系调节分量进行反坐标变换，得到三相初始调制波；
    获得所述三相初始调制波的调制波幅值，并根据所述调制波幅值和所述预置载波幅值的比值，得到变流器调制度。
  6. 一种脉冲宽度调制系统，应用于三相变流器，其特征在于，所述系统包括：
    获取模块，用于获取三相初始调制波及变流器调制度；
    计算模块，用于根据预设调制参数和所述变流器调制度计算得到共模分量变化速率调节系数，所述预设调制参数包括所述三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数；
    计算模块，还用于根据预设的调制波最大限幅值、调制波最小限幅值、所述三相初始调制波和所述共模分量变化速率调节系数，计算得到所述三相初始调制波对应的调制波集合，从所述调制波集合中选取绝对值最小的调制波作为共模调制波；
    调制模块，还用于将所述三相初始调制波与所述共模调制波进行波形叠加，得到三相输出调制波。
  7. 根据权利要求6所述的脉冲宽度调制系统，其特征在于，
    所述获取模块，还用于获取所述三相变流器的预置最大调制度Mmax、预置最小调制度Mmin、预置正负钳位状态之间共模量变化最小速率Kb、预置正负钳位状态之间共模量变化最大速率与最小速率的差值Ka及调制曲率参数N，N大于0；
    所述计算模块，还用于将所述Mmax、Mmin、Kb、Ka、N及所述变流器调制度M带入公 式
    Figure PCTCN2017081600-appb-100002
    计算得到共模分量变化速率调节系数K。
  8. 根据权利要求7所述的脉冲宽度调制系统，其特征在于，
    所述获取模块，还用于获取预设的调制波最大限幅值vmax和调制度最小限幅值vmin；
    所述计算模块，还用于根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第一相初始调制波va，计算得到所述va对应的调制波集合{vmax-va，K*(vmax/2+vmin/2-va)，vmin-va}，所述vmax-va为所述va对应的正钳位状态下的调制波变量，所述vmin-va为所述va对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-va)为所述va对应的切换状态下的调制波变量；
    所述计算模块，还用于根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第二相初始调制波vb，计算得到所述vb对应的调制波集合{vmax-vb，K*(vmax/2+vmin/2-vb)，vmin-vb}，所述vmax-vb为所述vb对应的正钳位状态下的调制波变量，所述vmin-vb为所述vb对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-vb)为所述vb对应的切换状态下的调制波变量；
    所述计算模块，还用于根据所述vmax、所述vmin、所述共模分量变化速率调节系数K及所述三相初始调制波中的第二相初始调制波vc，计算得到所述vc对应的调制波集合{vmax-vc，K*(vmax/2+vmin/2-vc)，vmin-vc}，所述vmax-vc为所述vc对应的正钳位状态下的调制波变量，所述vmin-vc为所述vc对应的负钳位状态下的调制波变量，所述K*(vmax/2+vmin/2-vc)为所述vc对应的切换状态下的调制波变量；
    所述计算模块，还用于根据所述va、所述vb和所述vc对应的调制波集合，得到所述三相初始调制波对应的调制波集合{vmax-va，vmax-vb，vmax-vc，K*(vmax/2+vmin/2-va)，K*(vmax/2+vmin/2-vb)，K*(vmax/2+vmin/2-vc)，vmin-va，vmin-vb，vmin-vc}；
    所述计算模块，还用于对所述共模调制波集合中每一个调制波变量取绝对值，选择绝对值最小的调制波变量作为共模调制波。
  9. 根据权利要求6至8中任一项所述的脉冲宽度调制系统，其特征在于，
    所述调制模块，具体用于将所述三相初始调制波中的每一相初始调制波分别与所述共模调制波进行一一对应的波形叠加，得到三相输出调制波。
  10. 根据权利要求9所述的脉冲宽度调制方法，其特征在于，
    所述获取模块，用于获取三相电网电压、三相电流及预设载波幅值；
    所述计算模块，还用于对所述三相电网电压进行锁相处理，得到旋转坐标系相位；
    所述计算模块，还用于根据所述旋转坐标系相位，对所述三相电流进行坐标变换，得到旋转坐标系电流；
    所述获取模块，还用于获取所述三相电流坐标变换的预设电流参考值，并计算所述预设电流参考值与所述旋转坐标系电流的差值，得到电流差值；
    所述计算模块，还用于通过比例积分PI调节器对所述电流差值进行处理，得到旋转坐标系调节分量；
    所述计算模块，还用于对所述旋转坐标系调节分量进行反坐标变换，得到三相初始调 制波；
    所述计算模块，还用于获得所述三相初始调制波的调制波幅值，并根据所述调制波幅值和所述预置载波幅值的比值，得到变流器调制度。
  11. 一种控制器，应用于三相变流器，其特征在于，所述控制器包括：
    处理器、存储器及信号接口之间互相连接，存储器中存储有处理器的运行指令，
    所述信号接口，用于获取三相初始调制波及变流器调制度；
    所述处理器，用于根据预设调制参数和所述变流器调制度计算得到共模分量变化速率调节系数，所述预设调制参数包括所述三相变流器的预置最大调制度、预置最小调制度、预置正负钳位状态之间共模量变化最小速率、预置正负钳位状态之间共模量变化最大速率与最小速率的差值及调制曲率参数；
    所述处理器，还用于根据预设的调制波最大限幅值、调制波最小限幅值、所述三相初始调制波和所述共模分量变化速率调节系数，计算得到所述三相初始调制波对应的调制波集合，从所述调制波集合中选取绝对值最小的调制波作为共模调制波；
    所述处理器，还用于将所述三相初始调制波与所述共模调制波进行波形叠加，得到三相输出调制波。
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