WO2018073874A1

WO2018073874A1 - Direct-current power supply device, motor drive device, fan, compressor, and air conditioner

Info

Publication number: WO2018073874A1
Application number: PCT/JP2016/080746
Authority: WO
Inventors: 啓介植村; 成雄梅原; 崇山川
Original assignee: 三菱電機株式会社
Priority date: 2016-10-17
Filing date: 2016-10-17
Publication date: 2018-04-26
Also published as: JPWO2018073874A1; JP6783867B2

Abstract

A direct-current power supply device 100 comprises: a bridge circuit 3 in which a first arm 31 and a second arm 32 are connected in parallel, the first arm 31 having a first switching element 311 and a second switching element 312 connected in series, and the second arm 32 having a third switching element 321 and a fourth switching element 322 connected in series; a reactor 2 provided between an alternating-current power supply 1 and the bridge circuit 3; a current detector 6 for detecting current flowing between the alternating-current power supply 1 and the bridge circuit 3 via the reactor 2; a first voltage detector 5 for detecting an output voltage of the alternating-current power supply 1; a second voltage detector 7 for detecting a voltage applied to a load 50; and a control unit 10 that generates a driving pulse for the first switching element 311, the second switching element 312, the third switching element 321, and the fourth switching element 322 on the basis of the detection values from the first voltage detector 5, the current detector 6, and the second voltage detector 7. When generating the driving pulses, the control unit 10 independently generates the driving pulse for the first arm 31 and the driving pulse for the second arm 32.

Description

DC power supply, motor drive, blower, compressor, and air conditioner

The present invention converts a DC power supplied from an AC power source into a DC power and supplies the load to a load, a motor driving device including the DC power device, a blower and a compressor including the motor driving device, In addition, the present invention relates to an air conditioner including the blower or the compressor.

In order to suppress disturbances due to harmonic components contained in the power supply current, international regulations have been established for electronic devices that generate harmonic currents. In order to clear the regulation, short-circuit the current path that flows out from the AC power source and returns to the AC power source by chopping with alternating current (AC: Alternate Current) or direct current (DC: Direct Current) at the converter, and is included in the power source current Measures to suppress harmonic currents are taken.

As a loss reduction technique by AC chopping technique, many technical studies have been made on a full bridge converter in which a rectifier circuit is configured by a switch element. As an example, in Patent Document 1 below, switching control of a full bridge circuit is realized by a logic circuit, power supply current is controlled by one arm, and synchronous rectification operation is performed by the other arm, thereby reducing loss. Techniques for realizing it are disclosed.

JP 2014-099946 A

However, in Patent Document 1, since the control system is configured using a logic circuit, there is a problem that the board area and the volume increase with the increase in the number of parts, and the apparatus becomes larger.

The present invention has been made in view of the above, and by configuring a control system that does not use a logic circuit, the number of parts can be reduced, and the DC power supply device, motor drive device, blower, An object is to provide a compressor and an air conditioner.

In order to solve the above-described problems and achieve the object, the present invention is a DC power supply device that converts an AC voltage applied from an AC power supply into a DC voltage and applies it to a load. The DC power supply device includes a first arm in which a first switching element and a second switching element are connected in series, and a second arm in which a third switching element and a fourth switching element are connected in series. And having a bridge circuit in which a first arm and a second arm are connected in parallel. Further, the DC power supply device includes a reactor provided between the AC power supply and the bridge circuit, a current detector that detects a current flowing between the AC power supply and the bridge circuit via the reactor, and a first detector that detects an output voltage of the AC power supply. One voltage detector and a second voltage detector for detecting a voltage applied to the load are provided. Further, the DC power supply device includes a first switching element, a second switching element, a third switching element, and a fourth switching element based on detection values of the first voltage detector, the current detector, and the second voltage detector. And an arithmetic unit for generating a driving pulse for the switching element. When generating the drive pulse, the arithmetic unit independently generates the drive pulse for the first arm and the drive pulse for the second arm.

According to the present invention, since a control system that does not use a logic circuit can be configured, the number of parts can be reduced and the apparatus can be miniaturized.

The figure which shows the structural example of the DC power supply device which concerns on Embodiment 1. FIG. Diagram showing current path when charging smoothing capacitor (when power supply voltage is positive) A diagram showing the current path when charging the smoothing capacitor (when the power supply voltage is negative) Diagram showing short circuit path of AC power supply via reactor (when power supply voltage is positive) Diagram showing short-circuit path of AC power supply via reactor (when power supply voltage is negative) The figure which shows the structural example of the control part in the DC power supply device of Embodiment 1. The figure which shows the operation example of a power supply voltage phase calculation part Block diagram showing a configuration example of the first arm pulse generator Diagram for explaining a method of generating a reference PWM signal The figure which serves for explanation of the dead time set in order to prevent the short circuit of the switching element Flow chart showing the operation of the first arm pulse generator Flow chart showing the operation of the second arm pulse generator The figure which shows an example of the operation waveform when a bridge circuit is controlled by the control part The figure which shows the example of a waveform when many ripple components are included in power supply current Diagram used to explain the phenomenon in which the direction of the short-circuit path is reversed in the power supply short-circuit switching pattern Diagram for explaining the phenomenon in which a power supply short circuit and discharge operation occur in a switching pattern that charges a smoothing capacitor The flowchart which shows operation | movement of the 1st arm pulse generation part different from FIG. 11 is a diagram for explaining the difference between the operation according to the control flow of FIG. 11 and the operation according to the control flow of FIG. Diagram showing current-loss characteristics of a typical switching element Schematic sectional view showing the schematic structure of MOSFET Diagram showing the current path when the fourth switching element is not controlled to be ON when charging the smoothing capacitor (when the power supply voltage is positive) Diagram showing the current path when the fourth switching element is controlled to be ON when charging the smoothing capacitor (when the power supply voltage is positive) Diagram showing the current path when the fourth switching element is not controlled to be ON in the power supply short-circuit operation mode (when the power supply voltage is positive) A diagram showing the current path when the fourth switching element is controlled to be ON in the power supply short-circuit operation mode (when the power supply voltage is positive) Diagram showing the current path when the third switching element is not controlled to be ON when charging the smoothing capacitor (when the power supply voltage is negative) Diagram showing the current path when the third switching element is controlled to be ON when charging the smoothing capacitor (when the power supply voltage is negative) Diagram showing the current path when the third switching element is not controlled to be ON in the power supply short-circuit operation mode (when the power supply voltage is negative) Diagram showing the current path when the third switching element is controlled to ON in the power supply short-circuit operation mode (when the power supply voltage is negative) The figure which shows the example of the drive pulse waveform when the cycle of the carrier and the cycle of the power supply voltage are not synchronized The figure which shows the structure which added the external interruption function to the control part of the direct-current power supply device shown in FIG. Diagram for explaining the operation of the power supply voltage polarity detector The figure which shows an operation | movement waveform when it controls by the control method demonstrated using FIG. The figure which shows the structural example of the DC power supply device which concerns on Embodiment 2. FIG. The figure which shows the other structural example different from FIG. 33 of the DC power supply device which concerns on Embodiment 2. FIG. The figure which shows the example which applied the direct-current power supply device shown in Embodiment 1 to the motor drive device The figure which shows the example which applied the motor drive device shown in FIG. 35 to the air conditioner

Referring to the accompanying drawings below, a DC power supply device according to an embodiment of the present invention, a motor drive device including the DC power supply device, a blower and a compressor including the motor drive device, and the blower or the compression An air conditioner equipped with a machine will be described. In addition, this invention is not limited by embodiment shown below.

Embodiment 1 FIG.
FIG. 1 is a diagram illustrating a configuration example of a DC power supply device according to the first embodiment. The DC power supply device 100 according to Embodiment 1 is a power supply device having an AC / DC conversion function of converting an AC voltage applied from the AC power supply 1 into a DC voltage and applying the DC voltage to a load 50. As shown in FIG. 1, a DC power supply device 100 according to Embodiment 1 includes an AC power supply 1, a reactor 2, a bridge circuit 3, a smoothing capacitor 4, a first voltage detector 5 that is power supply voltage detection means, A current detector 6 serving as a detection unit, a second voltage detector 7 serving as a bus voltage detection unit, and a control unit 10 serving as a control unit are provided.

The bridge circuit 3 includes a first arm 31 and a second arm 32. The first arm 31 and the second arm 32 are connected in parallel. In each of the first arm 31 and the second arm 32, two switching elements are connected in series. Specifically, in the first arm 31, the first switching element 311 and the second switching element 312 are connected in series, and in the second arm 32, the third switching element 321 and the fourth switching element are connected. 322 are connected in series.

In FIG. 1, MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistors) are illustrated as the first switching element 311, the second switching element 312, the third switching element 321, and the fourth switching element 322, respectively. However, it is not limited to MOSFET. A MOSFET is a switching element that allows a current to flow bidirectionally between a drain and a source, but allows a current to flow bidirectionally between a first terminal corresponding to the drain and a second terminal corresponding to the source. Any switching element can be used as long as it can be switched. The first switching element 311, the second switching element 312, the third switching element 321, and the fourth switching element 322 may be made of silicon (Si), silicon carbide (SiC), or gallium nitride (GaN). ) Is exemplified, but any material may be used and the material is not limited to these.

The reactor 2 is provided between the AC power source 1 and the bridge circuit 3. One end of the AC power supply 1 is connected to a connection point between the first switching element 311 and the second switching element 312 in the first arm 31 via the reactor 2. The other end of the AC power source 1 is connected to a connection point between the third switching element 321 and the fourth switching element 322.

The output voltage of the bridge circuit 3 is applied across the smoothing capacitor 4. The smoothing capacitor 4 smoothes the output voltage of the bridge circuit 3. The voltage smoothed by the smoothing capacitor 4 is referred to as “bus voltage”.

The control unit 10 generates a driving pulse for driving each switching element of the bridge circuit 3 based on detection values of the first voltage detector 5, the current detector 6, and the second voltage detector 7. The first voltage detector 5 detects the power supply voltage Vs that is the output voltage of the AC power supply 1 and outputs it to the control unit 10. The current detector 6 detects the power source current Is flowing between the AC power source 1 and the bridge circuit 3 via the reactor 2 and outputs the detected power source current Is to the control unit 10. The second voltage detector 7 detects the bus voltage Vdc, which is also the voltage applied to the load 50, and outputs it to the control unit 10.

Next, a basic circuit operation of the DC power supply device 100 according to the first embodiment will be described with reference to the drawings of FIGS. Each of FIG. 2 and FIG. 3 is a diagram showing a current path when charging the smoothing capacitor 4. 2 is when the polarity of the power supply voltage Vs is positive, that is, when the power supply voltage Vs is positive, and FIG. 3 is when the polarity of the power supply voltage Vs is negative, that is, when the power supply voltage Vs is negative. 4 and 5 are diagrams illustrating a short-circuit path of the AC power supply 1 through the reactor 2 when both ends of the AC power supply 1 are short-circuited through the reactor 2 without charging the smoothing capacitor 4. is there. The difference between the two is when the power supply voltage Vs is positive, and FIG. 5 is when the power supply voltage Vs is negative. As shown in FIGS. 2 and 4, when the upper terminal of the AC power supply 1 is a positive potential, the polarity of the power supply voltage Vs is defined as positive, and as shown in FIGS. 3 and 5, the AC power supply When the upper terminal in 1 is a negative potential, the polarity of the power supply voltage Vs is defined as negative.

When the first switching element 311, the second switching element 312, the third switching element 321, and the fourth switching element 322 are not switched, as shown in FIGS. 2 and 3, depending on the polarity of the power supply voltage A current for charging the smoothing capacitor 4 flows. The operation mode at this time is referred to as “normal mode”.

On the other hand, when the second switching element 312 and the fourth switching element 322 are turned on when the power supply voltage Vs is positive, as shown in FIG. 4, the AC power supply 1, the reactor 2, the second switching element The short circuit path can be formed by the path 312, the fourth switching element 322, and the AC power supply 1. Further, when the first switching element 311 and the third switching element 321 are turned on when the power supply voltage polarity is negative, as shown in FIG. 5, the AC power supply 1, the third switching element 321, the first switching element 321 The switching element 311, the reactor 2, and the AC power source 1 can form a short circuit path. Forming a short circuit path is called “power supply short circuit”, and an operation mode for controlling power supply short circuit is called “power supply short circuit mode”.

In the DC power supply device according to the first embodiment, the power supply current Is and the bus voltage Vdc can be controlled by switching control of these operation modes by switching control of the control unit 10.

FIG. 6 is a diagram illustrating a configuration example of the control unit 10 in the DC power supply device of the first embodiment. As shown in FIG. 6, the control unit 10 includes a power supply current command value control unit 21, an on-duty control unit 22, a power supply voltage phase calculation unit 23, a first arm pulse generation unit 24, and a second arm pulse generation unit 25. Have. The control part 10 is comprised using the calculating device which is a calculating means. An example of the arithmetic unit is a microcomputer, but other than this, a processor or processing device called CPU (Central Processing Unit), microprocessor, or DSP (Digital Signal Processor) may be used. .

The power supply current command value control unit 21 calculates a power supply current effective value command value Is_rms * from the bus voltage Vdc that is an output signal of the second voltage detector 7 and a preset bus voltage command value Vdc *. . The calculation of the power supply current effective value command value Is_rms * is realized by proportional-integral control of the difference between the bus voltage Vdc and the bus voltage command value Vdc *. The proportional-integral control is an example, and proportional control or proportional-differential-integral control may be employed instead of proportional-integral control.

The power supply voltage phase calculation unit 23 receives the power supply voltage Vs detected by the first voltage detector 5, generates a power supply voltage phase estimated value θs, and outputs a sine value sin θs of the power supply voltage phase estimated value θs. . The on-duty control unit 22 is calculated from the power supply current effective value command value Is_rms * output from the power supply current command value control unit 21 and the sine value sinθs of the power supply voltage phase estimation value θs output from the power supply voltage phase calculation unit 23. The reference on-duty DTs of the first switching element 311 and the second switching element 312 is calculated from the power supply current instantaneous value command value Is * and the power supply current Is detected by the current detector 6. The calculation of the reference on-duty DTs is performed by proportional-integral control of the difference between the power supply current effective value finger command value Is_rms * and the power supply current Is. Note that the control of the on-duty control unit 22 may employ proportional control or proportional differential integral control instead of proportional integral control.

FIG. 7 is a diagram illustrating an operation example of the power supply voltage phase calculation unit 23. FIG. 7 shows waveforms under ideal conditions that do not take into account the delay due to control or the delay due to detection processing. As shown in FIG. 7, the power supply voltage phase estimation value θs is 360 ° at the point where the power supply voltage Vs switches from negative polarity to positive polarity. The power supply voltage phase calculation unit 23 detects a point where the power supply voltage Vs switches from negative polarity to positive polarity, and resets the power supply voltage phase estimation value θs at this switching point, that is, returns to zero. When using the interrupt function of the microcomputer, a circuit for detecting a zero cross of the power supply voltage Vs may be added to FIG. In any case, any method may be used as long as the phase of the power supply voltage Vs can be detected.

FIG. 8 is a block diagram illustrating a configuration example of the first arm pulse generation unit 24. FIG. 9 is a diagram for explaining a method of generating the reference PWM signal Scom. FIG. 10 is a diagram for explaining a dead time set to prevent a short circuit of the switching element.

The first arm pulse generation unit 24 includes a carrier generation unit 241, a reference PWM signal generation unit 242, a dead time generation unit 243, and a pulse selector unit 244. The carrier generation unit 241 has a function of generating a carrier (carrier frequency) Cs for generating the reference PWM signal Scom.

As shown in FIG. 9, the reference PWM signal generator 242 generates a reference PWM signal Scom by comparing the magnitude relationship between the reference on-duty DTs and the carrier Cs. In the case of FIG. 9, the reference PWM signal Scom is generated by setting Scom as an ON signal when DTs> Cs, and by setting Scom as an OFF signal when DTs <Cs. In the case of FIG. 9, the ON signal has a higher level than the OFF signal and high active, but the ON signal may have a lower level than the OFF signal and low active.

The dead time generation unit 243 functions to generate a dead time td for the reference PWM signal Scom and the inverted signal Scom ′ obtained by inverting the reference PWM signal Scom. Based on the reference PWM signal Scom, the ON state or OFF state of the first switching element 311 and the ON state or OFF state of the second switching element 312 are controlled. In addition, based on the reference PWM signal Scom and the inverted signal Scom ′, the first switching element 311 and the second switching element 312 perform operations that are inverted from each other.

However, the switching element generally has a delay time in the transition from the ON state to the OFF state and in the transition from the OFF state to the ON state. As a result, only using the reference PWM signal Scom and the inverted signal Scom ′ causes a short circuit between the first switching element 311 and the second switching element 312 during the delay time. The dead time td is necessary to prevent such a short-circuit phenomenon. FIG. 10 shows an example of a signal waveform considering the dead time td in the case of high active.

In FIG. 10, S1 is the same signal as the reference PWM signal Scom. S2 is an inverted signal of S1, and is set to be short on both sides of the ON period by the dead time td. The period of the dead time td is a period in which both the first switching element 311 and the second switching element 312 are OFF. Hereinafter, S1 is referred to as a “reference PWM first signal”, and S2 is referred to as a “reference PWM second signal”. A known method other than that shown in FIG. 10 is also known, and there is no problem even if any method is used.

The pulse selector unit 244 functions to select which of the first switching element 311 and the second switching element 312 each of the reference PWM first signal S1 and the reference PWM second signal S2 is transmitted to. FIG. 11 is a flowchart showing the operation of the pulse selector unit 244.

In FIG. 11, when the power supply voltage Vs is positive (step ST11, Yes), the pulse selector unit 244 outputs the reference PWM first signal S1 as a drive pulse to the second switching element 312 (step ST12). The notation “S1 → pulse — 312” means this control. When the power supply voltage Vs is positive (step ST11, Yes), the pulse selector unit 244 outputs the reference PWM second signal S2 as a drive pulse to the first switching element 311 (step ST12). The notation “S2 → pulse — 311” means this control. These controls are because, as described in FIG. 4, when the power supply voltage Vs is positive, a short-circuit path can be formed by the switching operation of the second switching element 312, and the bus voltage Vdc and This is because the power supply current Is is controlled by the operation of the second switching element 312.

If the power supply voltage Vs is negative or zero (No in step ST11), the pulse selector unit 244 outputs the reference PWM first signal S1 as a drive pulse to the first switching element 311 (step ST13). The notation “S1 → pulse_311” means this control. If the power supply voltage Vs is negative or zero (step ST11, No), the reference PWM second signal S2 is output as a drive pulse to the second switching element 312 (step ST13). The notation “S2 → pulse — 312” means this control. These controls are because, as described in FIG. 5, when the power supply voltage Vs is negative, a short-circuit path can be formed by the switching operation of the first switching element 311, and the bus voltage Vdc and This is because the power supply current Is is controlled by operating the first switching element 311.

In the flow of FIG. 11, the output destinations of the reference PWM first signal S1 and the reference PWM second signal S2 are switched depending on the polarity of the power supply voltage Vs, but can be switched based on the power supply current Is. The operation in this case will be described later.

In the determination process of step ST11, the case where the power supply voltage Vs is zero is determined as “No”, but may be determined as “Yes”. That is, the case where the power supply voltage Vs is zero may be determined as “Yes” or “No”.

As described above, the first arm pulse generation unit 24 can generate the drive pulse pulse_311 to the first switching element 311 and the drive pulse pulse_312 to the second switching element 312.

Next, the operation of the second arm pulse generator 25 will be described. FIG. 12 is a flowchart showing the operation of the second arm pulse generator 25.

In FIG. 12, when the power supply voltage Vs is positive (step ST21, Yes), the second arm pulse generating unit 25 uses the drive pulse pulse_321 to the third switching element 321 as an OFF level signal, and the fourth switching element. The drive pulse pulse_322 to 322 is set to an ON level signal (step ST22). When the power supply voltage Vs is negative or zero, the drive pulse pulse_321 of the third switching element 321 is an ON level signal, and the drive pulse pulse_322 to the fourth switching element 322 is an OFF level signal (step ST23). ). These controls are aimed at reducing the loss by synchronous rectification using the characteristics of a MOSFET capable of flowing a current in both directions. Details of these controls will be described later.

In the determination process of step ST21, the case where the power supply voltage Vs is zero is determined as “No”, but may be determined as “Yes”. That is, the case where the power supply voltage Vs is zero may be determined as “Yes” or “No”.

The first switching element 311, the second switching element 312, the third switching element 321, and the fourth switching element 322 are operated by the control according to the flow of FIG. 11 and FIG. 12 described above, and the bus voltage Vdc And the power supply current Is are controlled.

FIG. 13 is a diagram illustrating an example of an operation waveform when the bridge circuit 3 is controlled by the control unit 10. In FIG. 13, the horizontal axis represents time, and the vertical axis represents the power supply voltage Vs, power supply current Is, carrier and reference on-duty, and drive pulses pulse_311, pulse_312, pulse_321, and pulse_322 from the upper stage side. What should be noted in FIG. 13 is that the power supply current Is repeats small fluctuations every carrier cycle, but the peak value or average value of the waveform repeating the small fluctuations is controlled in a sine wave form. . The peak value or average value of the waveform in the power supply current Is being controlled in a sine wave shape means that the harmonic component of the power supply current Is is suppressed.

Next, of the operations of the bridge circuit 3 under the control of the control unit 10, operations that require consideration will be described.

First, as described above, the pulse selector unit 244 of the first arm pulse generation unit 24 outputs the reference PWM first signal S1 to the drive pulse pulse_311 to the first switching element 311 according to the polarity of the power supply voltage Vs. Or the output of the drive pulse pulse_312 to the second switching element 312 and the reference PWM second signal S2 is output to the drive pulse pulse_311 to the first switching element 311 or to the second switching element 312. The output of the drive pulse pulse_312 is switched. Similarly, the second arm pulse generator 25 selects which of the third switching element 321 and the fourth switching element 322 is controlled to be ON according to the polarity of the power supply voltage Vs.

On the other hand, a certain operation in the bridge circuit 3 may include a lot of ripple components in the power supply current Is. FIG. 14 is a diagram illustrating a waveform example when the power supply current Is includes many ripple components. In FIG. 14, the enlarged waveform of the A part in the waveform of the power supply current Is shown on the lower side is shown on the upper side. Looking at the waveform at part A, the current fluctuates little by little. The vibration component seen in the waveform of the A part is a frequency component depending on the switching frequency of the first switching element 311 and the second switching element 312. Next, the operation of the bridge circuit 3 when such a waveform of the power supply current Is occurs will be described with reference to the drawings of FIGS. FIG. 15 is a diagram for explaining a phenomenon in which the direction of the short circuit path is reversed in the switching pattern of the power supply short circuit, and FIG. FIG.

Consider the case where the power supply voltage Vs is positive and the power supply current Is is negative. This is assumed as a phenomenon that occurs when the inductance value of the reactor 2 is very large, or when the power source power factor is extremely deteriorated due to a transient load fluctuation. As described with reference to FIG. 4, a short circuit path can be formed by turning on the second switching element 312. In the case of this switching pattern, as shown in FIG. 15, a short circuit path may be formed even in a state where the polarity of the power supply voltage Vs and the polarity of the power supply current Is are reversed. Further, when the first switching element 311 and the fourth switching element 322 are turned on to form the charging path of the smoothing capacitor 4 as shown in FIG. 2, a short circuit path is formed as shown in FIG. May end up. Further, in the case of FIG. 16, a discharge path from the smoothing capacitor 4 to the reactor 2 may be formed as indicated by a one-dot chain line. As a result, a discharge current flows toward the AC power source 1 and the power source current Is further increases. As a result, the switching pattern of FIG. 15 and the switching pattern of FIG. 16 are repeated, so that the power supply short circuit and the discharge of the charge charge are repeatedly performed, and the power supply current Is rapidly grows. The component will increase. Although the case where the power supply voltage Vs has a positive polarity and the power supply current Is has a negative polarity has been described here, the same phenomenon occurs when the power supply voltage Vs has a negative polarity and the power supply current Is has a positive polarity. May occur.

As a countermeasure against the above phenomenon, a method of switching the operation of the pulse selector unit 244 according to the polarities of both the power supply voltage Vs and the power supply current Is can be considered. FIG. 17 is a flowchart for implementing this technique.

In FIG. 17, when the power supply voltage Vs is positive (step ST31, Yes) and the power supply current Is is positive (step ST32, Yes), the pulse selector unit 244 performs the process α shown as step ST12 in FIG. Is executed (step ST34). When the power supply voltage Vs is positive (step ST31, Yes) and the power supply current Is is negative or zero (step ST32, No), the pulse selector unit 244 performs the process β shown as step ST13 in FIG. Is executed (step ST35). On the other hand, when the power supply voltage Vs is negative or zero (step ST31, No) and the power supply current Is is negative (step ST33, Yes), the pulse selector unit 244 executes the process β (step ST36). When the power supply voltage Vs is negative or zero (step ST31, No) and the power supply current Is is positive or zero (step ST33, No), the pulse selector unit 244 executes the process α (step ST37). ).

In summary, when the polarity of the power supply voltage Vs and the power supply current Is are both positive, the first process α is executed, and when the polarity of the power supply voltage Vs and the power supply current Is is both negative. Executes the second process, process β. When the polarities of the power supply voltage Vs and the power supply current Is are different, the process α is executed if the power supply current Is is positive, and the process β is executed if the power supply current Is is negative. Thereby, it can prevent that the path | route used as a power supply short circuit continues.

In the above processing, when the polarities of the power supply voltage Vs and the power supply current Is are different, the generation of drive pulses to the first arm 31 and the second arm 32 is stopped without executing the processing α and the processing β. You may make it do. In the case of this control, control according to the polarity of the power supply current Is cannot be performed, but since the generation of the drive pulse to the first arm 31 and the second arm 32 is stopped, the phenomenon that the power supply current Is rapidly grows. This can be suppressed and can contribute to suppression of harmonic components of the power supply current Is.

FIG. 18 is a diagram for explaining the difference between the operation according to the control flow of FIG. 11 and the operation according to the control flow of FIG. In FIG. 18, the waveform of the power supply voltage Vs is schematically shown in the upper stage portion, and the waveform of the power supply current Is is schematically shown in the middle stage portion. The lower part shows the difference between the operation according to the control flow of FIG. 11 and the operation according to the control flow of FIG. Here, the horizontal axis of FIG. 18 represents time, but the waveform of the power supply voltage Vs is approximated as linear in order to represent a waveform in a minute time. Further, a phase difference of Δt is provided between the power supply voltage Vs and the center line in the waveform of the power supply current Is. The phase difference Δt is a relationship in which the power supply current Is is delayed with respect to the power supply voltage Vs.

In the case of the operation according to the control flow in FIG. 11, since the determination is made based only on the polarity of the power supply voltage Vs, the process β is continued after the polarity of the power supply voltage Vs turns positive. In the portion of the process β, the portion where the power source current Is is negative has a switching pattern assuming that the power source current Is is positive, and therefore the power source short-circuit mode continues.

On the other hand, in the case of the operation according to the control flow of FIG. 17, the process α or the process β is appropriately selected according to not only the polarity of the power supply voltage Vs but also the polarity of the power supply current Is. That is, in the case of the operation according to the control flow of FIG. 17, the power supply short-circuit mode does not continue and distortion of the power supply current Is can be suppressed.

As described so far, in configuring the control, it is necessary to perform detailed control on the switching element driving method. When these controls are configured by an analog circuit or a logic circuit, the control method becomes large because the control method is complicated. In the case of an analog circuit or a logic circuit, there is a risk of malfunction due to the influence of noise from other hardware.

On the other hand, when the control system is implemented by software using an arithmetic unit, since only the arithmetic unit is used, the apparatus does not increase in size. In addition, when the control logic described so far is implemented in software, malfunctions due to the effects of noise are very rare, and a circuit that prevents malfunctions can be omitted, so that the system configuration can be reduced in size. .

So far, the pulse selector unit 244 in the control unit 10 has been described. Next, the details of the synchronous rectification operation in the second arm pulse generation unit 25 will be described. FIG. 19 is a diagram schematically showing current-loss characteristics in a general switching element. FIG. 19 shows the loss characteristic of the parasitic diode and the loss characteristic when the switching element is on. Here, the current value when the loss value in the loss characteristic of the parasitic diode and the loss value in the loss characteristic of the switching element are reversed is defined as the first current value. When the current value is smaller than the first current value, the loss characteristic of the switching element is smaller, and this region is referred to as a “low current region A”. Further, when the current value is larger than the first current value, the loss characteristic of the parasitic diode is smaller, and this region is referred to as a “high current region B”. Note that the first current value is held inside the calculator, or held in a memory that can be read by the calculator.

Generally, as shown in FIG. 19, the parasitic diode of the switching element has a large loss in the low current region A, and it is known that the loss when the switching element is on is smaller. Therefore, in the case of a MOSFET that is one of the switching elements, a synchronous rectification technique using switching characteristics may be used. The switching characteristic referred to here is a characteristic that is turned on in both the direction from the drain to the source and the direction from the source to the drain when an ON command is given to the gate of the MOSFET, that is, the switching element. There is a characteristic that allows current to flow in both directions. This characteristic is different from that in which a switching element such as a bipolar transistor or IGBT can conduct current only in one direction. In the case of utilizing this characteristic, in the low current region A shown in FIG. 19, the parasitic diode is not used, and the current is made to flow through the switching element, so that higher efficiency can be achieved than using the parasitic diode.

Next, the relationship between the operation of the switching element and the synchronous rectification in the first embodiment will be described. Prior to this description, the schematic structure of the MOSFET will be described with reference to FIG. FIG. 20 is a schematic cross-sectional view showing a schematic structure of a MOSFET. FIG. 20 illustrates an n-type MOSFET.

In the case of an n-type MOSFET, a p-type semiconductor substrate is used as shown in FIG. A source electrode (S), a drain electrode (D), and a gate electrode (G) are formed on a p-type semiconductor substrate. A high-concentration impurity is ion-implanted in a portion in contact with the source electrode (S) and the drain electrode (D) to form an n-type region. In the p-type semiconductor substrate, an oxide insulating film is formed between a portion where the n-type region is not formed and the gate electrode (G). That is, the oxide insulating film is interposed between the gate electrode (G) and the p-type region in the semiconductor substrate.

When a positive voltage is applied to the gate electrode (G), electrons are attracted to the interface between the p-type region and the oxide insulating film in the semiconductor substrate, and are negatively charged. Where the electrons gather, the electron density is higher than that of the holes and becomes n-type. This n-type portion becomes a current path and is called a channel (in the example of FIG. 20, an n-type channel).

When performing synchronous rectification, the MOSFET is controlled to be ON, so that a larger current flows in the channel side than in the parasitic diode side.

Next, a specific synchronous rectification operation will be described. First, a case where the polarity of the power supply voltage Vs is positive will be described. FIG. 21 is a diagram illustrating a current path when the fourth switching element 322 is not controlled to be turned ON when the smoothing capacitor 4 is charged. FIG. 22 is a diagram illustrating the fourth switching element when the smoothing capacitor 4 is charged. It is a figure which shows the electric current path | route when 322 is controlled to ON.

When the fourth switching element 322 is in the OFF state as shown in FIG. 21, the charging current flows through the parasitic diode side of the fourth switching element 322. On the other hand, when the fourth switching element 322 is controlled to be ON, the charging current flows through the channel side of the fourth switching element 322 as shown in FIG. Except for initial charging, the charging current does not become very large. For this reason, the loss in FIG. 22 in which the fourth switching element 322 is controlled to be ON and current is supplied to the channel side is lower than that in FIG. 21 in which current is supplied to the parasitic diode.

FIG. 23 is a diagram illustrating a current path when the fourth switching element 322 is not controlled to be ON in the power supply short-circuit operation mode, and FIG. 24 is a diagram illustrating the fourth switching element in the power supply short-circuit operation mode. It is a figure which shows the electric current path | route when 322 is controlled to ON. 23 and 24, the polarity of the power supply voltage is positive.

23, when the fourth switching element 322 is in the OFF state, the short-circuit current flows through the parasitic diode side of the fourth switching element 322. On the other hand, when the fourth switching element 322 is controlled to be ON as shown in FIG. 24, the short-circuit current flows through the channel side of the fourth switching element 322 as in FIG. Since there is the reactor 2 in the short circuit path, the magnitude of the short circuit current is suppressed by the reactor 2. Therefore, the loss in FIG. 24 in which the fourth switching element 322 is controlled to be ON and current is supplied to the channel side is lower than that in FIG. 23 in which current is supplied to the parasitic diode.

Next, the case where the polarity of the power supply voltage Vs is negative will be described. FIG. 25 is a diagram illustrating a current path when the third switching element 321 is not controlled to be ON when charging the smoothing capacitor 4. FIG. 26 is a diagram illustrating the third switching element when charging the smoothing capacitor 4. It is a figure which shows the electric current path when 321 is controlled to ON.

When the third switching element 321 is in the OFF state as shown in FIG. 25, the charging current flows through the parasitic diode side of the third switching element 321. On the other hand, when the third switching element 321 is controlled to be ON, the charging current flows through the channel side of the third switching element 321 as shown in FIG. However, the charging current does not become very large except for the initial charging. For this reason, the loss in FIG. 26 in which the third switching element 321 is controlled to be ON and current is supplied to the channel side is lower than that in FIG. 25 in which current is supplied to the parasitic diode.

FIG. 27 is a diagram illustrating a current path when the third switching element 321 is not controlled to be ON in the power supply short-circuit operation mode, and FIG. 28 is a diagram illustrating the third switching element in the power-supply short-circuit operation mode. It is a figure which shows the electric current path when 321 is controlled to ON. 27 and 28, the polarity of the power supply voltage is negative.

When the third switching element 321 is in the OFF state as shown in FIG. 27, the short-circuit current flows through the parasitic diode side of the third switching element 321. On the other hand, when the third switching element 321 is controlled to be ON as shown in FIG. 28, the short-circuit current flows on the channel side of the third switching element 321 as in FIG. Since there is the reactor 2 in the short circuit path, the magnitude of the short circuit current is suppressed by the reactor 2. Therefore, FIG. 28 in which the third switching element 321 is controlled to be ON and current is supplied to the channel side has a lower loss than that in FIG. 27 in which current is supplied to the parasitic diode.

The loss reduction can be achieved by the synchronous rectification technology as described above. This synchronous rectification technique can be realized by the function of the control unit 10 described so far.

Note that, as shown in FIG. 19, in the case of the high current region B, the loss of the parasitic diode is smaller than the loss characteristic when the switching element is ON. Therefore, in the case of the high current region B, the loss is reduced when the switching operation of the second arm 32 is stopped. For this reason, it is a preferred embodiment to switch the control method of the switching control according to the magnitude of the current flowing through the third switching element 321 and the fourth switching element 322 of the second arm 32. Note that the current value flowing through the third switching element 321 and the fourth switching element 322 can be calculated by the calculator of the control unit 10 if the detection value of the current detector 6 is used.

Even in the case where such processing is realized, it is possible to consolidate functions and to reduce the size of the entire apparatus by implementing software implementation using an arithmetic unit in the control unit 10.

Also, when the control unit 10 is configured using hardware, there is a risk of malfunction due to the influence of noise by other hardware. For example, consider a case where a switching element to be turned on during synchronous rectification is turned off due to the influence of noise. In this case, as described above, the loss in the switching element increases, and the amount of heat generated in the switching element increases. Further, when the ON state and OFF state of the switching element are continuously changed, not only the conduction loss of the switching element but also the switching loss is generated, so that the amount of heat generated in the switching element further increases. The risk of destruction of the switching element increases due to an increase in the amount of heat generated by the switching element. On the other hand, when the control system is implemented by software using an arithmetic unit, malfunctions due to noise are very rare. For this reason, configuring the control system using an arithmetic unit has an effect of reducing the risk of destruction of the switching element.

As can be understood from the above description, the first arm 31 contributes to boosting the charging voltage of the smoothing capacitor 4 and suppressing harmonic components of the power supply current Is. Further, the second arm 32 contributes to a reduction in loss due to synchronous rectification. The first arm 31 performs switching control with a period depending on the carrier frequency, whereas the second arm 32 depends on the period of the power supply voltage Vs or the power supply current Is, so that the switching control is performed around 50 to 60 Hz. Will be. Therefore, when the control is realized by the arithmetic unit, it is desirable that the control for the first arm 31 and the control for the second arm 32 use functions of independent arithmetic units, respectively.

FIG. 13 described above shows the waveforms of the power supply voltage Vs, the power supply current Is, the carrier, and the drive pulse to each switching element. Referring to FIG. 13, it can be seen that the third switching element 321 and the fourth switching element 322 are driven at a lower frequency than the first switching element 311 and the second switching element 312. Therefore, when the control unit 10 is configured using an arithmetic unit, the control of the first switching element 311 and the second switching element 312 is implemented by a control system using a timer, and the third switching element 321 and the second switching element 312 are controlled. The control of the fourth switching element 322 is implemented using a function different from the timer, such as a general-purpose output port or an external interrupt function. As a result, it is possible to use resources with few computing units, and it is possible to realize with a cheaper computing unit.

Also, the carrier cycle that determines the switching speed of the first arm 31 and the cycles of the power supply voltage Vs and the power supply current Is are not necessarily synchronized. FIG. 29 is a diagram illustrating an example of a drive pulse waveform when the carrier cycle and the cycle of the power supply voltage Vs are not synchronized. In FIG. 29, similarly to FIG. 18, the power supply voltage Vs is represented by a linear approximation waveform in order to represent a waveform in a minute time. According to the example of FIG. 29, the drive pulse pulse_311 to the first switching element 311 and the drive pulse pulse_312 to the second switching element 312 are generated based on the carrier, and also to the third switching element 321. The drive pulse pulse_321 and the drive pulse pulse_322 to the fourth switching element 322 are also generated in synchronization with the carrier.

From the viewpoint of general PWM control using a logic circuit, it is a normal idea that the update of the driving pulse of the switching element can be changed only at the peak of the carrier or at the valley of the carrier. In fact, in the example of FIG. 29, the duty command value is updated at the peaks and valleys of the carrier. Therefore, in the section of Δt in FIG. 29, the actual power supply voltage is positive, whereas the control system performs control when the power supply voltage Vs is negative, and the controllability deteriorates. End up. This is because the generation timing of the drive pulse for each switching element and the generation timing of the drive pulse for the power supply voltage Vs are determined by the carrier resolution, and a delay time occurs between the zero cross point of the power supply voltage Vs. It is caused by doing. Therefore, when an arithmetic unit is used, generation of drive pulses to the first switching element 311 and the second switching element 312 and generation of drive pulses to the third switching element 321 and the fourth switching element 322 are performed. It can be said that it is preferable to use an independent calculation function for generation. Thereby, when the arithmetic unit is used, the update of the duty command value of the first arm 31 may be performed at a periodic timing depending on the carrier frequency (switching frequency) other than the carrier peak or the carrier valley. Can change.

FIG. 30 is a diagram showing a configuration in which an external interrupt function is added to the controller 10 of the DC power supply device shown in FIG. In other words, independent generation functions for generating drive pulses to the first switching element 311 and the second switching element 312 and generating drive pulses to the third switching element 321 and the fourth switching element 322 are provided. It is an example configured by using.

30, a power supply voltage polarity detection unit 20 is provided between the first voltage detector 5 that detects the power supply voltage Vs and the control unit 10. The power supply voltage polarity detection unit 20 detects the polarity of the power supply voltage Vs, generates a power supply voltage polarity signal pulse_Vs representing the detected polarity, and outputs it to the external interrupt port 10 a of the control unit 10.

FIG. 31 is a diagram for explaining the operation of the power supply voltage polarity detection unit 20. In FIG. 31, the upper stage shows the waveform of the power supply voltage Vs, the middle stage shows the power supply voltage polarity signal pulse_Vs input to the external interrupt port 10a of the control unit 10, and the lower stage shows the inside of the control unit 10. The trigger signal of the control calculation used is shown. When the polarity of the power supply voltage Vs is positive, the power supply voltage polarity signal pulse_Vs is at a high level, and when the polarity of the power supply voltage Vs is negative, the power supply voltage polarity signal pulse_Vs is at a low level. However, these signal polarities are merely examples, and may be reversed. The power supply voltage polarity signal pulse_Vs is an external interrupt signal for the control unit 10 and may be in any signal format as long as it represents a change in the polarity of the power supply voltage Vs. The control unit 10 generates a trigger signal in response to the edge of the power supply voltage polarity signal pulse_Vs. When the trigger signal is generated, the control unit 10 generates drive pulses for the third switching element 321 and the fourth switching element 322 of the second arm 32.

FIG. 32 is a diagram showing operation waveforms when the control method described with reference to FIG. 31 is used. In FIG. 32, the upper stage shows the waveform of the power supply voltage Vs, the middle stage shows the waveform of the carrier, the lower stage shows the drive pulses pulse_311, pulse_312, pulse_321, pulse_322 to the four switching elements, and the power supply voltage. The polarity signal pulse_Vs is shown. In FIG. 32 as well, as in FIGS. 18 and 29, the power supply voltage Vs is represented by a linear approximation waveform in order to represent a waveform in a minute time.

32, the drive pulse pulse_311 to the first switching element 311 and the drive pulse pulse_312 to the second switching element 312 are generated based on the carrier. On the other hand, the drive pulse pulse_321 to the third switching element 321 and the drive pulse pulse_322 to the fourth switching element 322 are not synchronized with the carrier but are generated based on the power supply voltage polarity signal pulse_Vs. Accordingly, the driving states of the third switching element 321 and the fourth switching element 322 depend on frequencies other than the carrier (switching) frequency, and the drive pulse pulse_311 to the first switching element 311 and the second switching element It does not depend on the control state of 312. In the case of FIG. 29, the switching of the drive pulse pulse_322 to the third switching element 321 and the fourth switching element 322 is delayed by Δt with respect to the zero cross of the power supply voltage Vs. In the case of FIG. There is no delay. The reason why the delay does not occur is that the control for the third switching element 321 and the fourth switching element 322 does not depend on the control for the first switching element 311 and the second switching element 312.

In FIG. 32, the external interrupt according to the power supply voltage polarity signal pulse_Vs is the highest priority processing in the arithmetic unit. However, when emphasizing the power harmonic suppression control, the first switching of the first arm 31 is performed. A carrier wave interrupt for controlling the element 311 and the second switching element 312 can be set as the highest priority process. In any case, flexible handling can be performed by using an arithmetic unit.

As described above, in the first embodiment, the generation of drive pulses to the first switching element 311 and the second switching element 312 and the drive pulse to the third switching element 321 and the fourth switching element 322 are performed. Since the independent calculation function is used for each of the generations, it is possible to create an effect of improving controllability and an effect of reducing loss. Note that the control configuration described so far is merely an example, and any method may be used as long as there is independence of a calculation function with respect to generation of a drive pulse. Further, the update of the duty command value in the first arm is not a problem even when thinning is performed once every two carrier cycles.

Embodiment 2. FIG.
FIG. 33 is a diagram illustrating a configuration example of a DC power supply device according to the second embodiment. In the first and second embodiments, the first arm 31 configuring the bridge circuit 3 is configured by using one element pair including a pair of switching elements of the upper and lower arms, but the number of element pairs is plural. Also good. In FIG. 33, the first arm 31 is configured using two element pairs. Specifically, the first arm 31 includes a first switching element 311 and a second switching element 312 connected in series, as well as a fifth switching element 313 and a sixth switching element 314 connected in series. Is provided. Further, the first switching element 311 and the fifth switching element 313 are connected in parallel, and the second switching element 312 and the sixth switching element 314 are connected in parallel.

When driving the first arm 31 in which two pairs of switching elements are connected in parallel, the two switching elements constituting the upper arm are simultaneously driven, and the two switching elements constituting the lower arm are also driven. Are driven simultaneously. 33, the first switching element 311 and the fifth switching element 313 are simultaneously driven, and the second switching element 312 and the sixth switching element 314 are simultaneously driven. In addition, simultaneously driving each of the two switching elements connected in parallel is called “parallel driving”.

By driving in parallel two switching elements connected in parallel, the current flowing through the two switching elements is half that of one. As is clear from the characteristics of FIG. 19, if the current is reduced, the loss of the switching element is reduced. Therefore, the loss generated in the first arm 31 is reduced, and the first arm 31 and the second arm in the bridge circuit 3 are reduced. The bias of the loss between the first arm 31 and the second arm 32 can be reduced. As a result, the bias of the heat generation between the first arm 31 and the second arm 32 can be reduced.

In addition, in FIG. 33, although the structure which connects two element pairs in parallel is illustrated, it is not limited to two, You may comprise by connecting each of three or more element pairs in parallel. Good. That is, each of the fifth switching element 313 and the sixth switching element 314 may include a plurality of switching elements connected in parallel. In the case of using n (n is a natural number) element pairs, the current flowing through one element pair is 1 / n, so that the loss in the first arm 31 can be further reduced. Note that it is not necessary to completely suppress the bias of loss for each arm, and the number of element pairs connected in parallel may be selected within a range in which the bias of loss is allowed.

In the above-described example, the two switching elements connected in parallel in the first arm 31 are driven at the same time. However, the phase of the two switching elements connected in parallel is controlled by shifting by 180 °. Interleave control may be performed. FIG. 34 is a diagram illustrating another configuration example of the DC power supply device according to the second embodiment.

34, a first reactor 2a and a second reactor 2b are provided between the AC power source 1 and the first arm 31. One end of the first reactor 2a is connected to one output end of the AC power source 1, and the other end is connected between a connection point between the first switching element 311 and the second switching element 312. One end of the second reactor 2 b is connected to an output end on one side of the AC power supply 1, and the other end is connected to a connection point between the first switching element 311 and the second switching element 312.

In the DC power supply device 100 configured as shown in FIG. 34, the phase when turning on the first switching element 311 and the fifth switching element 313 connected in parallel is shifted by 180 °, and the parallel connection is performed. The second switching element 312 and the sixth switching element 314 that have been turned on can be driven in an interleaved manner by controlling the phase when the second switching element 312 and the sixth switching element 314 are turned on by shifting 180 degrees. By driving the first arm 31 in an interleaved manner, it is easy to increase the frequency, and the reactor can be downsized and the reactor loss can be reduced.

Embodiment 3 FIG.
The DC power supply apparatus described in Embodiment 1 and Embodiment 2 can be used as an apparatus that converts DC power into AC power and supplies a DC voltage to an inverter that drives a motor.

FIG. 35 is a diagram showing an example in which the DC power supply device shown in the first embodiment is applied to a motor drive device. In motor drive device 101 according to Embodiment 3 shown in FIG. 35, inverter 50a is connected as a load to the output side of DC power supply device 100 according to Embodiment 1. A motor 50b is connected to the output side of the inverter 50a. The inverter 50a drives the motor 50b by converting the output voltage of the DC power supply device 100 into an AC voltage and applying it to the motor 50b. 35 shows an example in which the DC power supply device 100 according to the first embodiment is applied as the DC power supply device 100 constituting the motor drive device 101, but the DC power supply device 100 according to the second embodiment is applied. May be.

35 can be applied to products such as a blower, a compressor, and an air conditioner.

FIG. 36 is a diagram showing an example in which the motor drive device 101 shown in FIG. 35 is applied to an air conditioner. A motor 50 b is connected to the output side of the motor drive device 101, and the motor 50 b is connected to the compression element 54. The compressor 55 includes a motor 50 b and a compression element 54. The refrigeration cycle unit 56 is configured to include a four-way valve 56a, an indoor heat exchanger 56b, an expansion valve 56c, and an outdoor heat exchanger 56d. The flow path of the refrigerant circulating inside the air conditioner passes from the compression element 54 through the four-way valve 56a, the indoor heat exchanger 56b, the expansion valve 56c, the outdoor heat exchanger 56d, and again through the four-way valve 56a. , In a manner returning to the compression element 54. The motor driving device 101 receives supply of AC voltage from the AC power source 1 and rotates the motor 50b. The compression element 54 performs a refrigerant compression operation by rotating the motor 50 b, and circulates the refrigerant inside the refrigeration cycle unit 56.

Note that the configurations shown in the above embodiments are examples of the contents of the present invention, and can be combined with other known techniques, and can be combined without departing from the gist of the present invention. It is also possible to omit or change a part of.

1 AC power supply, 2 reactor, 2a 1st reactor, 2b 2nd reactor, 3 bridge circuit, 4 smoothing capacitor, 5th voltage detector, 6 current detector, 7 second voltage detector, 10 control Unit, 10a external interrupt port, 20 power supply voltage polarity detection unit, 21 power supply current command value control unit, 22 on-duty control unit, 23 power supply voltage phase calculation unit, 24 first arm pulse generation unit, 25 second arm pulse generation unit , 31 1st arm, 32 2nd arm, 50 load, 50a inverter, 50b motor, 54 compression element, 55 compressor, 56 refrigeration cycle section, 56a four-way valve, 56b indoor heat exchanger, 56c expansion valve, 56d Outdoor heat exchanger, 100 DC power supply, 101 motor drive, 241 carrier life Part, 242 reference PWM signal generation part, 243 dead time generation part, 244 pulse selector part, 311 first switching element, 312 second switching element, 313 fifth switching element, 314 sixth switching element, 321st Three switching elements, 322 Fourth switching element.

Claims

A DC power supply device that converts an AC voltage applied from an AC power source into a DC voltage and applies it to a load,
A first arm in which a first switching element and a second switching element are connected in series; and a second arm in which a third switching element and a fourth switching element are connected in series; A bridge circuit in which the first arm and the second arm are connected in parallel;
A reactor provided between the AC power supply and the bridge circuit;
A current detector for detecting a current flowing between the AC power source and the bridge circuit via the reactor;
A first voltage detector for detecting an output voltage of the AC power supply;
A second voltage detector for detecting a voltage applied to the load;
Based on detection values of the first voltage detector, the current detector, and the second voltage detector, the first switching element, the second switching, the third switching element, and the first An arithmetic unit that independently generates a driving pulse for the first arm and a driving pulse for the second arm when generating the driving pulse for the four switching elements;
DC power supply with
The computing unit outputs a driving pulse whose duty is changed at a cycle depending on a switching frequency to the first arm, and outputs a driving pulse having a cycle depending on a frequency other than the switching frequency to the second arm. Item 4. The DC power supply device according to Item 1.
The said arithmetic unit outputs the drive pulse which controls the drive state of said 1st arm and said 2nd arm according to the polarity of the voltage which said 1st voltage detector detected. DC power supply.
The computing unit controls the driving state of the first arm and the second arm according to the polarity of the voltage detected by the first voltage detector and the polarity of the current detected by the current detector. The DC power supply device according to any one of claims 1 to 3, wherein a driving pulse is output.
When the arithmetic unit generates drive pulses to the first arm and the second arm,
If both the polarity of the voltage and the polarity of the current are positive, the first process is executed,
When both the polarity of the voltage and the polarity of the current are negative, a second process is executed,
5. If the polarity of the voltage and the polarity of the current are different, the first process is executed if the current is positive, and the second process is executed if the current is negative. The direct current power supply device described in 1.
When the arithmetic unit generates drive pulses to the first arm and the second arm,
If both the polarity of the voltage and the polarity of the current are positive, the first process is executed,
When both the polarity of the voltage and the polarity of the current are negative, a second process is executed,
The DC power supply device according to claim 4, wherein when the polarity of the voltage is different from the polarity of the current, the execution of the first process and the second process is stopped.
The computing unit stops the switching operation of the third switching element according to the magnitude of the current flowing through the third switching element, and according to the magnitude of the current flowing through the fourth switching element, The DC power supply device according to any one of claims 4 to 6, wherein the switching operation of the fourth switching element is stopped.
2. The computing unit outputs a driving pulse for controlling the first arm with a period depending on a carrier frequency, and outputs a driving pulse for controlling the second arm with a period of a power supply voltage or a power supply current. The direct current power supply device according to any one of 1 to 7.
In the computing unit, the control of the first arm is implemented by a control system using a timer, and the control of the second arm is implemented by using a general-purpose output port or an external interrupt function. The DC power supply device according to any one of the above.
The first arm further includes a fifth switching element and a sixth switching element connected in series,
The first switching element and the fifth switching element are connected in parallel, and the second switching element and the sixth switching element are connected in parallel. The direct-current power supply device according to item 1.
The DC power supply device according to claim 10, wherein each of the fifth switching element and the sixth switching element includes a plurality of switching elements connected in parallel.
The first arm includes a fifth switching element and a sixth switching element connected in series,
In the first arm, the element pair by the fifth switching element and the sixth switching element, and the element pair by the first switching element and the second switching element are connected in parallel,
A first reactor having one end connected to an output end on one side of the AC power supply and the other end connected between a connection point of the first switching element and the second switching element;
A second reactor having one end connected to an output end on one side of the AC power supply and the other end connected to a connection point between the fifth switching element and the sixth switching element;
The DC power supply device according to any one of claims 1 to 8, wherein the second switching element and the sixth switching element are interleaved.
A motor driving device for driving a motor,
The DC power supply device according to any one of claims 1 to 12,
An inverter that converts the output voltage of the DC power supply device into an AC voltage and applies it to the motor;
A motor drive device comprising:
A blower comprising the motor drive device according to claim 13.
A compressor provided with the motor drive device according to claim 13.
An air conditioner provided with the motor drive device according to claim 13.