JP2014099946A - Step-up pfc control unit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To largely reduce the power consumption of a step-up PFC control unit.SOLUTION: A step-up PFC control unit 100 includes an AC/DC converter 30 that rectifies and increases a voltage of a single-phase AC current. The AC/DC converter 30 includes: plural AC switches S1-S4; a step-up coil 4; a smoothing capacitor 7; and a drive logic circuit 53 that controls ON/OFF state of the AC switches S1-S4 to alternately perform magnetic energy storage to the step-up coil 4 and charge storage to the smoothing capacitor 7. When the gate-source voltage is higher than a threshold value voltage, the AC switches S1-S4 allow the current to flow from the drain to the source and from the source to the drain according to the polarity of the drain-source voltage; when the gate-source voltage is smaller than the threshold value voltage, the AC switches S1-S4 shut off the current from flowing from the drain to the source; and when the gate voltage with respect to the drain voltage is larger than the threshold value voltage, the AC switches S1-S4 allow current to flow from the source to the drain.

Description

  The present invention relates to a high-input power factor and high-efficiency step-up PFC (Power Factor Correction) controller that rectifies an input AC voltage and outputs a DC voltage larger than the peak value of the AC voltage.

  FIG. 16 is a circuit block diagram of a step-up PFC control device without a rectifier diode bridge described in Patent Document 1. The step-up PFC control device 500 without a rectifier diode bridge described in FIG. 16 is intended to reduce the size and cost of the high-frequency block filter 502.

  In order to realize this object, two sawtooth wave oscillators 514 and 515 whose phases are different from each other by 180 degrees are used, and the phase of the switching operation for boosting is shifted twice by 180 degrees. That is, the first switching operation by the switching elements T1 and T2 and the second switching operation by the switching elements T3 and T4. As a result, the ripple of the current flowing through the high-frequency blocking filter 502 has a second harmonic compared with the switching frequency. As a result, the high-frequency blocking filter 502 can be reduced in size and cost.

JP 2008-125310 A

  However, the above-described conventional step-up PFC control device 500 generates a loss due to a voltage drop due to the forward voltage VF of the diode D1 or D2 and the forward voltage VFP of the parasitic diode of the switching elements T1 to T4 made of silicon-based semiconductor. Have the problem of doing.

  Furthermore, there is a problem that the switching loss during the turn-on operation of the switching elements T1 to T4 is large due to the recovery current of the parasitic diodes of the switching elements T1 to T4 made of silicon-based semiconductor. Further, the switching loss increases in proportion to the switching frequency of the PFC control device. Therefore, increasing the switching frequency in order to reduce the size of the magnetic induction means 504 causes an increase in the switching loss.

  In view of the above problems, the present invention provides a step-up PFC control device that realizes low power consumption and miniaturization by eliminating the parasitic diodes of the switching elements and the diodes D1 and D2 that occupy most of the power consumption in the PFC control device. The purpose is to provide.

  In order to solve the above problems, a boost PFC control device according to an aspect of the present invention is a boost PFC control device including an AC / DC converter that rectifies and boosts an input single-phase AC voltage. The AC / DC converter unit includes a plurality of AC switch units that are bridge-connected, a boosting coil that stores magnetic energy corresponding to a bidirectional current that flows in response to the application of the single-phase AC voltage, and the boosting unit A smoothing capacitor that stores a charge corresponding to the magnetic energy stored in the coil, and switching a current path of the bidirectional current by controlling an on / off state of the plurality of AC switch units, to the boosting coil A drive logic circuit for alternately storing the magnetic energy and storing the charge in the smoothing capacitor, and the plurality of AC switches. Each of the units has a gate terminal, a drain terminal, and a source terminal. (1) When the gate-source voltage, which is the gate voltage with respect to the source voltage, is higher than the threshold voltage, depending on the polarity of the drain-source voltage Current flows from the drain terminal to the source terminal or from the source terminal to the drain terminal, and (2) when the gate-source voltage is equal to or lower than the threshold voltage, the drain terminal is changed to the source terminal. It is configured by a switching element that cuts off a flowing current and flows a current from the source terminal to the drain terminal when a gate voltage with respect to a drain voltage becomes equal to or higher than the threshold voltage.

  According to the above configuration, no diode is required in the AC / DC converter unit, and there is no parasitic diode in the switching element configuring the AC switch unit, so that power consumption due to a voltage drop in the forward voltage of the diode can be eliminated. . In addition, since there is no switching loss due to the recovery current of the parasitic diode that has existed in the past, a step-up type PFC control device with significantly reduced power consumption can be realized.

  In another aspect of the present invention, the plurality of AC switch units include first to fourth switching elements, and the first electrode of the boosting coil is applied with the single-phase AC voltage. The drain terminal of the first switching element and the drain terminal of the third switching element are connected to the first electrode of the smoothing capacitor, and are connected to the first AC terminal. The source terminal and the source terminal of the fourth switching element are connected to the second electrode of the smoothing capacitor, and the source terminal of the first switching element and the drain terminal of the second switching element are: The single-phase AC voltage is connected to the second electrode of the boosting coil, and the source terminal of the third switching element and the drain terminal of the fourth switching element are Connected to the second AC terminal to be applied, the gate terminals of the first to fourth switching elements are connected to each pre-drive circuit for level-shifting the control signal output from the drive logic circuit, The drive logic circuit may be controlled such that the first switching element and the second switching element or the third switching element and the fourth switching element are not simultaneously turned on. Good.

  According to this, the AC / DC converter unit is configured by a basic bridge circuit in which four bidirectional switching elements are bridge-connected. Therefore, the AC / DC converter unit can control and boost the bidirectional current path simply and with high efficiency.

  According to another aspect of the present invention, the drive logic circuit further includes the drive logic circuit when the current flowing between the AC power source that outputs the single-phase AC voltage and the AC / DC converter unit is substantially zero. It is preferable to control the gate signal of the switching element so that the gate-source voltage of the switching element constituting at least one AC switch part among the plurality of AC switch parts is equal to or lower than the threshold voltage.

  As a result, when the current flowing through the booster coil is substantially zero, it is possible to prevent the current from flowing backward from the output side of the booster type PFC control device to the input AC power supply even when the magnetic energy of the booster coil is lost. Become.

  In another aspect of the present invention, at least two AC switch units among the plurality of AC switch units may be configured by the two switching elements in which source terminals or drain terminals are connected in series. .

  According to this, similarly to the above-described step-up PFC control device, it is possible to reduce power consumption while maintaining the rectifying step-up operation of the conventional step-up PFC control device. Furthermore, a step-up PFC control device having high input voltage resistance can be realized. In addition, the step-up PFC control device can completely disconnect the load connected between the output terminal and the GND terminal of the step-up PFC control device from the input AC voltage source.

  According to another aspect of the present invention, the switching element includes a stacked body composed of a plurality of nitride semiconductor layers formed on a semiconductor substrate, the gate terminal formed on the stacked body, It is desirable to provide the drain terminal and the source terminal formed on both sides of the gate terminal.

  The switching element is generally known as a heterojunction field effect transistor using a gallium nitride semiconductor and is called a GaN transistor. The GaN transistor has FET characteristics and reverse FET characteristics when the gate / source voltage is higher than a certain threshold voltage, and has reverse conduction characteristics when the gate / source voltage is equal to or lower than the threshold voltage. It is also a bidirectional switching element having high withstand voltage characteristics, and is an FET transistor having a very low on-resistance value in FET characteristics and reverse FET characteristics. Further, there is no minority carrier effect as in a silicon-based semiconductor element, and there is almost no influence of an increase in switching loss due to a recovery current. Therefore, by using a GaN transistor as a bidirectional switching element included in the step-up PFC control device of the present invention, a step-down PFC control device with lower power consumption can be realized. In addition, the switching frequency can be increased by reducing the switching loss, whereby the size of the boosting coil can be reduced, and as a result, the device can be reduced in size.

  According to the present invention, it is possible to provide a step-up type PFC control device without a diode and a parasitic diode of a switching element. The power consumption of the conventional boost type PFC control device is largely due to the diode and the parasitic diode of the switching element, but the power consumption can be greatly reduced in the present invention. In addition, since the PFC control unit included in the boost type PFC control device can use the control unit of the conventional boost type PFC control device as it is, the conventional boost type PFC control device changes to the boost type PFC control device of the present invention. Is easy to replace.

1 is a circuit block diagram of a step-up PFC control device according to an embodiment of the present invention. It is an IV characteristic diagram showing the FET characteristic of the bidirectional | two-way switching element used for this invention. It is an IV characteristic diagram showing the reverse FET characteristic of the bidirectional | two-way switching element used for this invention. It is an IV characteristic view showing the reverse conduction characteristic of the bidirectional | two-way switching element used for this invention. It is a figure explaining the equivalent conversion in the ON operation state of a switching element. It is a figure explaining the equivalent conversion in the OFF operation state of a switching element. 1 is a circuit block diagram of a drive logic circuit according to an embodiment of the present invention. 4 is an operation timing chart of internal signals of the drive logic circuit according to the embodiment of the present invention. It is a figure showing the electric current path of the pressure | voltage rise operation | movement of the AC / DC converter part which concerns on embodiment when the voltage polarity of an input AC power supply is positive. It is a figure showing the current path of the pressure | voltage rise operation of the AC / DC converter part which concerns on embodiment when the voltage polarity of an input AC power supply is negative. It is an example of sectional drawing of the bidirectional | two-way switching element which the pressure | voltage rise type PFC control apparatus of this invention has. It is a circuit block diagram of the step-up PFC control device according to the first modification of the embodiment of the present invention. It is an equivalent circuit diagram showing the state of the AC switch according to the first modification of the embodiment when the voltage polarity of the input AC power supply is positive. It is an equivalent circuit diagram showing the state of the AC switch of the AC switch according to the first modification of the embodiment when the voltage polarity of the input AC power supply is negative. It is a circuit block diagram of the drive logic circuit concerning the 2nd modification of an embodiment of the invention. It is an operation | movement timing chart of the internal signal of the drive logic circuit which concerns on the 2nd modification of embodiment of this invention. It is a figure showing the electric current path of the pressure | voltage rise operation of the AC / DC converter part which concerns on the 2nd modification of embodiment when the voltage polarity of an input AC power supply is positive. It is a figure showing the electric current path of the pressure | voltage rise operation of the AC / DC converter part which concerns on the 2nd modification of embodiment when the voltage polarity of an input AC power supply is negative. It is a circuit block diagram of the pressure | voltage rise type PFC control apparatus which concerns on a comparative example. It is an equivalent circuit diagram showing the state of a switching element in case the voltage polarity of the input AC power supply which concerns on a comparative example is positive. It is an equivalent circuit diagram showing the state of a switching element in case the voltage polarity of the input AC power supply which concerns on a comparative example is negative. It is a figure showing the transient characteristic of a switching element in case the voltage polarity of the input AC power supply which concerns on a comparative example is positive. It is a figure showing the transient characteristic of a switching element in case the voltage polarity of the input alternating current power supply which concerns on a comparative example is negative. FIG. 10 is a circuit block diagram of a step-up PFC control device without a rectifier diode bridge described in Patent Document 1.

  Hereinafter, preferred embodiments of a step-up PFC control apparatus according to the present invention will be described in detail with reference to the accompanying drawings. The present invention is not limited to the specific configurations described in the following embodiments, and is based on the same technical idea as the technical idea described in the embodiment and the common general technical knowledge in this technical field. Is included.

(Embodiment)
FIG. 1 is a circuit block diagram of a step-up PFC control apparatus according to an embodiment of the present invention. The step-up PFC control device 100 shown in FIG. 1 includes an AC / DC converter unit 30, a current detection element 3, a second error amplifier 9, a multiplier circuit 10, a second absolute value circuit 11, and a PWM comparator. 12, a sawtooth wave oscillator 14, a differential amplifier 17, a reference voltage source 18, a first error amplifier 19, a comparator 20, and a first absolute value circuit 21. The AC / DC converter unit 30 has a function of rectifying and boosting the AC voltage input from the input AC power supply 1, and includes a boosting coil 4, a smoothing capacitor 7, AC switches S1 to S4, Four pre-drive circuits 51, three power supplies 52, and a drive logic circuit 53 are provided.

  The block configuration excluding the AC / DC converter unit 30 is the same as the block configuration of the conventional step-up PFC control device 500 shown in FIG. Therefore, in the description of the embodiment of the present invention, the AC / DC converter unit 30 will be mainly described.

  In the AC / DC converter unit 30, the silicon-based semiconductor switching elements T1 and T2 shown in FIG. 13 are replaced with AC switches S1 and S2 which are bidirectional switching elements, respectively. Further, the diodes D1 and D2 shown in FIG. 13 are replaced with AC switches S3 and S4 using bidirectional switching elements.

  The AC switches S1 to S4 are bridge-connected as follows. The input side electrode, which is the first electrode of the boosting coil 4, is connected to a first AC terminal to which a single-phase input AC voltage is applied. The drain terminal of the AC switch S1 and the drain terminal of the AC switch S3 are: The source terminal of the AC switch S2 and the source terminal of the AC switch S4 are connected to the ground electrode, which is the second electrode of the smoothing capacitor 7, and connected to the first electrode of the smoothing capacitor 7. The source terminal and the drain terminal of the AC switch S2 are connected to the output side electrode, which is the second electrode of the boosting coil 4, and the source terminal of the AC switch S3 and the drain terminal of the AC switch S4 have an input AC voltage. It is connected to the applied second AC terminal. The gate terminals of the AC switches S1 to S4 are connected to the respective predrive circuits 51 that shift the level of the control signal output from the drive logic circuit 53.

  The step-up coil 4 stores magnetic energy corresponding to a bidirectional current that flows in response to application of a single-phase input AC voltage.

  The smoothing capacitor 7 stores a charge corresponding to the magnetic energy stored in the boosting coil 4.

  Here, the characteristics of the bidirectional switching element will be described.

  The bidirectional switching element is a bidirectional switching element having the IV characteristic shown in FIGS. 2A, 2B, and 2C. This characteristic will be described below.

  FIG. 2A is an IV characteristic diagram showing FET characteristics of the bidirectional switching element used in the present invention. FIG. 2B is an IV characteristic diagram showing the reverse FET characteristic of the bidirectional switching element used in the present invention. FIG. 2C is an IV characteristic diagram showing reverse conduction characteristics of the bidirectional switching element used in the present invention.

  The bidirectional switching element has a gate terminal, a drain terminal, and a source terminal. When the gate / source voltage Vgs, which is a differential voltage of the gate terminal voltage with respect to the source terminal voltage, is higher than the threshold voltage Vth, The current IDS can flow from the drain terminal to the source terminal or the current IDS can flow from the source terminal to the drain terminal according to the polarity of the differential voltage VDS between the drain terminal and the source terminal.

  2A and 2B show the relationship between the current IDS and the differential voltage VDS when the switching element is in the on-operation state because the gate / source voltage Vgs is higher than the threshold voltage Vth. The current IDS is a positive value when flowing from the drain terminal to the source terminal.

  The IV characteristic has a triode region and a saturation region like the MOSFET IV characteristic. The triode region is a region in the vicinity of the zero voltage until the VDS reaches a certain voltage value from the zero voltage, and the saturation region is a characteristic similar to the constant current characteristic in which the IDS does not change much even if the VDS changes. It is an area | region which shows. In the triode region, the IV characteristic has linearity, and the slope of VDS with respect to IDS can be defined as the on-resistance Ron of the switching element. The on-resistance Ron in the triode region is sufficiently smaller than the on-resistance in the saturation region. The characteristics of the bidirectional switching element in the triode region are important as the operation characteristics of the AC switches S1 to S4. In the following description, the characteristics of the triode region are limited.

  FIG. 2A is an IV characteristic diagram when VDS is positive, that is, when the drain voltage is higher than the source voltage. As can be seen from the IV characteristic diagram, IDS has a positive value, and current flows from the drain terminal to the source terminal. The IV characteristics shown in FIG. 2A will be referred to as FET characteristics. FIG. 2B is an IV characteristic diagram when VDS is negative, that is, when the drain voltage is lower than the source voltage. As can be seen from this IV characteristic diagram, IDS has a negative value, and current flows from the source terminal to the drain terminal. The IV characteristics shown in FIG. 2B will be referred to as inverse FET characteristics.

  In the bidirectional switching element, when the gate / source voltage Vgs is lower than the threshold voltage Vth, the current IDS cannot flow from the drain terminal to the source terminal. However, in the bidirectional switching element, even when the gate-source voltage Vgs is lower than the threshold voltage Vth, the drain terminal voltage is lower than the gate terminal voltage, and the difference voltage (Vgs−VDS) is higher than the threshold voltage Vth. In some cases, the current IDS can flow from the source terminal to the drain terminal. This characteristic is called reverse conduction characteristic. FIG. 2C is an IV characteristic diagram showing the reverse conduction characteristic. As can be seen from FIG. 2C, in the state of Vgs = 0V, that is, in the state where the gate terminal and the source terminal are short-circuited, the diode I− is connected to the source terminal as the anode and the drain terminal as the cathode and the forward voltage is the threshold voltage Vth. It is the same as the V characteristic.

  As can be seen from FIGS. 2A, 2B, and 2C, the bidirectional switching element performs the FET operation and the reverse FET operation when the gate / source voltage Vgs is equal to or higher than the threshold voltage Vth, and the on-resistance It can be regarded as a resistor having the value Ron. On the other hand, when the gate-source voltage Vgs is equal to or lower than the threshold voltage Vth, the operation is performed according to the reverse conduction characteristic, and it can be regarded as a diode having the source terminal as an anode and the drain terminal as a cathode. The forward voltage of this diode is (Vth−Vgs).

  In the future, in the embodiment of the present invention, the bidirectional switching element is equivalently converted as follows.

  (1) In the ON operation state of the bidirectional switching element in which the gate / source voltage Vgs is higher than the threshold voltage Vth, the switching element is regarded as a resistance having an ON resistance value Ron.

  (2) In the off operation state of the bidirectional switching element in which the gate terminal and the source terminal are short-circuited, the switching element is regarded as a diode having the source terminal as an anode and the drain terminal as a cathode.

  FIG. 2D is a diagram for explaining equivalent conversion in the on-operation state of the switching element, and FIG. 2E is a diagram for explaining equivalent conversion in the off-operation state of the switching element.

  Next, the drive logic circuit 53 included in the AC / DC converter unit 30 will be described.

  FIG. 3A is a circuit block diagram of the drive logic circuit 53 according to the embodiment of the present invention, and FIG. 3B is an operation timing chart of internal signals of the drive logic circuit 53 according to the embodiment of the present invention.

  The drive logic circuit 53 shown in FIG. 3A receives a PWM signal, generates an LPWM signal for PWM driving one of the AC switches S1 and S2, and PWM-drives the other of the AC switches S1 and S2. A UPWM signal is generated. Here, the PWM signal is a signal output from the PWM comparator 12 shown in FIG. 1, and is a signal for PWM control of the boosting operation. One of the AC switches S1 and S2 drives the boosting coil 4 for boosting operation by the LPWM signal, and the other of the AC switches S1 and S2 is one of the AC switches S1 and S2 by the UPWM signal. Operates synchronously.

  As can be seen from FIG. 3B, the LPWM signal has the same waveform as the PWM signal for PWM control of the boosting operation, but is delayed from the PWM signal by a certain delay time DT. Similarly, the UPWM signal is a signal having a polarity opposite to that of the LPWM signal, and a waveform is generated so that a section in which the UPWM signal and the LPWM signal are simultaneously Low is provided only for the delay time DT. A section in which these two signals are Low at the same time is called a dead time. Due to this dead time, as long as the two AC switches S1 and S2 are operating normally, they cannot be in the ON operation state at the same time.

  As a result, it is avoided that the output of the step-up PFC control device and GND are short-circuited due to both of the AC switches S1 and S2 being in an on-operation state, so that the so-called AC switches S1 and S2 are not penetrated. Is done. The current flowing by the energy stored in the boosting coil 4 is transferred to the smoothing capacitor 7 by the LPWM signal and the UPWM signal via the switching operation of the AC switches S1 and S2, and a stable boosting operation is realized. The In this boosting operation, the AC switch S1 or S2 operates as a boosting diode in the dead time section (of course, not only in the dead time section but also in the section in which the UPWM signal is Low) according to the rules of equivalent conversion described above. The section in which the UPWM signal is High operates as a resistor having an on-resistance value Ron.

  The period in which current flows while the AC switches S1 and S2 operate as boosting diodes is only a very short period of time, which is a dead time period. Therefore, the power consumption of the AC switches S1 and S2 is very small compared to the switching elements T1 and T2 of the conventional step-up PFC control device.

  Four output signals G_S1 to G_S4 of the drive logic circuit 53 control the four AC switches S1 to S4. As shown in the timing waveform diagram of FIG. 3B, the output signals G_S1 and G_S2 output the LPWM signal or the UPWM signal, and the output signals G_S3 and G_S4 output the high or low level signal. As can be seen from FIG. 1, the DIR signal shown in FIG. 3B becomes a high level signal when the AC output of the input AC power supply 1 applies a positive voltage to the boosting coil 4, and the AC output is for boosting. When a negative voltage is applied to the coil 4, it becomes a Low level signal.

  FIG. 4A is a diagram illustrating a current path of the boost operation of the AC / DC converter unit according to the embodiment when the voltage polarity of the input AC power supply is positive, and FIG. 4B is a diagram illustrating that the voltage polarity of the input AC power supply is negative. It is a figure showing the electric current path of the pressure | voltage rise operation | movement of the AC / DC converter part which concerns on embodiment in the case of. The boosting operation described in FIGS. 4A and 4B is realized by the PWM operation of the drive logic circuit 53 shown in FIG. 3A. Here, it will be described that a current always flows through the smoothing capacitor 7 by the magnetic energy of the boosting coil 4 and the output voltage Vo is boosted regardless of whether the voltage polarity of the input AC power supply 1 is positive or negative. .

  When the input AC power supply 1 is positive, the DIR signal is at a high level. From the timing waveform diagram of FIG. 3B, the AC switch S2 receives the LPWM signal for driving the boosting coil 4, and the AC switch S1 is the UPWM signal. Receive. The AC switch S3 receives a Low level signal, and the AC switch S4 receives a High level signal. The AC switches S1 to S4 receive these signals and enter the operating state shown in FIG. 4A. In FIGS. 4A and 4B, the LPWM signal is described as a PWM signal for controlling the step-up PWM operation, and the UPWM signal that is a synchronization signal is described as a PWM signal with a bar that represents an inverted signal of the PWM signal. ing.

  In FIG. 4A, the AC switch S2 performs a PWM operation to drive the boosting coil 4, the AC switch S1 enters an operation state in which the PWM operation is inverted, and the AC switch S4 functions as a resistor having an on-resistance Ron to pass a current. Since the AC switch S3 is in an OFF operation state, it can be regarded as a diode in terms of an equivalent circuit, but no current flows. When the AC switch S2 is in the ON operation state by the PWM operation, a current for exciting the boosting coil 4 flows through the path (a) in FIG. 4A. On the other hand, when the AC switch S2 is in the OFF operation state by PWM operation, current flows through the AC switch S1 in the path (b) of FIG. A current is charged from a terminal of the capacitor 7 that is not grounded, and the output voltage Vo of the step-up PFC control device 100 is stepped up.

  In the PWM operation in which the AC switches S2 and S1 are operated synchronously, the above-described dead time interval is provided, and the AC switches S1 and S2 do not fall into the through state. In a very short time of the dead time interval, the AC switch S1 functions as a boosting diode. When the AC switch S2 is switched from ON to OFF, the current of the booster coil 4 is caused by the magnetic energy stored in the booster coil 4 so that the AC switch S1 is used as a booster diode and the AC switch S1 in the dead time period. In the ON operation state, the AC switch S1 flows as a resistor having an ON resistance Ron and flows into the ungrounded terminal of the smoothing capacitor 7 via the AC switch S1 through the path (b) of FIG. 4A.

  On the other hand, when the input AC power supply 1 has a negative polarity, the DIR signal is at a low level. From the timing waveform diagram of FIG. 3B, the movement of the AC switches S2 and S1 compared to the case where the input AC power supply 1 has a positive polarity. It turns out that the movements of the AC switches S4 and S3 are reversed.

  In FIG. 4B, the AC switch S1 performs a PWM operation to drive the boosting coil 4, the AC switch S2 enters an operation state in which the PWM operation is inverted, and the AC switch S3 functions as a resistor having an on-resistance Ron to pass a current. Since the AC switch S4 is in an OFF operation state, it can be regarded as a diode in terms of an equivalent circuit, but no current flows. When the AC switch S1 is in the ON operation state by PWM operation, a current for exciting the boosting coil 4 flows through the path (a) in FIG. 4B. On the other hand, when the AC switch S1 is in the OFF operation state by PWM operation, current flows through the AC switch S2 in the path (b) of FIG. A current is charged from a terminal of the capacitor 7 that is not grounded, and the output voltage Vo of the step-up PFC control device 100 is stepped up.

  Even in this case, in the PWM operation in which the AC switches S1 and S2 are operated synchronously, the above-described dead time interval is provided, and the AC switch S2 performs the same operation as the AC switch S1 when the input AC power supply 1 is positive. Then, the current of the boosting coil 4 flows into the ungrounded terminal of the smoothing capacitor 7 via the AC switch S2 through the path (b).

  That is, the drive logic circuit 53 switches the current path of the bidirectional current by controlling the on / off states of the AC switches S1 to S4 to accumulate magnetic energy in the boosting coil 4 and charge to the smoothing capacitor 7. Accumulate and alternate.

  From the above operation, a current always flows through the terminal of the smoothing capacitor 7 that is not grounded regardless of whether the voltage polarity of the input AC power supply 1 is positive or negative, by the magnetic energy of the boosting coil 4, and output It can be seen that the voltage Vo is boosted. Therefore, it can be seen that the step-up PFC control device 100 shown in FIG. 1 has a function of rectifying and stepping up the AC voltage of the input AC power supply 1 even though there is no rectifier diode bridge.

  When the current flowing through the boosting coil 4 shown in FIG. 1 is substantially zero, the hysteresis comparator shown in FIG. 3A outputs the output signal of the current detection element 3 to the positive value in the second absolute value circuit 11. By comparing the signal Sc converted into a voltage signal with the comparison reference voltage VRB, it is detected that almost no current flows. The AC switch S1 or S2 operating as a resistor having an on-resistance value Ron upon receiving a high-level gate signal changes the gate signal from the high level to the low level by the zero current detection operation of the hysteresis comparator. As a result, the operation changes to diode operation.

  In this function, when the current flowing through the boosting coil 4 is substantially zero, the magnetic energy stored in the boosting coil 4 is lost, so that the input AC power source is connected from the output side through the AC switch and the boosting coil 4. This is to prevent the current from flowing back to 1.

  If no current flows through the boosting coil 4, the voltage across the boosting coil 4 may cause a ringing phenomenon. In this case, although not shown in FIG. 3A, a function of maintaining the Low level for a certain period when the output signal of the hysteresis comparator becomes Low level may be added to the subsequent stage of the hysteresis comparator.

  As described above, the step-up PFC control apparatus 100 according to the embodiment of the present invention realizes a step-up PFC control apparatus without a diode and without a parasitic diode of a switching element.

  In the power consumption of the conventional boost type PFC control device, the conduction loss and the switching loss influenced by the diode and the parasitic diode of the switching element occupy a large proportion of the power loss of the entire device. In contrast, in the step-up PFC control device according to the present invention, the power consumption can be greatly reduced while maintaining the conventional rectification and step-up operation by eliminating the diode and the parasitic diode of the switching element.

  In addition, the bidirectional switching element used as an AC switch does not have a parasitic diode, and therefore there is no recovery current due to the parasitic diode. As a result, even if the switching frequency is increased, the switching loss of the step-up PFC controller does not increase significantly, and the step-up coil 4 can be made smaller by increasing the switching frequency.

  Further, since the PFC control unit included in the boost type PFC control device can use the control unit of the conventional boost type PFC control device as it is, the boost type PFC control device of the present invention is changed from the conventional boost type PFC control device. Replacement is also easy.

  Note that the switching element described above includes a stacked body formed of a nitride semiconductor layer formed on a semiconductor substrate, a drain terminal and a source terminal formed on the stacked body at intervals, and the drain terminal. And a gate terminal formed between the source terminals. This switching element will be described with reference to FIG.

  FIG. 5 is an example of a cross-sectional view of a bidirectional switching element included in the step-up PFC control device of the present invention. The bidirectional switching element shown in the figure is a normally-off type heterojunction FET made of a nitride semiconductor formed on a semiconductor substrate. Specifically, the switching element is realized by forming a stacked body 203 of semiconductor layers on a silicon substrate 201 via a buffer layer 202.

  The buffer layer 202 is formed by alternately stacking aluminum nitride and gallium nitride.

  In the stacked body 203, an n-type aluminum gallium nitride layer 205 is formed on an undoped gallium nitride layer 204, and an FET having a high carrier concentration called a two-dimensional electron gas is located in the vicinity of the heterointerface between the two layers. Channel regions are generated.

  On the stacked body 203, in order to form a source terminal and a drain terminal, a source terminal ohmic electrode 206a, a drain terminal ohmic electrode 206b, and a wiring 210 which are in ohmic contact with the channel region are arranged.

  In a region between the source terminal ohmic electrode 206a and the drain terminal ohmic electrode 206b, a control layer 209, which is a p-type semiconductor layer for controlling FET characteristics, is formed on the n-type aluminum gallium nitride layer 205.

  A gate electrode 208 is formed on the control layer 209 and is in ohmic contact with the control layer 209. The electric signal supplied to the gate electrode 208 controls the current flowing between the drain terminal and the source terminal of the normally-off type heterojunction FET, that is, the bidirectional switching element.

  In FIG. 5, the distance from the drain terminal ohmic electrode 206b to the gate electrode 208 is longer than the distance from the source terminal ohmic electrode 206a to the gate electrode 208 because the withstand voltage between the drain terminal and the gate terminal is higher than the source terminal. This is because it is required to be larger than the breakdown voltage between the gate terminals.

  The bidirectional switching element formed as shown in FIG. 5 is called a GaN transistor, and is a device that can be driven with a high current with a high breakdown voltage, such as an IGBT. Further, the current-voltage characteristics of the IGBT have a characteristic of flowing a current through both of them as shown in FIGS. 2A and 2B without having an offset voltage due to a PN junction. Furthermore, the on-resistance component Ron is very small with respect to the chip area of the device. In addition, the GaN transistor also has the reverse conduction characteristics shown in FIG. 2C.

  In addition to the above-described characteristics, the GaN transistor has almost no accumulation effect due to minority carriers, and almost no tail current effect during turn-off as in IGBTs and other silicon-based semiconductor devices.

  Therefore, the AC / DC converter unit 30 composed of an AC switch using the GaN transistor as a bidirectional switching element has a reduced conduction loss due to its very small on-resistance value Ron and almost no minority carrier storage effect. By reducing switching loss, power consumption can be greatly reduced.

  In addition, by applying a GaN transistor to the bidirectional switching element, there is almost no influence of an increase in switching loss due to the recovery current, and the switching frequency can be increased by reducing the switching loss. As a result, the apparatus can be miniaturized.

  Therefore, by using a GaN transistor as the bidirectional switching element of the step-up PFC control apparatus 100 according to the embodiment of the present invention, a step-down PFC control apparatus with lower power consumption can be realized.

  FIG. 6 is a circuit block diagram of a step-up PFC control device according to a first modification of the embodiment of the present invention. The step-up PFC control device 150 shown in the figure is different from the step-up PFC control device 100 shown in FIG. 1 in the configuration of a part of the AC switch of the AC / DC converter unit 31. Hereinafter, the description of the same points as the step-up PFC control apparatus 100 described in FIG.

  Each of the AC switches S2 and S4 is composed of two bidirectional switching elements in which drain terminals are connected in series. With this configuration, it is possible to increase the withstand voltage of the AC switch, and a higher input withstand voltage boost type PFC control device can be realized. Further, in the step-up PFC control device 150, the output side can be disconnected from the input AC power supply 1 of the step-up PFC control device 150 by giving a low level signal to the illustrated OFF signal.

  FIG. 7A is an equivalent circuit diagram showing the state of the AC switch according to the first modification of the embodiment when the voltage polarity of the input AC power supply is positive, and FIG. 7B shows the voltage polarity of the input AC power supply. It is an equivalent circuit diagram showing the state of the alternating current switch which concerns on the 1st modification of embodiment in the negative case. 7A and 7B, the PWM signal that is the output of the PWM comparator 12 is at the low level, and the OFF signal is also at the low level.

  In this modification, the AC switches S2 and S4 are each configured by connecting two bidirectional switching elements in series, but the two bidirectional switching connected in series to the AC switches S1 and S3. An element is applied, and the AC switches S2 and S4 may be normal AC switches.

  Moreover, although AC switch S2 and S4 which concern on this modification are set as the structure by which each drain terminal of two both switching elements was connected, the thing connected in series with the form which the source terminals were connected may be used.

  FIG. 8A is a circuit block diagram of a drive logic circuit 63 according to a second modification of the embodiment of the present invention, and FIG. 8B illustrates a drive logic circuit 63 according to the second modification of the embodiment of the present invention. It is an operation | movement timing chart of an internal signal.

  The drive logic circuit 63 shown in FIG. 8A receives a PWM signal, generates an LPWM signal for PWM driving one of the AC switches S2 and S4, and PWM drives one of the AC switches S1 and S3. A UPWM signal is generated. Here, the PWM signal is a signal output from the PWM comparator 12 shown in FIG. 1, and is a signal for PWM control of the boosting operation. One of the AC switches S2 and S4 drives the boosting coil 4 for boosting operation by the LPWM signal, and one of the AC switches S1 and S3 is one of the AC switches S2 and S4 by the UPWM signal. Operates synchronously.

  As can be seen from FIG. 8B, the relationship among the PWM signal, the UPWM signal, and the LPWM signal is the same as the relationship described in FIG. 3B. A section in which these two signals are Low at the same time is called a dead time. Due to the dead time when the UPWM signal and the LPWM signal are simultaneously Low, as long as the two AC switches S1 and S2 or the AC switches S3 and S4 are operating normally, they cannot be in the ON operation state at the same time. .

  As a result, when the AC switches S1 and S2 or the AC switches S3 and S4 are both turned on, the output of the step-up PFC control device 200 and the GND are short-circuited, resulting in a so-called through state. This is avoided. The current flowing due to the energy stored in the boosting coil 4 is switched by the smoothing capacitor 7 via the switching operation by the UPWM signal of the AC switch S1 or S3 when S2 or S4 performs the switching operation by the LPWM signal. And a stable boosting operation is realized. In this boosting operation, the AC switch S3 or S4 that performs switching operation with the UPWM signal is boosted in the dead time interval (of course, not only in the dead time interval but also in the interval in which the UPWM signal is Low) according to the rules of equivalent conversion described above. A section in which the UPWM signal is High operates as a resistor having an on-resistance value Ron.

  The period during which the current flows while the AC switches S1 and S3 are operating as boosting diodes is only a very short period of the dead time period. Therefore, the power consumption of the AC switches S1 and S3 is very small compared to the switching elements T3 and T4 of the conventional step-up PFC control device.

  There are four outputs of the drive logic circuit 63 as in the embodiment. The output signals G_S1 to G_S4 control the AC switches S1 to S4, respectively. As shown in the timing waveform diagram of FIG. 8B, the output signals G_S2 and G_S4 output an LPWM signal or a high level signal, and the output signals G_S1 and G_S3 output a UPWM signal or a low level signal. As can be seen from FIG. 1, the DIR signal shown in FIG. 8B becomes a high level signal when the AC output of the input AC power supply 1 applies a positive voltage to the boosting coil 4, and the AC output is for boosting. When a negative voltage is applied to the coil 4, it becomes a Low level signal.

  FIG. 9A is a diagram showing a current path of the boost operation of the AC / DC converter unit when the voltage polarity of the input AC power supply is positive, and FIG. 9B is an AC / DC when the voltage polarity of the input AC power supply is negative. It is a figure showing the electric current path of the pressure | voltage rise operation of a converter part. The boosting operation described in FIGS. 9A and 9B is realized by the PWM operation of the drive logic circuit 63 shown in FIG. 8A. Here, it will be described that a current always flows through the smoothing capacitor 7 by the magnetic energy of the boosting coil 4 and the output voltage Vo is boosted regardless of whether the voltage polarity of the input AC power supply 1 is positive or negative. .

  When the input AC power supply 1 is positive, the DIR signal is at a low level. From the timing waveform diagram of FIG. 8B, the AC switch S2 receives the LPWM signal for driving the boosting coil 4, and the AC switch S1 is the UPWM signal. Receive. The AC switch S3 receives a Low level signal, and the AC switch S4 receives a High level signal. The AC switches S1 to S4 receive these signals and enter the operating state shown in FIG. 9A. In FIGS. 9A and 9B, the LPWM signal is described as a PWM signal for controlling the step-up PWM operation, and the UPWM signal that is a synchronization signal is described as a PWM signal with a bar that represents an inverted signal of the PWM signal. ing.

  In FIG. 9A, the AC switch S2 performs a PWM operation to drive the boosting coil 4, and the AC switch S1 is in an operation state in which the PWM operation is inverted. The AC switch S4 allows current to flow as a resistor having an ON resistance Ron, and the AC switch S3 is in an OFF operation state, so that it can be regarded as a diode in terms of an equivalent circuit, but no current flows. When the AC switch S2 is in the ON operation state by the PWM operation, a current for exciting the boosting coil 4 flows through the path (a) in FIG. 9A. On the other hand, when the AC switch S2 is in the OFF operation state by the PWM operation, current flows through the AC switch S1 in the path (b) of FIG. 9A due to the magnetic energy stored in the boosting coil 4, and smoothing is performed. A current is charged from a terminal of the capacitor 7 that is not grounded, and the output voltage Vo of the step-up PFC control device 200 is stepped up.

  In the PWM operation in which the AC switches S2 and S1 are operated synchronously, the above-described dead time period is provided, and the AC switches S2 and S1 do not fall into the through state. When the AC switch S2 is switched from ON to OFF, the current in the booster coil 4 is caused by the magnetic energy stored in the booster coil 4 so that the AC switch S1 is used as a booster diode and the AC switch S1 during the dead time period. In the ON operation state, the AC switch S1 flows as a resistor having an ON resistance Ron and flows into the ungrounded terminal of the smoothing capacitor 7 via the AC switch S1 through the path (b) of FIG. 9A.

  On the other hand, when the input AC power supply 1 has a negative polarity, the DIR signal is at a high level. From the timing waveform diagram of FIG. 8B, the movement of the AC switches S2 and S4 compared to the case where the input AC power supply 1 has a positive polarity. Is reversed, and the movements of the AC switches S1 and S3 are reversed.

  In FIG. 9B, the AC switch S4 performs a PWM operation to drive the boosting coil 4, the AC switch S3 enters an operation state in which the PWM operation is inverted, and the AC switch S2 passes a current as a resistor having an on-resistance Ron. Since the switch S1 is in the OFF operation state, it can be regarded as a diode in terms of an equivalent circuit, but no current flows. When the AC switch S4 is in the ON operation state by the PWM operation, a current for exciting the boosting coil 4 flows through the path (a) in FIG. 9B. On the other hand, when the AC switch S4 is in the OFF operation state by the PWM operation, the magnetic energy stored in the boosting coil 4 causes a current to flow through the AC switch S3 in the path (b) of FIG. A current is charged from a terminal of the capacitor 7 that is not grounded, and the output voltage Vo of the step-up PFC control device 200 is stepped up.

  Even in this case, in the PWM operation in which the AC switches S4 and S3 are operated synchronously, the dead time interval described above is provided, and the AC switch S3 performs the same operation as the AC switch S1 when the input AC power source 1 is positive. Then, the current of the boosting coil 4 flows into the ungrounded terminal of the smoothing capacitor 7 through the path (b) via the AC switch S3.

  From the above operation, a current always flows through the terminal of the smoothing capacitor 7 that is not grounded regardless of whether the voltage polarity of the input AC power supply 1 is positive or negative, by the magnetic energy of the boosting coil 4, and output It can be seen that the voltage Vo is boosted. Therefore, it can be seen that even if the drive logic circuit 53 of FIG. 1 is replaced with the drive logic circuit 63 of another modification of the embodiment, the same boosting operation is realized as a result.

  When the current flowing through the boosting coil 4 shown in FIG. 1 is substantially zero, the hysteresis comparator shown in FIG. 8A causes the output signal of the current detection element 3 to be positive by the second absolute value circuit 11. By comparing the signal Sc converted into the voltage signal with the comparison reference voltage VRB, it is detected that there is almost no current. Further, as shown in FIG. 8B, the AC switch S1 or S3 that operates synchronously with the UPWM signal with respect to the AC switch S2 or S4 that switches the boosting coil 4 with the LPWM signal is the zero of the hysteresis comparator. The gate signal is changed to a low level by the current detection operation, and the AC switch S1 or S3 is changed to a diode operation.

  In this function, when the current flowing through the boosting coil 4 is substantially zero, the magnetic energy stored in the boosting coil 4 is lost, so that the input AC power source is connected from the output side through the AC switch and the boosting coil 4. This is to prevent the current from flowing back to 1.

  If no current flows through the boosting coil 4, the voltage across the boosting coil 4 may cause a ringing phenomenon. In this case, although not shown in FIG. 8A, a function of maintaining the Low level for a certain period when the output signal of the hysteresis comparator becomes Low level may be added to the subsequent stage of the hysteresis comparator.

  As described above, the drive logic circuit 63 according to the second modified example of the embodiment of the present invention realizes a step-up PFC control device without a diode and without a parasitic diode of a switching element.

(Comparative example)
Below, the comparative example with respect to embodiment of this invention and its modification is demonstrated.

  FIG. 10 is a circuit block diagram of a boost type PFC control device according to a comparative example. The step-up type PFC control device 300 shown in the figure is a step-up type PFC circuit without a rectifier diode bridge.

  The step-up PFC control apparatus 300 according to this comparative example includes the step-up PFC control apparatus 500 of Patent Document 1 described in FIG. A mechanism (a pair of switching elements T3 and T4) and a high-frequency block filter 502 are excluded. In addition, the pair of switching elements T1 and T2 that perform the switching operation included in the step-up PFC control device 500 are replaced with switching elements T5 and T6 of silicon-based semiconductor such as n-type MOSFET, and the switching element T5. And a pre-drive circuit 51 having a level shift function for driving T6 and a power source 52 are added. Hereinafter, the operation and problem of the step-up PFC control apparatus 300 without a rectifier diode bridge will be described with reference to FIG.

  In FIG. 10, the first error amplifier 19 generates an error voltage Ve that is proportional to the difference between the output voltage Vo of the step-up PFC controller 300 and the voltage value Vref of the reference voltage source 18 that sets the output voltage. A voltage signal Vina having a waveform similar to a voltage obtained by full-wave rectifying the AC voltage of the input AC power supply 1 is generated by the differential amplifier 17 and the first absolute value circuit 21. The multiplier circuit 10 multiplies Ve and Vina to generate a voltage signal Iref that is a current control command value. The voltage signal Iref is proportional to the error voltage Ve and has the same pulsation waveform as Vina that is similar to the voltage waveform obtained by full-wave rectifying the AC voltage of the input AC power supply 1.

  On the other hand, the current flowing from the input AC power source 1 to the boosting coil 4 by the switching operation of the switching elements T5 and T6 is detected by the current detection element 3, and converted to a positive voltage value signal by the second absolute value circuit 11. It is converted to a current-shaped waveform waveform Sc.

  The current waveform signal Sc includes a second error amplifier 9, a PWM comparator 12, a sawtooth oscillator 14, a switching element T5 or T6, a boosting coil 4, a diode D1 or D2, a smoothing capacitor 7, an output load 8, and a current detecting element 3. , And the current control negative feedback loop constituted by the second absolute value circuit 11 is controlled so as to follow the voltage signal Iref, and becomes almost the same value as the Iref.

  Since the voltage signal Iref is a pulsation waveform similar to the voltage waveform obtained by full-wave rectification of the AC voltage of the input AC power supply 1, the fact that the current waveform signal Sc follows Iref means that the AC voltage of the input AC power supply 1 is This means that the alternating current of the input AC power supply 1 has substantially the same phase and substantially the same waveform. Therefore, the power factor of the input AC power supply is approximately 1.

  The error voltage Ve is a voltage obtained by amplifying the difference between the output voltage Vo and the voltage signal Vref by the first error amplifier 19. The error voltage Ve that is the output of the first error amplifier 19 is controlled to be a finite value by the PFC control loop including the first error amplifier 19, the multiplier circuit 10, and the above-described current control negative feedback loop. . The error voltage Ve is described by the following equation, where A is the gain of the first error amplifier 19.

        Ve = A × (Vref−Vo) (Equation 1)

  Therefore, if the error voltage Ve is a finite value, if the gain A of the first error amplifier 19 is set sufficiently large, Vref = Vo. That is, the output voltage Vo of the step-up PFC control device 300 becomes a fixed voltage value determined by Vref.

  As can be seen from the above description, in the step-up PFC control device 300 according to this comparative example, the output voltage Vo maintains the desired voltage set by the voltage value Vref of the reference voltage, and the power factor of the input AC power supply. Thus, a step-up AC / DC converter with a high input power factor capable of reducing the power to approximately 1 is realized.

  The differential amplifier 17, the comparator 20, the NOT logic circuit 13, the two AND logic circuits 16, the switching elements T5 and T6, and the diodes D1 and D2 realize a rectified boost switching operation without using a rectifier diode bridge. . Hereinafter, this point will be described.

  The comparator 20 compares the output signal of the differential amplifier 17 to determine the polarity of the AC voltage of the input AC power supply 1 (the polarity of the AC voltage of the input AC power supply 1 is input to the boosting coil 4). When the AC power supply 1 applies a positive voltage, the polarity is positive). If the polarity of the alternating voltage is positive, the output signal of the comparator 20 is at a high level, and if the polarity of the alternating voltage is negative, the output is low.

  When the output signal of the comparator 20 is received, the NOT logic circuit 13 and the two AND logic circuits 16 are PWM signals output from the PWM comparator 12 for the boosting switching operation if the polarity of the input AC voltage is positive. The switching element T6 is PWM driven, the gate voltage of the switching element T5 is set to the low level, and the switching element T5 is turned off. Similarly, if the polarity of the input AC voltage is negative, the NOT logic circuit 13 and the two AND logic circuits 16 drive the switching element T5 by PWM with the PWM signal, and switch the gate voltage of the switching element T6 to the low level. The element T6 is turned off.

  FIG. 11A is an equivalent circuit diagram showing the state of the switching element when the voltage polarity of the input AC power supply according to the comparative example is positive, and FIG. 11B is when the voltage polarity of the input AC power supply according to the comparative example is negative. It is an equivalent circuit diagram showing the state of a switching element. That is, FIG. 11A and FIG. 11B show which is different depending on the PWM operation by the operation of the comparator 20, the NOT logic circuit 13, and the two AND logic circuits 16, depending on whether the voltage polarity of the input AC power supply 1 is positive or negative. It is a figure explaining how voltage | pressure | voltage rise operation | movement is made in this way.

  When the switching element T6 is in the ON operation state by the PWM operation, a current for exciting the boosting coil 4 flows through the path (a) in FIG. 11A. Further, when the switching element T6 is turned off by the PWM operation, a current flows due to the magnetic energy stored in the boosting coil 4 in the path (b) of FIG. 11A, and the smoothing capacitor 7 is not grounded. Current is charged, and the output voltage Vo of the step-up PFC control device 300 is stepped up. Since the switching elements T5 and T6 are silicon semiconductor switching elements such as n-type MOSFETs, there are parasitic diodes between the source terminal and the drain terminal as shown in FIG. Therefore, as shown in FIG. 11A, the current generated by the magnetic energy of the boosting coil 4 in the path (b) flows via the parasitic diode of the switching element T5.

  On the other hand, when the switching element T5 is in the ON operation state by the PWM operation, a current for exciting the boosting coil 4 flows through the path (a) in FIG. 11B. Further, when the switching element T5 is turned off by PWM operation, a current flows due to the magnetic energy stored in the boosting coil 4 in the path (b) of FIG. 11B, and even in this case, the smoothing capacitor 7 is grounded. A current is charged from a terminal that is not connected, and the output voltage Vo of the boost PFC control device 300 is boosted. Therefore, as shown in FIG. 11B, the current that flows due to the magnetic energy of the boosting coil 4 in the path (b) flows through the parasitic diode of the switching element T6.

  However, the boost PFC control device 300 according to the comparative example has the following two problems.

  (1) A loss due to a voltage drop due to the forward voltage VF of the diode D1 or D2 and the forward voltage VFP of the parasitic diode of the switching element T5 or T6 occurs.

  (2) The switching current during the turn-on operation of the switching element T5 or T6 is large due to the recovery current of the parasitic diode of the switching element T5 or T6.

  These two problems will be described.

  First, the problem (1) related to the forward voltage VF of the diode and the forward voltage VFP of the parasitic diode will be described. When the AC voltage of the input AC power supply 1 is positive, the forward voltage VF of the diode D2 is generated when a current flows through the path (a) in FIG. 11A. When a current flows through the path (b) in FIG. 11A, a voltage drop due to the forward voltage VFP of the switching element T5 in the off operation state occurs in addition to the forward voltage VF of the diode D2. Power consumption, that is, loss occurs in the diode D2 and the switching element T5 due to these voltage drops and the current flowing through these diodes. Similarly, when the AC voltage of the input AC power supply 1 is negative and the current flows through the path (a) in FIG. 11B, the forward voltage VF of the diode D1 is generated, and the path (b) in FIG. When current flows, a voltage drop occurs due to the forward voltage VF of the diode D1 and the forward voltage VFP of the switching element T6, and loss occurs in the diode D1 and the switching element T6.

  The power consumption of the element due to the constant flow of current is called conduction loss. The conduction loss in the case where current flows when the switching element T5 or T6 is in the on-operation state is the forward direction of the diode because the resistance of the channel portion through which the current of the switching element flows (hereinafter referred to as on-resistance) is small. This is very small compared to the conduction loss due to the voltage drop of the voltage VF or VFP. Therefore, as can be seen from the current paths shown in FIGS. 11A and 11B, it can be seen that most of the conduction loss of the conventional boost type PFC control device is due to the forward voltages VF and VFP of the diode. That is, if the diodes D1 and D2 of the boost type PFC control device 300 according to the comparative example and the parasitic diodes of the switching elements T5 and T6 can be eliminated, the conduction loss of the boost type PFC control device 300 can be greatly reduced.

  Next, the problem (2) of switching loss will be described. The anode of the parasitic diode existing between the source terminal and the drain terminal of the switching elements T5 and T6 is at the source of the switching element, and the cathode of the parasitic diode is at the drain of the switching element. In FIG. 11A, when the switching element T5 is in the OFF operation state, and in FIG. 11B, when the current flows through each path (b) when the switching element T6 is in the OFF operation state, the parasitic diode is switched from the anode to the cathode. Current is flowing.

  At this time, a recovery current component due to the minority carrier accumulation effect in the diode is generated in the parasitic diode. As a result, from the path (b) state in which one of the switching elements T5 and T6 is in the off operation state and the current flows through the parasitic diode, the other of the switching elements T5 and T6 is turned on and the current in the path (a) state. When the current starts to flow, the other of the switching elements T5 and T6 drives not only a current for driving the boosting coil 4 but also a recovery current of an unnecessary parasitic diode by a turn-on operation.

  FIG. 12A is a diagram illustrating transient characteristics of the switching element when the voltage polarity of the input AC power supply according to the comparative example is positive, and FIG. 12B illustrates switching when the voltage polarity of the input AC power supply according to the comparative example is negative. It is a figure showing the transient characteristic of an element. As shown in FIG. 12A, the current IDS (T6) flowing through the switching element T6 has a turn-on operation that is a change from the state (b) to the state (a), in addition to the driving current IL4 of the boosting coil 4 and the switching element T5. The recovery current of the parasitic diode is superimposed. On the other hand, as shown in FIG. 12B, the current IDS (T5) flowing through the switching element T5 includes a switching element in addition to the drive current IL4 of the boosting coil 4 by a turn-on operation that is a change from the state (b) to the state (a). The recovery current of the parasitic diode of T6 is superimposed.

  The recovery current is a large one that cannot be ignored, and affects the increase in switching loss, which means power consumption during the switching operation of the switching element. This switching loss increases in proportion to the switching frequency of the step-up PFC control device. Therefore, increasing the switching frequency for the purpose of reducing the size of the boosting coil 4 causes an increase in switching loss of the switching element. That is, it is difficult for the boost type PFC control apparatus 300 according to this comparative example to downsize the boost coil in the future. In addition, as a current problem, a considerably large switching loss is generated, so that it is not suitable as a step-up PFC control device in the current continuous mode. Appropriate conditions for using this boosting method must be based on the premise of so-called soft switching in which the switching element performs switching operation when no current flows. That is, in this boosting method, there is a limitation on the PFC control method.

  As described above, in the step-up PFC control device 300 according to this comparative example, the parasitic diode of the switching element is not only involved in the increase in power consumption due to the forward voltage VF of the parasitic diode itself, but also different switching There is a problem of affecting the switching loss of the element.

(Summary)
As described in the embodiments and the modifications thereof, the step-up PFC control apparatus according to the present invention includes an AC / DC converter unit that rectifies and boosts an input single-phase AC voltage, and the AC / DC converter. The unit includes a plurality of bridge-connected AC switch units, a boosting coil that stores magnetic energy corresponding to a bidirectional current that flows in response to application of a single-phase AC voltage, and magnetic energy stored in the boosting coil. The smoothing capacitor that stores the corresponding charge and the current path of the bidirectional current can be switched by controlling the on / off state of multiple AC switch units, and the magnetic energy is stored in the boosting coil and the charge is stored in the smoothing capacitor. Each of the plurality of AC switch sections has a gate terminal, a drain terminal, and a source terminal. (1) When the gate / source voltage, which is the differential voltage of the gate terminal voltage with respect to the source terminal voltage, is higher than the threshold voltage, depending on the polarity of the drain / source voltage, or from the drain terminal to the source terminal, or the source terminal FET characteristics and reverse FET characteristics that allow current to flow from the drain terminal to the drain terminal. (2) When the gate-source voltage is less than the above threshold voltage, the current from the drain terminal to the source terminal is cut off. However, when the gate terminal voltage becomes equal to or higher than the threshold voltage with respect to the drain terminal voltage, the bidirectional switching element has a reverse conduction characteristic that allows current to flow from the source terminal to the drain terminal.

  According to this configuration, no diode is required in the AC / DC converter unit, and there is no parasitic diode in the switching element, so that power consumption due to a voltage drop in the forward voltage of the diode can be eliminated. In addition, since there is no switching loss due to the recovery current of the parasitic diode of the switching element, it is possible to realize a step-up PFC control device that significantly reduces power consumption.

  Further, the boosting PFC control unit that controls the bidirectional current path can use the boosting PFC control unit of the conventional boosting PFC control device as it is, and the control operation as the boosting PFC control device is the conventional one. It is almost the same as the thing.

  In addition, when the current flowing between the AC / DC converter unit and the input AC power supply is substantially zero, the drive logic circuit receives a control signal from a hysteresis comparator that detects the absence of the current, Control is performed so that the gate-source voltage of at least one of the AC switch units is equal to or lower than the threshold voltage.

  As a result, when the current flowing through the booster coil is substantially zero, it is possible to prevent the current from flowing backward from the output side of the booster type PFC control device to the input AC power supply even when the magnetic energy of the booster coil is lost. Become.

  Further, according to the first modification of the embodiment, at least two of the AC switch units are two bidirectional switching elements in which source terminals or drain terminals are connected in series. Composed.

  According to this, similarly to the above-described step-up PFC control device, it is possible to reduce power consumption while maintaining the rectifying step-up operation of the conventional step-up PFC control device. Furthermore, a step-up PFC control device having high input voltage resistance can be realized. In addition, the step-up PFC control device can completely disconnect the load connected between the output terminal and the GND terminal of the PFC control device from the input AC voltage source.

  Further, in the step-up PFC control device according to the embodiment and the modification thereof, the bidirectional switching element is formed on the stacked body including the nitride semiconductor layer formed on the semiconductor substrate, and on the stacked body. And a drain terminal and a source terminal formed on both sides of the gate terminal. This bidirectional switching element is generally known as a heterojunction field effect transistor using a gallium nitride semiconductor, and is called a GaN transistor.

  The GaN transistor has FET characteristics and reverse FET characteristics when the gate-source voltage is higher than a certain threshold voltage, and has reverse conduction characteristics when the gate-source voltage is equal to or lower than the threshold voltage. In addition, it is also a bidirectional switching element having a high breakdown voltage characteristic, and is an FET transistor having a very low on-resistance value in FET characteristics and reverse FET characteristics. Further, there is no minority carrier effect as in a silicon-based semiconductor element, and there is almost no influence of an increase in switching loss due to a recovery current. Therefore, by using a GaN transistor as a bidirectional switching element included in the step-up PFC control device of the present invention, a step-down PFC control device with lower power consumption can be realized.

  As mentioned above, although the boost type PFC control device of the present invention has been described based on the embodiment, the boost type PFC control device according to the present invention is not limited to the above embodiment and its modifications. Other embodiments realized by combining arbitrary components in the embodiment and its modifications, and various modifications conceivable by those skilled in the art without departing from the gist of the present invention to the embodiments and their modifications. Modifications obtained by applying the above and various devices incorporating the step-up PFC control device according to the present invention are also included in the present invention.

  The present invention can be applied to an AC / DC converter that outputs a DC voltage from an AC input, and is particularly useful as an AC / DC converter having a high-input power factor and high-efficiency step-up PFC controller that does not require a rectifier diode bridge. It is.

DESCRIPTION OF SYMBOLS 1 Input AC power source 3 Current detection element 4 Boosting coil 7 Smoothing capacitor 8 Output load 9 Second error amplifier 10 Multiplier circuit 11 Second absolute value circuit 12 PWM comparator 14, 514, 515 Sawtooth wave oscillator 17 Differential amplifier 18 Reference Voltage source 19 First error amplifier 20 Comparator 21 First absolute value circuit 30, 31, 32, 33 AC / DC converter unit 51 Pre-drive circuit 52 Power supply 100, 150, 200, 250, 300, 500 Step-up PFC controller 201 Silicon substrate 202 Buffer layer 203 Stack 204 Undoped gallium nitride layer 205 N-type aluminum gallium nitride layer 206a Ohmic electrode for source terminal 206b Ohmic electrode for drain terminal 208 Gate electrode 209 Control layer 210 Wiring 502 Pass blocking filter 504 magnetic induction means S1, S2, S3, S4 AC switch T1, T2, T3, T4, T5, T6 switching elements D1, D2 diode

Claims (5)

A step-up PFC control device including an AC / DC converter unit that rectifies and boosts an input single-phase AC voltage,
The AC / DC converter unit includes:
A plurality of AC switches connected in a bridge;
A boosting coil that stores magnetic energy according to a bidirectional current that flows in response to the application of the single-phase AC voltage;
A smoothing capacitor for storing a charge corresponding to the magnetic energy stored in the boosting coil;
The current path of the bidirectional current is switched by controlling the on / off states of the plurality of AC switch units, and the accumulation of the magnetic energy in the boosting coil and the charge accumulation in the smoothing capacitor are alternately performed. Drive logic circuit
Each of the plurality of AC switch units is
A gate terminal, a drain terminal, and a source terminal; (1) when the gate-source voltage, which is the gate voltage with respect to the source voltage, is higher than the threshold voltage, the drain terminal depends on the polarity of the drain-source voltage. A current is passed to the source terminal or from the source terminal to the drain terminal; and (2) when the gate-source voltage is equal to or lower than the threshold voltage, the current flowing from the drain terminal to the source terminal is cut off. A step-up PFC control device comprising a switching element that causes a current to flow from the source terminal to the drain terminal when the gate voltage with respect to the drain voltage becomes equal to or higher than the threshold voltage. The plurality of AC switch units are
Comprising the first to fourth switching elements,
A first electrode of the boosting coil is connected to a first AC terminal to which the single-phase AC voltage is applied;
The drain terminal of the first switching element and the drain terminal of the third switching element are connected to the first electrode of the smoothing capacitor,
The source terminal of the second switching element and the source terminal of the fourth switching element are connected to the second electrode of the smoothing capacitor,
The source terminal of the first switching element and the drain terminal of the second switching element are connected to the second electrode of the boosting coil,
The source terminal of the third switching element and the drain terminal of the fourth switching element are connected to a second AC terminal to which the single-phase AC voltage is applied,
The gate terminals of the first to fourth switching elements are connected to respective pre-drive circuits that level-shift control signals output from the drive logic circuit,
The drive logic circuit controls so that the first switching element and the second switching element or the third switching element and the fourth switching element are not turned on at the same time. 2. The step-up PFC control device according to 1. The drive logic circuit further includes at least one of the plurality of AC switch units when the current flowing between the AC power source that outputs the single-phase AC voltage and the AC / DC converter unit is substantially zero. The step-up PFC control device according to claim 1, wherein a gate signal of the switching element is controlled so that a voltage between the gate and the source of the switching element constituting the AC switch unit is equal to or lower than the threshold voltage. At least two AC switch units among the plurality of AC switch units are:
The step-up PFC control device according to claim 1, comprising two switching elements in which source terminals or drain terminals are connected in series. The switching element is
A laminate composed of a plurality of nitride semiconductor layers formed on a semiconductor substrate;
The gate terminal formed on the laminate;
The step-up PFC control device according to claim 1, further comprising: the drain terminal and the source terminal formed on both sides of the gate terminal.

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