WO2018059330A1 - 数据传输方法及通信设备 - Google Patents

数据传输方法及通信设备 Download PDF

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Publication number
WO2018059330A1
WO2018059330A1 PCT/CN2017/103012 CN2017103012W WO2018059330A1 WO 2018059330 A1 WO2018059330 A1 WO 2018059330A1 CN 2017103012 W CN2017103012 W CN 2017103012W WO 2018059330 A1 WO2018059330 A1 WO 2018059330A1
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Prior art keywords
signal sequence
interpolation
signal
time domain
pilot
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PCT/CN2017/103012
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English (en)
French (fr)
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董朋朋
丁梦颖
胡远洲
王宗杰
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华为技术有限公司
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Priority to EP17854780.8A priority Critical patent/EP3515028A4/en
Publication of WO2018059330A1 publication Critical patent/WO2018059330A1/zh
Priority to US16/371,801 priority patent/US10924315B2/en

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2649—Demodulators
    • H04L27/26524—Fast Fourier transform [FFT] or discrete Fourier transform [DFT] demodulators in combination with other circuits for demodulation
    • H04L27/26526—Fast Fourier transform [FFT] or discrete Fourier transform [DFT] demodulators in combination with other circuits for demodulation with inverse FFT [IFFT] or inverse DFT [IDFT] demodulators, e.g. standard single-carrier frequency-division multiple access [SC-FDMA] receiver or DFT spread orthogonal frequency division multiplexing [DFT-SOFDM]
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2668—Details of algorithms
    • H04L27/2673—Details of algorithms characterised by synchronisation parameters
    • H04L27/2676—Blind, i.e. without using known symbols
    • H04L27/2678—Blind, i.e. without using known symbols using cyclostationarities, e.g. cyclic prefix or postfix
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2602—Signal structure
    • H04L27/2605—Symbol extensions, e.g. Zero Tail, Unique Word [UW]
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2626—Arrangements specific to the transmitter only
    • H04L27/2627—Modulators
    • H04L27/2634—Inverse fast Fourier transform [IFFT] or inverse discrete Fourier transform [IDFT] modulators in combination with other circuits for modulation
    • H04L27/2636—Inverse fast Fourier transform [IFFT] or inverse discrete Fourier transform [IDFT] modulators in combination with other circuits for modulation with FFT or DFT modulators, e.g. standard single-carrier frequency-division multiple access [SC-FDMA] transmitter or DFT spread orthogonal frequency division multiplexing [DFT-SOFDM]
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2668—Details of algorithms
    • H04L27/2669—Details of algorithms characterised by the domain of operation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2668—Details of algorithms
    • H04L27/2673—Details of algorithms characterised by synchronisation parameters
    • H04L27/2675—Pilot or known symbols

Definitions

  • the present invention relates to the field of communications technologies, and in particular, to a data transmission method and a communication device.
  • Orthogonal Frequency Division Multiplexing (OFDM)-based waveforms are recognized as baseline waveforms, and their advantages are low-complexity frequency-domain equalization algorithms using multiple inputs.
  • Multiple Input Multiple Output (MIMO) technology for flexible multi-layer data space multiplexing.
  • DL CoMP Downlink Coordinated Multipoint Transmission
  • the existing LTE protocol adopts the CP method in order to combat channel delay spread.
  • the extended CP is a cell-level configuration, so the throughput of all users in the entire cell is affected, and the throughput is directly reduced by 13%.
  • the existing protocol only extends the CP format, which is a long CP format, so it is not well adapted to time-varying channel delay variations, or is too wasteful of resources. Or ISI will be introduced.
  • the definition of the CP of the existing protocol cannot flexibly meet the requirements of various scenarios in 5G, and an OFDM waveform scheme that can adapt to various delay deviation scenarios is needed.
  • the embodiment of the invention provides a data transmission method and a communication device, so as to solve the problem that the definition of the CP in the prior art cannot flexibly meet the requirements of various scenarios in the 5G.
  • an embodiment of the present invention provides a data transmission method, where the method includes: performing an interpolation operation on a first signal sequence to obtain a second signal sequence; and mapping the second signal sequence to a subcarrier to obtain a second signal sequence on the subcarrier; performing an inverse fast Fourier transform on the second signal sequence on the subcarrier IFFT, obtaining a time domain signal; transmitting the time domain signal.
  • the effect of the OFDM symbol adaptive zero power ZT is embodied in the time domain by the frequency domain interpolation operation, so that the transmitted time domain signal can better resist the delay deviation.
  • the method before performing the interpolation operation on the first signal sequence, further includes: determining an interpolation parameter; performing an interpolation operation on the first signal sequence to obtain a second signal sequence, including: according to the The interpolation parameter performs an interpolation operation on the first signal sequence to obtain the second signal sequence, and the length of the second signal sequence is greater than the length of the first signal sequence.
  • the length of the zero-power ZT can be adjusted by the interpolation parameter, and different interpolation parameters can be combined to flexibly cope with different channel delay changes.
  • the method before the determining the interpolation parameter, further includes: acquiring a maximum delay deviation of a signal of the terminal device, where the signal of the terminal device is a wireless signal sent to the terminal device or the terminal a wireless signal sent by the device, where the maximum delay deviation is a difference between a time when the signal of the terminal device is sent from the transmitting end, and a time when the wireless channel passes the earliest arrival time to the receiving end and the time of reaching the receiving end at the latest; the determining the interpolation parameter includes Determining the interpolation parameter according to a maximum delay deviation of a signal of the terminal device.
  • the embodiment of the invention can take into account multiple possible delay deviations of the terminal device signal, and can better resist the channel delay.
  • the maximum delay deviation includes a delay deviation of data signals transmitted by different transmission points when downlink multi-point transmission occurs.
  • the maximum delay deviation may further include a delay deviation of data transmitted by users having different distances when uplink multi-user asynchronous access is used.
  • the interpolating operation specifically includes: performing an interpolation operation on the first signal sequence to obtain a third signal sequence; performing a first phase rotation on the third signal sequence to obtain a second signal sequence .
  • the interpolating operation specifically includes: performing an inverse discrete Fourier transform IDFT on the first signal sequence to obtain a fourth signal sequence; adding ZH at the head of the fourth signal sequence Zero, and adding ZT zeros at the tail of the fourth signal sequence to obtain a fifth signal sequence, wherein ZH and ZT are both integers greater than zero; performing discrete Fourier transform DFT on the fifth signal sequence Obtaining the second signal sequence, wherein the length of the second signal sequence is equal to the sum of the length of the first signal sequence, ZH and ZT.
  • the embodiment of the present invention can embody the effect of the OFDM symbol adaptive zero power ZT in the time domain by a generalized interpolation operation or a DFT interpolation operation, so that the transmitted time domain signal can better resist the delay deviation.
  • the first signal sequence includes at least one first pilot symbol, and the first pilot symbol is obtained by rotating a second pilot symbol by a second phase, the second guide The frequency symbol is used by the receiving end to perform at least one of channel measurement and channel estimation.
  • the first signal sequence includes at least one first pilot symbol, and the first pilot symbol is obtained by rotating a second pilot symbol by a third phase, the second guide The frequency symbol is used by the receiving end to perform at least one of channel measurement and channel estimation.
  • mapping the second signal sequence to the subcarrier to obtain the second signal sequence on the subcarrier comprises: mapping at least one group of the second signal sequence to the subcarrier, to obtain at least A second sequence of signals on a set of subcarriers.
  • the resource partitioning of a user is discretely distributed in the transmission bandwidth by the interpolation operation based on the blocking mode, so that the target user can obtain better frequency domain diversity effect, and for other purposes.
  • the scheduling of the user can also be more flexible.
  • the first signal sequence includes at least one first pilot symbol and at least one data symbol, wherein the at least one first pilot symbol and the at least one data symbol are in accordance with a first pre- Defining a rule to form the first signal sequence;
  • the first predefined rule is: in the first signal sequence, selecting at least one of every first predefined number of data symbols is a candidate first pilot symbol position Inserting, by the candidate first pilot symbol position, the first pilot symbol, wherein the first predefined number is determined according to the interpolation parameter.
  • Embodiments of the present invention may implement a multi-carrier based interpolation operation, and may insert pilot symbols on certain subcarriers of an OFDM symbol.
  • the position before the pilot symbol interpolation is preset according to the interpolation parameter, so that the number, amplitude and phase of the pilot symbols before and after the interpolation remain unchanged. Therefore, the pilot symbol does not change with respect to the receiving end, that is, the interpolation scheme is adopted for the transmitting end and the interpolation scheme is not used for the transmitting end, and the processing of the receiving pilot is completely the same.
  • the first signal sequence includes at least one zero and at least one data symbol, wherein the at least one zero and the at least one data symbol form the first signal according to a second predefined rule
  • the second predefined rule is: in the first signal sequence, every first predefined number of data symbols is a candidate zero insertion position, and at least one of the candidate zero insertion positions is selected to be inserted into the zero, wherein The first predefined number is determined according to the interpolation parameter.
  • first phase rotation and the second phase rotation mentioned in the embodiments of the present invention are both for keeping the pilot symbols unchanged from the receiving end. That is to say, the interpolation scheme is adopted for the transmitting end and the interpolation scheme is not adopted for the transmitting end, and the processing of the pilot at the receiving end is exactly the same.
  • every second predefined number of subcarriers is a candidate second pilot symbol position, and at least one candidate second pilot symbol position is replaced by a second pilot symbol.
  • a symbol in a second signal sequence on the carrier the second predefined number being determined according to the interpolation parameter, wherein the second predefined number is greater than the first predefined number, the second In the signal sequence, the replacement position of the second pilot symbol is related to a position in the first signal sequence in which the zero is inserted, and the second pilot symbol is used by the receiving end to perform at least one of channel measurement and channel estimation.
  • the pilot symbol in the interpolation process, does not change relative to the receiving end, and the pilot position in the first signal sequence before interpolation may be reset to zero, and finally the pilot symbol is used to replace the corresponding guide in the second signal sequence.
  • the symbol at the frequency position is an alternative to inserting pilot symbol valid data directly in the first signal sequence.
  • the method further includes: sending information related to the interpolation parameter, where the information related to the interpolation parameter is used by the receiving end to determine the interpolation parameter.
  • the data transmission method provided by the foregoing first aspect may further include the following possible implementation manners:
  • the interpolation parameter is a preset fixed value.
  • the interpolation parameters of the pilot symbols can be fixed to avoid different situations in which the intervals of the pilot insertable positions are different under different interpolation parameters caused by changing the interpolation parameters according to the channel delay variation.
  • different interpolation parameters can be changed.
  • the interpolation parameter is one of a plurality of interpolation parameters; when the first signal sequence includes at least one first pilot Symbol or At least one zero time, the first predetermined number is determined according to the plurality of interpolation parameters.
  • the embodiment of the present invention may adopt a pilot symbol design of a variable interpolation parameter, and determine a position where a pilot symbol is inserted according to an optional plurality of interpolation parameters in a certain delay range, so that a pilot symbol insertion interval is performed under multiple interpolation parameters. the same.
  • the embodiment of the present invention implements a pattern design of multiple pilots similar to the 5G standard.
  • For the DMRS pilot two unified pilot patterns can be used to adaptively switch different interpolation parameters, thereby avoiding When different embedding points are used in CoMP, different interpolation parameters are used, and DMRS pilots can maintain orthogonal effects.
  • the first signal sequence further includes at least one fourth pilot symbol, wherein the at least one fourth pilot symbol, the at least one first pilot symbol, and the at least A data symbol forms the first signal sequence in accordance with a third predefined rule.
  • the third predefined rule is that every first predefined number of data symbols is a candidate first pilot symbol position, and at least one candidate first pilot symbol position and its neighbor are selected. Positioning the first pilot symbol and the fourth pilot symbol respectively; selecting adjacent positions of the at least one candidate first pilot symbol position to insert the fourth pilot symbol respectively; wherein The first predefined number is determined according to the interpolation parameter.
  • the pilot is also inserted for the position next to the integer multiple point, and the density of the pilot symbol is enhanced. Avoiding relatively low density of pilot symbols can affect the quality of channel estimation, noise estimation, etc., as well as affecting the final throughput rate. By increasing the density of available pilot locations to support more port ports, or by swapping in the same number of pilot symbols, more data symbols can be transmitted.
  • the fourth pilot symbol is obtained by rotating a third pilot symbol by using a third phase, where the third pilot symbol is used by the receiving end to perform at least channel measurement and channel estimation.
  • the third pilot symbol is used by the receiving end to perform at least channel measurement and channel estimation.
  • the neighboring position of the second predefined number of subcarriers is the candidate third pilot symbol position, and the at least one candidate third pilot is replaced by the third pilot symbol.
  • the symbol in the second signal sequence on the subcarrier where the symbol position is located, the second predefined number is determined according to the interpolation parameter, wherein the second predefined number is greater than the first predefined number,
  • the replacement position of the third pilot symbol is related to a position in the first signal sequence in which the fourth pilot symbol is inserted.
  • the at least one transmitting device when the at least one transmitting device cooperates to transmit data to the receiving end, the at least one transmitting device comprises a serving transmitting device and a cooperative transmitting device, the cooperative transmitting device according to its relative to the serving transmitting device The delay deviation is determined by the interpolation parameter, so that the receiving end performs joint MIMO reception on the signal of the at least one transmitting device.
  • the embodiment of the invention can ensure that a plurality of transmission points with different timings select appropriate interpolation parameters to transmit time domain signals according to respective delay deviations, and can ensure that the receiving end uses the joint MIMO method for receiving, and the processing delay is shorter.
  • the head and tail of the second sequence of signals respectively comprise a plurality of data that are less than a predetermined threshold.
  • the data smaller than the preset data may be approximate zero data.
  • the frequency domain second signal sequence is obtained by performing a generalized interpolation operation on the frequency domain data of the first signal sequence.
  • the head of the second signal sequence in the frequency domain has ZH data of approximately zero, and the tail has ZT data of approximately zero.
  • the first signal sequence is subjected to a DFT interpolation operation, and the first signal sequence is performed.
  • the IDFT transform obtains the first time domain symbol, and adds ZH zeros and ZT zeros to the head and the tail of the first time domain symbol respectively to obtain a second time domain symbol, and performs DFT transformation on the second time domain symbol to obtain a frequency.
  • the second signal sequence of the domain achieves the effect of interpolation in the frequency domain by zero-padding the two sides of the time domain signal.
  • the ratio of the length of the second signal sequence to the length of the first signal sequence is an interpolation rate, and when the interpolation rate is 2, the transmitting the time domain signal further includes:
  • the time domain signal includes a first time domain signal and a second time domain signal, wherein the first time domain signal occupies the same length of time as the second time domain signal; and the first time domain signal and The second time domain signals are respectively intercepted to obtain a third time domain signal and a fourth time domain signal, wherein the second time domain signal is delayed by half a cycle in the time domain compared to the first time domain signal,
  • the length of time occupied by the third time domain signal is half of the length of time of the first time domain signal, and the length of time occupied by the fourth time domain signal is half of the length of time of the second time domain signal;
  • the third time domain signal and the fourth time domain signal are mixed to obtain a fifth time domain signal; and the fifth time domain signal is transmitted, where the half period is the length of time of the first time domain signal half.
  • the first time domain signal and the second time domain signal are respectively intercepted by using an IFFT/2 window function, and the first time domain signal or the second time domain signal header and tail are respectively A plurality of data less than a preset threshold are truncated.
  • the transmitting the time domain signal further includes: the time domain signal includes a sixth time domain signal, a seventh time domain signal, and an eighth time a domain signal and a ninth time domain signal, a length of time occupied by the sixth time domain signal, a length of time occupied by the seventh time domain signal, a length of time occupied by the eighth time domain signal, and the The ninth time domain signal occupies the same length of time; the sixth time domain signal, the seventh time domain signal, the eighth time domain signal and the ninth time domain signal are respectively intercepted to obtain the tenth time domain signal, the eleventh a time domain signal, a twelfth time domain signal, and a thirteenth time domain signal, wherein the seventh time domain signal is delayed by a quarter cycle and the eighth time in the time domain compared to the sixth time domain signal The domain signal is delayed by a quarter cycle in the time domain compared to the seventh time domain signal, and the ninth time domain signal is delayed by a quarter cycle in the time domain compared to
  • the length of time occupied by the tenth time domain signal is four quarters of the length of time of the sixth time domain signal
  • the time length of the eleventh time domain signal is one quarter of the length of time of the seventh time domain signal
  • the time length of the twelfth time domain signal is the eighth time.
  • the time period of the domain signal is one quarter
  • the length of time of the thirteenth time domain signal is one quarter of the length of time of the ninth time domain signal
  • the tenth time domain signal is to be And the eleventh time domain signal, the twelfth time domain signal, and the thirteenth time domain signal are mixed to obtain a fourteenth time domain signal; and the fourteenth time domain signal is transmitted, wherein the quarter period is The sixth time domain signal accounts for a quarter of the length of time.
  • the sixth time domain signal, the seventh time domain signal, the eighth time domain signal, and the ninth time domain signal are respectively intercepted by using an IFFT/4 window function, and the sixth time is The domain signal, the seventh time domain signal, the eighth time domain signal, and the plurality of data of the ninth time domain signal header and tail that are less than a preset threshold are truncated.
  • Embodiments of the present invention can construct a shorter time interval in the time domain to achieve a wider subcarrier spacing effect. At the same time, there is no mutual ICI interference between different services. It can save the frequency domain protection interval and effectively improve the throughput.
  • the interpolation operation includes one or more of the following: discrete Fourier transform DFT interpolation, spline interpolation, first order interpolation, and higher order interpolation.
  • an embodiment of the present invention provides another data transmission method, where the method includes: receiving a time domain signal Performing a fast Fourier transform FFT on the time domain signal to obtain a sixth signal sequence on the subcarrier; demodulating the sixth signal sequence to obtain a seventh signal sequence; and solving the seventh signal sequence Interpolating operation, obtaining an eighth signal sequence, the eighth signal sequence comprising soft information of the data symbol, wherein the length of the eighth signal sequence is smaller than the length of the seventh signal sequence; and decoding the soft information of the data symbol, Get the data symbol.
  • the method before the demodulating the sixth signal sequence, further includes: acquiring a pilot symbol; performing channel estimation according to the pilot symbol to obtain channel related information; and the sixth signal The sequence is demodulated to obtain a seventh signal sequence, comprising: demodulating the sixth signal sequence according to the channel related information to obtain the seventh signal sequence.
  • the method before performing the de-interpolating operation on the seventh signal sequence, the method further includes: determining an interpolation parameter; performing de-interpolation operation on the seventh signal sequence to obtain an eighth signal sequence The method includes: performing an interpolating operation on the seventh signal sequence according to the interpolation parameter to obtain an eighth signal sequence.
  • the de-interpolating operation specifically includes: performing a first phase rotation operation on the seventh signal sequence to obtain a ninth signal sequence; and performing a de-interpolation operation on the ninth signal sequence to obtain The eighth signal sequence.
  • the de-interpolating operation specifically includes: performing an inverse discrete Fourier transform (IDFT) on the seventh signal sequence to obtain a tenth signal sequence; deleting the ZH of the tenth signal sequence header Zero, and deleting ZT zeros at the tail of the tenth signal sequence to obtain an eleventh signal sequence; performing discrete Fourier transform DFT on the eleventh signal sequence to obtain the eighth signal sequence, where The length of the eighth signal sequence is equal to the length of the seventh signal sequence minus the values obtained by ZH and ZT.
  • IDFT inverse discrete Fourier transform
  • the demodulating the sixth signal sequence to obtain the seventh signal sequence includes: demodulating the sixth signal sequence to obtain at least one group of the seventh signal sequence.
  • the method further includes: receiving information related to the interpolation parameter, where the information related to the interpolation parameter is used to determine the interpolation parameter.
  • the seventh signal sequence further includes a pilot symbol, wherein the pilot symbol meets a position of a fourth predefined rule in the seventh signal sequence; the fourth predefined The rule is that the pilot symbol is located every fourth predefined number of positions in at least one of the seventh signal sequences, and the fourth predefined number is determined according to the interpolation parameter.
  • the de-interpolation operation includes one or more of the following: de-discrete Fourier transform DFT interpolation, strip spline interpolation, solution first order interpolation, and solution high order interpolation.
  • an embodiment of the present invention provides a communication device, including: an interpolation unit, configured to perform an interpolation operation on a first signal sequence to obtain a second signal sequence, and a mapping unit, configured to map the second signal sequence Going to the subcarrier, obtaining a second signal sequence on the subcarrier; an IFFT unit, configured to perform an inverse fast Fourier transform IFFT on the second signal sequence on the subcarrier to obtain a time domain signal; and a transmitting unit, configured to transmit The time domain signal.
  • the method further includes: a determining unit, configured to determine an interpolation parameter, where the interpolation unit is configured to perform an interpolation operation on the first signal sequence according to the interpolation parameter to obtain the second signal a sequence, the length of the second signal sequence being greater than the length of the first signal sequence.
  • the method further includes: an acquiring unit, configured to acquire a maximum delay of the signal of the terminal device Deviation, the signal of the terminal device is a wireless signal sent to the terminal device or a wireless signal sent by the terminal device, and the maximum delay deviation is that the signal of the terminal device is sent from a transmitting end, and passes through a wireless channel.
  • the difference between the time of the earliest arrival of the receiving end and the time of the latest arrival of the receiving end; the determining unit is specifically configured to determine the interpolation parameter according to the maximum delay deviation of the signal of the terminal device.
  • the interpolation unit is configured to perform an interpolation operation on the first signal sequence to obtain a third signal sequence, and perform a first phase rotation on the third signal sequence to obtain a second Signal sequence.
  • the interpolation unit is specifically configured to perform an inverse discrete Fourier transform (IDFT) on the first signal sequence to obtain a fourth signal sequence; and increase the header of the fourth signal sequence.
  • IFT inverse discrete Fourier transform
  • ZH zeros, and adding ZT zeros at the tail of the fourth signal sequence to obtain a fifth signal sequence, wherein ZH and ZT are integers greater than or equal to zero, but at least one of ZH and ZT is a positive integer
  • the fifth signal sequence performs a discrete Fourier transform DFT to obtain the second signal sequence, wherein the length of the second signal sequence is equal to the length of the first signal sequence, the sum of ZH and ZT.
  • the mapping unit is specifically configured to: map at least one set of the second signal sequence to a subcarrier to obtain a second signal sequence on at least one group of subcarriers.
  • the first signal sequence includes at least one first pilot symbol and at least one data symbol, wherein the at least one first pilot symbol and the at least one data symbol are in accordance with a first pre- Defining a rule to form the first signal sequence;
  • the first predefined rule is: in the first signal sequence, selecting at least one of every first predefined number of data symbols is a candidate first pilot symbol position Inserting, by the candidate first pilot symbol position, the first pilot symbol, wherein the first predefined number is determined according to the interpolation parameter.
  • the first signal sequence includes at least one zero and at least one data symbol, wherein the at least one zero and the at least one data symbol form the first signal according to a second predefined rule
  • the second predefined rule is: in the first signal sequence, every first predefined number of data symbols is a candidate zero insertion position, and at least one of the candidate zero insertion positions is selected to be inserted into the zero, wherein The first predefined number is determined according to the interpolation parameter.
  • the first pilot symbol is obtained by rotating a second pilot symbol by using a second phase
  • the second pilot symbol is used by the receiving end to perform at least channel measurement and channel estimation.
  • the mapping unit is further configured to replace the symbol in the second signal sequence on the subcarrier with a second pilot symbol every second predefined number of subcarriers.
  • the second predefined number is determined according to the interpolation parameter, where the second predefined number is greater than the first predefined number, and the second pilot symbol is used by the receiving end to perform channel measurement. And at least one of the channel estimates.
  • the method further includes: a sending unit, configured to send information related to the interpolation parameter, where the information related to the interpolation parameter is used by the receiving end to determine the interpolation parameter.
  • the embodiment of the present invention provides a communication device, including: a receiving unit, configured to receive a time domain signal; and an FFT unit, configured to perform fast Fourier transform FFT on the time domain signal to obtain a subcarrier. a sixth signal sequence; a demodulation unit configured to demodulate the sixth signal sequence to obtain a seventh signal sequence; and a de-interpolation unit configured to perform an interpolation operation on the seventh signal sequence to obtain an eighth signal Sequence, the eighth letter
  • the sequence of numbers includes soft information of the data symbols, wherein the length of the eighth signal sequence is smaller than the length of the seventh signal sequence, and the decoding unit is configured to decode the soft information of the data symbols to obtain data symbols.
  • the method further includes: an acquiring unit, configured to acquire a pilot symbol; a channel estimating unit, configured to perform channel estimation according to the pilot symbol, to obtain channel related information; And demodulating the sixth signal sequence according to the channel related information to obtain the seventh signal sequence.
  • the method further includes: a determining unit, configured to determine an interpolation parameter, where the de-interpolating unit is configured to perform an interpolating operation on the seventh signal sequence according to the interpolation parameter to obtain an eighth signal sequence.
  • the de-interpolating operation specifically includes: performing a first phase rotation operation on the seventh signal sequence to obtain a ninth signal sequence; and performing a de-interpolation operation on the ninth signal sequence to obtain The eighth signal sequence.
  • the de-interpolating unit is specifically configured to perform an inverse discrete Fourier transform IDFT on the seventh signal sequence to obtain a tenth signal sequence, and delete the tenth signal sequence header. ZH zeros, and deleting ZT zeros at the tail of the tenth signal sequence to obtain an eleventh signal sequence; performing discrete Fourier transform DFT on the eleventh signal sequence to obtain the eighth signal sequence, wherein The length of the eighth signal sequence is equal to the length of the seventh signal sequence minus the values obtained by ZH and ZT.
  • the de-interpolating unit is specifically configured to: demodulate the sixth signal sequence to obtain at least one set of the seventh signal sequence.
  • the method further includes: a receiving unit, configured to receive information related to the interpolation parameter, where the information related to the interpolation parameter is used by the receiving end to determine the interpolation parameter.
  • the transmitting end may interpolate the modulation symbols mapped to the frequency domain, and embody the effect of the OFDM symbol adaptive zero power ZT in the time domain, so that the transmission is performed.
  • the time domain signal is better able to counter the delay skew.
  • FIG. 1 is a structural diagram of a communication system according to an embodiment of the present invention
  • FIG. 2 is a schematic flowchart of a data transmission method according to an embodiment of the present invention.
  • FIG. 3 is a schematic diagram of a data transmission method and apparatus based on frequency domain generalized interpolation according to an embodiment of the present invention
  • FIG. 4 is a schematic diagram of an interpolation matrix according to an embodiment of the present invention.
  • FIG. 5 is a schematic diagram of a data transmission method and apparatus based on frequency domain DFT interpolation according to an embodiment of the present invention
  • FIG. 7 is an equivalent data transmission method and apparatus based on frequency domain DFT interpolation according to an embodiment of the present invention.
  • FIG. 9 is a flowchart of another data transmission method according to an embodiment of the present invention.
  • 11 is a data transmission method and apparatus based on frequency domain DFT interpolation according to an embodiment of the present invention.
  • FIG. 12 is a schematic diagram of a pilot design pattern of a CSI-RS according to an embodiment of the present invention.
  • FIG. 13 is a schematic diagram of a pilot design of a DMRS according to an embodiment of the present invention.
  • FIG. 14 is a schematic diagram of a pilot design of another DMRS according to an embodiment of the present invention.
  • 16 is a data transmission method and apparatus based on a frequency domain DFT interpolation pilot enhancement scheme according to an embodiment of the present invention
  • FIG. 17 is a schematic diagram of an enhanced pilot design of a DMRS according to an embodiment of the present invention.
  • FIG. 18 is a schematic diagram of another enhanced pilot design of a DMRS according to an embodiment of the present invention.
  • FIG. 19 is a schematic diagram of a method for asynchronous data transmission applied to multiple transmission points according to an embodiment of the present invention.
  • FIG. 20 is a schematic diagram of a data transmission method for a shorter time interval according to an embodiment of the present invention.
  • FIG. 21 is a structural diagram of a communication device according to an embodiment of the present invention.
  • FIG. 22 is a schematic structural diagram of another communication device according to an embodiment of the present invention.
  • FIG. 23 is a structural diagram of another communication device according to an embodiment of the present invention.
  • FIG. 24 is a structural diagram of still another communication device according to an embodiment of the present invention.
  • the network architecture and the service scenario described in the embodiments of the present invention are used to more clearly illustrate the technical solutions of the embodiments of the present invention, and do not constitute a limitation of the technical solutions provided by the embodiments of the present invention.
  • the technical solutions provided by the embodiments of the present invention are equally applicable to similar technical problems.
  • the technology described in the embodiments of the present invention may be applied to a subsequent evolution system of the LTE system, such as a fifth generation 5G system.
  • a subsequent evolution system of the LTE system such as a fifth generation 5G system.
  • the terms “network” and “system” are often used interchangeably, but those skilled in the art can understand the meaning thereof.
  • FIG. 1 is a structural diagram of a communication system according to an embodiment of the present invention. As shown in FIG. 1, a plurality of transmitting devices and a plurality of receiving devices are included. Uplink or downlink data transmission is performed between a plurality of transmitting devices and receiving devices.
  • the transmitting apparatus and the receiving apparatus in the embodiments of the present invention may be any one of a transmitting end device and a receiving device that performs data transmission in a wireless manner.
  • the transmitting device and the receiving device may be any device having a wireless transceiving function, including but not limited to: a base station NodeB, an evolved base station eNodeB, a base station in a fifth generation (5th) communication system, and a WiFi system.
  • the terminal device may also be referred to as a terminal terminal, a mobile station (MS), a mobile terminal (MT), etc., and the terminal device may be connected to one or more via a radio access network (RAN).
  • the core network communicates, and the terminal device can also directly communicate wirelessly with other terminal devices.
  • Embodiments of the present invention can be applied to base station to terminal equipment downlink data transmission, terminal equipment to base station uplink data transmission, device to device (D2D) data transmission, and none in a wireless communication system.
  • the application scenario is not limited in the embodiment of the present invention.
  • the application scenario of the embodiment of the present invention is described by using the terminal device 1 to the terminal device 3 and the base station 1 and the base station 2 as an example.
  • the base station 1 and the base station 2 perform downlink coordinated multipoint transmission on the terminal device 1.
  • the base station 1 and the base station 2 use different clock sources, there is a certain delay variation between the signals transmitted by the base station 1 and the base station 2.
  • two devices communicate with base station 1.
  • the wireless environment in which the terminal device 1 and the terminal device 2 are located is very different, and the maximum delay deviation (delay extension) of the signals of the terminal device 1 and the terminal device 2 is greatly different.
  • the embodiment of the invention provides a data transmission method, which configures a certain interpolation parameter for a signal sent by a base station or a terminal device according to a delay of a signal of the terminal device, and forms a zero-power tail with different lengths, thereby overcoming the above time. Delay deviation.
  • the signal of the terminal device is a wireless signal sent to the terminal device or a wireless signal sent by the terminal device.
  • the delay variation of the signal of the terminal device may also include a delay deviation caused by signal propagation of different transmitting ends with respect to the receiving end, and a delay spread of the wireless channel itself.
  • the transmitting end can configure the interpolation parameters in an adaptive manner. For example, the transmitting end configures corresponding interpolation parameters according to the obtained maximum channel delay deviation. Or when the transmitting end initially accesses the wireless network, the transmitting end adopts a certain interpolation parameter to configure a long zero power tail, which can resist a large delay deviation. In the subsequent process, the transmitting end can reconfigure the interpolation parameters according to the obtained maximum channel delay deviation, thereby ensuring data transmission efficiency.
  • the receiving end measures the maximum delay deviation of the channel at the current time, and feeds back the quantized delay deviation to the transmitting end through the uplink control message. Or the transmitter directly measures to obtain the channel maximum delay deviation.
  • the transmitting end selects an appropriate interpolation parameter according to the obtained maximum delay deviation for ZT-OFDM modulation.
  • the transmitting end sends the selected interpolation parameter to the receiving end through the control message, so as to ensure that the receiving end performs the correct solution interpolation operation.
  • the base station and the terminal device perform data transmission based on the OFDM symbol
  • the data transmission method provided by the embodiment of the present invention can be understood as an adaptive ZT OFDM symbol transmission method.
  • FIG. 2 is a schematic flowchart of a data transmission method according to an embodiment of the present invention.
  • the main body is a transmitting device. As shown in FIG. 2, this embodiment includes the following steps:
  • Step S101 The transmitting device performs an interpolation operation on the first signal sequence to obtain a second signal sequence, where the length of the second signal sequence is greater than the length of the first signal sequence.
  • the transmitting device for downlink data transmission from the base station to the terminal device, the transmitting device is a base station; for uplink data transmission from the terminal device to the base station, the transmitting device is a terminal device; for D2D data transmission, the transmitting device is a terminal device; The transmitted data is transmitted by the wireless backhaul node.
  • the method before performing the interpolation operation on the first signal sequence, the method further includes: determining an interpolation parameter; performing an interpolation operation on the first signal sequence to obtain a second signal sequence, including: performing, according to the interpolation parameter The first signal sequence is subjected to an interpolation operation to obtain the second signal sequence.
  • the method further includes: acquiring a maximum delay skew of the signal of the terminal device Poor, the signal of the terminal device is a wireless signal sent to the terminal device or a wireless signal sent by the terminal device, and the maximum delay deviation is that the signal of the terminal device is sent from the transmitting end and passes through the wireless channel.
  • the difference between the time when the receiving device reaches the receiving device and the time when the receiving device arrives at the latest; the determining the interpolation parameter comprises: determining the interpolation parameter according to the maximum delay deviation of the signal of the terminal device.
  • the maximum delay deviation herein may include at least one of the following delay deviations: a delay deviation caused by time synchronization between different transmission points; a signal propagation delay deviation of different transmission points reaching the receiving device; the same transmission point The signal reaches the delay spread of the receiving device.
  • the maximum delay deviation can be obtained by measuring the signal sent to the terminal device by the terminal device, and then reported to the network device; or the signal from the terminal device is measured by the network device to obtain the maximum delay deviation; or For the communication scenario of Device to Device (D2D), the signal from another terminal device is measured by the terminal device to obtain the maximum delay deviation, and then reported to the network device.
  • the network device referred to herein may be a base station.
  • the interpolation operation may specifically include: a typical interpolation algorithm such as discrete Fourier transform DFT interpolation, spline interpolation, first-order interpolation, and high-order interpolation.
  • a typical interpolation algorithm such as discrete Fourier transform DFT interpolation, spline interpolation, first-order interpolation, and high-order interpolation.
  • the interpolating operation specifically includes: performing an inverse discrete Fourier transform (IDFT) on the first signal sequence to obtain a fourth signal sequence; adding ZH zeros to the head of the fourth signal sequence, and Adding ZT zeros to the tail of the fourth signal sequence, to obtain a fifth signal sequence, wherein ZH and ZT are integers greater than zero; performing discrete Fourier transform DFT on the fifth signal sequence to obtain the second a signal sequence, wherein the length of the second signal sequence is equal to the length of the first signal sequence, ZH and the sum of ZT.
  • IDFT inverse discrete Fourier transform
  • the first sequence of signals comprises at least one data symbol.
  • said first signal sequence comprises at least one first pilot symbol, said first pilot symbol being obtained by a second phase rotation of said second pilot symbol, said second pilot symbol being for receiving
  • the apparatus performs at least one of channel measurement and channel estimation.
  • the first signal sequence further comprises at least one data symbol, wherein the at least one first pilot symbol and the at least one data symbol form the first signal sequence according to a first predefined rule;
  • the first predefined rule is: in the first signal sequence, every first predefined number of data symbols is a candidate first pilot symbol position, and at least one candidate first pilot symbol position insertion station is selected.
  • the first pilot symbol is described, wherein the first predefined number is determined according to the interpolation parameter.
  • Step S102 mapping the second signal sequence to a subcarrier to obtain a second signal sequence on the subcarrier.
  • mapping at least one of said second signal sequences to subcarriers results in a second sequence of signals on at least one set of subcarriers.
  • Step S103 performing an inverse fast Fourier transform IFFT on the second signal sequence on the subcarrier to obtain a time domain signal.
  • the operations of layer mapping, pre-coding, and the like are also included, and the related art may be referred to, and details are not described herein.
  • the time domain signal may be referred to as an OFDM symbol, and the frequency resource occupied by the OFDM symbol is a system bandwidth of the cell.
  • the interpolated and time-domain signals after IFFT can be added with ZP or CP to make the OFDM symbols reach a predefined length of time and further eliminate inter-symbol interference.
  • Step S104 transmitting the time domain signal.
  • the time domain signal is transmitted. Or transmit a time domain signal with ZP or CP.
  • information related to the interpolation parameter is transmitted, and the information related to the interpolation parameter is used by the receiving device to determine the interpolation parameter.
  • FIG. 3 is a schematic diagram of a data transmission method and apparatus based on frequency domain generalized interpolation according to an embodiment of the present invention.
  • the interpolation parameter in this embodiment be (m, n). If the resource allocated by the system for a certain user is N subcarriers, the scheme needs to generate M symbols before the interpolation, and M symbols are interpolated in the frequency domain by the fractional interpolation filter of (m, n). Symbol, at the same time T is the number of interpolation points corresponding to integer multiples. Then the embodiment mainly includes the following steps:
  • Step 301 The user data bit is subjected to channel coding, rate matching, scrambling, and modulation to generate at least one data symbol.
  • the at least one data symbol may be: M-L QAM symbols.
  • Step 302 Generate at least one second pilot symbol according to the cell number, the frame number, and the like.
  • the second pilot symbol is used by the receiving device to perform at least one of channel measurement and channel estimation.
  • the at least one first pilot symbol is obtained by ensuring that the receiving device can transparently receive the pilot symbols and performing the second phase rotation on the at least one second pilot symbol.
  • the at least one second pilot symbol may be L (L ⁇ T) pilot symbols
  • the second phase rotation may be represented by multiplying a phase rotation factor Where i ⁇ [0, T), i is determined according to which first pilot symbol positions of the first signal sequence in step 303 are specifically inserted according to the first pilot symbol, and ZH is the front end of N data corresponding to the time domain after the interpolation process The number that is approximately zero.
  • the transparent receiving pilot symbol of the receiving device refers to whether the transmitting end uses the interpolation scheme to transparently receive the pilot by the receiving apparatus, that is, the interpolation scheme is adopted for the transmitting end and the interpolation scheme is not adopted by the transmitting end. In this scenario, the processing of the receiving device pilots is exactly the same.
  • Step 303 The at least one first pilot symbol and the at least one data symbol are combined into the first signal sequence according to a first predefined rule.
  • the first predefined rule is: in the first signal sequence, every first predefined number of data symbols is a candidate first pilot symbol position, and at least one candidate first pilot is selected. The symbol position is inserted into the first pilot symbol, wherein the first predefined number is determined according to the interpolation parameter.
  • the first predefined number is m.
  • the at least one data symbol is first serial-converted (Serial/Parallel, S/P) with the first pilot symbol.
  • the S/P-converted at least one data symbol is interleaved with the first pilot symbol into a first signal sequence of M length.
  • the fractional multiple of n) corresponds to the position of the i*n of the sequence of N lengths, and the value remains unchanged.
  • the number of first pilot symbols is less than or equal to T, and the integer multiple interpolation point is the candidate first pilot symbol position.
  • Step 304 Perform an interpolation operation on the first signal sequence to obtain a third signal sequence.
  • the first signal sequence of M length is subjected to a fractional interpolation filter of (m, n), and an interpolated third signal sequence of length N is output.
  • the interpolation filter can be various typical interpolation algorithms, such as DFT interpolation, spline interpolation, first order Interpolation, high-order interpolation, etc.
  • the operation of the interpolation filter provided by the embodiment of the present invention is equivalent to multiplying a matrix in front of the unit matrix of the multi-carrier modulation of the existing LTE system. Specifically, since the multicarrier is directly mapped, the precoding matrix can be considered as an identity matrix.
  • the matrix corresponding to the interpolation operation according to the embodiment of the present invention can be referred to FIG. 4.
  • the M data before the interpolation is arranged into a matrix of 20 ⁇ 1, and the 20 ⁇ 1 matrix corresponding to the M data is multiplied by the 24 ⁇ 20 interpolation matrix to obtain a 24 ⁇ 1 matrix corresponding to N data. .
  • the elements of the 1st to 5th rows are respectively multiplied by the 20 ⁇ 1 matrix composed of M data symbols to obtain the corresponding 6 data.
  • the 6 data is only related to the first 5 data of the M data. And so on. If in the M data, the position of the first pilot symbol is corresponding to the element in the interpolation matrix is always 1, the amplitude of the pilot symbol before and after the interpolation does not change.
  • FIG. 4 only shows one possible way of the interpolation matrix.
  • the position of a certain value on both sides of the diagonal line may include not only the case shown in FIG. 4, but also the data having a certain value on both sides of the diagonal line may further include Other cases.
  • the size of the matrix and the minimum cyclic unit depend on the interpolation parameters (m, n), and the values of M and N.
  • the first signal sequence of the M length before interpolation can be multiplied by the amplification factor.
  • Step 305 Perform a first phase rotation on the third signal sequence to obtain a second signal sequence.
  • the third signal sequence has a length of N and the second signal sequence has a length of N. That is, the second signal sequence includes N symbols.
  • the third phase sequence after the interpolation is rotated in the first phase to obtain a second signal sequence.
  • ZH and ZT are integers greater than or equal to zero, but at least one of ZH and ZT is a positive integer.
  • the interpolation operation can be intuitively understood as generating ZH+ZT approximate zero data based on the first signal sequence composed of M data.
  • first phase rotation and the second phase rotation cooperate to achieve the same amplitude and phase of the pilot symbols and the second pilot symbols in the second signal sequence after interpolation.
  • the receiving device may identify the second pilot symbol generated by the transmitting end according to the cell number, the frame number, and the like, so the first phase rotation and the second phase rotation make the second pilot symbol transparent to the receiving device.
  • Figure 3 shows the generalized interpolation method.
  • the second pilot symbol can be made transparent with respect to the receiving device according to specific needs, such as the first phase rotation, the second phase rotation, or the other phase rotation.
  • Step 306a mapping the second signal sequence to a subcarrier to obtain a second signal sequence on the subcarrier.
  • the second signal sequence corresponding to the N symbols after the first phase rotation is continuously mapped to the N subcarriers in the frequency domain.
  • Step 306b Perform an inverse fast Fourier transform IFFT on the second signal sequence on the subcarrier to obtain a time domain signal.
  • Step 307 performing a ZP or CP operation on the time domain signal that is finally subjected to Parallel/Serail (P/S), wherein adding ZP is to add N zp zero values after the time domain signal, and adding CP
  • P/S Parallel/Serail
  • adding ZP is to add N zp zero values after the time domain signal
  • adding CP The operation is to copy the last N cp values of the time domain signal to the forefront of the time domain signal of the string, and the CP operation needs to be satisfied.
  • FFTSize is the FFT size.
  • the data transmission method shown in FIG. 2 above can also be implemented by other interpolation methods.
  • the DFT fast algorithm has a mature chip implementation.
  • the embodiment of the present invention further provides a data transmission method and apparatus based on frequency domain DFT interpolation. As shown in FIG. 5, the method includes steps 501 to 506.
  • the interpolation parameter and the system allocate resources to the user, and the number of symbols to be generated before the interpolation can be referred to the description in FIG. 3 .
  • Step 501, Step 503, Step 505a, Step 505b, and Step 506 can also refer to the descriptions in Step 301, Step 303, Step 306a, Step 306b, and Step 307, respectively. To simplify the description, the details are not described below.
  • Step 502 Generate at least one second pilot symbol according to the cell number, the frame number, and the like.
  • the at least one first pilot symbol is obtained by ensuring that the receiving end can transparently receive the pilot symbol and performing the third phase rotation on the at least one second pilot symbol.
  • the at least one second pilot symbol may be L (L ⁇ T) pilot symbols
  • the third phase rotation may be represented by multiplying a phase rotation factor Where i ⁇ [0, T), i is determined according to which first pilot symbol positions of the first signal sequence in step 503 are specifically inserted in the first pilot symbol.
  • Step 504 performing an inverse discrete Fourier transform (IDFT) on the first signal sequence to obtain a fourth signal sequence; adding ZH zeros to the head of the fourth signal sequence, and at the tail of the fourth signal sequence Adding ZT zeros, obtaining a fifth signal sequence, wherein ZH and ZT are integers greater than or equal to zero, but at least one of ZH and ZT is a positive integer; performing discrete Fourier transform DFT on the fifth signal sequence, The second signal sequence is obtained, wherein the length of the second signal sequence is equal to the sum of the length of the first signal sequence, ZH and ZT.
  • IDFT inverse discrete Fourier transform
  • ZH when ZH is equal to zero and ZT is greater than zero, it is equivalent to adding ZT zeros at the tail of the fourth signal sequence to obtain a fifth signal sequence; when ZH is greater than zero, ZT is equal to zero, equivalent to fourth The head of the signal sequence is increased by ZH zeros to obtain a fifth signal sequence; when both ZH and ZT are greater than zero, it is equivalent to adding ZH zeros at the head of the fourth signal sequence and increasing at the tail of the fourth signal sequence. ZT zeros, resulting in a fifth signal sequence.
  • the first signal sequence of M length is multiplied by the amplification factor before interpolation.
  • the fourth signal sequence is a time domain sequence obtained by IDFT conversion of the M point to the time domain. Then, ZH zeros are added to the time domain sequence header of the M point, ZT zeros are added to the tail, and the time domain sequence length of the M point is extended to N to obtain a fifth signal sequence. Finally, the fifth signal sequence is transformed into the frequency domain sequence by N-point DFT to obtain a second signal. sequence.
  • the first signal sequence when L is equal to zero, the first signal sequence only includes data symbols; when L is equal to M, then the first signal sequence includes only the first pilot symbol; when L is greater than zero and less than M, then A sequence of signals includes at least one data symbol and at least one first pilot symbol.
  • the drawings and embodiments of the present invention are described by taking only the first signal sequence including at least one first pilot symbol and at least one data symbol as an example, but the method and the processing flow of the embodiment of the present invention are equally applicable to the first signal sequence.
  • the data transmission method provided by the embodiment of the present invention utilizes the effect of adding zeros to the two sides of the signal in the frequency domain interpolation equivalent to the time domain, that is, performing interpolation in one transform domain and obtaining zero signal addition effect in another transform domain.
  • the data transmission method provided by the embodiment of the present invention can be applied to an OFDM system.
  • the scheme against the channel delay deviation can also be applied to other systems. It should be understood by those skilled in the art that other equivalents that are similar to the embodiments of the present invention are within the scope of the embodiments of the present invention.
  • the data transmission method provided by the embodiment of the present invention may also be referred to as an adaptive ZT waveform generation method based on an OFDM system.
  • the transmitting end according to the obtained maximum delay of the channel, performs a certain ratio of fractional interpolation filtering operation on the modulation symbols mapped to the frequency domain, and achieves approximation of the ZH data and the tail ZT data of the time domain signal sequence. Zero power effect, thus achieving the goal of reducing ISI.
  • the transmitting end can also be implemented by another equivalent manner.
  • FIG. 6 and FIG. 7 Two other specific examples of the interpolation operation provided by the embodiment of the present invention are described below by taking FIG. 6 and FIG. 7 as examples.
  • FIG. 6 is an equivalent data transmission method and apparatus based on frequency domain generalized interpolation according to an embodiment of the present invention.
  • the first signal sequence comprises at least one zero and at least one data symbol, wherein the at least one zero and the at least one data symbol form the first signal sequence according to a second predefined rule
  • the second predefined rule is: in the first signal sequence, every first predefined number of data symbols is a candidate zero insertion position, and at least one candidate zero insertion position is selected to be inserted into the zero, wherein
  • the first predefined number is determined according to the interpolation parameter.
  • the first predefined number may be equal to m-1.
  • step 601 a value of zero is inserted in the position of the i**m of the pilot symbol that needs to be inserted before the interpolation, so that the value of the position corresponding to the integer multiple of the corresponding i*n is also zero after the interpolation.
  • mapping the at least one set of the second signal sequence to the subcarrier to obtain the second signal sequence on the at least one set of subcarriers further includes: every second predefined number of subcarriers being candidate a second pilot symbol position, wherein the second pilot symbol is used to replace a symbol in a second signal sequence on a subcarrier where the at least one candidate second pilot symbol position is located, the second predefined number according to the The interpolation parameter is determined, wherein the second predefined number is greater than the first predefined number, and the second signal sequence, the replacement position of the second pilot symbol is in the first signal sequence Inserting a position of the zero correlation, the second pilot symbol is used by the receiving end to perform at least one of channel measurement and channel estimation.
  • the second predefined number may be equal to n-1.
  • the second pilot symbol can be directly inserted at the position of the i**n after the interpolation.
  • FIG. 7 is an equivalent data transmission method and apparatus based on frequency domain DFT interpolation according to an embodiment of the present invention.
  • Step 701 and step 702 can refer to the detailed description in step 601 and step 602.
  • other steps not shown in FIG. 7 can be referred to FIG. 5, and details are not described herein.
  • the interpolation algorithm shown in FIG. 5 or FIG. 7 is based on DFT interpolation, in order to facilitate the adoption of a mature DFT chip, it is necessary to ensure that the interpolation parameters (m, n) satisfy the requirements of the bases 2, 3, and 5. That is, the parameters m and n are multiples of 2, 3, and 5. Also define the ratio The larger the value, the higher the interpolation rate, and the longer the corresponding ZT, the stronger the ability to fight against asynchronous. Conversely, the smaller the value, the lower the interpolation rate, and the shorter the corresponding ZT, the weaker the ability to fight against asynchronous, but the more effective data is transmitted. Therefore, a certain amount of asynchronous range can be quantized.
  • Table 1 is a multi-carrier OFDM interpolation parameter configuration table based on frequency domain DFT interpolation, and a possible interpolation parameter combination is given.
  • Table 1 Corresponding to the table is a scenario in which a normal ZP/CP is added to the time domain signal after frequency domain interpolation. It can be understood that, according to the same method, the difference parameter of the scene in which the time domain signal is not added with ZP/CP can be given, and details are not described herein.
  • the overhead and the maximum delay are divided into 14 OFDM symbols based on 1 ms, and the ZP/CP of each symbol is about 4.7 us.
  • the calculation formula of the overhead Overhead Ratio is:
  • the maximum deviation delay deviation MaxDelay Deviation is calculated as:
  • the data transmission method provided by the embodiment of the present invention further includes: sending information related to the interpolation parameter, where the information related to the interpolation parameter is used by the receiving end to determine the interpolation parameter.
  • the interpolation parameter indication information may be sent to the receiving end through a control channel or a data channel. As shown in Table 1, the indication information may be represented by 3 bits. In addition, the interpolation parameter indication information may not be transmitted, and the receiving end performs blind detection based on the difference of the pattern of the pilots in different interpolation configurations.
  • the performance of the adaptive ZT-OFDM scheme is compared with the performance of the conventional Normal CP/Extended CP adaptive switching.
  • the normal CP exceeds 4 us, the delay deviation cannot work, and the extended CP exceeds 16 us. jobs.
  • the adaptive ZT-OFDM scheme provided by the embodiment of the present invention can work in the range of 0 s to 24 s delay deviation, and configuring appropriate interpolation parameters under different delay deviations can make the present invention
  • the working point of the example is stable. And it is better than the Extended CP operating point in the 0us to 12us delay deviation range.
  • the working point of the adaptive ZT-OFDM scheme provided by the embodiment of the present invention has a lower Signal to Noise Ratio (SNR) than the operation of the Extended CP.
  • SNR Signal to Noise Ratio
  • Different delay biases correspond to different optimal interpolation parameters, and their working points can support a longer range than Extended CP.
  • different interpolation parameters divide the delay granularity into smaller, it
  • FIG. 3, FIG. 5, FIG. 6, and FIG. 7 need to maintain the interpolation characteristics, and the resources of the target user are required to be continuous in the frequency domain to ensure the ZT effect in the time domain.
  • the embodiment of the present invention further provides a block interpolation. Ways to get a certain degree of flexible scheduling effect.
  • the mapping the second signal sequence to the subcarrier to obtain the second signal sequence on the subcarrier may include: mapping at least one group of the second signal sequence to the subcarrier to obtain at least one group of subcarriers The second signal sequence on.
  • the signal sequence before the interpolation includes the M0 signal sequence, the M1 signal sequence, and the Mk signal sequence.
  • the M0 signal sequence, the M1 signal sequence, the Mk signal sequence may be subjected to interpolation operations as described above for the interpolation parameters (m, n), respectively, to obtain a N0 signal sequence, an N1 signal sequence, and a Nk signal sequence, respectively.
  • at least one set of second signal sequences may include one or more of a sequence of N0 signals, N1 signal sequences, ... Nk signals.
  • any one or more of the schemes shown in FIG. 3, FIG. 5, FIG. 6, and FIG. 7 may be used, and details are not described herein.
  • FIG. 8 is a schematic diagram of a data transmission method and apparatus based on frequency domain block generalized interpolation according to an embodiment of the present invention.
  • Figure 8 only shows the implementation of the method based on frequency domain generalized interpolation.
  • the difference between block interpolation and blockless interpolation is that the user-modulated symbols are divided into K groups, and the pilot symbols are also divided into K groups, each group
  • the data size can vary and is determined by the scheduler.
  • the data symbols and pilot symbols of each group are subsequently implemented in exactly the manner shown in Figures 3, 5, 6, and 7.
  • the interpolated data is independently mapped to the frequency domain subcarriers according to the resources allocated by the scheduler, and the subcarriers of each group are continuous, but the different groups may be discontinuous.
  • the resource partitioning of one user can be discretely distributed in the transmission bandwidth by means of block interpolation, so that the target user can obtain better frequency domain diversity effect, and the scheduling for other users can be more flexible. support.
  • block difference value can also be combined with other difference schemes.
  • the solution provided by the embodiment of the present invention has more options than the existing Normal CP/Extended CP by configuring a plurality of different interpolation parameters.
  • the solution provided by the embodiment of the present invention can support different optimal interpolation parameters under different delay deviations, and can support a longer delay deviation range than the Extended CP.
  • the solution provided by the embodiment of the present invention has the advantage that the pilot can be interleaved in the frequency domain, so that the multi-antenna multiplexing of the MIMO transmission can be more flexibly supported.
  • one advantage of the multi-carrier OFDM system over the single-carrier S-OFDM system is that it can be flexibly scheduled, that is, a user's resources can be discretely distributed in the transmission bandwidth, so that the target user can obtain better.
  • the frequency domain diversity effect can be more flexibly supported for other users' scheduling.
  • the solution provided by the embodiment of the present invention can adaptively configure interpolation parameters under different delay deviations to improve Data transmission efficiency increases throughput. Further, in the embodiment of the present invention, since the pilot can be interleaved in the frequency domain, the multi-antenna multiplexing of the MIMO transmission can be more flexibly supported.
  • FIG. 9 is a flowchart of another data transmission method according to an embodiment of the present invention. The embodiment includes the following steps:
  • Step S201 the receiving device receives the time domain signal.
  • the method further comprises receiving information related to the interpolation parameter, the information related to the interpolation parameter being used to determine an interpolation parameter used by the de-interpolation operation.
  • Step S202 performing fast Fourier transform FFT on the time domain signal to obtain a sixth signal sequence on the subcarrier.
  • Step S203 demodulating the sixth signal sequence to obtain a seventh signal sequence.
  • the method before demodulating the sixth signal sequence, the method further includes: acquiring pilot symbols; performing channel estimation according to the pilot symbols to obtain channel related information.
  • the channel related information may include information such as a channel factor and interference noise.
  • the received time domain signal is a signal after the transmitted signal is superimposed by channel weighting and interference noise.
  • R be the received signal
  • h be the channel factor
  • S be the transmitted signal
  • n be the interference noise
  • R h * S + n.
  • the pilot symbol in the transmission signal S may be Sp
  • the data symbol is S d .
  • the sixth received signal including a signal sequence Rp received signal and pilot data symbols R d S d of pilot symbols Sp.
  • the receiving end and the transmitting end can negotiate information such as the transmitted and received pilot sequence Sp and the pilot pattern position in advance through control messages.
  • the receiving end estimates channel related information according to the pilot symbol Sp and the received signal Rp of the pilot symbol: a channel factor And interference noise matrix
  • the demodulating the sixth signal sequence to obtain the seventh signal sequence comprises: demodulating the sixth signal sequence according to the channel related information to obtain the seventh signal sequence.
  • the channel factor is estimated using the pilot symbol Sp and the received signal Rp of the pilot symbol And interference noise matrix After that, you can use R d , with Demodulate the corresponding data symbol
  • the demodulated seventh signal sequence includes soft information of the data symbol.
  • the receiving device is further configured according to the channel factor. And interference noise matrix Channel measurement is performed, and the quality information of the channel of the transmitting device is fed back, and the information related to the channel delay is used. In the embodiment of the present invention, this will not be specifically described.
  • Step S204 performing a de-interpolation operation on the seventh signal sequence to obtain an eighth signal sequence, where the eighth signal sequence includes soft information of the data symbol, wherein the length of the eighth signal sequence is smaller than the length of the seventh signal sequence.
  • the method before performing the de-interpolating operation on the seventh signal sequence, the method further includes: determining an interpolation parameter; performing de-interpolation operation on the seventh signal sequence to obtain an eighth signal sequence, including: according to The interpolation parameter performs a de-interpolation operation on the seventh signal sequence to obtain an eighth signal sequence.
  • the solution interpolation operation includes one or more of the following: solution DFT interpolation, solution strip interpolation, solution first order interpolation, and solution high order interpolation.
  • the seventh signal sequence corresponds to the soft information of the second signal sequence of the transmitting device
  • the eighth signal sequence corresponds to the soft information of the first signal sequence of the transmitting device.
  • the seventh signal sequence includes soft information of the data symbols.
  • the eighth signal sequence includes soft information of the corresponding deinterpolated data symbols.
  • the soft information of the data symbols in the eighth signal sequence is extracted for the next processing.
  • Step S205 decoding soft information of the data symbol to obtain a data symbol.
  • the pilot symbols are removed in the eighth signal sequence, and the soft information and channel related information of all the data symbols except the pilot symbols are extracted, and operations such as decoding are performed to obtain data symbols.
  • the data symbol corresponds to user data.
  • FIG. 10 is a schematic diagram of a data transmission method and apparatus based on frequency domain generalized interpolation according to an embodiment of the present invention. As shown in FIG. 10, this embodiment mainly includes the following steps:
  • Step 1001 receiving a time domain signal.
  • the operation of removing the ZP or CP is performed on the received time domain signal.
  • the removal of ZP can be referred to the simplest overlap and add OLA (OverLap and Add) method, that is, the tail N zp signals of a time domain symbol are added back to the frontmost N zp data to form the symbol.
  • OLA OverLap and Add
  • the self-loop feature of the domain signal within the FFTSize size removes the CP and directly deletes the N cp data at the front end.
  • Step 1002 Perform fast Fourier transform FFT on the time domain signal to obtain a sixth signal sequence on the subcarrier.
  • the time domain signal after removing the ZP or the CP is subjected to S/P conversion before the FFT of the time domain signal after the ZP or CP is removed.
  • Step 1003 Demodulate the sixth signal sequence to obtain a seventh signal sequence.
  • the receiving device may perform the pilot position according to a conventional manner, for example, in the manner shown in step S203.
  • Frequency domain channel estimation, noise estimation, interference noise covariance matrix estimation and other measurement operations, and the estimated frequency domain channel related information is used for decoding operation to obtain a seventh signal sequence corresponding to the frequency domain sixth signal sequence.
  • the seventh signal sequence is the decoded N subcarrier soft information:
  • the demodulation coefficient is:
  • s(i) denotes a data symbol
  • r(i) denotes a received signal of a data symbol
  • q(i) denotes an equivalent channel factor
  • n(i) denotes interference noise
  • h H (i) denotes a channel factor
  • Step 1004 Perform a first phase rotation operation on the seventh signal sequence to obtain a ninth signal sequence.
  • the ninth signal sequence is N subcarrier soft information after the first phase rotation is solved.
  • the equivalent channel factor is a real value that characterizes the amplitude, it is not necessary to solve the first phase rotation for the equivalent channel factor.
  • Figure 10 illustrates the generalized solution interpolation method.
  • de-interpolation methods When other de-interpolation methods are used, other dephasing rotations may be performed correspondingly, or de-interpolation may be performed directly without de-phase rotation.
  • the receiving device can select a de-interpolation manner corresponding to the interpolating manner of the transmitting device according to the relevant indication transmitted by the transmitting device.
  • the data processing can also be performed by using a fixed interpolation method and a corresponding de-interpolation method by pre-arranging the transmitting device and the receiving end.
  • Step 1005 Perform a de-interpolation operation on the ninth signal sequence to obtain an eighth signal sequence.
  • the ninth signal sequence is subjected to a (n, m) de-interpolation operation opposite to the origin.
  • the original eighth signal sequence is the soft information ⁇ (i) of the QAM symbol:
  • ⁇ is the regulatory factor
  • the eighth signal sequence includes soft information of the data symbol, wherein the length of the eighth signal sequence is less than the length of the seventh signal sequence.
  • the soft information of the data symbols in the eighth signal sequence is extracted for further processing.
  • Step 1006 Decode the soft information of the data symbol to obtain a data symbol.
  • pilot symbols are removed in the eighth signal sequence, and the soft information and the equivalent channel factor of all the data symbols except the pilot symbols are extracted, and after P/S conversion, the subsequent QAM demodulation, descrambling, and decoding rate are sent. Matching, channel decoding, etc., to obtain data symbols.
  • the signal processing flow of the receiving apparatus based on the generalized interpolation method shown in FIG. 10 corresponds to the signal flow of the transmitting apparatus shown in FIG. 3.
  • FIG. 11 is a schematic diagram of a data transmission method and apparatus based on frequency domain DFT interpolation according to an embodiment of the present invention. As shown in FIG. 11, the steps 1101 to 1106 are mainly included.
  • step 1101, step 1102, step 1103, and step 1105 can refer to 1001, step 1002, step 1003, and step 1006, respectively.
  • step 1104 The following mainly describes the DFT interpolation operation in step 1104:
  • Step 1004 Perform an inverse discrete Fourier transform IDFT on the seventh signal sequence to obtain a tenth signal sequence; delete ZH zeros of the tenth signal sequence header, and delete a ZT at the tail of the tenth signal sequence Zero, obtaining an eleventh signal sequence; performing a discrete Fourier transform DFT on the eleventh signal sequence to obtain the eighth signal sequence, wherein the length of the eighth signal sequence is equal to the seventh signal The length of the sequence is subtracted from the values obtained by ZH and ZT.
  • the (n, m) DFT solution interpolation operation opposite to the origin is performed on the seventh signal sequence to obtain an eighth signal sequence.
  • the seventh signal sequence is first transformed into the time domain by N-point IDFT to obtain a tenth signal sequence; then the ZH values of the head corresponding to the originating end in the tenth signal sequence and the ZT values of the tail are obtained to obtain the M length.
  • the original eighth signal sequence is the soft information ⁇ (i) of the QAM symbol:
  • F M represents the DFT of the M point
  • ⁇ is the regulatory factor
  • the ZH+ZT values of the tail can be directly deleted to obtain a sequence of M length.
  • the eighth signal sequence includes soft information of the data symbol, wherein the length of the eighth signal sequence is less than the length of the seventh signal sequence.
  • the signal processing flow of the receiving apparatus based on the DFT interpolation method shown in FIG. 11 corresponds to the signal flow of the transmitting apparatus shown in FIG. 5.
  • the receiving device operation of the multi-carrier OFDM system based on the frequency domain block generalized interpolation can be performed with reference to FIG. 8 and the signal processing of the receiving device such as the de-interpolation shown in FIG. 10 and FIG. The opposite operation is fine. I will not repeat them here.
  • the interpolation parameter (m, n) of the embodiment of the present invention can theoretically select any parameter, and the selection principle can be determined according to the granularity of different delays and the density requirement of the frequency domain pilot.
  • the interpolation parameter reflects the size of the final OFDM symbol to a certain extent, and the maximum delay information that the OFDM symbol can support.
  • the embodiment of the present invention introduces an interpolation operation in the process of modulating an OFDM symbol, and the interpolation operation is equivalent to introducing two parts of data of ZH and ZT.
  • the embodiment of the present invention can control the lengths of ZT and ZH by selecting appropriate interpolation parameters according to the delay to ensure that valid data in the OFDM symbols received by the receiving device falls within the effective interval, effectively combating channel delay and inter-symbol interference.
  • the embodiment of the present invention may perform interpolation modulation on an OFDM symbol including only data symbols, and may also perform interpolation modulation on an OFDM symbol including only pilot symbols, and may also perform interpolation modulation on an OFDM symbol including at least one data symbol and at least one pilot symbol.
  • the OFDM symbol includes a pilot
  • the data transmission method provided by the embodiment of the present invention can be applied to a multi-carrier system. Because of the direct mapping characteristics of multiple carriers, multiple carriers can insert pilot symbols at any subcarrier position. Wherein, the pilot symbols are frequency domain symbols. In conjunction with the interpolation operation provided by the embodiment of the present invention, the pilot symbol can be inserted at an integer multiple of the data symbol interval, and the amplitude of the data after the interpolation of the integer multiple interpolation position is controlled by the interpolation matrix. In addition, the embodiment of the present invention also needs to perform a phase rotation operation on the pre-interpolation or interpolated pilot symbols, and cancel the phase rotation caused by the interpolation symbols to the pilot symbols to ensure that the pilot symbols are transparent to the receiving end.
  • the integer multiple interpolation point satisfies the characteristic that the signal amplitude does not change before and after the interpolation. Therefore, it is possible to flexibly insert pilot symbols at integer multiple interpolation points, so that measurement operations such as channel estimation and noise estimation can be performed at the receiving apparatus.
  • the pilot symbols are also called Reference Signals (RS).
  • pilot symbols such as Channel State Information RS (CSI-RS), Demodulation Reference Signal (Demodulation RS, DMRS), and Phase Noise RS (PNRS).
  • CSI-RS Channel State Information RS
  • Demodulation RS Demodulation Reference Signal
  • PNRS Phase Noise RS
  • the CSI-RS is a channel state information measurement pilot, and is used for performing beam selection measurement, channel quality indicator (CQI) measurement, rank Rank measurement, and precoding matrix indicator (PMI) measurement, and the like. It also supports measurement of multi-transmitted point cooperation, so CSI-RS requires support for as many beam and antenna port measurements as possible, but can be triggered for a certain period of time.
  • CQI channel quality indicator
  • PMI precoding matrix indicator
  • FIG. 12 is a schematic diagram of a pilot design pattern of a CSI-RS according to an embodiment of the present invention.
  • Figure 12 shows a way to apply higher-order interpolation (2, 3), which can also transmit some immediate control messages while transmitting pilots.
  • Figure 12 shows three sets of pilot positions corresponding to Port0 to Port3 of the three Beams. Due to the time requirement of analog beam switching, different beams can be separated by one symbol, and each port supports four port ports.
  • the specific pilot multiplexing mode can be frequency division multiplexing (Freq terminal equipment ncydivision multiplexing, FDM). Time Division Multiplexing (TDM), code division multiplexing (CDM), and the like.
  • FIG. 12 only shows an example of CSI-RS pilot interpolation parameters. Those skilled in the art can understand that different CSI-RS or other pilot insertion patterns can be designed according to actual frequency domain and time domain pilot density requirements.
  • the DMRS is a signal demodulation pilot, which is generally configured at the user level, is in the signal bandwidth, and represents the joint equivalent channel of the precoding of the transmitting device and the wireless signal, so the number of port ports is less than that of the CSI-RS. many.
  • FIG. 13 is a schematic diagram of a pilot design pattern of a DMRS according to an embodiment of the present invention.
  • the interpolation parameter is a preset fixed value.
  • the DMRS pilot design provided by the embodiment shown in FIG. 13 is a way of fixing the pilot symbol interpolation rate. Since the adaptive ZT configuration is considered, the user changes the interpolation rate according to the change of the channel delay, but the pilot insertion position interval is different under different interpolation rates, so the pilot symbol design can be designed in order to unify different interpolation rates. For a fixed interpolation rate, instead of a pilot symbol, different interpolation rates can be varied. Two sets of pilot positions are designed in Figure 13. Each set of pilot positions corresponds to 4 ports.
  • Port1 to Port 3 are multiplexed by using a set of pilot positions, where Port0 and Port1 adopt FDM or TDM mode or OCC orthogonal mode. Reuse.
  • Port2 and Port3 can be multiplexed with Port0 and Port1 by cyclic shift Cyclic Shift.
  • the Cyclic Shift technology has been applied to the uplink of LTE protocol, specifically Port2 and Port0.
  • pilot sequence such as ZC (Zadoff-Chu) sequence, random sequence, etc.
  • Port2 is based on the pilot sequence
  • the symbol-by-symbol phase rotation makes it possible to separately estimate the channel times of the two port ports in the time domain channel estimation.
  • FIG. 14 is a schematic diagram of another pilot design pattern of the DMRS according to an embodiment of the present invention.
  • the interpolation parameter is multiple One of the interpolation parameters; when the first signal sequence includes at least one first pilot symbol or at least one zero, the first predetermined number is determined according to the plurality of interpolation parameters.
  • the pilot insertable position interval is different under different interpolation parameters. Multiple pilot insertable position intervals are included under multiple interpolation parameters. Further, the pilot insertion position of the plurality of interpolation parameters is set to be the least common multiple of the multiple insertion intervals, and the patterns of the pilot insertable positions under different interpolation parameters are unified, so that the receiving end receives the pilot symbols.
  • n i when a set of interpolation parameters (m i , n i ) satisfy the following two conditions: 1.
  • the least common multiple of n i of the set of interpolation parameters is as small as possible, which determines the pilot frequency domain interval of each symbol; 2.
  • the set of interpolation parameters can cover sufficient channel delay and uniformly quantize the delay range as much as possible.
  • the least common multiple of the set of interpolation parameters n i can be selected as the first predefined position.
  • the set of interpolation parameters selected in Figure 14 are (1, 2), (2, 3), (5, 6), (8, 9), and the least common multiple of n i is 18, so the pilot of each DMRS symbol
  • the interval is 18 subcarriers. Because the interval is large, relatively accurate channel estimation cannot be performed. Considering more DMRS symbols, different symbols are subjected to different frequency domain cyclic shifts to achieve better pilot density.
  • each RB has 4 DMRS pilot symbols, so generally it can support the estimation of 4 port ports, wherein the first group of pilot positions multiplexes Port0 and Port2, and the multiplexing mode adopts Cyclic Shift mode, and the second The group pilot positions are multiplexed with Port and Port3, and are also multiplexed by Cyclic Shift.
  • FIG. 15 is a schematic diagram of a pilot design pattern of a PNRS according to an embodiment of the present invention.
  • PNRS is used for the estimation of random phase noise introduced in medium and high frequency PA devices. It requires very low frequency domain density and can be placed in more than a dozen RBs, but each symbol is required in the time domain. Therefore, n i of a set of interpolation parameters can be taken as the least common multiple, and the phase noise pilot can be set at a position N times the least common multiple.
  • the PNSR pilot pattern design shown in Fig. 15 can be referred to in a manner in which the pilot symbol interpolation rate is fixed or the pilot pattern interpolation rate is variable (least common multiple). In addition, for symbols of existing DMRS, it may not be necessary to additionally set the PNSR pilot.
  • the PNSR pilot pattern shown in FIG. 15 is only an example. In practical applications, the PNSR pilot can be used for each symbol, that is, at least one of the OFDM symbols represented by each column shown in FIG. 16 is a PNSR pilot.
  • the embodiment of the present invention implements a pattern design of multiple pilots similar to the 5G standard.
  • DMRS pilot two unified pilot patterns can be used to adaptively switch different interpolation parameters, thereby avoiding different transmission points under CoMP. Different interpolation parameters, while DMRS pilots can maintain orthogonal effects.
  • the pilot pattern of the OFDM system of the existing 5G standard is very flexible, and the pilot pattern design scheme of the frequency domain interval insertion pilot provided by the embodiment of the present invention has a frequency domain interval constraint of the pilot of the ZT-OFDM, and a design is designed. A pilot pattern that approximates the OFDM pilot effect.
  • pilot symbols only at integer multiple interpolation points may result in relatively low density of pilot symbols, which may affect the quality of channel estimation, noise estimation, etc., and thus affect the final throughput rate.
  • the pilot considering the insertion of the pilot at the integer multiple interpolation point, the pilot is also inserted for the position next to the integer multiple point, and the density of the pilot symbol is enhanced.
  • interpolation matrix shown in FIG. 4 there is only a certain value at the positions on both sides of the diagonal, so that the interpolated symbols are basically only strongly correlated with several nearby symbols before interpolation, and in the matrix at integer multiple interpolation positions.
  • the element is 1, and the interpolated i*n values are only related to the i*mth values before interpolation (invariant amplitude, phase rotation).
  • the first column element is 1, the elements on both sides of the diagonal of the second column can be set to 0.9 and 0.1, respectively, and the elements of the third column are set to 0.7 and 0.3, respectively. Set to 0.5 and 0.5, respectively, and the fifth column elements are set to 0.1 and 0.9, respectively. Therefore, the second column element only introduces interference of 0.1 symbols of other symbols. Therefore, two locations around the integer multiple interpolation point can also be used to insert pilots (also including zero power pilots).
  • FIG. 16 is a data transmission method and apparatus based on a frequency domain DFT interpolation pilot enhancement scheme according to an embodiment of the present invention. This embodiment includes steps 1601 through 1606:
  • the interpolation parameter and the system allocate resources to the user, and the number of symbols to be generated before the interpolation can be referred to the description in FIG. 3 .
  • step 1601, step 1604, and step 1606 can also refer to the descriptions in step 301, step 304, and step 307, respectively. To simplify the description, the details are not described below.
  • Step 1602 Generate at least one second pilot symbol and at least one third pilot symbol according to the cell number, the frame number, and the like, respectively.
  • the number of the second pilot symbols is the same as the number of the third pilot symbols.
  • the number of second pilot symbols and the number of third pilot symbols are L (L ⁇ T).
  • steps 1602a and 1602b are included:
  • Step 1602a performing third phase rotation on the at least one second pilot symbol to obtain at least one first pilot symbol.
  • the third phase rotation may be expressed as multiplied by a phase rotation factor Where i ⁇ [0, T), i is determined according to which first pilot symbol positions of the first signal sequence in step 1603 are specifically inserted according to the first pilot symbol.
  • Step 1602b performing fourth phase rotation and amplitude adjustment on the at least one third pilot symbol to obtain at least one fourth pilot symbol.
  • Step 1603 the at least one fourth pilot symbol, the at least one first pilot symbol, and the at least one data symbol are formed into the first signal sequence according to a third predefined rule.
  • the third predefined rule is that every first predefined number of data symbols is a candidate first pilot symbol position, and at least one candidate first pilot symbol position is selected to be inserted into the first Pilot symbol; selection Adjacent positions of the at least one candidate first pilot symbol position are inserted into the fourth pilot symbol, respectively; wherein the first predefined number is determined according to the interpolation parameter.
  • the first predefined number may be equal to m-1.
  • the at least one data symbol and the at least one first pilot symbol and the at least one fourth pilot symbol are interpolated by S/P conversion into an M length first signal sequence.
  • the fractional-fold DFT interpolation corresponds to the position of the i*n of the sequence of N lengths.
  • Step 1605a the mapping at least one set of the second signal sequence to a subcarrier, to obtain a second signal sequence on at least one group of subcarriers.
  • the neighboring position of the second predefined number of subcarriers is the candidate third pilot symbol position, and the third pilot symbol is used to replace the subcarrier of the at least one candidate third pilot symbol position.
  • a symbol in the second signal sequence, the second predefined number is determined according to the interpolation parameter, wherein the second predefined number is greater than the first predefined number, the second signal
  • the replacement position of the third pilot symbol is related to a position in the first signal sequence in which the fourth pilot symbol is inserted.
  • Step 1605b Perform an inverse fast Fourier transform IFFT on the second signal sequence on the subcarrier to obtain a time domain signal.
  • the traditional IFFT operation is performed, and the multi-layer MIMO transmission also includes layer mapping, pre-coding and the like before the IFFT.
  • the pilot enhancement scheme provided by the embodiment of the present invention adds some configurable pilot positions on both sides of the original pilot position, so there is a certain enhancement regardless of the orthogonality or the maximum supported port port.
  • the design of the pilot pattern of the present invention can refer to the two schemes of the fixed interpolation rate and the variable interpolation rate (the least common multiple interpolation position) provided by the foregoing embodiments. I will not repeat them here.
  • FIG. 17 is a schematic diagram of an enhanced pilot design of a DMRS according to an embodiment of the present invention. As shown in FIG. 17, since the pilot position is increased, the DMRS pilot scheme in this embodiment can adopt a lower interpolation rate such as an interpolation parameter of (5, 6), thereby ensuring a higher data transmission rate of the symbol.
  • an interpolation parameter of (5, 6) such as an interpolation parameter of (5, 6)
  • Port0 to Port3 in Fig. 17 is the same as that of Fig. 13, and Port4 to Port7 can be placed next to the pilot symbols of Port0 to Port3.
  • the orthogonality between the two sets of port ports can be configured by using the technique shown in FIG. 13 to configure at least one third pilot symbol with zero power to achieve no interference between the two port ports, or to configure at least one third pilot symbol for normal power.
  • the OCC is orthogonal to achieve OCC orthogonality between the two sets of port ports.
  • the third pilot symbol zero power or the normal power OCC orthogonality can be configured according to different port ports to ensure that different groups of port ports have no interference or OCC orthogonal.
  • FIG. 18 is a schematic diagram of another enhanced pilot design pattern of a DMRS according to an embodiment of the present invention.
  • a group of ports Port4 to Port7 are added next to the ports Port0 to Port4, and the orthogonality of the two port ports can also be configured with at least one third guide.
  • Frequency Symbol zero power or normal power OCC is implemented in an orthogonal manner.
  • the third pilot symbol is specifically inserted into which of the candidate locations and is not associated with the location at which the first pilot symbol is inserted.
  • Figures 17 and 18 show only one of the pilot enhancements.
  • the technical solution provided by the embodiment of the present invention can improve the density of the available pilot positions by using the symbols on both sides of the integer multiple interpolation point position to transmit the pilot signals. In turn, it can support more port ports, or switch to pilot symbols to transmit more data symbols.
  • FIG. 19 is a schematic diagram of a method for transmitting asynchronous data in multiple transmission points according to an embodiment of the present invention.
  • the at least one transmitting device when the at least one transmitting device cooperates to transmit data to the receiving end, the at least one transmitting device includes a serving transmitting device and a cooperative transmitting device, and the cooperative transmitting device determines according to a delay deviation thereof with respect to the serving transmitting device. Interpolating parameters to cause the receiving device to perform joint MIMO reception on signals of the at least one transmitting device.
  • the head and the tail of the second signal sequence respectively comprise a plurality of data smaller than a preset threshold.
  • the data smaller than the preset data may be approximate zero data.
  • downlink multi-cooperation point non-coherent mode transmission to an end user or uplink multi-user simultaneous frequency asynchronous contention access, can be applied in the manner of FIG. 19, and the following two transmission point transmissions are taken as an example:
  • the terminal device may receive data transmitted by at least two Transmission Point (TP) cooperatively. Let the two transmission points be TP0 and TP1.
  • TP Transmission Point
  • the terminal device accesses the TP0 of the target cell and performs timing according to TP0. Therefore, the TP0 can be transmitted in a manner without ZT (no interpolation, the interpolation rate is 1). TP0 can be called a service transmission point.
  • the terminal device measures the signal receiving power of each TP point in the downlink, and selects TP1 as the coordinated transmission point.
  • TP1 and TP0 are not co-timed, there is a delay deviation between the signals transmitted by the terminal device and TP1 and TP0.
  • the terminal device measures the delay deviation of the TP1, and feeds back to the TP1 delay indication of the delay deviation by the uplink control channel.
  • TP1 selects an appropriate interpolation parameter of ZT-OFDM, interpolates the data, and sends a signal to the terminal device.
  • the terminal device performs joint MIMO reception on the signals of the two TPs.
  • TP1 has a certain delay deviation (TP1delay) with respect to TP0.
  • the length of ZT in the TP1 signal is controlled by selecting an appropriate interpolation parameter such that the effective signal of TP1 falls completely within one cycle of the receiving end, or the effective signal of TP1 is completely within the IFFTsize of TP0.
  • the valid signal refers to the data in the IFFTsize to remove the ZT and ZH parts. The proportion of ZH is small and therefore is not shown in FIG.
  • the application mode shown in FIG. 19 may further include other cooperative TPs, and the delay deviation of each cooperative TP relative to TP0 of the terminal device is fed back to the coordinated control TP by the uplink control channel.
  • the cooperative TP selects an appropriate interpolation parameter of the ZT-OFDM according to the received delay deviation value, and interpolates the data to send a signal to the terminal device. To ensure that the valid signals of multiple TPs are completely within the IFFTsize of TP0. So that the terminal device can perform joint MIMO reception on signals of multiple TPs.
  • the prior art can only be configured to transmit in the manner of Extended CP, so that TPs without delay skew or short delay skew are transmitted according to a longer CP format, which will reduce these.
  • the signal rate of the TP is not limited to the manner of Extended CP, so that TPs without delay skew or short delay skew are transmitted according to a longer CP format, which will reduce these.
  • the signal rate of the TP is not limited to transmit in the manner of Extended CP, so that TPs without delay skew or short delay skew are transmitted according to a longer CP format, which will reduce these.
  • the signal rate of the TP is not limited to transmit in the manner of Extended CP, so that TPs without delay skew or short delay skew are transmitted according to a longer CP format, which will reduce these.
  • the signal rate of the TP is not limited to the manner of Extended CP, so that TPs without delay skew or short delay skew are transmitted according to a longer CP
  • the multi-emission point asynchronous transmission technology provided by the embodiment of the present invention can ensure that a plurality of transmission points of different timings select appropriate ZT-OFDM interpolation parameter transmission signals according to respective delay deviations, and can ensure that the receiving apparatus adopts joint MIMO mode for reception. Compared with simple serial interference cancellation (SIC), the processing delay is shorter.
  • SIC serial interference cancellation
  • the embodiment of the present invention utilizes the characteristics of ZT-OFDM to construct a shorter time interval in the time domain to achieve a wider subcarrier spacing effect.
  • the transmitting the time domain signal further includes: the time domain signal includes a first time domain signal and a second time domain signal, the first time domain signal The length of time occupied by the second time domain signal is the same; the first time domain signal and the second time domain signal are respectively intercepted to obtain a third time domain signal and a fourth time domain signal, The second time domain signal is delayed by half a period in the time domain compared to the first time domain signal, and the time length of the third time domain signal is half of the length of time of the first time domain signal.
  • the time length of the fourth time domain signal is half of the length of time of the second time domain signal; mixing the third time domain signal and the fourth time domain signal to obtain a fifth time domain signal;
  • the fifth time domain signal, the half period is half of the length of time of the first time domain signal.
  • the first time domain signal and the second time domain signal are respectively intercepted by using an IFFT/2 window function, and the first time domain signal or the second time domain signal header and the tail part are smaller than the pre Set the threshold data to be truncated.
  • the transmitting the time domain signal further includes: the time domain signal includes a sixth time domain signal, a seventh time domain signal, and an eighth time domain signal. And a ninth time domain signal, a length of time occupied by the sixth time domain signal, a length of time occupied by the seventh time domain signal, a length of time occupied by the eighth time domain signal, and the ninth The time domain signal takes the same length of time; the sixth time domain signal, the seventh time domain signal, the eighth time domain signal and the ninth time domain signal are respectively intercepted to obtain a tenth time domain signal and an eleventh time domain.
  • the seventh time domain signal is delayed by a quarter cycle and the eighth time domain signal in the time domain compared to the sixth time domain signal Delaying a quarter period in the time domain compared to the seventh time domain signal, the ninth time domain signal being delayed by a quarter period in the time domain compared to the eighth time domain signal, the first The length of time occupied by the ten-time domain signal is four minutes of the length of time of the sixth time domain signal In one of the first time domain signals, the length of time occupied by the seventh time domain signal is one quarter of the length of time, and the time length of the twelfth time domain signal is the eighth time.
  • the time domain signal occupies a quarter of the length of time, and the time length of the thirteenth time domain signal is one quarter of the length of time of the ninth time domain signal; Mixing the signal, the eleventh time domain signal, the twelfth time domain signal, and the thirteenth time domain signal to obtain a fourteenth time domain signal; transmitting the fourteenth time domain signal, the quarter period It is a quarter of the length of time of the sixth time domain signal.
  • the sixth time domain signal, the seventh time domain signal, the eighth time domain signal, and the ninth time domain signal are respectively intercepted by using an IFFT/4 window function, and the sixth time domain signal, the seventh The time domain signal, the eighth time domain signal, and the plurality of data of the ninth time domain signal header and tail that are less than a preset threshold are truncated.
  • Another example of the above achieves the effect of an equivalent 60K subcarrier spacing by constructing a time interval of T/4.
  • FIG. 20 is a schematic diagram of a data transmission method for a shorter time interval according to an embodiment of the present invention.
  • the transmitting end uses the interpolation parameter of (1, 2) to obtain the first time domain signal and the second time domain signal.
  • the length of time occupied by the first time domain signal is the same as the length of time occupied by the second time domain signal, and the length of time occupied by the first time domain signal is FFTsize.
  • the ZT close to the IFFT/2 length in the first time domain signal and the second time domain signal is approximately zero power.
  • the time delay between the signals of the first time domain signal and the second time domain signal is T/2 (Delay 0 and Delay T/2).
  • the window function of the IFFT/2 is used to intercept the effective signal portions of the first time domain signal and the second time domain signal respectively to obtain a third time domain signal and a fourth time domain signal, as shown in FIG. 20, 2003 and 2004. Show.
  • the window function may be a rectangular window or other window function with a certain roll-off.
  • the third time domain signal of the zero power tail with the ZP/2 length added and the fourth time domain signal are mixed to obtain a fifth time domain signal, as shown in the 2007 section of FIG.
  • the fifth time domain signal is transmitted.
  • the length of the fifth time domain signal is the same as the length of the original first time domain signal or the second time domain signal, and is FFTsize.
  • the fifth time domain signal constructs a time domain time interval of T/2, which achieves an effect of twice the subcarrier spacing.
  • the receiving device receives the signal in the T/2 window, and processes the ZP portion of the tail in an OLA manner; adds a FFT/2 zero power signal to the tail at the FFT/2 length signal; and passes the final FFT length signal through the normal
  • the (1, 2) interpolation parameter is processed at the receiving end of ZT-OFDM.
  • signals of a plurality of terminal devices can also be time-division multiplexed in the manner of FIG.
  • the signals of a plurality of terminal devices are time-division multiplexed by using (1, 2) or (1, 4) interpolation parameters, and intercepting and mixing.
  • the solution provided by the embodiment of the present invention utilizes the characteristics of ZT-OFDM to construct a shorter time interval in the time domain to achieve a wider subcarrier spacing effect.
  • the air interface between different services is still 15K, there is no mutual ICI interference problem.
  • this embodiment consists of The air interface is 15k, there is no ICI interference problem, so you can save the frequency domain GB and effectively improve the throughput.
  • the transmitting end performs a certain ratio of fractional interpolation filtering operation on the modulation symbol mapped to the frequency domain, and reaches the final IFFT signal head ZH.
  • the data and the tail ZT data approximate the effect of zero power, and the frequency domain RS pilot signal can be inserted at the integer multiple interpolation position before interpolation, in order to keep the receiving device RS transparent, the inserted frequency domain RS according to the time domain offset of the signal Perform a phase rotation operation.
  • the OFDM time domain symbols formed by interpolation can directly form a continuous signal in series.
  • the length of the ZT can cover the total channel delay deviation; and each symbol after the IFFT can be added with a ZP at the tail.
  • the zero-power signal, or the CP zero-power signals at the end of the header copy symbol form a cyclic prefix signal to achieve the effect of the zero-power signal in the symbol being connected to the ZP or CP zero-power signal.
  • the interpolation filter in the embodiment of the present invention is not limited to the DFT interpolation filter, and may also be a plurality of interpolation algorithms such as linear interpolation and spline interpolation, wherein the DFT interpolation filter includes an IDFT transform, a time domain before and after, and a zero power head end. , DFT transformation, in which the length of the zero power head and tail ZH and ZT can be adjusted.
  • the embodiment of the present invention provides an OFDM time domain symbol formed by interpolation, and the greater the corresponding interpolation rate, the stronger the ability to resist channel delay deviation.
  • the maximum delay deviation is set at a certain interval to set the gear positions of the plurality of interpolation ratios, and the transmitting device selects an interpolation ratio according to the measured maximum delay deviation, and notifies the receiving device by the control message, or the receiving device according to the pilot format or zero
  • the length of the power is blindly detected.
  • the data symbols and pilot symbols of the user may be divided into multiple blocks, each block is independently interpolated, and the last blocks may be discretely distributed in the frequency domain, thereby obtaining better scheduling gain and frequency. Domain diversity gain.
  • the pilot symbol insertion may be a fixed interpolation rate, and the non-pilot symbols may change different interpolation rates, so that the same pilot pattern can be multiplexed under different interpolation parameter configurations.
  • the pilot insertion symbol may also be a variable interpolation rate.
  • a set of interpolation parameters (m i , n i ) needs to be selected.
  • the selected criterion needs to satisfy the minimum common multiple of n i of the set of parameters as small as possible to ensure sufficient pilot frequency domain density, and the set of parameters can cover sufficient channel delay and uniformly quantize the delay range as much as possible.
  • the solution provided by the embodiment of the present invention can insert a pilot symbol at an integer multiple interpolation point.
  • a pilot or a zero power symbol can be inserted at a finite number of points around the integer multiple interpolation point.
  • the symbol is subjected to a certain phase rotation and amplitude adjustment at the time of insertion, and the symbol of the corresponding position is replaced with the inserted pilot or zero power symbol before interpolation when mapped to the subcarrier after interpolation.
  • the embodiment of the present invention provides a communication device, which is used to implement the data transmission method provided in the foregoing embodiment.
  • the communication device includes: an interpolation unit 2101, a mapping unit 2102, and an IFFT unit 2103. , a transmitting unit 2104, a determining unit 2105, and an obtaining unit 2106.
  • the interpolation unit 2101 of the communication device is configured to perform an interpolation operation on the first signal sequence to obtain a second signal sequence, the length of the second signal sequence being greater than the length of the first signal sequence.
  • the mapping unit 2102 is configured to map the second signal sequence to the subcarriers to obtain a second signal sequence on the subcarriers.
  • the IFFT unit 2103 is configured to perform an inverse fast Fourier transform IFFT on the second signal sequence on the subcarrier to obtain a time domain signal.
  • a transmitting unit 2104 is configured to transmit the time domain signal.
  • the determining unit 2105 is configured to determine an interpolation parameter; the interpolation unit 2101 is specifically configured to perform an interpolation operation on the first signal sequence according to the interpolation parameter to obtain the second signal sequence.
  • the method further includes: an obtaining unit 2106, configured to acquire a maximum delay deviation of a signal of the terminal device, where the signal of the terminal device is a wireless signal sent to the terminal device or a wireless signal sent by the terminal device,
  • the maximum delay is the difference between the time when the signal of the terminal device is sent from the transmitting end and the time when the wireless channel is the earliest to reach the receiving device and the time when the receiving device arrives at the receiving device.
  • the determining unit 2105 is specifically configured to be used according to the terminal. The maximum delay deviation of the signal of the device determines the interpolation parameter.
  • the interpolation unit 2101 is specifically configured to: perform an inverse discrete Fourier transform IDFT on the first signal sequence to obtain a fourth signal sequence; add ZH zeros in a header of the fourth signal sequence, and Adding ZT zeros at the tail of the fourth signal sequence to obtain a fifth signal sequence, wherein ZH and ZT are integers greater than zero; performing discrete Fourier transform DFT on the fifth signal sequence to obtain the A second sequence of signals, wherein the length of the second sequence of signals is equal to the sum of the length of the first sequence of signals, ZH and ZT.
  • the interpolation unit 2101 can also be implemented by the generalized interpolation and the last replacement of the pilot position interpolation.
  • the interpolation unit 2101 can also be implemented by the generalized interpolation and the last replacement of the pilot position interpolation.
  • FIG. 3, FIG. 5, FIG. 6, and FIG. 7, which are not described herein.
  • the first sequence of signals comprises at least one data symbol.
  • said first signal sequence comprises at least one first pilot symbol, said first pilot symbol being obtained by a second phase rotation of said second pilot symbol, said second pilot symbol being for receiving
  • the apparatus performs at least one of channel measurement and channel estimation.
  • the first signal sequence further comprises at least one data symbol, wherein the at least one first pilot symbol and the at least one data symbol form the first signal sequence according to a first predefined rule;
  • the first predefined rule is: in the first signal sequence, every first predefined number of data symbols is a candidate first pilot symbol position, and at least one candidate first pilot symbol position insertion station is selected.
  • the first pilot symbol is described, wherein the first predefined number is determined according to the interpolation parameter.
  • the plurality of symbols that the first signal sequence can include and the arrangement of the plurality of symbols can be referred to FIG. 3,
  • the mapping unit 2102 is specifically configured to: map at least one group of the second signal sequence to a subcarrier to obtain a second signal sequence on at least one group of subcarriers.
  • block interpolation mapping scheme can be referred to the description in FIG. 8 and will not be described here.
  • the transmitting unit 2104 is further configured to send information related to the interpolation parameter, where the information related to the interpolation parameter is used by the receiving device to determine the interpolation parameter.
  • the communication device shown in FIG. 21 can also adopt the following implementation manner. Specifically as shown in Figure 22 As shown, the communication device includes a processor 2201 and a transmitter 2202.
  • the processor 2201 of the communication device is configured to perform an interpolation operation on the first signal sequence to obtain a second signal sequence, where the length of the second signal sequence is greater than the length of the first signal sequence.
  • the processor 2201 is further configured to map the second signal sequence to the subcarriers to obtain a second signal sequence on the subcarriers.
  • the processor 2201 is further configured to perform an inverse fast Fourier transform IFFT on the second signal sequence on the subcarrier to obtain a time domain signal.
  • a transmitter 2202 is configured to transmit the time domain signal.
  • the processor 2201 is further configured to determine an interpolation parameter; the processor 2201 is specifically configured to perform an interpolation operation on the first signal sequence according to the interpolation parameter to obtain the second signal sequence.
  • the processor 2201 is further configured to acquire a maximum delay deviation of the signal of the terminal device, where the signal of the terminal device is a wireless signal sent to the terminal device or a wireless signal sent by the terminal device, where the maximum time
  • the delay is the difference between the time when the signal of the terminal device is sent from the transmitting end, the time when the wireless channel is reached, and the time when the device reaches the receiving device at the earliest; the processor 2201 is specifically configured to use the signal according to the terminal device.
  • the maximum delay deviation determines the interpolation parameter.
  • the processor 2201 is specifically configured to: perform an inverse discrete Fourier transform IDFT on the first signal sequence to obtain a fourth signal sequence; add ZH zeros in a header of the fourth signal sequence, and Adding ZT zeros at the tail of the fourth signal sequence to obtain a fifth signal sequence, wherein ZH and ZT are integers greater than zero; performing discrete Fourier transform DFT on the fifth signal sequence to obtain the A second sequence of signals, wherein the length of the second sequence of signals is equal to the sum of the length of the first sequence of signals, ZH and ZT.
  • the processor 2201 can also be implemented by the generalized interpolation and the last replacement of the pilot position interpolation. For details, refer to the descriptions in FIG. 3, FIG. 5, FIG. 6, and FIG. 7, which are not described herein.
  • the first sequence of signals comprises at least one data symbol.
  • said first signal sequence comprises at least one first pilot symbol, said first pilot symbol being obtained by a second phase rotation of said second pilot symbol, said second pilot symbol being for receiving
  • the apparatus performs at least one of channel measurement and channel estimation.
  • the first signal sequence further comprises at least one data symbol, wherein the at least one first pilot symbol and the at least one data symbol form the first signal sequence according to a first predefined rule;
  • the first predefined rule is: in the first signal sequence, every first predefined number of data symbols is a candidate first pilot symbol position, and at least one candidate first pilot symbol position insertion station is selected.
  • the first pilot symbol is described, wherein the first predefined number is determined according to the interpolation parameter.
  • the multiple symbols that can be included in the first signal sequence and the arrangement of the multiple symbols can be referred to the descriptions in FIG. 3 and FIG. 5, and details are not described herein.
  • the processor 2201 is specifically configured to: map at least one set of the second signal sequence to a subcarrier to obtain a second signal sequence on at least one group of subcarriers.
  • block interpolation mapping scheme can be referred to the description in FIG. 8 and will not be described here.
  • the transmitter 2202 is further configured to send information related to the interpolation parameter, where the information related to the interpolation parameter is used by the receiving device to determine the interpolation parameter.
  • the embodiment of the present invention provides a communication device for implementing the data transmission method provided in the foregoing embodiment.
  • the communication device includes: a receiving unit 2301, an FFT unit 2302, and a demodulation. Unit 2303, deinterpolation unit 2304, decoding unit 2305, acquisition unit 2306, channel estimation unit 2307, and determination unit 2308.
  • the receiving unit 2301 of the communication device is configured to receive a time domain signal.
  • the FFT unit 2302 is configured to perform fast Fourier transform FFT on the time domain signal to obtain a sixth signal sequence on the subcarrier.
  • the demodulation unit 2303 is configured to demodulate the sixth signal sequence to obtain a seventh signal sequence.
  • the de-interpolation unit 2304 is configured to perform a de-interpolation operation on the seventh signal sequence to obtain an eighth signal sequence, where the eighth signal sequence includes soft information of the data symbol, wherein the length of the eighth signal sequence is smaller than the seventh signal sequence. length.
  • the decoding unit 2305 is configured to decode the soft information of the data symbol to obtain a data symbol.
  • the method further includes: an obtaining unit 2306, configured to acquire pilot symbols; a channel estimating unit 2307, configured to perform channel estimation according to the pilot symbols, to obtain channel related information; and the demodulating unit 2303 is specifically configured to use the channel according to the channel
  • the correlation information demodulates the sixth signal sequence to obtain the seventh signal sequence.
  • the determining unit 2308 is configured to determine an interpolation parameter.
  • the de-interpolation unit 2304 is specifically configured to perform a de-interpolation operation on the seventh signal sequence according to the interpolation parameter to obtain an eighth signal sequence.
  • the de-interpolation unit 2304 is specifically configured to: perform an inverse discrete Fourier transform IDFT on the seventh signal sequence to obtain a tenth signal sequence; delete ZH zeros of the tenth signal sequence header, and Deleting ZT zeros at the tail of the tenth signal sequence to obtain an eleventh signal sequence; performing discrete Fourier transform DFT on the eleventh signal sequence to obtain the eighth signal sequence, wherein the eighth The length of the signal sequence is equal to the length of the seventh signal sequence minus the values obtained by ZH and ZT.
  • the de-interpolation unit 2304 can also be implemented by a generalized solution interpolation.
  • the de-interpolation unit 2304 can also be implemented by a generalized solution interpolation.
  • FIG. 10 and FIG. 11 and details are not described herein.
  • the de-interpolation unit 2304 is specifically configured to: demodulate the sixth signal sequence to obtain at least one group of the seventh signal sequence.
  • the receiving unit 2301 is further configured to receive information related to the interpolation parameter, where the information related to the interpolation parameter is used by the receiving device to determine the interpolation parameter.
  • the communication device shown in FIG. 23 can also adopt the following implementation manner. Specifically, as shown in FIG. 24, the communication device includes: a receiver 2401 and a processor 2402.
  • the receiver 2401 of the communication device is configured to receive a time domain signal.
  • the processor 2402 is further configured to perform fast Fourier transform FFT on the time domain signal to obtain a sixth signal sequence on the subcarrier.
  • the processor 2402 is further configured to demodulate the sixth signal sequence to obtain a seventh signal sequence.
  • the processor 2402 is further configured to perform a de-interpolation operation on the seventh signal sequence to obtain an eighth signal sequence, where the eighth signal sequence includes soft information of the data symbol, where the length of the eighth signal sequence is smaller than the seventh signal sequence. length.
  • the processor 2402 is further configured to decode the soft information of the data symbol to obtain a data symbol.
  • the processor 2402 is further configured to acquire a pilot symbol; the processor 2402 is further configured to perform channel estimation according to the pilot symbol to obtain channel-related information; and the processor 2402 is specifically configured to perform sixth according to the channel-related information.
  • the signal sequence is demodulated to obtain the seventh signal sequence.
  • the processor 2402 is further configured to determine an interpolation parameter; the processor 2402 is specifically configured to perform a de-interpolation operation on the seventh signal sequence according to the interpolation parameter to obtain an eighth signal sequence.
  • the processor 2402 is specifically configured to: perform an inverse discrete Fourier transform IDFT on the seventh signal sequence to obtain a tenth signal sequence; delete ZH zeros of the tenth signal sequence header, and delete the a ZT zeros at the tail of the tenth signal sequence, to obtain an eleventh signal sequence; performing a discrete Fourier transform DFT on the eleventh signal sequence to obtain the eighth signal sequence, wherein the eighth signal sequence
  • the length is equal to the length of the seventh signal sequence minus the values obtained by ZH and ZT.
  • the processor 2402 can also be implemented by a generalized solution interpolation.
  • the processor 2402 can also be implemented by a generalized solution interpolation.
  • FIG. 10 and FIG. 11 the descriptions in FIG. 10 and FIG. 11 , and details are not described herein.
  • the processor 2402 is specifically configured to: demodulate the sixth signal sequence to obtain at least one group of the seventh signal sequence.
  • the receiver 2401 is further configured to receive information related to the interpolation parameter, where the information related to the interpolation parameter is used by the receiving device to determine the interpolation parameter.
  • the functions described herein can be implemented in hardware, software, firmware, or any combination thereof.
  • the functions may be stored in a computer readable medium or transmitted as one or more instructions or code on a computer readable medium.
  • Computer readable media includes both computer storage media and communication media including any medium that facilitates transfer of a computer program from one location to another.
  • a storage medium may be any available media that can be accessed by a general purpose or special purpose computer.

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Abstract

本发明实施例涉及一种数据传输方法及通信设备，该方法包括：对第一信号序列进行插值操作，得到第二信号序列，所述第二信号序列的长度大于所述第一信号序列的长度；将所述第二信号序列映射到子载波，得到子载波上的第二信号序列；对所述子载波上的第二信号序列进行快速傅里叶逆变换IFFT，得到时域信号；发射所述时域信号。本发明实施例可更好地对抗时延偏差。

Description

数据传输方法及通信设备
本申请要求于2016年09月30日提交中国专利局、申请号为201610877323.8、申请名称为“数据传输方法及通信设备”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本发明涉及通信技术领域，尤其涉及一种数据传输方法及通信设备。
背景技术
在LTE系统以及目前正在讨论的5G标准中，基于正交频分复用(Orthogonal Frequency Division Multiplexing，OFDM)的波形被公认为基线波形，其优势有低复杂度的频域均衡算法，采用多输入多输出(Multiple Input Multiple Output，MIMO)技术进行灵活的多层数据空间复用等。这些优点来自于OFDM符号间插入了循环前缀(Cyclic Prefix，CP)，保证了每个符号理想的自循环特性，从而确保在CP能够覆盖信道最大时延扩展的前提下，不引入任何符号间的干扰(Inter Symbol Interference，ISI)。
但是在实际的网络中，用户与基站的距离时刻在变化，同时对于超远覆盖等特殊情况，现有的LTE协议还定义了扩展CP(Extended CP)的波形。另外，下行协作多点传输(Downlink Coordinated Multipoint Transmission，DL CoMP)也是目前5G讨论的热点。由于不同的发射点属于不同的基站，不同基站使用的时钟源可能不同，则发射点间很难保证定时同步。因此在无线信道本身的时延扩展的基础上又引入了额外的发送点之间的时延偏差，导致不同发射点之间不再满足准共址QCL(QuasiCo-Located)的要求，这对CP长度的挑战会更大。
现有的LTE协议为了对抗信道时延扩展，均采用CP方式。现有扩展CP方案，扩展CP为小区级配置，因此整个小区的所有用户的吞吐量都会受影响，直观上直接缩减13％的吞吐量；对于CoMP的每个发送端，都需要配置为扩展CP，这就导致每个发送端信号的吞吐量都直接缩减13％；现有协议只有扩展CP这一种长CP格式，因此对于时变的信道时延偏差不能很好的适应，或者太浪费资源或者会引入ISI。
综合来看，现有协议的CP的定义不能很灵活的满足5G中各种场景的需求，需要一种可以自适应各种时延偏差场景的OFDM波形方案。
发明内容
本发明实施例提供一种数据传输方法及通信设备，以解决现有技术中CP的定义不能很灵活的满足5G中各种场景的需求的问题。
在第一方面，本发明实施例提供了一种数据传输方法，所述方法包括：对第一信号序列进行插值操作，得到第二信号序列；将所述第二信号序列映射到子载波，得到子载波上的第二信号序列；对所述子载波上的第二信号序列进行快速傅里叶逆变换 IFFT，得到时域信号；发射所述时域信号。
本发明实施例通过频域插值操作，在时域上体现了OFDM符号自适应零功率ZT的效果，使得发射的时域信号可以更好地对抗时延偏差。
在一个可选的实现中，在所述对第一信号序列进行插值操作之前，还包括：确定插值参数；所述对第一信号序列进行插值操作，得到第二信号序列，包括：根据所述插值参数对所述第一信号序列进行插值操作，得到所述第二信号序列，所述第二信号序列的长度大于所述第一信号序列的长度。
本发明实施例可通过插值参数调控零功率ZT的长度，结合不同的插值参数，可以灵活的应对不同的信道时延变化。
在一个可选的实现中，在所述确定插值参数之前，还包括：获取终端设备的信号的最大时延偏差，所述终端设备的信号为发给所述终端设备的无线信号或所述终端设备发送的无线信号，所述最大时延偏差为所述终端设备的信号从发送端发出，经过无线信道，最早到达接收端的时间与最晚到达接收端的时间之差；所述确定插值参数，包括：根据所述终端设备的信号的最大时延偏差确定所述插值参数。
本发明实施例兼顾终端设备信号多种可能的时延偏差，可以更好地对抗信道时延。所述最大时延偏差包括下行多点传输时，不同发射点传输的数据信号的时延偏差。所述最大时延偏差还可包括上行多用户异步接入时远近距离不同的用户发射的数据的时延偏差。
在一个可选的实现中，所述插值操作具体包括：对所述第一信号序列进行插值操作，得到第三信号序列；对所述第三信号序列进行第一相位旋转，得到第二信号序列。
在一个可选的实现中，所述插值操作具体包括：对所述第一信号序列进行离散傅里叶逆变换IDFT，得到第四信号序列；在所述第四信号序列的头部增加ZH个零，以及在所述第四信号序列的尾部增加ZT个零，得到第五信号序列，其中，ZH和ZT均为大于零的整数；；对所述第五信号序列进行离散傅里叶变换DFT，得到所述第二信号序列，其中，第二信号序列的长度等于所述第一信号序列的长度、ZH以及ZT的总和。
本发明实施例可以通过广义插值操作或DFT插值操作，在时域上体现了OFDM符号自适应零功率ZT的效果，使得发射的时域信号可以更好地对抗时延偏差。
在一个可选的实现中，所述第一信号序列包括至少一个第一导频符号，所述第一导频符号为第二导频符号经过第二相位旋转而得到的，所述第二导频符号用于接收端进行信道测量和信道估计中的至少一个。
在一个可选的实现中，所述第一信号序列包括至少一个第一导频符号，所述第一导频符号为第二导频符号经过第三相位旋转而得到的，所述第二导频符号用于接收端进行信道测量和信道估计中的至少一个。
在一个可选的实现中，所述将第二信号序列映射到子载波，得到子载波上的第二信号序列，具体包括：将至少一组所述第二信号序列映射到子载波，得到至少一组子载波上的第二信号序列。
本发明实施例通过基于分块方式的插值操作，使得一个用户的资源分块离散地分布在传输带宽内，这样对于目标用户可以获得更好的频域分集效果，同时对于其它用 户的调度也可以更灵活的支持。
在一个可选的实现中，所述第一信号序列包括至少一个第一导频符号以及至少一个数据符号，其中，所述至少一个第一导频符号与所述至少一个数据符号按照第一预定义规则组成所述第一信号序列；所述第一预定义规则为：在第一信号序列中，每隔第一预定义个数的数据符号为候选的第一导频符号位置，选择至少一个所述候选的第一导频符号位置插入所述第一导频符号，其中，所述第一预定义个数根据所述插值参数确定。
本发明实施例可实现基于多载波的插值操作，可在OFDM符号的某些子载波上插入导频符号。根据插值参数预设导频符号插值前的位置，使得插值前后导频符号的个数、幅值、相位保持不变。以便导频符号相对接收端不发生任何变化，即对于发送端采用插值方案与发送端不采用插值方案两种场景，接收端导频的处理是完全一样的。
在一个可选的实现中，所述第一信号序列包括至少一个零以及至少一个数据符号，其中，所述至少一个零与所述至少一个数据符号按照第二预定义规则组成所述第一信号序列；所述第二预定义规则为：在第一信号序列中，每隔第一预定义个数的数据符号为候选的插零位置，选择至少一个所述候选的插零位置插入零，其中，所述第一预定义个数根据所述插值参数确定。
需要说明的是，本发明实施例中提到的第一相位旋转、第二相位旋转，均是为了使得导频符号相对接收端保持不变。即对于发送端采用插值方案与发送端不采用插值方案两种场景，接收端导频的处理是完全一样的。
在一个可选的实现中，每隔第二预定义个数的子载波为候选的第二导频符号位置，用第二导频符号替换至少一个所述候选的第二导频符号位置所在子载波上的第二信号序列中的符号，所述第二预定义个数根据所述插值参数确定，其中，所述第二预定义个数大于所述第一预定义个数，所述第二信号序列中，所述第二导频符号的替换位置与所述第一信号序列中插入所述零的位置相关，所述第二导频符号用于接收端进行信道测量和信道估计中的至少一个。
本发明实施例为实现在插值过程中，导频符号相对接收端不发生变化，可通过将插值前第一信号序列中导频位置放零，最后使用导频符号替换第二信号序列中对应导频位置上的符号。该实现方式为直接在第一信号序列插入导频符号有效数据的替换方案。
在一个可选的实现中，还包括：发送所述插值参数相关的信息，所述插值参数相关的信息用于接收端确定所述插值参数。
进一步地，上述第一方面提供的数据传输方法还可包括以下几种可能的实现方式：
在一个可选的实现中，当所述第一信号序列包括至少一个第一导频符号或至少一个零时，所述插值参数为预设的固定值。
本发明实施例可以固定导频符号的插值参数，避免根据信道时延变化改变插值参数导致的不同插值参数下导频可插入位置的间隔不同的情况。当时域信号不包括导频符号时，可以变化不同的插值参数。
在一个可选的实现中，在所述第一信号序列仅包括至少一个数据符号时，所述插值参数为多个插值参数中的一个；当所述第一信号序列包括至少一个第一导频符号或 至少一个零时，所述第一预定个数根据所述多个插值参数确定。
本发明实施例可采用可变插值参数的导频符号设计，并根据一定时延范围下可选的多个插值参数确定导频符号插入的位置，使得多个插值参数下，导频符号插入间隔相同。在一个具体示例中，本发明实施例实现了与5G标准类似的多种导频的图样设计，对于DMRS导频可以采用两种统一的导频图样进行不同插值参数的自适应切换，从而可以避免CoMP下不同发射点采用不同插值参数时，而DMRS导频可以保持正交的效果。
在一个可选的实现中，所述第一信号序列还包括至少一个第四导频符号，其中，将所述至少一个第四导频符号、所述至少一个第一导频符号以及所述至少一个数据符号按照第三预定义规则组成所述第一信号序列。
具体地，所述第三预定义规则为：每隔第一预定义个数的数据符号为候选的第一导频符号位置，选择至少一个所述候选的第一导频符号位置及其相邻位置分别插入所述第一导频符号和所述第四导频符号；选择至少一个所述候选的第一导频符号位置的相邻位置分别插入所述第四导频符号；其中，所述第一预定义个数根据所述插值参数确定。
本发明实施例在整数倍插值点插入导频的基础上，对于整数倍点旁边的位置也插入导频，增强导频符号的密度。避免导频符号的密度相对较低，会影响信道估计、噪声估计等的质量，以及影响最终的吞吐率。通过提升可利用导频位置的密度，以支持更多的Port口，或者换来相同数量导频符号时可以传输更多的数据符号的效果。
在一个可选的实现中，所述第四导频符号为第三导频符号经过第三相位旋转而得到的，所述第三导频符号用于接收端进行信道测量和信道估计中的至少一个。
在一个可选的实现中，每隔第二预定义个数的子载波的相邻位置为候选的第三导频符号位置，用第三导频符号替换至少一个所述候选的第三导频符号位置所在子载波上的第二信号序列中的符号，所述第二预定义个数根据所述插值参数确定，其中，所述第二预定义个数大于所述第一预定义个数，所述第二信号序列中，所述第三导频符号的替换位置与所述第一信号序列中插入所述第四导频符号的位置相关。
在一个可选的实现中，当至少一个发射装置协作向接收端发送数据时，所述至少一个发射装置包括服务发射装置和协作发射装置，所述协作发射装置根据其相对所述服务发射装置的时延偏差，确定插值参数，以使接收端对所述至少一个发射装置的信号进行联合MIMO接收。
本发明实施例可以保证多个不同定时的传输点按照各自的时延偏差选择合适的插值参数发送时域信号，同时可以保证接收端采用联合MIMO方式进行接收，处理时延更短。
在一个可选的实现中，第二信号序列的头部和尾部分别包括多个小于预设阈值的数据。具体地，所述小于预设数据的数据可以为近似零数据。
在一个可能的示例中，通过对第一信号序列的频域数据进行广义的插值操作，得到的频域第二信号序列。所述频域第二信号序列的头部有ZH个近似零的数据，尾部有ZT个近似零的数据。
在一个可能的示例中，对第一信号序列进行DFT插值操作，将第一信号序列进行 IDFT变换，得到第一时域符号，在第一时域符号的头部和尾部分别增加ZH个零和ZT个零，得到第二时域符号，将第二时域符号进行DFT变换，得到频域第二信号序列，通过对时域信号两边补零达到在频域插值的效果。
在一个可选的实现中，所述第二信号序列长度与所述第一信号序列长度的比值为插值率，当插值率为2时，所述发射所述时域信号，进一步包括：所述时域信号包括第一时域信号和第二时域信号，所述第一时域信号所占的时间长度与所述第二时域信号所占的时间长度相同；对第一时域信号和第二时域信号分别进行截取，得到第三时域信号和第四时域信号，所述第二时域信号在时域上相比所述第一时域信号延迟半个周期，所述第三时域信号所占时间长度为所述第一时域信号所占时间长度的一半，所述第四时域信号所占时间长度为所述第二时域信号所占时间长度的一半；将所述第三时域信号和第四时域信号进行混合，得到第五时域信号；发射所述第五时域信号，所述半个周期为所述第一时域信号所占时间长度的一半。
在一个可选的实现中，采用IFFT/2窗函数对所述第一时域信号和第二时域信号分别进行截取，将所述第一时域信号或第二时域信号头部和尾部的多个小于预设阈值的数据截掉。
在一个可选的实现中，当所述插值率为4时，所述发射所述时域信号，进一步包括：所述时域信号包括第六时域信号、第七时域信号、第八时域信号和第九时域信号，所述第六时域信号所占的时间长度、所述第七时域信号所占的时间长度、所述第八时域信号所占的时间长度以及所述第九时域信号所占的时间长度相同；对第六时域信号、第七时域信号、第八时域信号和第九时域信号分别进行截取，得到第十时域信号、第十一时域信号、第十二时域信号和第十三时域信号，所述第七时域信号在时域上相比所述第六时域信号延迟四分之一周期、所述第八时域信号在时域上相比所述第七时域信号延迟四分之一周期、所述第九时域信号在时域上相比所述第八时域信号延迟四分之一周期，所述第十时域信号所占时间长度为所述第六时域信号所占时间长度的四分之一，所述第十一时域信号所占时间长度为所述第七时域信号所占时间长度的四分之一，所述第十二时域信号所占时间长度为所述第八时域信号所占时间长度的四分之一，所述第十三时域信号所占时间长度为所述第九时域信号所占时间长度的四分之一；将所述第十时域信号、第十一时域信号、第十二时域信号和第十三时域信号进行混合，得到第十四时域信号；发射所述第十四时域信号，所述四分之一周期为所述第六时域信号所占时间长度的四分之一。
在一个可选的实现中，采用IFFT/4窗函数对所述第六时域信号、第七时域信号、第八时域信号和第九时域信号分别进行截取，将所述第六时域信号、第七时域信号、第八时域信号和第九时域信号头部和尾部的多个小于预设阈值的数据截掉。
本发明实施例可以构造时域更短的时间间隔，达到更宽子载波间隔的效果。同时不同业务之间不存在互相ICI干扰问题。可以节省频域保护间隔，有效提高吞吐量。
在一个可选的实现中，所述插值操作包括下述中的一种或多种：离散傅里叶变换DFT插值、样条插值、一阶插值和高阶插值。
第二方面，本发明实施例提供了又一种数据传输方法，该方法包括：接收时域信 号；对所述时域信号进行快速傅里叶变换FFT，得到子载波上的第六信号序列；对第六信号序列进行解调，得到第七信号序列；对所述第七信号序列进行解插值操作，得到第八信号序列，所述第八信号序列包括数据符号的软信息，其中，第八信号序列的长度小于第七信号序列的长度；对所述数据符号的软信息进行译码，得到数据符号。
在一个可选的实现中，在所述对第六信号序列进行解调之前，还包括：获取导频符号；根据所述导频符号进行信道估计，得到信道相关信息；所述对第六信号序列进行解调，得到第七信号序列，包括：根据所述信道相关信息对第六信号序列进行解调，得到所述第七信号序列。
在一个可选的实现中，在所述对所述第七信号序列进行解插值操作之前，还包括：确定插值参数；所述对所述第七信号序列进行解插值操作，得到第八信号序列，包括：根据所述插值参数对所述第七信号序列进行解插值操作，得到第八信号序列。
在一个可选的实现中，所述解插值操作具体包括：对所述第七信号序列进行解第一相位旋转操作，得到第九信号序列；对所述第九信号序列进行解插值操作，得到第八信号序列。
在一个可选的实现中，所述解插值操作具体包括：对所述第七信号序列进行离散傅里叶逆变换IDFT，得到第十信号序列；删除所述第十信号序列头部的ZH个零，以及删除所述第十信号序列尾部的ZT个零，得到第十一信号序列；对所述第十一信号序列进行离散傅里叶变换DFT，得到所述第八信号序列，其中，所述第八信号序列的长度等于所述第七信号序列的长度减去ZH和ZT得到的数值。
在一个可选的实现中，所述对第六信号序列进行解调，得到第七信号序列，具体包括：对所述第六信号序列进行解调，得到至少一组所述第七信号序列。
在一个可选的实现中，还包括：接收所述插值参数相关的信息，所述插值参数相关的信息用于确定所述插值参数。
在一个可选的实现中，所述第七信号序列还包括导频符号，其中，所述导频符号在所述第七信号序列中符合第四预定义规则的位置；所述第四预定义规则为：所述导频符号在至少一个所述第七信号序列中每隔第四预定义个数的位置上，所述第四预定义个数根据所述插值参数确定。
在一个可选的实现中，所述解插值操作包括下述中的一种或多种：解离散傅里叶变换DFT插值、解样条插值、解一阶插值和解高阶插值。
第三方面，本发明实施例提供了一种通信设备，包括：插值单元，用于对第一信号序列进行插值操作，得到第二信号序列；映射单元，用于将所述第二信号序列映射到子载波，得到子载波上的第二信号序列；IFFT单元，用于对所述子载波上的第二信号序列进行快速傅里叶逆变换IFFT，得到时域信号；发射单元，用于发射所述时域信号。
在一个可选的实现中，还包括：确定单元，用于确定插值参数；所述插值单元，具体用于根据所述插值参数对所述第一信号序列进行插值操作，得到所述第二信号序列，所述第二信号序列的长度大于所述第一信号序列的长度。
在一个可选的实现中，还包括：获取单元，用于获取终端设备的信号的最大时延 偏差，所述终端设备的信号为发给所述终端设备的无线信号或所述终端设备发送的无线信号，所述最大时延偏差为所述终端设备的信号从发送端发出，经过无线信道，最早到达接收端的时间与最晚到达接收端的时间之差；所述确定单元，具体用于根据所述终端设备的信号的最大时延偏差确定所述插值参数。
在一个可选的实现中，所述插值单元，具体用于：对所述第一信号序列进行插值操作，得到第三信号序列；对所述第三信号序列进行第一相位旋转，得到第二信号序列。
在一个可选的实现中，所述插值单元，具体用于：对所述第一信号序列进行离散傅里叶逆变换IDFT，得到第四信号序列；在所述第四信号序列的头部增加ZH个零，以及在所述第四信号序列的尾部增加ZT个零，得到第五信号序列，其中，ZH和ZT均为大于等于零的整数，但ZH和ZT中至少有一个为正整数；对所述第五信号序列进行离散傅里叶变换DFT，得到所述第二信号序列，其中，第二信号序列的长度等于所述第一信号序列的长度、ZH以及ZT的总和。
在一个可选的实现中，所述映射单元，具体用于：将至少一组所述第二信号序列映射到子载波，得到至少一组子载波上的第二信号序列。
在一个可选的实现中，所述第一信号序列包括至少一个第一导频符号以及至少一个数据符号，其中，所述至少一个第一导频符号与所述至少一个数据符号按照第一预定义规则组成所述第一信号序列；所述第一预定义规则为：在第一信号序列中，每隔第一预定义个数的数据符号为候选的第一导频符号位置，选择至少一个所述候选的第一导频符号位置插入所述第一导频符号，其中，所述第一预定义个数根据所述插值参数确定。
在一个可选的实现中，所述第一信号序列包括至少一个零以及至少一个数据符号，其中，所述至少一个零与所述至少一个数据符号按照第二预定义规则组成所述第一信号序列；所述第二预定义规则为：在第一信号序列中，每隔第一预定义个数的数据符号为候选的插零位置，选择至少一个所述候选的插零位置插入零，其中，所述第一预定义个数根据所述插值参数确定。
在一个可选的实现中，所述第一导频符号是第二导频符号经过第二相位旋转而得到的，所述第二导频符号用于接收端进行信道测量和信道估计中的至少一个。
在一个可选的实现中，所述映射单元，具体还用于：每隔第二预定义个数的子载波，用第二导频符号替换所述子载波上的第二信号序列中的符号，所述第二预定义个数根据所述插值参数确定，其中，所述第二预定义个数大于所述第一预定义个数，所述第二导频符号用于接收端进行信道测量和信道估计中的至少一个。
在一个可选的实现中，还包括：发送单元，用于发送所述插值参数相关的信息，所述插值参数相关的信息用于接收端确定所述插值参数。
第四方面，本发明实施例提供了又一种通信设备，包括：接收单元，用于接收时域信号；FFT单元，用于对所述时域信号进行快速傅里叶变换FFT，得到子载波上的第六信号序列；解调单元，用于对第六信号序列进行解调，得到第七信号序列；解插值单元，用于对所述第七信号序列进行解插值操作，得到第八信号序列，所述第八信 号序列包括数据符号的软信息，其中，第八信号序列的长度小于第七信号序列的长度；译码单元，用于对所述数据符号的软信息进行译码，得到数据符号。
在一个可选的实现中，还包括：获取单元，用于获取导频符号；信道估计单元，用于根据所述导频符号进行信道估计，得到信道相关信息；所述解调单元，具体用于根据所述信道相关信息对第六信号序列进行解调，得到所述第七信号序列。
在一个可选的实现中，还包括：确定单元，用于确定插值参数；所述解插值单元，具体用于根据所述插值参数对所述第七信号序列进行解插值操作，得到第八信号序列。
在一个可选的实现中，所述解插值操作具体包括：对所述第七信号序列进行解第一相位旋转操作，得到第九信号序列；对所述第九信号序列进行解插值操作，得到第八信号序列。
在一个可选的实现中，所述解插值单元，具体用于：对所述第七信号序列进行离散傅里叶逆变换IDFT，得到第十信号序列；删除所述第十信号序列头部的ZH个零，以及删除所述第十信号序列尾部的ZT个零，得到第十一信号序列；对所述第十一信号序列进行离散傅里叶变换DFT，得到所述第八信号序列，其中，所述第八信号序列的长度等于所述第七信号序列的长度减去ZH和ZT得到的数值。
在一个可选的实现中，所述解插值单元，具体用于：对所述第六信号序列进行解调，得到至少一组所述第七信号序列。
在一个可选的实现中，还包括：接收单元，用于接收所述插值参数相关的信息，所述插值参数相关的信息用于接收端确定所述插值参数。
基于上述技术方案，本发明实施例提供的数据传输方法及通信设备，发送端可以对映射到频域的调制符号进行插值，在时域上体现了OFDM符号自适应零功率ZT的效果，使得发射的时域信号可以更好地对抗时延偏差。
附图说明
为了更清楚地说明本发明实施例中的技术方案，下面将对实施例中所需要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本发明的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据这些附图获得其他的附图。
图1为本发明实施例提供的通信系统架构图；
图2为本发明实施例提供的数据传输方法流程示意图；
图3为本发明实施例提供的基于频域广义插值的数据传输方法与装置；
图4为本发明实施例提供的插值矩阵示意图；
图5为本发明实施例提供的基于频域DFT插值的数据传输方法与装置；
图6为本发明实施例提供的等效的基于频域广义插值的数据传输方法与装置；
图7为本发明实施例提供的等效的基于频域DFT插值的数据传输方法与装置；
图8为本发明实施例提供的基于频域分块广义插值的的数据传输方法与装置；
图9为本发明实施例提供的另一种数据传输方法流程图；
图10为本发明实施例提供的基于频域广义插值的数据传输方法与装置；
图11为本发明实施例提供的基于频域DFT插值的数据传输方法与装置；
图12为本发明实施例提供的CSI-RS的导频设计图样示意图；
图13为本发明实施例提供的一种DMRS的导频设计图样示意图；
图14为本发明实施例提供的又一种DMRS的导频设计图样示意图；
图15为本发明实施例提供的PNRS的导频设计图样示意图；
图16为本发明实施例提供的基于频域DFT插值导频增强方案的数据传输方法和装置；
图17为本发明实施例提供的一种DMRS的增强导频设计图样示意图；
图18为本发明实施例提供的又一种DMRS的增强导频设计图样示意图；
图19为本发明实施例提供的应用于多发射点异步数据传输方法示意图；
图20为本发明实施例提供的更短时间间隔的数据传输方法示意图；
图21为本发明实施例提供的一种通信设备架构图；
图22为本发明实施例提供的又一种通信设备架构图；
图23为本发明实施例提供的另一种通信设备架构图；
图24为本发明实施例提供的再一种通信设备架构图。
具体实施方式
下面将结合附图，对本发明实施例中的技术方案进行清楚、完整地描述。显然，所描述的实施例仅仅是本发明一部分实施例，而不是全部的实施例。基于本发明中的实施例，本领域普通技术人员在没有付出创造性劳动前提下所获得的所有其他实施例，都属于本发明保护的范围。
本发明实施例描述的网络架构以及业务场景是为了更加清楚的说明本发明实施例的技术方案，并不构成对于本发明实施例提供的技术方案的限定，本领域普通技术人员可知，随着网络架构的演变和新业务场景的出现，本发明实施例提供的技术方案对于类似的技术问题，同样适用。
本发明实施例描述的技术可以适用于LTE系统后续的演进系统，如第五代5G系统等。本发明实施例中，名词“网络”和“系统”经常交替使用，但本领域的技术人员可以理解其含义。
图1为本发明实施例提供的通信系统架构图。如图1所示，包括多个发送装置和多个接收装置。多个发送装置和接收装置之间进行上行或下行的数据传输。
本发明各实施例中的发送装置和接收装置可以为以无线方式进行数据传输的任意一种发送端的装置和接收装置的装置。发送装置和接收装置可以是任意一种具有无线收发功能的装置，包括但不限于：基站NodeB、演进型基站eNodeB、未来第五代(the fifth generation，5G)通信系统中的基站、WiFi系统中的接入节点、无线中继节点、无线回传节点以及用户设备(user equipment，终端设备)。其中，终端设备也可以称之为终端Terminal、移动台(mobile station，MS)、移动终端(mobile terminal，MT)等，终端设备可以经无线接入网(radio access network，RAN)与一个或多个核心网进行通信，终端设备也可以与其它终端设备直接进行无线通信。
本发明各实施例可以应用于无线通信系统中的基站到终端设备下行数据传输、终端设备到基站的上行数据传输、设备到设备(device to device，D2D)数据传输以及无 线回传的数据传输等场景，本发明实施例对应用场景不做限定。
以图1中所示的终端设备1至终端设备3，以及基站1、基站2为例，说明本发明实施例的应用场景。
在一个可能的场景中，基站1和基站2对终端设备1进行下行协作多点传输。当基站1与基站2使用的时钟源不同时，导致基站1和基站2发送的信号之间存在一定的时延偏差。
在另一个可能的场景中，终端设备1和终端设备2两个设备与基站1进行通信。终端设备1和终端设备2所处的无线环境差异很大，导致终端设备1和终端设备2的信号的最大时延偏差(时延扩展)相差很大。
本发明实施例提供一种数据传输方法，根据终端设备的信号的时延偏差，对基站或者终端设备发射端发出的信号配置一定的插值参数，形成不同长度的零功率尾部，从而克服以上的时延偏差。终端设备的信号为发给所述终端设备的无线信号或所述终端设备发送的无线信号。
具体地，终端设备的信号的时延偏差还可能包括不同发射端相对于接收端的信号传播带来的时延偏差，以及无线信道本身的时延扩展。
在一种可能的实施方式中，发射端可以通过自适应的方式配置插值参数。如，发射端根据获取到的最大信道时延偏差，配置相应的插值参数。或者发射端初始接入无线网络时，发射端采用一定的插值参数以配置较长的零功率尾部，可对抗较大的时延偏差。后续过程中，发射端可以根据获取到的最大信道时延偏差，重配插值参数，进而保证数据传输效率。
本发明实施例提供的数据传输方法，为了自适应调整零功率尾部(Zero Tail，ZT)，接收端测量当前时刻的信道最大时延偏差，通过上行控制消息将量化的时延偏差反馈给发送端；或者发送端直接测量获得信道最大时延偏差。发送端根据获得的最大时延偏差选择合适的插值参数进行ZT-OFDM调制。同时发送端将选择的插值参数通过控制消息发送给接收端，保证接收端进行正确的解插值操作。
可以理解的是，基站和终端设备基于OFDM符号进行数据传输，本发明实施例提供的数据传输方法又可理解为自适应ZT的OFDM符号传输方法。
图2为本发明实施例提供的数据传输方法流程示意图。在本实施例中实施主体为发射装置。如图2所示，该实施例包括以下步骤：
步骤S101，发送装置对第一信号序列进行插值操作，得到第二信号序列，所述第二信号序列的长度大于所述第一信号序列的长度。
可以理解的是，对于基站到终端设备的下行数据传输，发送装置为基站；对于终端设备到基站的上行数据传输，发送装置为终端设备；对于D2D数据传输，发送装置为终端设备；对于无线回传的数据传输，发送装置为无线回传节点。
优选地，在所述对第一信号序列进行插值操作之前，还包括：确定插值参数；所述对第一信号序列进行插值操作，得到第二信号序列，包括：根据所述插值参数对所述第一信号序列进行插值操作，得到所述第二信号序列。
优选地，在所述确定插值参数之前，还包括：获取终端设备的信号的最大时延偏 差，所述终端设备的信号为发给所述终端设备的无线信号或所述终端设备发送的无线信号，所述最大时延偏差为所述终端设备的信号从发送端发出，经过无线信道，最早到达接收装置的时间与最晚到达接收装置的时间之差；所述确定插值参数，包括：根据所述终端设备的信号的最大时延偏差确定所述插值参数。这里的最大时延偏差可以包括以下时延偏差中的至少一个：不同的发射点之间的时间不同步导致的时延偏差；不同发射点到达接收装置的信号传播时延偏差；同一个发射点的信号到达接收装置的时延扩展。
可以理解的是，可以通过终端设备对发给终端设备的信号进行测量获得最大时延偏差，然后报告给网络设备；或者，通过网络设备对来自终端设备的信号进行测量获得最大时延偏差；或者，对于设备到设备(Device to Device，D2D)的通信场景，通过终端设备对来自另一个终端设备的信号进行测量获得最大时延偏差，然后报告给网络设备。这里所说的网络设备可以是基站。
所述插值操作具体可包括：离散傅里叶变换DFT插值、样条插值、一阶插值和高阶插值等典型插值算法。
优选地，所述插值操作具体包括：对所述第一信号序列进行离散傅里叶逆变换IDFT，得到第四信号序列；在所述第四信号序列的头部增加ZH个零，以及在所述第四信号序列的尾部增加ZT个零，得到第五信号序列，其中，ZH和ZT均为大于零的整数；对所述第五信号序列进行离散傅里叶变换DFT，得到所述第二信号序列，其中，第二信号序列的长度等于所述第一信号序列的长度、ZH以及ZT的总和。
优选地，所述第一信号序列包括至少一个数据符号。
优选地，所述第一信号序列包括至少一个第一导频符号，所述第一导频符号为第二导频符号经过第三相位旋转而得到的，所述第二导频符号用于接收装置进行信道测量和信道估计中的至少一个。
优选地，所述第一信号序列还包括至少一个数据符号，其中，所述至少一个第一导频符号与所述至少一个数据符号按照第一预定义规则组成所述第一信号序列；所述第一预定义规则为：在第一信号序列中，每隔第一预定义个数的数据符号为候选的第一导频符号位置，选择至少一个所述候选的第一导频符号位置插入所述第一导频符号，其中，所述第一预定义个数根据所述插值参数确定。
步骤S102，将所述第二信号序列映射到子载波，得到子载波上的第二信号序列。
优选地，将至少一组所述第二信号序列映射到子载波，得到至少一组子载波上的第二信号序列。
步骤S103，对所述子载波上的第二信号序列进行快速傅里叶逆变换IFFT，得到时域信号。
具体地，对于多层MIMO传输，在对所述子载波上的第二信号序列进行IFFT之前，还包括层映射、预编码等操作，可参照相关现有技术，在此不做赘述。
所述时域信号可称作OFDM符号，该OFDM符号占用的频率资源为小区的系统带宽。经过插值处理及IFFT后的时域信号可以进行加ZP或加CP操作，以使得OFDM符号达到预定义的时间长度，并进一步消除符号间干扰。
步骤S104，发射所述时域信号。
具体地，发射所述时域信号。或者发射加ZP或CP的时域信号。
进一步地，发送所述插值参数相关的信息，所述插值参数相关的信息用于接收装置确定所述插值参数。
下面以图3为例，说明本发明实施例提供的插值操作的具体示例。
图3为本发明实施例提供的基于频域广义插值的数据传输方法与装置。设该实施例中的插值参数为(m,n)。假如系统为某个用户分配的资源为N个子载波，该方案需要在插值之前共产生M个符号，通过(m,n)的分数倍插值滤波器将M个符号在频域插值为N个符号，同时令
Figure PCTCN2017103012-appb-000001
T为对应整数倍插值点的个数。则该实施例主要包括以下步骤：
步骤301，用户数据比特经过信道编码、速率匹配、加扰、调制后产生至少一个数据符号。
具体地，所述至少一个数据符号可为：M-L个QAM符号。
步骤302，根据小区号、帧号等产生至少一个第二导频符号。第二导频符号用于接收装置进行信道测量和信道估计中的至少一个。其中，为了保证接收装置可以透明的接收导频符号，并对所述至少一个第二导频符号进行第二相位旋转得到至少一个第一导频符号。
具体地，所述至少一个第二导频符号可以为L(L≤T)个导频符号，所述第二相位旋转可以表示为乘上相位旋转因子
Figure PCTCN2017103012-appb-000002
其中i∈[0,T)，i根据第一导频符号具体插入步骤303中第一信号序列哪些候选第一导频符号位置来确定，ZH为插值处理之后时域对应的N个数据的前端近似为零的个数。
需要说明的是，接收装置透明的接收导频符号指的是，发送端是否采用插值方案对接收装置接收导频来说是透明的，即对于发送端采用插值方案与发送端不采用插值方案两种场景，接收装置导频的处理是完全一样的。
步骤303，将所述至少一个第一导频符号与所述至少一个数据符号按照第一预定义规则组成所述第一信号序列。
其中，所述第一预定义规则为：在第一信号序列中，每隔第一预定义个数的数据符号为候选的第一导频符号位置，选择至少一个所述候选的第一导频符号位置插入所述第一导频符号，其中，所述第一预定义个数根据所述插值参数确定。
具体地，所述第一预定义个数为m。先对至少一个数据符号与第一导频符号进行串并转换(Serial/Parallel，S/P)。将经过S/P转换的至少一个数据符号与第一导频符号混插排列为M长度的第一信号序列。其中，第一导频符号需要插入在整数倍插值点位置，该位置对应M长度的第一信号序列的第i*m的位置，其中i＝0,1,…,T-1，经过(m,n)的分数倍插值后则对应N长度的序列的第i*n的位置，并且保持值不变。
需要说明的是，第一导频符号的个数小于或等于T，整数倍插值点为所述候选第一导频符号位置。
步骤304，对所述第一信号序列进行插值操作，得到第三信号序列。
具体地，将M长度的第一信号序列经过(m,n)的分数倍插值(Interpolation)滤波器，输出长度为N的插值后第三信号序列。
其中，该插值滤波器可以为各种典型的插值算法，如DFT插值、样条插值、一阶 插值、高阶插值等。
需要说明的是，本发明实施例提供的插值滤波器的操作相当于在现有LTE系统多载波调制的单位矩阵前面又乘了一个矩阵。具体地，多载波由于直接映射，因此可以认为预编码矩阵为单位矩阵。
本发明实施例涉及的插值操作对应的矩阵可参照图4所示。图4为插值率为(5,6)，m＝5,n＝6,M＝20,N＝24对应的24×20插值矩阵。如图4所示，将插值前的M个数据排列为20×1的矩阵，使用24×20插值矩阵乘以M个数据对应的20×1矩阵，得到N个数据对应的24×1的矩阵。
在图4中，每5列，对应的列中出现一个1。其他列上仅在对角线两边的位置有一定数值，其他位置上的矩阵元素均为0。图4所示的矩阵以5×6的矩阵单元在对角线上循环了4次。
根据矩阵算法可知，将第1行至第5行的元素分别与M个数据符号组成的20×1的矩阵相乘，得到相应的6个数据。且该6个数据仅仅与M个数据中前5个数据相关。以此类推循环。如果在M个数据中，第一导频符号放置的位置对应插值矩阵中的元素始终为1，那么插值前后的导频符号幅值不变。
需要说明的是，图4仅示出了插值矩阵的一种可能方式，对角线两边有一定数值的位置可不止包括图4示出的情况，对角线两边有一定数值的数据还可包括其他情况。
可以理解的是，该矩阵的大小以及最小循环单元取决与插值参数(m,n)、以及M、N的数值。
另外，为了保持插值前后的功率谱密度归一，插值前的M长度的第一信号序列可以乘上放大因子
Figure PCTCN2017103012-appb-000003
步骤305，对所述第三信号序列进行第一相位旋转，得到第二信号序列。
具体地，第一相位旋转可以表示为乘上相位旋转因子
Figure PCTCN2017103012-appb-000004
其中i＝0,1,…,N-1。第三信号序列的长度为N，第二信号序列的长度为N。即第二信号序列包括N个符号。
需要说明的是，将插值后的第三信号序列进行第一相位旋转，得到第二信号序列。该操作在时域体现为信号的时移，ZH为时域对应的N个数据的前端近似为零的个数，尾端近似为零的个数为ZT＝N-M-ZH。其中，ZH和ZT均为大于等于零的整数，但ZH和ZT中至少有一个为正整数。
其中，插值操作可直观上理解为在M个数据组成的第一信号序列的基础上产生了ZH+ZT个近似零的数据。
需要说明的是，第一相位旋转、第二相位旋转配合，以达到插值后第二信号序列中的导频符号与第二导频符号的幅值和相位相同。进一步地，接收装置可以识别发送端根据小区号、帧号等产生的第二导频符号，因此第一相位旋转、第二相位旋转使得第二导频符号相对接收装置透明。
图3所示为广义的插值方式。当使用具体如DFT插值、样条插值等方式插值时，可根据具体需要，配合第一相位旋转、第二相位旋转或其他方式的相位旋转等操作使得第二导频符号相对接收装置透明。
步骤306a，将所述第二信号序列映射到子载波，得到子载波上的第二信号序列。
具体地，将第一相位旋转后的N个符号对应的第二信号序列连续映射到频域的N个子载波上。
步骤306b，对所述子载波上的第二信号序列进行快速傅里叶逆变换IFFT，得到时域信号。
步骤307，对最后经过并串转换(Parallel/Serail，P/S)的时域信号进行加ZP或者CP的操作，其中,加ZP为在该时域信号后面添加Nzp个零值，加CP的操作为将时域信号最后Ncp个值拷贝到该串时域信号的最前端，加CP操作需要满足
Figure PCTCN2017103012-appb-000005
其中，FFTSize为FFT大小。
上述图2所示的数据传输方法，还可采用其他插值方式实现。
由于DFT快速算法有成熟的芯片实现，本发明实施例还提供了一种基于频域DFT插值的数据传输方法与装置，如图5所示，包括步骤501至步骤506。
其中，插值参数以及系统为用户分配资源、插值前需产生符号个数可参见图3中的描述。
需要说明的是，步骤501、步骤503、步骤505a、步骤505b、步骤506同样可分别参照步骤301、步骤303、步骤306a、步骤306b、步骤307中的描述。为简化说明，以下不再赘述。
步骤502，根据小区号、帧号等产生至少一个第二导频符号。其中，为了保证接收端可以透明的接收导频符号，并对所述至少一个第二导频符号进行第三相位旋转得到至少一个第一导频符号。
具体地，所述至少一个第二导频符号可以为L(L≤T)个导频符号，所述第三相位旋转可以表示为乘上相位旋转因子
Figure PCTCN2017103012-appb-000006
其中i∈[0,T)，i根据第一导频符号具体插入步骤503中第一信号序列哪些候选第一导频符号位置来确定。
步骤504，对所述第一信号序列进行离散傅里叶逆变换IDFT，得到第四信号序列；在所述第四信号序列的头部增加ZH个零，以及在所述第四信号序列的尾部增加ZT个零，得到第五信号序列，其中，ZH和ZT均为大于等于零的整数，但ZH和ZT中至少有一个为正整数；对所述第五信号序列进行离散傅里叶变换DFT，得到所述第二信号序列，其中，第二信号序列的长度等于所述第一信号序列的长度、ZH以及ZT的总和。
可以理解的是，当ZH等于零，ZT大于零时，等效于在第四信号序列的尾部增加ZT个零，得到第五信号序列；当ZH大于零，ZT等于零时，等效于在第四信号序列的头部增加ZH个零，得到第五信号序列；当ZH和ZT都大于零时，等效于在第四信号序列的头部增加ZH个零，以及在第四信号序列的尾部增加ZT个零，得到第五信号序列。
其中，为了保持插值后的功率谱密度归一，将M长度的第一信号序列在插值前乘上放大因子
Figure PCTCN2017103012-appb-000007
具体地，第四信号序列为经过M点的IDFT变换到时域得到的时域序列。然后对M点的时域序列头部增加ZH个零，尾部增加ZT个零，将M点的时域序列长度扩展到N，得到第五信号序列。最后对第五信号序列经过N点DFT变换回频域序列得到第二信号 序列。
可以理解的是，当L等于零时，则第一信号序列仅包括数据符号；当L等于M时，则第一信号序列仅包括第一导频符号；当L大于零且小于M时，则第一信号序列包括至少一个数据符号以及至少一个第一导频符号。本发明的附图及实施例仅以第一信号序列包括至少一个第一导频符号以及至少一个数据符号为例进行说明，但本发明实施例的方法和处理流程同样适用于第一信号序列仅包括第一导频符号以及第一信号序列仅包括数据符号的场景。
本发明实施例提供的数据传输方法，利用在频域插值等效到时域即为信号两边添零的效果，即在一个变换域进行插值，在另一个变换域获得信号添零的效果。本发明实施例提供的数据传输方法，可适用于OFDM系统。另外，结合本发明插值的方式，对抗信道时延偏差的方案，还可适用于其他系统。本领域技术人员应当理解的是，其他类似本发明实施例的等价方案，均应属于本发明实施例的保护范围内。
本发明实施例提供的数据传输方法又可称为一种基于OFDM系统的自适应ZT波形生成方法。本发明实施例根据获得的信道最大时延偏差，发送端对映射到频域的调制符号进行一定比率的分数倍插值滤波操作，达到时域信号序列头部ZH个数据和尾部ZT个数据近似零功率的效果，从而达到降低ISI的目的。
相应地，由于整数倍插值位置的值在插值前后可以保持幅度不变，因此在发送端也可以通过另一种等效方式来实现。
下面以图6、图7为例，说明本发明实施例提供的插值操作的另外两个等效的具体示例。
图6为本发明实施例提供的一种等效的基于频域广义插值的数据传输方法与装置。
在一个可能的设计中，所述第一信号序列包括至少一个零以及至少一个数据符号，其中，所述至少一个零与所述至少一个数据符号按照第二预定义规则组成所述第一信号序列；所述第二预定义规则为：在第一信号序列中，每隔第一预定义个数的数据符号为候选的插零位置，选择至少一个所述候选的插零位置插入零，其中，所述第一预定义个数根据所述插值参数确定。在本实施例中，第一预定义个数可以等于m-1。
步骤601，在插值前本来需要插入导频符号的第i*m的位置插入零值，这样在插值后对应的整数倍插值第i*n的位置的值也是零。
进一步地，所述将至少一组所述第二信号序列映射到子载波，得到至少一组子载波上的第二信号序列，进一步包括：每隔第二预定义个数的子载波为候选的第二导频符号位置，用第二导频符号替换至少一个所述候选的第二导频符号位置所在子载波上的第二信号序列中的符号，所述第二预定义个数根据所述插值参数确定，其中，所述第二预定义个数大于所述第一预定义个数，所述第二信号序列中，所述第二导频符号的替换位置与所述第一信号序列中插入所述零的位置相关，所述第二导频符号用于接收端进行信道测量和信道估计中的至少一个。在本实施例中，第二预定义个数可以等于n-1。
步骤602，可以在插值后第i*n的位置直接插入第二导频符号。
另外，图6中其他未指出的步骤可参见图3中描述，在此不做赘述。
相应地，图7为本发明实施例提供的一种等效的基于频域DFT插值的数据传输方法与装置。步骤701、步骤702可参见步骤601、步骤602中的详细介绍。另外，图7中其他未指出的步骤可参见图5中描述，在此不做赘述。
进一步地，由于图5或图7所示的插值算法基于DFT插值，为了便于采用成熟的DFT芯片，需要保证插值参数(m,n)均满足基2、3、5的要求。即参数m和n均为2、3、5的倍数。另外定义比值
Figure PCTCN2017103012-appb-000008
该值越大则插值率越高，对应的ZT越长，对抗异步的能力也越强。反之该值越小则插值率越低，对应的ZT越短，对抗异步的能力也越弱，但是传输的有效数据越多。因此可以对一定的异步范围内进行一定量化，为了满足以上两个需求，表1为基于频域DFT插值的多载波OFDM插值参数配置表，给出了一种可能的插值参数组合。表中对应的是在频域插值后还对时域信号加了正常ZP/CP的场景。可以理解的是，按照相同的方法，可以给出时域信号没有加ZP/CP的场景的差值参数，在此不做赘述。
表1基于频域DFT插值的多载波OFDM插值参数配置表
Figure PCTCN2017103012-appb-000009
其中开销和最大时延基于1ms划分为14个OFDM符号，每个符号的ZP/CP约为4.7us，开销Overhead Ratio的计算公式为：
Figure PCTCN2017103012-appb-000010
最大对抗时延偏差MaxDelay Deviation计算公式为：
Figure PCTCN2017103012-appb-000011
需要说明的是，本发明实施例提供的数据传输方法还包括：发送所述插值参数相关的信息，所述插值参数相关的信息用于接收端确定所述插值参数。
具体地，可以通过控制信道或者数据信道发送插值参数指示信息给接收端，如表1所示，该指示信息可以用3比特表示。另外，也可以不发送插值参数指示信息，由接收端根据不同插值配置下导频的图样差异进行盲检来识别。
相应地，本发明实施例将自适应ZT-OFDM方案与传统的Normal CP/Extended CP自适应切换的性能进行了对比，Normal CP超过4us时延偏差不能工作，且Extended CP超过16us时延偏差不能工作。本发明实施例提供的自适应ZT-OFDM方案可在0us至24us时延偏差的范围内工作，且在不同时延偏差下配置合适的插值参数可使本发明实 施例的工作点稳定变化。并且在0us至12us时延偏差范围内比Extended CP工作点要好。本发明实施例提供的自适应ZT-OFDM方案工作点相比Extended CP的工作的具有更低的信噪比(Signal Noise Rate，SNR)。不同的时延偏差下对应不同的最佳插值参数，其工作点可以支持相比Extended CP更长的范围。另外，由于不同插值参数将时延粒度划分更细，相比传统自适应切换工作点变化更平滑。
前述图3、图5、图6及图7的方案为了保持插值的特性，需要目标用户的资源在频域连续来保证时域上ZT的效果，本发明实施例还提供一种分块插值的方式，来获得一定程度的灵活调度的效果。
优选地，所述将第二信号序列映射到子载波，得到子载波上的第二信号序列，具体可包括：将至少一组所述第二信号序列映射到子载波，得到至少一组子载波上的第二信号序列。
设插值前的信号序列包括M0信号序列、M1信号序列…Mk信号序列。可对M0信号序列、M1信号序列…Mk信号序列分别进行如前述插值参数(m，n)的插值操作，分别得到N0信号序列、N1信号序列…Nk信号序列。其中，至少一组第二信号序列可包括N0信号序列、N1信号序列…Nk信号序列中的一个或多个。
具体对每个信号序列的插值方式可采用图3、图5、图6及图7所示方案中的任意一种或多种，在此不做赘述。
下面以图8为例，说明本发明实施例提供的插值操作的又一具体示例。图8为本发明实施例提供的基于频域分块广义插值的的数据传输方法与装置。
图8仅给出基于频域广义插值的方式的实现，分块插值与不分块插值的差异在于：将用户调制后的符号分成K组，同时导频符号也对应分成K组，每组的数据大小可以不同，由调度器决定。每一组的数据符号和导频符号后续完全按照图3、图5、图6及图7所示的方式实现。插值后的数据根据调度器分配的资源分别独立的映射到频域子载波上，每一组的子载波连续，但是不同组之间可以是不连续的。
本发明实施例通过分块插值的方式可以使得一个用户的资源分块离散地分布在传输带宽内，这样对于目标用户可以获得更好的频域分集效果，同时对于其它用户的调度也可以更灵活的支持。
可以理解的是，分块差值也可以与其它差值方案进行结合，例如可以有基于频域DFT差值的分块差值方案，在此不做赘述。
本发明实施例提供的方案通过配置多个不同的插值参数，相比现有的Normal CP/Extended CP有更多的选择。本发明实施例提供的方案在不同的时延偏差下对应不同的最佳插值参数，可以支持相比Extended CP更长时延偏差范围。本发明实施例提供的方案相比ZT-s-OFDM方案具备可以在频域上间隔插导频的优势，因此可以更灵活地支持MIMO传输的多天线口复用。
需要说明的是，多载波OFDM系统相比于单载波S-OFDM系统的一个优势是可以灵活的调度，也就是一个用户的资源可以离散地分布在传输带宽内，这样对于目标用户可以获得更好的频域分集效果，对于其它用户的调度也可以更灵活的支持。
同时本发明实施例提供的方案可以自适应配置不同时延偏差下的插值参数，提高 数据传输效率，增加吞吐率。进一步地，本发明实施例由于可以在频域上间隔插导频，因此可以更灵活地支持MIMO传输的多天线口复用。
需要说明的是，本发明实施例中，接收装置需要对接收到的时域信号转换到频域，并进行频域解插值等操作，得到用户的数据。下面结合附图9，详细说明本发明实施例提供的方案。图9为本发明实施例提供的另一种数据传输方法流程图，该实施例包括以下步骤：
步骤S201，接收装置接收时域信号。
优选地，还包括接收插值参数相关的信息，所述插值参数相关的信息用于确定解插值操作使用的插值参数。
步骤S202，对所述时域信号进行快速傅里叶变换FFT，得到子载波上的第六信号序列。
步骤S203，对第六信号序列进行解调，得到第七信号序列。
优选地，在所述对第六信号序列进行解调之前，还包括：获取导频符号；根据所述导频符号进行信道估计，得到信道相关信息。
具体地，所述信道相关信息可包括信道因子和干扰噪声等信息。
需要说明的是，接收的时域信号为发射的信号经过信道加权和干扰噪声叠加后的信号。设R为接收信号，h为信道因子，S为发送信号，n为干扰噪声，则R＝h*S+n。其中，为方便区分，可设发送信号S中的导频符号为Sp，数据符号为Sd。
具体地，第六信号序列包括数据符号Sd的接收信号Rd和导频符号Sp的接收信号Rp。
需要说明的是，接收端和发射端可通过控制消息事先协商好收发的导频序列Sp和导频图样位置等信息。接收端最终接收的导频符号的接收信号为Rp＝h*Sp+n。
进一步，接收端根据导频符号Sp和导频符号的接收信号Rp估计信道相关信息：信道因子
Figure PCTCN2017103012-appb-000012
和干扰噪声矩阵
Figure PCTCN2017103012-appb-000013
优选地，所述对第六信号序列进行解调，得到第七信号序列，包括：根据所述信道相关信息对第六信号序列进行解调，得到所述第七信号序列。
具体地，在使用导频符号Sp及导频符号的接收信号Rp估计出信道因子
Figure PCTCN2017103012-appb-000014
和干扰噪声矩阵
Figure PCTCN2017103012-appb-000015
后，可使用Rd、
Figure PCTCN2017103012-appb-000016
和
Figure PCTCN2017103012-appb-000017
解调出对应数据符号
Figure PCTCN2017103012-appb-000018
需要说明的是，由于
Figure PCTCN2017103012-appb-000019
和
Figure PCTCN2017103012-appb-000020
是估计值，故根据
Figure PCTCN2017103012-appb-000021
和
Figure PCTCN2017103012-appb-000022
解调出来的数据符号
Figure PCTCN2017103012-appb-000023
也属于估计值，估计值可能会与Sd的真实值存在一定的误差，因此将上述解调出的信息称为软信息。也就是说，解调的数据符号
Figure PCTCN2017103012-appb-000024
为对应发射端发出的数据符号的软信息。
可以理解的是，解调得到的第七信号序列包括数据符号的软信息
Figure PCTCN2017103012-appb-000025
需要说明的是，接收装置还根据所述信道因子
Figure PCTCN2017103012-appb-000026
和干扰噪声矩阵
Figure PCTCN2017103012-appb-000027
进行信道测量，并反馈发射装置该信道的质量信息、以信道时延相关信息。在本发明实施例中，将不再对此做特别说明。
步骤S204，对所述第七信号序列进行解插值操作，得到第八信号序列，所述第八信号序列包括数据符号的软信息，其中，第八信号序列的长度小于第七信号序列的长度。
优选地，在所述对所述第七信号序列进行解插值操作之前，还包括：确定插值参数；所述对所述第七信号序列进行解插值操作，得到第八信号序列，包括：根据所述插值参数对所述第七信号序列进行解插值操作，得到第八信号序列。
所述解插值操作包括下述中的一种或多种：解DFT插值、解样条插值、解一阶插值和解高阶插值。
可以理解的是，第七信号序列对应发射装置的第二信号序列的软信息，第八信号序列对应发射装置第一信号序列的软信息。第七信号序列包括数据符号的软信息。经过接插值操作后，所述第八信号序列包括对应的解插值后的数据符号的软信息。
进一步地，抽取第八信号序列中的数据符号的软信息，以进行下一步处理。
步骤S205，对所述数据符号的软信息进行译码，得到数据符号。
具体地，在第八信号序列中除去导频符号，抽取除去导频符号的所有数据符号的软信息和信道相关信息，进行译码等操作，得到数据符号。所述数据符号对应用户的数据。
下面以图10为例，说明本发明实施例提供的解插值操作的具体示例。
图10为本发明实施例提供的一种基于频域广义插值的数据传输方法与装置。如图10所示，该实施例主要包括以下步骤：
步骤1001，接收时域信号。
进一步地，对接收的时域信号进行去除ZP或者CP的操作。
其中，去除ZP可以参用最简单的交叠和相加OLA(OverLap and Add)方式，即将一个时域符号的尾部Nzp个信号加回到最前端的Nzp个数据上，构成本符号时域信号在FFTSize大小内的自循环特性，去除CP则直接将最前端的Ncp个数据删除。
步骤1002，对所述时域信号进行快速傅里叶变换FFT，得到子载波上的第六信号序列。
具体地，在对去除ZP或者CP后的时域信号FFT之前，先将去除ZP或者CP后的时域信号经过S/P转换。
步骤1003，对第六信号序列进行解调，得到第七信号序列。
具体地，抽取出该用户的所有N个子载波信号，由于导频符号放置在整数倍插值位置而数值保持不变，接收装置可以按照传统方式，例如步骤S203所示的方式根据导频的位置进行频域信道估计、噪声估计、干扰噪声协方差矩阵估计等测量操作，同时利用估计出来的频域信道相关信息进行解码操作，得到频域第六信号序列对应的第七信号序列。其中，第七信号序列为解码后的N个子载波软信息：
Figure PCTCN2017103012-appb-000028
其中解调系数为：
Figure PCTCN2017103012-appb-000029
解调后等效信道因子为：
q(i)＝ωH(i)h(i)
其中，s(i)表示数据符号，r(i)表示数据符号的接收信号，q(i)表示等效信道因子，n(i)表示干扰噪声,hH(i)表示信道因子，Ruu-1表示干扰噪声矩阵，i＝0,1,…,N-1。
步骤1004，对所述第七信号序列进行解第一相位旋转操作，得到第九信号序列。
具体地，对第七信号序列进行与发端反向的解第一相位旋转操作。第七信号序列中每个子载波软信息乘以相位旋转因子
Figure PCTCN2017103012-appb-000030
其中i＝0,1,…,N-1，得到第九信号序列。第九信号序列为解第一相位旋转之后的N个子载波软信息。
需要说明的是，由于等效信道因子是一个表征幅度的实数值，故对等效信道因子不需要解第一相位旋转。
图10所述为广义的解插值方式。当采用其他解插值方式时，可能对应进行其他解相位旋转，或者不进行解相位旋转，直接进行解插值操作。
可以理解的是，接收装置可以根据发射装置发射的相关指示选择具体与发射装置插值方式对应的解插值方式。另外，也可通过接发射装置与收端预先约定，采用固定的插值方式和对应的解插值方式进行数据处理。
步骤1005，对所述第九信号序列进行解插值操作，得到第八信号序列。
对第九信号序列进行与发端逆向的(n,m)的解插值操作。
假设发端的插值矩阵为AN×M,收端的解插值矩阵为AH,则有解插值后的原始第八信号序列。其中，原始第八信号序列为QAM符号的软信息α(i)：
Figure PCTCN2017103012-appb-000031
对均衡后的N个等效信道因子ρ(i)进行(n,m)的解插值操作得到：
Figure PCTCN2017103012-appb-000032
其中i＝0,1,…,M-1。
同时有信道补偿因子：
Figure PCTCN2017103012-appb-000033
其中σ为调节因子。
将该信道补偿因子与每个原始第八信号序列和每个解插值后的等效信道因子相乘，得到归一化的第八信号序列和等效信道因子ρ′(j)，其中，第八信号序列为归一化的QAM符号软信息α′(j)：
α′(j)＝δ(j)α(j)
ρ′(j)＝δ(j)ρ(j)
其中j＝0,1,…,M-1。
具体地，第八信号序列包括数据符号的软信息，其中，第八信号序列的长度小于第七信号序列的长度。
具体地，抽取第八信号序列中的数据符号的软信息，以进行下一步处理。
步骤1006，对所述数据符号的软信息进行译码，得到数据符号。
具体地，在第八信号序列中除去导频符号，抽取除去导频符号的所有数据符号的软信息和等效信道因子，经过P/S转换后送入后续QAM解调、解扰、解速率匹配、信道译码等操作，得到数据符号。
需要说明的是，图10所示的基于广义插值方式的接收装置信号处理流程与图3所示的发射装置的信号流程是相对应的。
另外，上述图9所示的数据传输方法，还可采用其他解插值方式实现。下面以图11为例，说明本发明实施例提供的插值操作的另一具体示例。
图11为本发明实施例提供的一种基于频域DFT插值的数据传输方法与装置。如图11所示，主要包括步骤1101至步骤1106。
需要说明的是，步骤1101、步骤1102、步骤1103、步骤1105可分别参照1001、步骤1002、步骤1003、步骤1006。
以下主要对步骤1104解DFT插值操作进行说明：
步骤1004，对所述第七信号序列进行离散傅里叶逆变换IDFT，得到第十信号序列；删除所述第十信号序列头部的ZH个零，以及删除所述第十信号序列尾部的ZT个零，得到第十一信号序列；对所述第十一信号序列进行离散傅里叶变换DFT，得到所述第八信号序列，其中，所述第八信号序列的长度等于所述第七信号序列的长度减去ZH和ZT得到的数值。
具体地，对第七信号序列进行与发端逆向的(n,m)的DFT解插值操作，得到第八信号序列。
其中，先对第七信号序列经过N点IDFT变换到时域，得到第十信号序列；然后删除第十信号序列中与发端对应的头部的ZH个值以及尾部的ZT个值得到M长度的第十一信号序列；最后再对第十一信号序列经过M点DFT变换回频域得到原始第八信号序列。所述原始第八信号序列为QAM符号的软信息α(i)：
Figure PCTCN2017103012-appb-000034
其中i＝0,1,…,M-1。FM表示M点的DFT，
Figure PCTCN2017103012-appb-000035
表示N点的IDFT，[0ZH×M IM×M 0ZT×M]表示头部删除ZH个零以及尾部删除ZT个零。
对解调后的N个等效信道因子进行(n,m)的DFT解插值操作，即先经过N点IDFT变换到时域，然后删除尾部的ZH+ZT个值得到M长度的序列，最后再经过M点DFT变换回频域，得到：
Figure PCTCN2017103012-appb-000036
同时有信道补偿因子δ(j)：
Figure PCTCN2017103012-appb-000037
其中σ为调节因子。
同样地，由于等效信道因子是一个表征幅度的实数值，故可以直接删除尾部的ZH+ZT个值得到M长度的序列。
将该信道补偿因子与每个原始第八信号序列和每个解插值后的等效信道因子相乘，得到归一化的第八信号序列和等效信道因子ρ′(j)，其中，第八信号序列为归一化的QAM符号软信息α′(j)：
α′(j)＝δ(j)α(j)
ρ′(j)＝δ(j)ρ(j)
具体地，第八信号序列包括数据符号的软信息，其中，第八信号序列的长度小于第七信号序列的长度。
需要说明的是，图11所示的基于DFT插值方式的接收装置信号处理流程与图5所示的发射装置的信号流程是相对应的。
需要说明的是，基于频域分块广义插值的多载波OFDM系统的接收装置操作，可参照图8，以及以上图10、图11所示的解插值等接收装置信号流程处理，进行与发送端相反的操作即可。在此不做赘述。
需要说明的是，本发明实施例的插值参数(m,n)理论上可以选择任意的参数，选取的原则可以根据不同时延的粒度和频域导频的密度要求来确定。其中，插值参数一定程度上反映了最终的OFDM符号的大小，以及该OFDM符号可支持的最大时延信息。
本发明实施例在调制OFDM符号的过程中引入插值操作，插值操作等效于引入ZH和ZT两部分数据。本发明实施例可通过根据时延选择合适的插值参数控制ZT以及ZH的长度，以保证接收装置接收到的OFDM符号中有效数据落在有效区间，有效地对抗信道时延和符号间干扰。
本发明实施例可以对仅包括数据符号的OFDM符号插值调制，也可以对仅包括导频符号的OFDM符号插值调制，还可对包括至少一个数据符号和至少一个导频符号的OFDM符号进行插值调制。其中，当OFDM符号包括导频时，需考虑发射的OFDM符号内部的导频符号经过插值操作未引起相位和幅值的变化。
本发明实施例提供的数据传输方法，可适用于多载波系统。因为多载波的直接映射特性，多载波可以在任意子载波位置插导频符号。其中，导频符号为频域符号。结合本发明实施例提供的插值操作，本发明实施例可以在数据符号间隔整数倍位置插入导频符号，通过插值矩阵，控制整数倍插值位置的数据插值之后幅值不变。另外，本发明实施例还需要对插值前或插值后的导频符号进行相位旋转操作，抵消插值等过程给导频符号带来的相位旋转，以保证导频符号相对接收端透明。
需要说明的是，由于ZT-OFDM基于频域插值实现，在整数倍插值点满足插值前后信号幅度不变的特性。因此可以灵活的在整数倍插值点插入导频符号，从而在接收装置可以进行信道估计、噪声估计等测量操作。其中，导频符号又称作参考信号(Reference Signals，RS)。
在5G研究中，主要涉及信道状态信息参考信号(Channel State Information RS，CSI-RS)、解调参考信号(Demodulation RS，DMRS)、相噪导频(Phase Noise RS，PNRS)等导频符号。
其中CSI-RS为信道状态信息测量导频，用于进行波束选择测量、信道质量指示(Channel Quality Indicator，CQI)测量、秩Rank测量、预编码矩阵指示(Precoding Matrix Indicator，PMI)测量等，同时还会支持多发射点协作的测量，因此CSI-RS要求支持尽量多的波束Beam和天线口Port的测量，但是可以按照一定的时间周期触发。
图12为本发明实施例提供的CSI-RS的导频设计图样示意图。图12给出一个应用较高倍插值(2,3)的方式，在导频传输的同时还可以传输一些即时的控制消息。
图12示出了三组导频位置，分别对应三个Beam中的Port0至Port3。由于模拟波束切换的时间要求，不同的波束可以间隔一个符号，每个波束内支持4个Port口的测量，具体的导频复用方式可以是频分复用(Freq终端设备ncydivision multiplexing，FDM)、时分复用(Time Division Multiplexing，TDM)、码分复用(code division multiplexing，CDM)等。
需要说明的是，图12仅示出了CSI-RS导频插值参数的一种示例。本领域技术人员可以理解的是，可以根据实际频域、时域导频密度需求，设计不同的CSI-RS或其他导频的插入图样。
DMRS为信号解调导频，其一般配置为用户级，随路在信号带宽内，并且表征了发射装置预编码与无线信号的联合等效信道，因此其Port端口数要比CSI-RS少的多。图13为本发明实施例提供的一种DMRS的导频设计图样示意图。
优选地，当所述第一信号序列包括至少一个第一导频符号和至少一个数据符号时，所述插值参数为预设的固定值。
图13所示的实施例提供的DMRS导频设计方案为一种固定导频符号插值率的方式。由于考虑自适应的ZT配置，用户会根据信道时延的变化改变插值率，但是不同插值率下的导频可插入位置间隔是不同的，因此为了能够统一不同插值率，可以将导频符号设计为固定插值率，而非导频符号则可以变化不同的插值率。图13中设计了两组导频位置。每组导频位置对应4个Port。
需要说明的是，图13所示的方案中为了尽可能节省导频的开销，Port1至Port 3均采用一组导频位置进行复用，其中Port0和Port1采用FDM或者TDM方式或者OCC正交方式复用。
为了不增加额外的导频位置开销，Port2和Port3可以采用循环移位Cyclic Shift的CDM方式与Port0和Port1进行复用，Cyclic Shift技术在现在LTE协议的上行已经有应用，具体则为Port2与Port0采用相同的导频序列(比如ZC(Zadoff-Chu)序列，随机序列等)，同时Port2在该导频序列基础上进行
Figure PCTCN2017103012-appb-000038
的逐符号相位旋转，从而可以在时域信道估计中将两个Port口的信道时分开来分别进行估计。
另外，对于需要支持到8Port口的情况，启用另一组导频位置进行复用。
需要说明的是，DMRS导频设计方案还可为一种可变插值率的导频符号设计方式，图14为本发明实施例提供的又一种DMRS的导频设计图样示意图。
优选地，在所述第一信号序列仅包括至少一个数据符号时，所述插值参数为多个 插值参数中的一个；当所述第一信号序列包括至少一个第一导频符号或至少一个零时，所述第一预定个数根据所述多个插值参数确定。
需要说明的是，在一定时延内，插值参数有多个选择。不同插值参数下，导频可插入位置间隔是不同的。多个插值参数下包括多个导频可插入位置间隔。进一步设置多个插值参数下导频可插入位置为多个插入间隔的最小公倍数，将不同插值参数下导频可插入位置的图样统一，方便接收端接收导频符号。
具体地，当一组插值参数(mi,ni)满足以下两点条件时：1、该组插值参数的ni的最小公倍数尽量的小，这决定了每一个符号的导频频域间隔；2、该组插值参数能够覆盖足够的信道时延并尽量的均匀量化该时延范围。可选取该组插值参数ni的最小公倍数做为第一预定义位置。
图14选取的一组插值参数为(1,2),(2,3),(5,6),(8,9)，其ni的最小公倍数为18，因此每个DMRS符号的导频间隔为18个子载波，由于该间隔较大，不能进行较为准确的信道估计，考虑采用较多的DMRS符号，同时不同符号进行不同的频域循环移位来达到较好的导频密度的效果。
图14中每个RB有4个DMRS导频符号，因此一般来说可以支持4个Port口的估计，其中第一组导频位置复用Port0和Port2，复用方式采用Cyclic Shift方式，第二组导频位置复用Port和Port3，同样采用Cyclic Shift方式复用。
图15为本发明实施例提供的PNRS的导频设计图样示意图。PNRS用于在中高频的PA器件引入的随机相位噪声的估计，其对频域密度的要求很低，可以十几个RB放置一个，但是时域上要求每个符号均有。因此可以将一组插值参数的ni取最小公倍数，并以最小公倍数的N倍的位置设置相噪导频。
图15中所示的PNSR导频图样设计可参照导频符号插值率固定的方式或导频图样插值率可变(最小公倍数)的方式。另外，对于已有DMRS的符号，可以不需要另外设置PNSR导频。
需要说明的是，图15所示PNSR导频图样，仅为一种举例示意。实际应用中，PNSR导频满足每个符号均有即可，即图16所示的每列代表的一个OFDM符号中至少有一个RE上为PNSR导频。
本发明实施例实现了与5G标准类似的多种导频的图样设计，对于DMRS导频可以采用两种统一的导频图样进行不同插值参数的自适应切换，从而可以避免CoMP下不同发射点采用不同插值参数时，而DMRS导频可以保持正交的效果。
现有5G标准的OFDM系统的导频图样非常灵活，而本发明实施例提供的频域间隔插导频的导频图样设计方案，ZT-OFDM的导频有频域间隔的约束，设计了一种可以近似达到OFDM导频效果的导频图样。
需要说明的是，仅在整数倍插值点插入导频符号，可能会导致导频符号的密度相对较低，会影响信道估计、噪声估计等的质量，从而影响最终的吞吐率。本发明实施例考虑在整数倍插值点插入导频的基础上，对于整数倍点旁边的位置也插入导频，增强导频符号的密度。
为了便于理解，下面对该方法的原理进行简单的解释，考虑基于DFT插值方式， 其插值矩阵表示为：
Figure PCTCN2017103012-appb-000039
参照图4所示的插值矩阵，仅在对角线两边的位置有一定数值，使得插值后的符号基本上只与插值前的几个附近的符号强相关，并且在整数倍插值位置，矩阵中的元素为1，插值后的第i*n个值只与插值前的第i*m个值相关(幅度不变，相位旋转)。
可以理解的是，越靠近整数倍插值点的系数能量越高，表征了基本与插值前的一个符号强相关，再叠加上周围几个符号的较低的符号间干扰。
例如，图4所示的矩阵中，第一列元素为1，第二列对角线两边的元素分别可设为0.9和0.1，第三列的元素分别设为0.7和0.3，第四列元素分别设为0.5和0.5，第五列元素分别设为0.1和0.9等。故第二列元素仅引入了0.1比例的其他符号的干扰。因此，整数倍插值点左右的两个位置也可以利用来插入导频(也包含零功率导频)。
本发明以基于频域的DFT插值为例，说明导频增强方案，图16为本发明实施例提供的基于频域DFT插值导频增强方案的数据传输方法和装置。该实施例包括步骤1601至步骤1606：
其中，插值参数以及系统为用户分配资源、插值前需产生符号个数可参见图3中的描述。
需要说明的是，步骤1601、步骤1604、步骤1606同样可分别参照步骤301、步骤304、步骤307中的描述。为简化说明，以下不再赘述。
步骤1602，根据小区号、帧号等分别产生至少一个第二导频符号和至少一个第三导频符号。其中，第二导频符号的个数与第三导频符号的个数相同。
在图16所示的实施例中，第二导频符号的个数与第三导频符号的个数为L(L≤T)。
进一步地，为了保证接收装置可以透明的接收导频，包括步骤1602a和步骤1602b:
步骤1602a，对至少一个第二导频符号进行第三相位旋转得到至少一个第一导频符号。
具体地，所述第三相位旋转可以表示为乘上相位旋转因子
Figure PCTCN2017103012-appb-000040
其中i∈[0,T)，i根据第一导频符号具体插入步骤1603中第一信号序列哪些候选第一导频符号位置来确定。
步骤1602b，对至少一个第三导频符号进行第四相位旋转和幅度调整，得到至少一个第四导频符号。
具体地，所述第三相位旋转和幅度调制可以表示为乘上
Figure PCTCN2017103012-appb-000041
其中
Figure PCTCN2017103012-appb-000042
Figure PCTCN2017103012-appb-000043
表征了由于相对于整数倍插值点偏移d,d＝1,-1带来的相位变化和幅度的衰减，
Figure PCTCN2017103012-appb-000044
表征了由于第四导频符号插入位置本身带来的相位旋转，i∈[0,T)，根据第四导频符号具体插入步骤1603中第一信号序列哪些候选第一导频符号位置来确定。
步骤1603，将所述至少一个第四导频符号、所述至少一个第一导频符号以及所述至少一个数据符号按照第三预定义规则组成所述第一信号序列。
其中，所述第三预定义规则为：每隔第一预定义个数的数据符号为候选的第一导频符号位置，选择至少一个所述候选的第一导频符号位置插入所述第一导频符号；选 择至少一个所述候选的第一导频符号位置的相邻位置分别插入所述第四导频符号；其中，所述第一预定义个数根据所述插值参数确定。在本实施例中，第一预定义个数可以等于m-1。
具体地，将至少一个数据符号与至少一个第一导频符号、至少一个第四导频符号经过S/P转换混插排列为M长度第一信号序列。其中第一导频符号需要插入在整数倍插值点位置，该位置对应M长度的序列的第i*m的位置，其中i＝0,1,…,T-1，经过(m,n)的分数倍DFT插值后则对应N长度的序列的第i*n的位置。而第四导频符号需要插入在整数倍插值点偏移d,d＝1,-1的位置im+d。
步骤1605a，所述将至少一组所述第二信号序列映射到子载波，得到至少一组子载波上的第二信号序列。
进一步地：每隔第二预定义个数的子载波的相邻位置为候选的第三导频符号位置，用第三导频符号替换至少一个所述候选的第三导频符号位置所在子载波上的第二信号序列中的符号，所述第二预定义个数根据所述插值参数确定，其中，所述第二预定义个数大于所述第一预定义个数，所述第二信号序列中，所述第三导频符号的替换位置与所述第一信号序列中插入所述第四导频符号的位置相关。
将N个符号组成的第二信号序列连续映射到频域的N个子载波上，然后将至少一个第四导频符号根据插值前插入位置im+d替换对应的子载波位置in+d上的值，该操作保证了该子载波的导频信息保存，同时去除了其它子载波引入的插值间干扰。
步骤1605b，对所述子载波上的第二信号序列进行快速傅里叶逆变换IFFT，得到时域信号。
最后进行传统的IFFT操作，对于多层的MIMO传输在IFFT之前还包括层映射，预编码等操作。
本发明实施例提供的导频增强方案，在原有导频位置两边又增加了一些可配置导频的位置，因此不管从正交性上还是可支持的最大Port口上都有了一定的增强。
可以理解的是，本发明导频图样的设计，可参照前述实施例提供的固定插值率和可变插值率(最小公倍数插值位置)的两种方案。在此不做赘述。
图17为本发明实施例提供的一种DMRS的增强导频设计图样示意图。如图17所示，由于多了导频位置，该实施例中DMRS导频方案可以采用更低的插值率如插值参数为(5,6)，从而保证该符号较高的数据传输率。
图17中Port0至Port3的方式与图13相同，而Port4至Port7可以在Port0至Port3的导频符号旁边放置。两组Port口间的正交性可以通过图13所示的技术，配置至少一个第三导频符号零功率实现两组Port口间无干扰，也可以通过配置至少一个第三导频符号正常功率OCC正交，以实现两组Port口间的OCC正交。
需要说明的是，可根据不同组Port口具体设置配置哪些第三导频符号零功率或正常功率OCC正交，以保证不同组Port口无干扰或OCC正交。
图18为本发明实施例提供的又一种DMRS的增强导频设计图样示意图。该实施例中DMRS增强导频图样方案图14所示的方案基础上，在Port0至Port4端口旁边再增加一组端口Port4至Port7，两组Port口的正交同样可以采用配置至少一个第三导频 符号零功率或者正常功率OCC正交的方式来实现。
需要说明的是，第三导频符号的候选位置在第一导频符号候选位置的偏移d,d＝1,-1的位置im+d。第三导频符号具体插入候选位置中的哪个与第一导频符号插入的位置之间无关联。图17、图18仅示出了导频增强的其中一种图样。
本发明实施例提供的技术方案通过将整数倍插值点位置两边的符号利用起来发送导频信号，可以提升可利用导频位置的密度。进而能支持更多的Port口，或者换来导频符号可以传输更多的数据符号的效果。
在一个示例中，假设N均为12个子载波，导频符号均占4个子载波。不采用导频增强方案时，插值参数可为(2,3)，则插值参数(2,3)，m＝2，n＝3对应的M为8个符号，其中导频符号为4个，则数据符号为4个。采用导频增强方案时，插值参数为(5,6)，m＝5，n＝6对应的M为10个符号，其中导频符号为4个，则数据符号为6个。即，同样数量的导频符号下，采用导频增强方案达到了传输更多的数据符号的效果。
相应地，本发明实施例提供一种应用在多发射点异步传输的方案，以使接收端可使用联合MIMO方式接收信号。图19为本发明实施例提供的应用于多发射点异步数据传输方法示意图。
具体地，当至少一个发射装置协作向接收端发送数据时，所述至少一个发射装置包括服务发射装置和协作发射装置，所述协作发射装置根据其相对所述服务发射装置的时延偏差，确定插值参数，以使接收装置对所述至少一个发射装置的信号进行联合MIMO接收。
参见图3、图5等实施例中的描述，第二信号序列的头部和尾部分别包括多个小于预设阈值的数据。具体地，所述小于预设数据的数据可以为近似零数据。
在下行多协作点非相干方式传输给一个终端用户，或者上行多用户的同时频异步竞争接入，均可以按照图19的方式应用，图19以下行两个传输点传输为例：
终端设备可能接收至少两个传输点(Transmission Point，TP)协作传输的数据。设两个传输点为TP0和TP1。
终端设备接入目标小区的TP0，按照TP0进行定时，因此TP0可以按照无ZT的方式进行传输(不进行插值，插值率为1)。其中TP0可称为服务发射点。
终端设备测量下行各TP点的信号接收功率，选择TP1作为协作传输点。当TP1与TP0不共定时，终端设备接收到的TP1、TP0发射的信号之间会存在时延偏差。终端设备测量TP1的时延偏差，通过上行控制信道反馈给TP1时延偏差的量化指示。TP1根据接收到的时延偏差值，选择合适的ZT-OFDM的插值参数，对数据进行插值后发送信号给终端设备。终端设备对于两个TP的信号进行联合MIMO接收。
具体地，如图19所示，TP1相对TP0有一定的时延偏差(TP1delay)。通过选择合适的插值参数控制TP1信号中ZT的长度，以使得TP1的有效信号完全落在接收端的一个周期内，或使得TP1的有效信号完全在TP0的IFFTsize内。参照图19所示。其中，有效信号指的是IFFTsize中除去ZT和ZH部分的数据。ZH所占比例较小，因此未在图19中示出。
图19中采用相对简单的OLA方式，对于两个TP的信号进行联合MIMO接收。
可以理解的是，实际应用中，图19所示的应用方式还可包括其他协作TP，终端设备每个协作TP相对TP0的时延偏差，通过上行控制信道反馈给协作TP时延偏差的量化指示。协作TP根据接收到的时延偏差值，选择合适的ZT-OFDM的插值参数，对数据进行插值后发送信号给终端设备。以保证多个TP的有效信号均完全在TP0的IFFTsize内。以使终端设备可对多个TP的信号进行联合MIMO接收。
为了保证异步下多传输点联合传输，现有技术只能配置为Extended CP的方式进行传输，这样无时延偏差或者短时延偏差的TP都要按照较长的CP的格式发送，会降低这些TP的信号速率。
本发明实施例提供的多发射点异步传输技术可以保证多个不同定时的传输点按照各自的时延偏差选择合适的ZT-OFDM插值参数发送信号，同时可以保证接收装置采用联合MIMO方式进行接收，相比进行简单的串行干扰对消(successive interference cancellation，SIC)性能更好，处理时延更短。
在5G标准中，由于考虑多种业务的共存，会出现多numerology的场景，也就是不同子载波间隔的业务(如15K/30K/60K)会共存，由于不同子载波间隔的频域Sinc窗函数不同，因此不同子载波间隔业务之间需要预留一定的保护间隔(Guard Band，GB)子载波进行保护。
本发明实施例利用ZT-OFDM的特性，构造时域更短的时间间隔，达到更宽子载波间隔的效果。
在一个示例中，当插值率为2时，所述发射所述时域信号，进一步包括：所述时域信号包括第一时域信号和第二时域信号，所述第一时域信号所占的时间长度与所述第二时域信号所占的时间长度相同；对第一时域信号和第二时域信号分别进行截取，得到第三时域信号和第四时域信号，所述第二时域信号在时域上相比所述第一时域信号延迟半个周期，所述第三时域信号所占时间长度为所述第一时域信号所占时间长度的一半，所述第四时域信号所占时间长度为所述第二时域信号所占时间长度的一半；将所述第三时域信号和第四时域信号进行混合，得到第五时域信号；发射所述第五时域信号，所述半个周期为所述第一时域信号所占时间长度的一半。
具体地，采用IFFT/2窗函数对所述第一时域信号和第二时域信号分别进行截取，将所述第一时域信号或第二时域信号头部和尾部的多个小于预设阈值的数据截掉。
上述一个示例通过构造T/2的时间间隔，达到等效30K子载波间隔的效果。
在另一个示例中，当所述插值率为4时，所述发射所述时域信号，进一步包括：所述时域信号包括第六时域信号、第七时域信号、第八时域信号和第九时域信号，所述第六时域信号所占的时间长度、所述第七时域信号所占的时间长度、所述第八时域信号所占的时间长度以及所述第九时域信号所占的时间长度相同；对第六时域信号、第七时域信号、第八时域信号和第九时域信号分别进行截取，得到第十时域信号、第十一时域信号、第十二时域信号和第十三时域信号，所述第七时域信号在时域上相比所述第六时域信号延迟四分之一周期、所述第八时域信号在时域上相比所述第七时域信号延迟四分之一周期、所述第九时域信号在时域上相比所述第八时域信号延迟四分之一周期，所述第十时域信号所占时间长度为所述第六时域信号所占时间长度的四分 之一，所述第十一时域信号所占时间长度为所述第七时域信号所占时间长度的四分之一，所述第十二时域信号所占时间长度为所述第八时域信号所占时间长度的四分之一，所述第十三时域信号所占时间长度为所述第九时域信号所占时间长度的四分之一；将所述第十时域信号、第十一时域信号、第十二时域信号和第十三时域信号进行混合，得到第十四时域信号；发射所述第十四时域信号，所述四分之一周期为所述第六时域信号所占时间长度的四分之一。
具体地，采用IFFT/4窗函数对所述第六时域信号、第七时域信号、第八时域信号和第九时域信号分别进行截取，将所述第六时域信号、第七时域信号、第八时域信号和第九时域信号头部和尾部的多个小于预设阈值的数据截掉。
上述另一个示例通过构造T/4的时间间隔，达到等效60K子载波间隔的效果。
本发明实施例以构造T/2为例，详细进行说明。图20为本发明实施例提供的更短时间间隔的数据传输方法示意图。
如图20所示，发送端采用(1,2)的插值参数，得到第一时域信号和第二时域信号。所述第一时域信号所占的时间长度与所述第二时域信号所占的时间长度相同，设第一时域信号所占的时间长度为FFTsize。设FFTsize对应的时域周期为T，设时间周期T对应的子载波间隔为15K。
其中，第一时域信号和第二时域信号中接近IFFT/2长度的ZT为近似零功率。如如图20中2001、2002部分所示，第一时域信号和第二时域信号的信号之间时间延迟为T/2(Delay 0和Delay T/2)。
采用IFFT/2的窗函数对对第一时域信号和第二时域信号中有效信号部分分别进行截取，得到第三时域信号和第四时域信号，如图20中2003、2004部分所示。其中，窗函数可以是矩形窗，也可以是有一定滚降的其它窗函数。
分别在所述第三时域信号和第四时域信号最后添加ZP/2长度的零功率尾部，分别形成时域长度为T/2的更短的时间间隔的两个时域信号，如图20中2005、2006部分所示。
将添加ZP/2长度的零功率尾部的第三时域信号和第四时域信号进行混合，得到第五时域信号，如图20中2007部分所示。
发射所述第五时域信号。其中，第五时域信号的长度与原第一时域信号或第二时域信号长度相同，为FFTsize。但第五时域信号构造时域时间间隔为T/2，达到了子载波间隔两倍的效果。
相应地，接收装置接收T/2窗内的信号，采用OLA方式处理尾部的ZP部分；将FFT/2长度的信号在尾部添加FFT/2的零功率信号；将最终的FFT长度的信号经过正常的(1,2)的插值参数的ZT-OFDM的接收端处理。
可以理解的是，如果采用(1,4)插值参数，则形成了类似于60k的效果，另外，也可以把多个终端设备的信号采用图20的方式进行时分复用。即将多个终端设备的信号采用(1,2)或(1,4)插值参数、以及截取、混合的方式，进行时分复用。
本发明实施例提供的方案，利用ZT-OFDM的特性，构造时域更短的时间间隔，达到更宽子载波间隔的效果。同时不同业务之间在空口还都是15K，不存在互相ICI干扰问题。相对于5G中多numerology场景下15k/30k/60k复用的场景，该实施例由 于空口都是15k，不存在ICI干扰问题，因此可以节省频域GB，有效提高吞吐量。
本发明实施例提供的数据传输方法，根据接收端测量的信道最大时延偏差，发送端对映射到频域的调制符号进行一定比率的分数倍插值滤波操作，达到最终IFFT后信号头部ZH个数据和尾部ZT个数据近似零功率的效果，并且在插值前可以在整数倍插值位置插入频域RS导频信号，为保持接收装置RS透明，插入的频域RS根据信号的时域偏移进行加相位旋转操作。
本发明实施例通过插值形成的OFDM时域符号可以直接串联形成连续的信号，此时需要保证ZT的长度能够覆盖总的信道时延偏差；也可以对IFFT后的每个符号可以在尾部添加ZP个零功率信号，或者在头部拷贝符号尾部的CP个零功率信号形成循环前缀信号，达到符号内零功率信号与ZP或者CP零功率信号连接的效果。本发明实施例提供的多个方案的选择可以最终以5G标准的帧格式要求为准。
本发明实施例中的插值滤波器不限于DFT插值滤波器，还可以为线性插值、样条插值等多种插值算法，其中对于DFT插值滤波器，包含IDFT变换、时域前后填充零功率头尾、DFT变换，其中零功率头尾ZH和ZT的长度大小可调节。
本发明实施例提供插值形成的OFDM时域符号，对应的插值率越大则对抗信道时延偏差的能力越强。将最大时延偏差按照一定间隔设定多个插值比率的档位，发射装置根据测量的最大时延偏差选择某个插值比率，并通过控制消息通知接收装置，或者接收装置根据导频格式或者零功率的长度进行盲检测。
本发明实施例可以将用户的数据符号和导频符号分为多个块，每个块独立的进行插值操作，最后块之间可以在频域离散的分布，从而获得更好的调度增益和频域分集增益。
本发明实施例提供的方案，导频符号插入可以为固定插值率的方式，而非导频符号则可以变化不同的插值率，这样可以保证不同插值参数配置下可以复用相同的导频图样。
本发明实施例提供的方案，导频插入符号也可以为可变插值率的方式，为了满足不同插值参数配置下复用相同的导频图样，需要挑选一组插值参数(mi,ni)，选取的准则需要满足该组参数的ni的最小公倍数尽量的小，来保证足够的导频频域密度，同时该组参数能够覆盖足够的信道时延并尽量的均匀量化该时延范围。
本发明实施例提供的方案，除了在整数倍插值点可以插入导频符号，对于较低的插值率，还可以在整数倍插值点左右的有限个点插入导频或者零功率符号，此时需要在插入时对符号进行一定的相位旋转和幅度调整，同时在插值后映射到子载波时对对应的位置的符号用插值前的插入导频或者零功率符号替换。
相应地，本发明实施例提供了一种通信设备，用于实现前述实施例中提供的数据传输方法，如图21所示，所述通信设备包括：插值单元2101、映射单元2102、IFFT单元2103、发射单元2104、确定单元2105和获取单元2106。
所述通信设备的插值单元2101用于对第一信号序列进行插值操作，得到第二信号序列，所述第二信号序列的长度大于所述第一信号序列的长度。
映射单元2102用于将所述第二信号序列映射到子载波，得到子载波上的第二信号序列。
IFFT单元2103用于对所述子载波上的第二信号序列进行快速傅里叶逆变换IFFT，得到时域信号。
发射单元2104用于发射所述时域信号。
优选地，还包括：确定单元2105用于确定插值参数；所述插值单元2101具体用于根据所述插值参数对所述第一信号序列进行插值操作，得到所述第二信号序列。
优选地，还包括：获取单元2106用于获取终端设备的信号的最大时延偏差，所述终端设备的信号为发给所述终端设备的无线信号或所述终端设备发送的无线信号，所述最大时延偏差为所述终端设备的信号从发送端发出，经过无线信道，最早到达接收装置的时间与最晚到达接收装置的时间之差；所述确定单元2105，具体用于根据所述终端设备的信号的最大时延偏差确定所述插值参数。
优选地，所述插值单元2101具体用于：对所述第一信号序列进行离散傅里叶逆变换IDFT，得到第四信号序列；在所述第四信号序列的头部增加ZH个零，以及在所述第四信号序列的尾部增加ZT个零，得到第五信号序列，其中，ZH和ZT均为大于零的整数；对所述第五信号序列进行离散傅里叶变换DFT，得到所述第二信号序列，其中，第二信号序列的长度等于所述第一信号序列的长度、ZH以及ZT的总和。
具体地，插值单元2101还可通过广义插值、以及导频位置插零值最后替换方案实现，具体可参见图3、图5、图6及图7中的介绍，在此不做赘述。
优选地，所述第一信号序列包括至少一个数据符号。
优选地，所述第一信号序列包括至少一个第一导频符号，所述第一导频符号为第二导频符号经过第三相位旋转而得到的，所述第二导频符号用于接收装置进行信道测量和信道估计中的至少一个。
优选地，所述第一信号序列还包括至少一个数据符号，其中，所述至少一个第一导频符号与所述至少一个数据符号按照第一预定义规则组成所述第一信号序列；所述第一预定义规则为：在第一信号序列中，每隔第一预定义个数的数据符号为候选的第一导频符号位置，选择至少一个所述候选的第一导频符号位置插入所述第一导频符号，其中，所述第一预定义个数根据所述插值参数确定。
具体地，第一信号序列可包括的多个符号以及多个符号的排列，可参照见图3、
图5中的介绍，在此不做赘述。
优选地，所述映射单元2102具体用于：将至少一组所述第二信号序列映射到子载波，得到至少一组子载波上的第二信号序列。
具体地，分块插值映射方案，可参照图8中的介绍，在此不做赘述。
优选地，所述发射单元2104还用于发送所述插值参数相关的信息，所述插值参数相关的信息用于接收装置确定所述插值参数。
本发明实施例提供的通信设备的工作流程具体可参见前述图3至图20的详细说明，在此不做赘述。
需要说明的是，图21所示的通信设备，还可采用如下的实现方式。具体如图22 所示，所述通信设备包括：处理器2201、发射器2202。
所述通信设备的处理器2201用于对第一信号序列进行插值操作，得到第二信号序列，所述第二信号序列的长度大于所述第一信号序列的长度。
处理器2201还用于将所述第二信号序列映射到子载波，得到子载波上的第二信号序列。
处理器2201还用于对所述子载波上的第二信号序列进行快速傅里叶逆变换IFFT，得到时域信号。
发射器2202用于发射所述时域信号。
优选地，处理器2201还用于确定插值参数；处理器2201具体用于根据所述插值参数对所述第一信号序列进行插值操作，得到所述第二信号序列。
优选地，处理器2201还用于获取终端设备的信号的最大时延偏差，所述终端设备的信号为发给所述终端设备的无线信号或所述终端设备发送的无线信号，所述最大时延偏差为所述终端设备的信号从发送端发出，经过无线信道，最早到达接收装置的时间与最晚到达接收装置的时间之差；所述处理器2201具体用于根据所述终端设备的信号的最大时延偏差确定所述插值参数。
优选地，所述处理器2201具体用于：对所述第一信号序列进行离散傅里叶逆变换IDFT，得到第四信号序列；在所述第四信号序列的头部增加ZH个零，以及在所述第四信号序列的尾部增加ZT个零，得到第五信号序列，其中，ZH和ZT均为大于零的整数；对所述第五信号序列进行离散傅里叶变换DFT，得到所述第二信号序列，其中，第二信号序列的长度等于所述第一信号序列的长度、ZH以及ZT的总和。
具体地，处理器2201还可通过广义插值、以及导频位置插零值最后替换方案实现，具体可参见图3、图5、图6及图7中的介绍，在此不做赘述。
优选地，所述第一信号序列包括至少一个数据符号。
优选地，所述第一信号序列包括至少一个第一导频符号，所述第一导频符号为第二导频符号经过第三相位旋转而得到的，所述第二导频符号用于接收装置进行信道测量和信道估计中的至少一个。
优选地，所述第一信号序列还包括至少一个数据符号，其中，所述至少一个第一导频符号与所述至少一个数据符号按照第一预定义规则组成所述第一信号序列；所述第一预定义规则为：在第一信号序列中，每隔第一预定义个数的数据符号为候选的第一导频符号位置，选择至少一个所述候选的第一导频符号位置插入所述第一导频符号，其中，所述第一预定义个数根据所述插值参数确定。
具体地，第一信号序列可包括的多个符号以及多个符号的排列，可参照见图3、图5中的介绍，在此不做赘述。
优选地，处理器2201具体用于：将至少一组所述第二信号序列映射到子载波，得到至少一组子载波上的第二信号序列。
具体地，分块插值映射方案，可参照图8中的介绍，在此不做赘述。
优选地，发射器2202还用于发送所述插值参数相关的信息，所述插值参数相关的信息用于接收装置确定所述插值参数。
本发明实施例提供的通信设备的工作流程具体可参见前述图3至图20的详细说明， 在此不做赘述。
相应地，本发明实施例提供了又一种通信设备，用于实现前述实施例中提供的数据传输方法，如图23所示，所述通信设备包括：接收单元2301、FFT单元2302、解调单元2303、解插值单元2304、译码单元2305、获取单元2306、信道估计单元2307和确定单元2308。
所述通信设备的接收单元2301用于接收时域信号。
FFT单元2302用于对所述时域信号进行快速傅里叶变换FFT，得到子载波上的第六信号序列。
解调单元2303用于对第六信号序列进行解调，得到第七信号序列。
解插值单元2304用于对所述第七信号序列进行解插值操作，得到第八信号序列，所述第八信号序列包括数据符号的软信息，其中，第八信号序列的长度小于第七信号序列的长度。
译码单元2305用于对所述数据符号的软信息进行译码，得到数据符号。
优选地，还包括：获取单元2306用于获取导频符号；信道估计单元2307用于根据所述导频符号进行信道估计，得到信道相关信息；所述解调单元2303具体用于根据所述信道相关信息对第六信号序列进行解调，得到所述第七信号序列。
优选地，还包括：确定单元2308用于确定插值参数；所述解插值单元2304具体用于根据所述插值参数对所述第七信号序列进行解插值操作，得到第八信号序列。
优选地，所述解插值单元2304具体用于：对所述第七信号序列进行离散傅里叶逆变换IDFT，得到第十信号序列；删除所述第十信号序列头部的ZH个零，以及删除所述第十信号序列尾部的ZT个零，得到第十一信号序列；对所述第十一信号序列进行离散傅里叶变换DFT，得到所述第八信号序列，其中，所述第八信号序列的长度等于所述第七信号序列的长度减去ZH和ZT得到的数值。
具体地，解插值单元2304还可通过广义解插值实现，具体可参见图10、图11中的介绍，在此不做赘述。
优选地，所述解插值单元2304具体用于：具体用于：对所述第六信号序列进行解调，得到至少一组所述第七信号序列。
优选地，所述接收单元2301还用于接收所述插值参数相关的信息，所述插值参数相关的信息用于接收装置确定所述插值参数。
本发明实施例提供的通信设备的工作流程具体可参见前述图3至图20的详细说明，在此不做赘述。
需要说明的是，图23所示的通信设备，还可采用如下的实现方式。具体如图24所示，所述通信设备包括：接收器2401、处理器2402。
所述通信设备的接收器2401用于接收时域信号。
处理器2402还用于对所述时域信号进行快速傅里叶变换FFT，得到子载波上的第六信号序列。
处理器2402还用于对第六信号序列进行解调，得到第七信号序列。
处理器2402还用于对所述第七信号序列进行解插值操作，得到第八信号序列，所述第八信号序列包括数据符号的软信息，其中，第八信号序列的长度小于第七信号序列的长度。
处理器2402还用于对所述数据符号的软信息进行译码，得到数据符号。
优选地，处理器2402还用于获取导频符号；处理器2402还用于根据所述导频符号进行信道估计，得到信道相关信息；处理器2402具体用于根据所述信道相关信息对第六信号序列进行解调，得到所述第七信号序列。
优选地，处理器2402还用于确定插值参数；处理器2402具体用于根据所述插值参数对所述第七信号序列进行解插值操作，得到第八信号序列。
优选地，处理器2402具体用于：对所述第七信号序列进行离散傅里叶逆变换IDFT，得到第十信号序列；删除所述第十信号序列头部的ZH个零，以及删除所述第十信号序列尾部的ZT个零，得到第十一信号序列；对所述第十一信号序列进行离散傅里叶变换DFT，得到所述第八信号序列，其中，所述第八信号序列的长度等于所述第七信号序列的长度减去ZH和ZT得到的数值。
具体地，处理器2402还可通过广义解插值实现，具体可参见图10、图11中的介绍，在此不做赘述。
优选地，所述处理器2402具体用于：具体用于：对所述第六信号序列进行解调，得到至少一组所述第七信号序列。
优选地，所述接收器2401还用于接收所述插值参数相关的信息，所述插值参数相关的信息用于接收装置确定所述插值参数。
本发明实施例提供的通信设备的工作流程具体可参见前述图3至图20的详细说明，在此不做赘述。
本领域技术人员应该可以意识到，在上述一个或多个示例中，本发明所描述的功能可以用硬件、软件、固件或它们的任意组合来实现。当使用软件实现时，可以将这些功能存储在计算机可读介质中或者作为计算机可读介质上的一个或多个指令或代码进行传输。计算机可读介质包括计算机存储介质和通信介质，其中通信介质包括便于从一个地方向另一个地方传送计算机程序的任何介质。存储介质可以是通用或专用计算机能够存取的任何可用介质。
以上所述的具体实施方式，对本发明的目的、技术方案和有益效果进行了进一步详细说明，所应理解的是，以上所述仅为本发明的具体实施方式而已，并不用于限定本发明的保护范围，凡在本发明的技术方案的基础之上，所做的任何修改、等同替换、改进等，均应包括在本发明的保护范围之内。

Claims (30)

  1. 一种数据传输方法，其特征在于，包括：
    对第一信号序列进行插值操作，得到第二信号序列，所述第二信号序列的长度大于所述第一信号序列的长度；
    将所述第二信号序列映射到子载波，得到子载波上的第二信号序列；
    对所述子载波上的第二信号序列进行快速傅里叶逆变换IFFT，得到时域信号；
    发射所述时域信号。
  2. 根据权利要求1所述的方法，其特征在于，在所述对第一信号序列进行插值操作之前，还包括：
    确定插值参数；
    所述对第一信号序列进行插值操作，得到第二信号序列，包括：
    根据所述插值参数对所述第一信号序列进行插值操作，得到所述第二信号序列。
  3. 根据权利要求2所述的方法，其特征在于，在所述确定插值参数之前，还包括：
    获取终端设备的信号的最大时延偏差，所述终端设备的信号为发给所述终端设备的无线信号或所述终端设备发送的无线信号，所述最大时延偏差为所述终端设备的信号从发送端发出，经过无线信道，最早到达接收装置的时间与最晚到达接收装置的时间之差；
    所述确定插值参数，包括：
    根据所述终端设备的信号的最大时延偏差确定所述插值参数。
  4. 根据权利要求1至3任一项所述的方法，其特征在于，所述插值操作具体包括：
    对所述第一信号序列进行离散傅里叶逆变换IDFT，得到第四信号序列；
    在所述第四信号序列的头部增加ZH个零，以及在所述第四信号序列的尾部增加ZT个零，得到第五信号序列，其中，ZH和ZT均为大于零的整数；
    对所述第五信号序列进行离散傅里叶变换DFT，得到所述第二信号序列，其中，第二信号序列的长度等于所述第一信号序列的长度、ZH以及ZT的总和。
  5. 根据权利要求4所述的方法，其特征在于，所述第一信号序列包括至少一个数据符号。
  6. 根据权利要求4所述的方法，其特征在于，所述第一信号序列包括至少一个第一导频符号，所述第一导频符号为第二导频符号经过第三相位旋转而得到的，所述第二导频符号用于接收装置进行信道测量和信道估计中的至少一个。
  7. 根据权利要求6所述的方法，其特征在于，所述第一信号序列还包括至少一个数据符号，其中，所述至少一个第一导频符号与所述至少一个数据符号按照第一预定义规则组成所述第一信号序列；
    所述第一预定义规则为：在第一信号序列中，每隔第一预定义个数的数据符号为候选的第一导频符号位置，选择至少一个所述候选的第一导频符号位置插入所述第一导频符号，其中，所述第一预定义个数根据所述插值参数确定。
  8. 根据权利要求1至7任一项所述的方法，其特征在于，所述将第二信号序列映射到子载波，得到子载波上的第二信号序列，具体包括：
    将至少一组所述第二信号序列映射到子载波，得到至少一组子载波上的第二信号 序列。
  9. 根据权利要求2至8任一项所述的方法，其特征在于，还包括：
    发送所述插值参数相关的信息，所述插值参数相关的信息用于接收装置确定所述插值参数。
  10. 一种数据传输方法，其特征在于，包括：
    接收时域信号；
    对所述时域信号进行快速傅里叶变换FFT，得到子载波上的第六信号序列；
    对第六信号序列进行解调，得到第七信号序列；
    对所述第七信号序列进行解插值操作，得到第八信号序列，所述第八信号序列包括数据符号的软信息，其中，第八信号序列的长度小于第七信号序列的长度；
    对所述数据符号的软信息进行译码，得到数据符号。
  11. 根据权利要求10所述的方法，其特征在于，在所述对第六信号序列进行解调之前，还包括：
    获取导频符号；
    根据所述导频符号进行信道估计，得到信道相关信息；
    所述对第六信号序列进行解调，得到第七信号序列，包括：
    根据所述信道相关信息对第六信号序列进行解调，得到所述第七信号序列。
  12. 根据权利要求10或11所述的方法，其特征在于，在所述对所述第七信号序列进行解插值操作之前，还包括：
    确定插值参数；
    所述对所述第七信号序列进行解插值操作，得到第八信号序列，包括：
    根据所述插值参数对所述第七信号序列进行解插值操作，得到第八信号序列。
  13. 根据权利要求10至12任一项所述的方法，其特征在于，所述解插值操作具体包括：
    对所述第七信号序列进行离散傅里叶逆变换IDFT，得到第十信号序列；
    删除所述第十信号序列头部的ZH个零，以及删除所述第十信号序列尾部的ZT个零，得到第十一信号序列；
    对所述第十一信号序列进行离散傅里叶变换DFT，得到所述第八信号序列，其中，所述第八信号序列的长度等于所述第七信号序列的长度减去ZH和ZT得到的数值。
  14. 根据权利要求10至13任一项所述的方法，其特征在于，所述对第六信号序列进行解调，得到第七信号序列，具体包括：
    对所述第六信号序列进行解调，得到至少一组所述第七信号序列。
  15. 根据权利要求12至14任一项所述的方法，其特征在于，还包括：
    接收所述插值参数相关的信息，所述插值参数相关的信息用于确定所述插值参数。
  16. 一种通信设备，其特征在于，包括：
    插值单元，用于对第一信号序列进行插值操作，得到第二信号序列，所述第二信号序列的长度大于所述第一信号序列的长度；
    映射单元，用于将所述第二信号序列映射到子载波，得到子载波上的第二信号序列；
    IFFT单元，用于对所述子载波上的第二信号序列进行快速傅里叶逆变换IFFT，得到时域信号；
    发射单元，用于发射所述时域信号。
  17. 根据权利要求16所述的通信设备，其特征在于，还包括：
    确定单元，用于确定插值参数；
    所述插值单元，具体用于根据所述插值参数对所述第一信号序列进行插值操作，得到所述第二信号序列。
  18. 根据权利要求17所述的通信设备，其特征在于，还包括：
    获取单元，用于获取终端设备的信号的最大时延偏差，所述终端设备的信号为发给所述终端设备的无线信号或所述终端设备发送的无线信号，所述最大时延偏差为所述终端设备的信号从发送端发出，经过无线信道，最早到达接收装置的时间与最晚到达接收装置的时间之差；
    所述确定单元，具体用于根据所述终端设备的信号的最大时延偏差确定所述插值参数。
  19. 根据权利要求16至18任一项所述的通信设备，其特征在于，所述插值单元，具体用于：对所述第一信号序列进行离散傅里叶逆变换IDFT，得到第四信号序列；在所述第四信号序列的头部增加ZH个零，以及在所述第四信号序列的尾部增加ZT个零，得到第五信号序列，其中，ZH和ZT均为大于零的整数；
    对所述第五信号序列进行离散傅里叶变换DFT，得到所述第二信号序列，其中，第二信号序列的长度等于所述第一信号序列的长度、ZH以及ZT的总和。
  20. 根据权利要求19所述的通信设备，其特征在于，所述第一信号序列包括至少一个数据符号。
  21. 根据权利要求19所述的通信设备，其特征在于，所述第一信号序列包括至少一个第一导频符号，所述第一导频符号为第二导频符号经过第三相位旋转而得到的，所述第二导频符号用于接收装置进行信道测量和信道估计中的至少一个。
  22. 根据权利要求21所述的通信设备，其特征在于，所述第一信号序列还包括至少一个数据符号，其中，所述至少一个第一导频符号与所述至少一个数据符号按照第一预定义规则组成所述第一信号序列；
    所述第一预定义规则为：在第一信号序列中，每隔第一预定义个数的数据符号为候选的第一导频符号位置，选择至少一个所述候选的第一导频符号位置插入所述第一导频符号，其中，所述第一预定义个数根据所述插值参数确定。
  23. 根据权利要求16至22任一项所述的通信设备，其特征在于，所述映射单元，具体用于：将至少一组所述第二信号序列映射到子载波，得到至少一组子载波上的第二信号序列。
  24. 根据权利要求17至23任一项所述的通信设备，其特征在于，所述发射单元，还用于发送所述插值参数相关的信息，所述插值参数相关的信息用于接收装置确定所述插值参数。
  25. 一种通信设备，其特征在于，包括：
    接收单元，用于接收时域信号；
    FFT单元，用于对所述时域信号进行快速傅里叶变换FFT，得到子载波上的第六信号序列；
    解调单元，用于对第六信号序列进行解调，得到第七信号序列；
    解插值单元，用于对所述第七信号序列进行解插值操作，得到第八信号序列，所述第八信号序列包括数据符号的软信息，其中，第八信号序列的长度小于第七信号序列的长度；
    译码单元，用于对所述数据符号的软信息进行译码，得到数据符号。
  26. 根据权利要求25所述的通信设备，其特征在于，还包括：
    获取单元，用于获取导频符号；
    信道估计单元，用于根据所述导频符号进行信道估计，得到信道相关信息；
    所述解调单元，具体用于根据所述信道相关信息对第六信号序列进行解调，得到所述第七信号序列。
  27. 根据权利要求25或26所述的通信设备，其特征在于，还包括：
    确定单元，用于确定插值参数；
    所述解插值单元，具体用于根据所述插值参数对所述第七信号序列进行解插值操作，得到第八信号序列。
  28. 根据权利要求25至27任一项所述的通信设备，其特征在于，所述解插值单元，具体用于：对所述第七信号序列进行离散傅里叶逆变换IDFT，得到第十信号序列；删除所述第十信号序列头部的ZH个零，以及删除所述第十信号序列尾部的ZT个零，得到第十一信号序列；对所述第十一信号序列进行离散傅里叶变换DFT，得到所述第八信号序列，其中，所述第八信号序列的长度等于所述第七信号序列的长度减去ZH和ZT得到的数值。
  29. 根据权利要求25至28任一项所述的通信设备，其特征在于，所述解插值单元，具体用于：对所述第六信号序列进行解调，得到至少一组所述第七信号序列。
  30. 根据权利要求27至29任一项所述的通信设备，其特征在于，所述接收单元，还用于接收所述插值参数相关的信息，所述插值参数相关的信息用于接收装置确定所述插值参数。
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