WO2017098762A1

WO2017098762A1 - Power conversion device, power supply device, and method for controlling same

Info

Publication number: WO2017098762A1
Application number: PCT/JP2016/076154
Authority: WO
Inventors: 泰明乗松; 叶田　玲彦; 馬淵　雄一; 尊衛嶋田; 充弘門田; 祐樹河口; 輝米川; 瑞紀中原
Original assignee: 株式会社日立製作所
Priority date: 2015-12-10
Filing date: 2016-09-06
Publication date: 2017-06-15
Also published as: JP2019041427A

Abstract

Provided are a power conversion device in which an insulating transformer and peripheral circuits thereof can be reduced in size, a power supply device, and a method for controlling the same. Disclosed are a power conversion device and a power supply device, comprising: a first inverter unit that obtains a direct-current input and provides a high-frequency output; an LLC transformer that transduces the voltage of the high-frequency output of the first inverter unit; a rectifier unit that converts the output of the LLC transformer into a direct current; a second inverter unit that converts the direct-current output of the rectifier unit into an alternating current; and a power supply circuit that obtains, via a rectifying circuit from a control-circuit power supply transformer connected in parallel to the secondary circuit of the LLC transformer, power for a control circuit which provides gate power for semiconductor elements constituting the second inverter unit and a gate signal for said semiconductor elements. A direct-current voltage of the rectifying circuit is divided and is introduced to the control circuit as a direct-current output of the rectifier unit.

Description

Power conversion device, power supply device and control method thereof

The present invention relates to a power conversion device, a power supply device, and a control method thereof for DC power and AC power.

Many isolation transformers are used in the power system, but it is difficult to reduce the size and weight because it is driven at a low frequency of several tens of Hz (50/60 Hz in Japan). There was a problem.

On the other hand, by adopting SST (Solid State Transformer: hereinafter simply referred to as SST) technology that has been studied for application to high-voltage and high-power applications in recent years, it is expected to contribute to reduction in size and weight. ing. The SST is composed of a high-frequency transformer and a power conversion circuit, and the output or input of the SST is AC power having the same frequency as the conventional one. In SST, a high frequency is generated by a power conversion circuit (DC / DC converter or inverter) and a high frequency transformer is driven to connect an input or output to an AC power system having the same frequency as the conventional one. It replaces the isolation transformer. According to this configuration, the high-frequency transformer can be driven at a high frequency of several tens to several hundreds of kHz, so that a significant reduction in size and weight can be realized as compared with a conventional insulation transformer alone.

For example, as a new application of power converters for power systems, high-performance power conversion that controls the power of natural energy and outputs it to the power system with the global expansion of natural energy introduction such as solar power generation and wind power generation. PCS (Power Conditioning System: hereinafter simply referred to as PCS) is required. Since the PCS is connected to the power system and used, a high-voltage insulation transformer is used on its output side, and it must be driven at a low frequency of several tens of Hz, which is the same as the frequency of the power system. There is a problem that the equipment becomes larger.

In addition to power grid interconnections, there are some power converters that use high voltage power, such as for high voltage motors and pumps, and railways, that use a high voltage isolation transformer on the input side. Even when the output side is the same as the output side, the high voltage isolation transformer is driven at a low frequency of several tens of Hz that is the same as the frequency of the power system by receiving power from the power system. Have

However, to apply SST for the realization of miniaturization and weight reduction of the power converter for high voltage, even in the power supply of the control circuit of the power converter, between the primary side and the secondary side of the high frequency isolation transformer. Insulation is necessary.

On the other hand, as an example of a countermeasure against insulation between the primary side and the secondary side of an insulation transformer employed in a general high-voltage power converter and its control circuit power supply, the configuration of FIG. Things are known. FIG. 2 shows a multi-level high-voltage inverter using multiple transformers as insulation transformers. The multilevel in FIG. 2 shows an example of a one-stage configuration.

2, the electric power of the main circuit inverter 11 is input from the three-phase AC multiple transformer 12 and is output from the main circuit inverter 11 as a single-phase AC. As a result, the voltage value or frequency is appropriately adjusted and obtained from the main circuit inverter 11. The main circuit inverter 11 is composed of a rectifier section R, a smoothing capacitor C2, and an inverter section In2.

The gate power supply 13 for supplying power for starting the semiconductor elements constituting the inverter unit In2 in the main circuit inverter 11 is provided for each of the single-phase transformer 4, the single-phase bridge rectifier circuit 16, the DC reactor 14, and the semiconductor elements. The power supply circuit 17 is configured. The starting power is supplied by extracting a single phase from the output of the three-phase AC multiple transformer 12.

In the multi-level high-voltage inverter shown in FIG. 2, insulation between the power system (several kV to several tens of kV) is achieved by the above-described connection configuration using the multiple transformer 12, the main circuit inverter 11, and the gate power supply 13. Is ensured by the multiple transformer 12. For this reason, the single-phase transformer 15 for the gate power supply 13 can be reduced to a voltage (about several hundred volts) required for one multilevel stage, and a control circuit for controlling voltage conversion can be eliminated.

Patent Document 1 discloses a high-voltage inverter power supply apparatus basically having the same configuration as that shown in FIG. 2, although there is a difference in that the multiple transformer 12 is a single phase.

JP 2007-151224 A

However, since the single-phase transformer 15 used in Patent Document 1 is driven at 50/60 Hz which is the same as the frequency of the power system, it is particularly difficult to reduce the size of the transformer core. Further, since the conversion to the DC voltage for the gate power supply 13 is performed by the single-phase bridge rectifier circuit 16, the voltage fluctuation is increased by the load of the control circuit, and a DC / DC converter or voltage fluctuation corresponding to the voltage fluctuation is obtained. It is necessary to add a DC reactor 14 for suppressing the power supply as necessary, and the power supply for the gate power supply 13 is enlarged.

In addition, when applying SST for the realization of a compact and lightweight power converter for high voltage, insulation is required between the high voltage power system and the low voltage control system, and the voltage of the high voltage power system is reduced. It needs to be converted to a low pressure for detection. For example, when a DC voltage of a main circuit is detected and introduced as a feedback signal into a control circuit that controls the DC voltage, the DC voltage is generally divided by a voltage dividing resistor and detected. However, in detecting a DC voltage of 600 V to 1200 V in the main circuit, an element having a breakdown voltage of 100 V or more is required for each voltage dividing resistor, and a creepage distance must be secured, so that the circuit becomes large.

In addition, as a configuration of the main circuit inverter 11, a control power source configuration in which the DC voltage is stepped down by a DC / DC converter is conceivable. In this case as well, insulation between the power system can be ensured by the multiple transformer 12. Further, the transformer and the reactor necessary for the DC / DC converter are driven at the driving frequency of the DC / DC converter, so that the size can be reduced. However, in this configuration, since the DC voltage of the main circuit inverter 11 is several hundred volts, the application corresponding to the several hundred volts used for the main circuit inverter 11 is also applied to the DC / DC converter for the gate power supply 13. It is necessary to secure an insulation distance of several hundred volts for each element, and a gate power supply 13 for a DC / DC converter corresponding to several hundred volts input is also required. Therefore, as a result, the power source for the gate power source 13 is increased in size even in the configuration to which the DC / DC converter is applied.

In view of the above, an object of the present invention is to provide a power conversion device, a power supply device, and a control method thereof that can reduce the size of an insulating transformer and its peripheral circuits.

In order to solve the above problems, in the present invention, a first inverter unit that obtains a DC input and gives a high-frequency output, an LLC transformer that converts the high-frequency output of the first inverter unit, and an output of the LLC transformer A rectifier unit for conversion, a second inverter unit for converting the direct current output of the rectifier unit to alternating current, and a control circuit power supply transformer connected in parallel to the secondary circuit of the LLC transformer through the rectifier circuit, A power supply circuit that obtains the gate power of the semiconductor element constituting the inverter unit and the power of the control circuit that supplies the gate signal of the semiconductor element, and divides the DC voltage of the rectifier circuit and introduces it to the control circuit as the DC output of the rectifier unit. This is a power conversion device and a power supply device.

Also, a control method for a power conversion device or a power supply device, in which the resonance frequency in the LLC transformer and the drive frequency in the first inverter unit are an operation mode in which the drive frequency and the resonance frequency are equal, and the drive frequency is greater than the resonance frequency The control method is characterized in that the first inverter unit is controlled by switching between a higher operation mode and an operation mode in which the dynamic frequency is lower than the resonance frequency.

According to the present invention, it is possible to provide a power conversion device, a power supply device, and a control method thereof that can reduce the size of the peripheral circuit including the isolation transformer.

Further, according to the embodiment of the present invention, the volume of the control circuit power transformer is reduced by increasing the frequency inputted to the control circuit power transformer, and the control circuit for voltage conversion becomes unnecessary.
In addition to the power transformer for the control circuit, a low-voltage element can be used. As a result, the entire control circuit can be reduced in size and weight, so that the high-voltage power converter can be reduced in size and weight.

The figure which shows the structure of the power converter device which concerns on Example 1 of this invention. The circuit diagram which shows the structure of the conventional multilevel high voltage inverter. The figure which shows the structure of the three-phase power supply device by the power converter device which concerns on Example 1 of this invention. The figure which shows a voltage and electric current waveform when LLC resonant frequency is equal to a drive frequency. The figure which shows the voltage and electric current waveform when LLC resonance frequency> drive frequency. The figure which shows a voltage and electric current waveform at the time of LLC resonance frequency <drive frequency. The figure which shows the relationship of DC voltage Vdc with respect to the input voltage by photovoltaic power generation, photovoltaic power generation output, and a driving frequency. FIG. 10 is a diagram illustrating a specific circuit configuration example of the gate power supply according to the third embodiment. The figure which shows the specific circuit structural example of the power supply for gates which concerns on Example 3 which employ | adopted IGBT. The figure which shows the specific circuit structural example of the power supply for gates concerning Example 3 which used the power transformer for control circuits as the single output. The figure which shows the structure at the time of dividing | segmenting the power transformer for control circuits into 4 outputs for power supplies for gates, and the single output for power supplies for controls. The figure which shows the structure of the power converter device which concerns on Example 4. FIG. FIG. 10 is a diagram illustrating a specific circuit configuration example of a gate power supply according to a fifth embodiment. FIG. 10 is a diagram illustrating a specific circuit configuration example of a gate power supply according to a fifth embodiment. FIG. 10 is a diagram illustrating a specific circuit configuration example of a gate power supply according to a sixth embodiment.

Hereinafter, embodiments of the power conversion device, the power supply device, and the control method thereof according to the present invention will be described with reference to the drawings.

First, the configuration of the power conversion apparatus according to the first embodiment will be described with reference to FIG.

In the power conversion apparatus of the first embodiment shown in FIG. 1, a power conversion apparatus for a PCS of several hundred kW to several MW class whose input is DC 600 to 1000 V and whose output is connected to a high voltage system of 6.6 kV system. Assumed.

1, the main circuit 100 is configured by a series circuit of a full-bridge type LLC resonant converter 1 and a single-phase inverter 2. The full-bridge type LLC resonant converter 1 is connected to a DC power source by, for example, photovoltaic power generation at an input terminal Ti, and provides an AC output to an output terminal To of the single-phase inverter 2.

The full-bridge type LLC resonant converter 1 includes a smoothing capacitor C1, a first inverter unit In1, an LLC transformer 3, a rectifier unit R, and a smoothing capacitor C2 from the input side. Here, the LLC transformer 3 is a transformer primary winding side of a first reactor 31, a second reactor 32 as a primary winding, and a capacitor 33 arranged in series. It is a vessel.

In FIG. 1, the single-phase inverter 2 includes a second inverter unit In2. In the configuration of the main circuit 100, the first inverter unit In1 and the second inverter unit In2 have a single-phase full bridge configuration, and these inverter units In1 and In2 have their semiconductor elements configured by MOS-FETs. An example is shown.

According to the configuration of the main circuit 100, in the first inverter unit In1, a direct current input by photovoltaic power generation applied to the input terminal Ti is converted into a high frequency alternating current and adjusted to an arbitrary voltage in the LLC transformer 3. Then, after converting into direct current in the rectifier section R, the second inverter section In2 converts to a frequency (50/60 Hz) that can be connected to, for example, a commercial AC power source and outputs it to the output terminal To.

1, the power supply circuit 17 obtains power from the secondary winding 34 of the LLC transformer 3 via the control circuit power supply transformer 4. In the illustrated example, the control circuit power transformer 4 includes a primary winding 41 and a secondary winding 42 electromagnetically coupled to the primary winding 41. The secondary winding 42 supplies DC power to the power supply circuit 17 via the single-phase bridge rectifier circuit 16. The DC power of the power supply circuit 17 is used as gate power given to the gates of the semiconductor elements Q1, Q2, Q3, and Q4 of the second inverter unit In2, or as control circuit power.

Further, voltage dividing resistors R1 and R2 are connected in series on the output side of the single-phase bridge rectifier circuit 16, and the connection point potential is used as a detection signal for the DC voltage Vdc of the main circuit.

According to this configuration, since the LLC transformer 3 operates at a high frequency, the device configuration can be reduced in size. Since the control circuit power transformer 4 also operates at a high frequency, it can be miniaturized. Further, the voltage dividing circuit for detecting the main circuit DC voltage Vdc is also lower in potential than the case where it is installed in the main circuit and is insulated, so that the circuit is miniaturized.

In this figure, the power supply circuit 17 supplies the gate power of the MOS-FET semiconductor elements Q1, Q2, Q3, Q4 constituting the second inverter section In2 and the power supply for the gate control circuit. The power supply circuit for one inverter unit In1 is not shown. A power supply circuit for the first inverter unit In1 is separately installed but is not described here. The reason for this is that the voltage level to be insulated (usually about 100 volts AC) is low on the first inverter unit In1 side, so that a high degree of insulation measure that is applied to the power supply circuit 17 is not required. This is because it can be handled by technology.

As described above, the configuration of the main circuit 100 in FIG. 1 is a configuration in which the DC output after full-bridge diode rectification in the full-bridge type LLC resonant converter 1 is AC-output to the power system via the single-phase inverter 2. In the embodiment of FIG. 1, a single-phase inverter 2 is assumed, but a three-level inverter configuration may be applied.

FIG. 1 shows a single-phase output, but three-phase is required for connection to an AC system. Therefore, practical applications suitable for the three-phase configuration of FIG. 3 are made. FIG. 3 is a diagram illustrating the configuration of the three-phase power supply device using the power conversion device according to the first embodiment. In this application, the output side of the single-phase output main circuit 100 is connected in multiple stages for each of the three phases U, V, and W.

FIG. 1 shows a detailed configuration of the power converter, and FIG. 3 shows the configuration of a three-phase power supply device as an example of deployment to a three-phase power system. A series multi-level configuration connected in series can support high-voltage output. For example, although the input voltage of each main circuit 100 is 600 to 1000 VDC, 6.6 kV can be obtained as the voltage between the three-phase lines. This means that relatively low-voltage semiconductor elements Q1, Q2, Q3, and Q4 such as 1700V, 1200V, and 650V can be used when viewed as a single-phase inverter 2. Further, since the terminal voltage Vdc of the second capacitor C2 is also a voltage according to the semiconductor elements Q1, Q2, Q3, and Q4, it means that the DC capacitor C2 can use a capacitor having a lower voltage than the power system voltage. is doing.

The specifications and functions of the components constituting the power conversion device or power supply device shown in FIGS. 1 and 3 may be as follows, and the following effects can be exhibited.

First, since the full-bridge type LLC resonant converter 1 has a DC voltage of 1000 V or less, it is assumed that a MOS-FET suitable for high-frequency driving is applied as the first inverter unit In1. The switching frequency of the semiconductor element of the first inverter unit In1 is assumed to be several tens kHz to several hundreds kHz. As the MOS-FET to be used, a SiC MOS-FET suitable for high withstand voltage / high frequency switching may be applied, and any other one having a similar function may be used.

The secondary side rectifier section R of the LLC resonant converter 1 is assumed to be smoothed by a diode. In addition to Si-type diodes, Si-type Schottky barrier diodes or SiC Schottky barrier diodes may be applied to reduce conduction loss, and loss can be reduced by using SiC MOS-FETs in synchronization. It does not matter if it has other similar functions.

The LLC transformer 3 has an insulating function from the power system voltage, and the first reactor 31 on the transformer primary winding side, the second reactor 32 as the primary winding, and the capacitor 33 are arranged in series. It is an insulation transformer called as follows. Of these circuit components, the first reactor 31 defines the leakage inductance Lr, the second reactor 32 as the primary winding defines the exciting inductance Lm of the high-frequency transformer, and the capacitor 33 defines the resonant capacitor capacitance Cr. is doing. The LLC resonant transformer 3 has a configuration in which a leakage inductance Lr and a resonant capacitor capacitance Cr are connected to resonate with the exciting inductance Lm of the high-frequency transformer to achieve LLC resonance. The leakage inductance Lr may be integrated in the high frequency transformer as a structure that can adjust the constant of the leakage magnetic flux in the high frequency transformer. The resonance capacitor capacity Cr is assumed to use a film capacitor, but may have any similar function.

Although FIG. 1 shows an example in which the semiconductor element of the second inverter unit In2 is composed of a MOS-FET, this may be an IGBT. The second inverter unit In2 may employ an IGBT because the switching frequency of the series multiplex PWM is as low as several kHz or less as a whole compared to the drive frequency of the LLC resonant converter 1.

In addition, when applied to the three-phase power system of FIG. In the series multiplexing configuration of the power conversion device in FIG. 3, the number of single-phase inverters 2 per phase is assumed to be about 8 to 6 stages in series. In FIG. 3, in order to reduce the number of stages of the single-phase inverter 2, a Y-connection configuration is assumed, but a Δ-connection configuration is also possible. In the case of Y connection, the phase voltage is 1 / √3 with respect to the line voltage of 6.6 kV, and the DC voltage of the entire phase is based on √2 times, so the second capacitor C2 in the case of eight stages The terminal voltage Vdc is about 600 to 700V. As a result, a voltage at which a 1200-V MOS-FET can be used for the single-phase inverter 2 is obtained, and a high-voltage output can be realized by a low-voltage element. Since the LLC resonant converter 1 is 1000 V or less with respect to the ground voltage, the single-phase inverter 2 is in a floating connection, and therefore, it is assumed that the LLC transformer 3 has an insulation function corresponding to 6.6 kV of the power system. . By adopting such a serial multi-level configuration as described above, it is possible to realize a configuration that can flexibly cope with changes in the number of stages and the insulation of the LLC transformer 3 depending on circumstances even when the output voltage is other than 6.6 kV.

Furthermore, regarding the specifications and functions of the power transformer 4 for the control circuit, these specifications and functions may be as follows, and the following effects can be exhibited.

The power transformer 4 for the control circuit shown in FIG. 1 is configured to be connected to the secondary side of the LLC transformer 3, and is driven at several tens to several hundreds of kHz by the LLC control, so that several hundreds of volts on the primary side is secondary. The side can be converted to several tens to several V.

Here, since the power required for the power supply circuit 17 is less than 1% with respect to the 100% output of the single-phase inverter 2, the excitation inductance of the power transformer 4 for the control circuit has a sufficiently large value with respect to the LLC transformer 3. Assumes that For example, it is assumed that the excitation inductance Lm of the LLC transformer 3 is several hundred μH, whereas the excitation inductance of the power transformer 4 for the control circuit is set to be a value of several hundred mH or more. It is not limited.

Although details of the drive voltage will be described later, since the control circuit power transformer 4 is driven by the LLC control, a controller for controlling voltage conversion becomes unnecessary.

As described above, since the LLC transformer 3 secures insulation from the power system, the power transformer 4 for the control circuit can be reduced to an insulation function of about several hundred volts that is equal to or higher than the terminal voltage Vdc of the second capacitor C2. As a result, the voltage dividing resistors R1 and R2 installed as the detection circuit for the DC voltage Vdc of the main circuit on the output side of the single-phase bridge rectifier circuit 16 can be applied to a low voltage circuit. The size can be greatly reduced compared to the case where it is provided.

In the embodiment of FIG. 1, since the outline of the present invention has been described, the power supply circuit 17 is simply described, but actually, as shown in the third and subsequent embodiments, the secondary side of the power transformer 4 for the control circuit has many secondary sides. Assuming an output configuration, each semiconductor element Q1, Q2, Q3, Q4 supplies isolated output to

power supply circuits

17a, 17b, 17c, 17d for the gate and a controller (such as a microcomputer) for controlling the gate. is doing. A configuration assuming that the gates of the semiconductor elements Q1, Q2, Q3, and Q4 have a voltage stabilizing circuit (such as a linear regulator) and a driving photoMOS, and the controller is controlled by driving a light emitting diode of the photoMOS. It is. The power required for each output is assumed to be about several tens to several hundreds mW, but is not limited thereto.

From the above assumed configuration, it is assumed that a pulse transformer is used for the power transformer 4 for the control circuit. However, the configuration is not limited to the above as long as it has a similar function.

In the power conversion device of the present invention illustrated in FIG. 1, an LLC transformer 3 is employed in the LLC resonant converter 1. In the second embodiment, a control method peculiar to the LLC resonant converter 1 including the LLC transformer 3 and its effect will be described.

In the control of the first inverter section In1 in the LLC resonant converter 1 of the present invention, frequency control of duty 50% with dead time added is performed instead of PWM. In this case, the resonance frequency is determined by the values of the excitation inductance Lm, leakage inductance Lr, and resonance capacitor capacitance Cr of the high-frequency transformer described above for the first inverter unit In1, and the resonance frequency is set to several tens to several hundreds kHz. Assumed.

The control of the LLC resonant converter 1 is executed by a control circuit that controls on / off of four sets of semiconductor elements constituting the full-bridge first inverter unit In1. The control circuit itself has a circuit configuration that is usually performed, and a specific circuit configuration is omitted. In short, the semiconductor element is controlled as follows.

For example, among the four full-bridge semiconductor elements, the upper right and lower left semiconductor elements Q9 and Q8 in FIG. 1 are the first set, and the lower right and upper left semiconductor elements Q10 and Q7 are the second set. The first conductive state in which the first set is ON and the second set is OFF, and the second conductive state in which the second set is ON and the first set is OFF are alternately formed. The ON / OFF control is performed with a drive frequency having a period from the first conduction state to the first conduction state again through the second conduction state.

4, 5, and 6, the magnitude relationship between the LLC resonance frequency and the drive frequency, and the relationship between the voltage and current waveform of the first inverter unit In <b> 1 will be described. In these figures, the horizontal axis represents time, and the vertical axis represents the magnitude of voltage and current during one cycle.

FIG. 4 shows voltage and current waveforms when the LLC resonance frequency is equal to the drive frequency. In this case, the drive frequency is controlled by a control circuit that controls on / off of the four sets of semiconductor elements constituting the full-bridge first inverter unit In1, and the drive frequency is a high-frequency transformer. This coincides with the resonance frequency determined by the values of the excitation inductance Lm, leakage inductance Lr, and resonance capacitor capacitance Cr of a certain LLC transformer 3.

In the state where the LLC resonance frequency is equal to the drive frequency, as shown in the waveform of the primary side current ITin of the LLC transformer 3, when the MOS-FET which is a semiconductor element is ON, the current flowing through the MOS-FET is MOS− Since it flows in the reverse direction through the body diode of the FET, it becomes ZVS (zero volt switching), and no switching loss occurs when it is ON. When the MOS-FET is OFF, the current flowing through the MOS-FET peaks out and is kept low enough, and the switching loss when OFF is also small. Therefore, high-efficiency switching can be realized by LLC resonance control, and the power device is cooled. The device can be downsized.

When the LLC resonance frequency is controlled to be equal to the drive frequency by the LLC control, the voltage on the secondary side of the LLC transformer 3 is VTout, and in short, a rectangular voltage output is obtained.
Therefore, a rectangular wave voltage is input to the control circuit power transformer 4. For this reason, since the voltage corresponding to the turn ratio with respect to the terminal voltage Vdc of the second capacitor C2 is output to the secondary side output of the power transformer 4 for the control circuit, the voltage fluctuation is the same as in the LLC resonant converter 1. There is no need for a DC reactor.

FIG. 5 shows voltage and current waveforms when the drive frequency is lowered below the LLC resonance frequency and the boost operation is performed.

In a state where the drive frequency is lower than the LLC resonance frequency, as shown in the waveform of the primary side current ITin of the LLC transformer 3, when the MOS-FET is ON, the current flowing through the MOS-FET is equal to that of the MOS-FET. Since it flows in the reverse direction through the body diode, it becomes ZVS (zero volt switching) and no switching loss occurs when it is ON. Since the current flowing through the MOS-FET is sufficiently low and leveled off after peaking out when it is OFF, the switching loss when OFF is small as with resonant frequency driving.

When the LLC resonance frequency is controlled to be higher than the drive frequency by the LLC control, the secondary side voltage of the LLC transformer 3 becomes a rectangular wave voltage such as VTout. This rectangular wave voltage is a waveform with a slight delay between rising and falling with respect to the power transformer 4 for the control circuit, but basically a rectangular wave is input. For this reason, since the voltage corresponding to the turn ratio with respect to the terminal voltage Vdc of the second capacitor C2 is output to the secondary side output of the power transformer 4 for the control circuit, the voltage fluctuation is the same as in the LLC resonant converter 1. There is no need for a DC reactor.

FIG. 6 shows voltage and current waveforms when the drive frequency is raised above the LLC resonance frequency and the step-down operation is performed.

In the state where the drive frequency is higher than the LLC resonance frequency, as shown in the waveform of the primary side current ITin of the LLC transformer 3, when the MOS-FET is ON, the current flowing through the MOS-FET is equal to that of the MOS-FET. Since it flows in the reverse direction through the body diode, it becomes ZVS (zero volt switching) and no switching loss occurs when it is ON. Since the current flowing through the MOS-FET does not decrease at the time of OFF, the switching loss at the time of OFF increases. However, in the case of the control operation by the PCS, the reduction in efficiency can be minimized by limiting the voltage range for the step-down control.

When the drive frequency is raised above the LLC resonance frequency by the LLC control, the voltage on the secondary side of the LLC transformer 3 is like VTout, and in short, a rectangular voltage output is obtained. Therefore, a rectangular wave voltage is input to the control circuit power transformer 4. For this reason, since the voltage corresponding to the turn ratio with respect to the terminal voltage Vdc of the second capacitor C2 is output to the secondary side output of the power transformer 4 for the control circuit, the voltage fluctuation is the same as in the LLC resonant converter 1. There is no need for a DC reactor.

The above three operation modes in the LLC control (LLC resonance frequency = drive frequency, LLC resonance frequency> drive frequency, LLC resonance frequency <drive frequency) are appropriately selected and executed in the actual operation scene of the power converter.

For example, in the case of a system configuration in which the DC input of the power conversion device in FIG. 1 is obtained from solar power generation, the switching operation is preferably performed as shown in FIG. 7 according to the weather conditions. FIG. 7 is a diagram illustrating a case example of LLC control when applied to solar power generation.

FIG. 7 shows the relationship between the terminal voltage Vdc of the second capacitor C2, the photovoltaic power generation output, and the driving frequency with respect to the input voltage Vin by photovoltaic power generation.

In FIG. 7, regarding the relationship between the input voltage Vin and the photovoltaic power generation output by the solar power generation in the middle stage, the magnitude of the input voltage Vin and the photovoltaic power generation output by the photovoltaic power generation is variable depending on the weather (fine weather, cloudy weather). . In the operation of the second embodiment, the range of the voltage input to the PCS (that is, the input voltage Vin by solar power generation) is divided into a first region D1, a second region D2, and a third region D3.

In the first region D1 where the input voltage Vin is lower than the first threshold value Vin1, the drive frequency is made lower than the LLC resonance frequency by LLC control so that the terminal voltage Vdc of the second capacitor C2 is boosted so as not to be lower than the lower limit value. Implement control. In the third region D3 where the input voltage Vin is higher than the second threshold value Vin2, the drive frequency is made higher than the LLC resonance frequency by LLC control so that the terminal voltage Vdc of the second capacitor C2 is stepped down so as not to exceed the upper limit value. Implement control. In the second region D2 between the first threshold value Vin1 and the second threshold value Vin2, the drive frequency is made equal to the LLC resonance frequency by the LLC control and is made constant, thereby enabling high-efficiency operation.

The region may be set based on the upper and lower limits of the terminal voltage Vdc of the second capacitor C2. In this case, the second threshold value Vin2 that defines the upper limit region is set to be equal to or higher than the maximum output point voltage in normal sunny weather. It is good. By doing so, the probability of the step-down operation in the third region is reduced.

The voltage on the secondary side of the LLC transformer 3 by this LLC control is VTout. Since a rectangular wave is input to the power transformer 4 for the control circuit, a voltage corresponding to the turn ratio with respect to the terminal voltage Vdc of the second capacitor C2 is output to the secondary output, so that the same as the LLC converter There is no voltage fluctuation and no DC reactor is required.

Even if the load is changed in the above three operations, since VTout is a rectangular wave, the control voltage can be generated with the turn ratio of the terminal voltage Vdc of the second capacitor C2 regardless of the load.

For the above reasons, connecting the power transformer 4 for the control circuit to the secondary side of the LLC transformer 3 reduces the dielectric breakdown voltage, eliminates the need for voltage conversion drive control, suppresses voltage fluctuations due to the load, and reduces the size of the transformer due to higher frequencies. It becomes possible to satisfy the requirements, and the entire power supply device can be reduced in size and weight.

Example 3 shows a specific circuit configuration example of the gate power supply 13 of FIG.

In the specific configuration of the gate power supply 13 shown in FIG. 8, a gate

power supply circuit

17a, 17b, 17c, 17d and a single-phase bridge rectifier circuit are provided for each of the semiconductor elements Q1, Q2, Q3, Q4 of the second inverter unit In2. 16a, 16b, 16c, and 16d are provided, and a control power supply circuit 17j and a single-phase bridge rectifier circuit 16j are provided. The control circuit power transformer 4 is configured so that the secondary side of the control circuit transformer 4 has multiple outputs by

secondary windings

42a, 42b, 42c, 42d, and 42j, so that each single-phase

bridge rectifier circuit

16a, 16b, 16c, 16d and 16j. In the configuration of FIG. 8, the control circuit transformer 4 shows an example of a one-to-n connection method in which the primary side is a single winding and the secondary side is a plurality of windings.

Further, voltage dividing resistors R1 and R2 are connected in series to the output side of the single-phase bridge rectifier circuit 16j, and the connection point potential is introduced into the control power supply circuit 17j as a detection signal of the DC voltage Vdc of the main circuit and used for control. ing.

The configuration of the third embodiment in FIG. 8 uses the secondary side of the control circuit transformer 4 as multiple outputs, so that it can be used for the voltage detection of the DC voltage Vdc and the microcomputer in the control power supply circuit 17j shown in the first embodiment. In addition, the power source for driving the gates of the semiconductor elements Q1, Q2, Q3, and Q4 for driving the inverter is generated.

The gates of the semiconductor elements Q1, Q2, Q3, and Q4 have a voltage stabilizing circuit (such as a linear regulator) and a driving photo MOS, and the microcomputer controls the driving element by driving the light emitting diode of the photo MOS. This is an assumed configuration. The power required for each output is assumed to be about several tens to several hundreds mW, but is not limited thereto.

From the configuration assumed above, it is assumed that a pulse transformer is used for the power transformer 4 for the control circuit. However, any structure having the same function may be used. As shown in FIG. 9, the inverter driving semiconductor elements Q1, Q2, Q3, and Q4 may use IGBTs instead of MOS FETs. Since the inverter section In2 for output has a switching frequency of the serial multiplex PWM as low as several kHz or less as a whole compared with the drive frequency of the LLC resonant converter, problems such as heat generation are small even when the IGBT is applied.

FIG. 10 shows a configuration when the control circuit power transformer 4 has a single output. It is assumed that the present invention is applied to a case where a plurality of control

circuit power transformers

4a, 4b, 4c, 4d, and 4j can be individually arranged on the secondary side of the LLC transformer 3 so that downsizing in a distributed arrangement becomes possible. is doing.

FIG. 11 shows a configuration when the control circuit power transformer 4 is divided into four outputs for the

gate power circuits

17a, 17b, 17c and 17d and a single output for the control power circuit 17j. The turns ratio of the four outputs for the gate

power supply circuits

17a, 17b, 17c and 17d is the same, and the turn ratio of one output for the control power supply circuit 17j is the four outputs for the gate

power supply circuits

17a, 17b, 17c and 17d. By enabling output at a lower voltage, productivity can be improved and overall loss can be reduced. In addition, it is assumed that the size can be reduced by dividing the output.

The configuration of FIG. 11 is a configuration in which the voltage detection of the DC voltage Vdc is divided on the side of the gate power supply circuit 17d and supplied to the control power supply circuit 17j. It is assumed that an error such as an influence of a voltage drop of the rectifying element of the control circuit power transformer 1 is reduced by dividing the voltage from a voltage higher than that on the control power circuit 17j side to improve detection accuracy.

The control method in the third embodiment is basically the same as that described above. The third embodiment is also the control of the LLC resonant converter 1 and the duty control of 50% duty with a dead time. It is assumed that the resonance frequency is determined by the values of the excitation inductance Lm, the leakage inductance Lr, and the resonance capacitor capacitance Cr, and is set to several tens to several hundreds kHz.

In Example 3 as well, the waveform when the voltage is changed is the same as in FIGS. 4, 5, and 6. Even if the load is changed, VTout is a rectangular wave, so the Vdc turns ratio is independent of the load. A control voltage can be generated.

Also in this embodiment, by connecting the control circuit transformer 4 to the secondary side of the LLC transformer 3, downsizing due to a reduction in dielectric strength, no need for voltage conversion drive control, suppression of voltage fluctuation due to load, downsizing of the transformer due to higher frequency It is possible to satisfy all of the above, and the power supply as a whole can be reduced in size and weight.

FIG. 12 shows the configuration of the power conversion apparatus according to the fourth embodiment. The configuration of the third embodiment is a configuration in which the number of semiconductor elements constituting the rectifier unit R of the LLC resonant converter 1 is halved. The voltage width input to the primary-side LLC transformer 3 is ½ of the full-bridge configuration of FIG. 1, but can be similarly adjusted by the turn ratio of the LLC transformer 3.

The control method of Example 3 is a frequency control of Duty 50% performed by controlling the LLC resonant converter 1 and giving a dead time. It is assumed that the resonance frequency is determined by the values of the excitation inductance Lm, the leakage inductance Lr, and the resonance capacitor capacitance Cr, and is set to several tens to several hundreds kHz.

Also in the third embodiment, the waveforms when the voltage is changed are the same as those in FIGS. 4, 5, and 6, and even if the load is changed, VTout is a rectangular wave. Therefore, the second capacitor C2 does not depend on the load. The control voltage can be generated with the turn ratio of the terminal voltage Vdc.

In the fourth embodiment as well, the control circuit transformer 1 is connected to the secondary side of the LLC transformer 2 to reduce the size due to a decrease in the withstand voltage, no need for voltage conversion drive control, the suppression of voltage fluctuation due to the load, and the downsizing of the transformer due to the higher frequency. It is possible to satisfy all of the above, and the power supply as a whole can be reduced in size and weight.

13 and 14 show the configuration of the power conversion apparatus according to the fifth embodiment. The configuration of the fifth embodiment assumes a system in which output is connected to three phases of U, V, and W phases in a single-stage configuration, and a motor and a pump are driven to output.

In FIG. 13, the output of the power transformer 3 for the control circuit is assumed to be a total of seven power circuits, including six for driving the semiconductor elements and one for the controller in accordance with the three-phase driving of the U, V, and W phases. However, the number is not limited as long as the configuration has the same function.

FIG. 14 shows a configuration in which the number of outputs of the control circuit transformer 4 is reduced. Since the source voltages of the semiconductor elements Q2, Q4, and Q6 are the same, the number of outputs of the control circuit transformer 1 can be reduced by using the same output.

The control method of Example 5 is frequency control of Duty 50% performed by control of the LLC resonant converter 1 and provided with dead time. It is assumed that the resonance frequency is determined by the values of the excitation inductance Lm, the leakage inductance Lr, and the resonance capacitor capacitance Cr, and is set to several tens to several hundreds kHz.

Also in the fifth embodiment, the waveform when the voltage is changed is the same as that of FIGS. 4, 5, and 6. Even if the load is changed, VTout is a rectangular wave, so the second capacitor C2 does not depend on the load. The control voltage can be generated with the turn ratio of the terminal voltage Vdc.

As described above, the four examples have been described, but it goes without saying that the contents described in the examples may be used in combination depending on the application.

FIG. 15 shows a configuration in which the control circuit power transformer 4 of FIG. 1 is divided and distributed in series.
In FIG. 1, the primary winding and the secondary winding are configured to 1: n. However, in FIG. 15, n = 5 controls are provided as the control circuit power transformer 4 on the secondary side of the LLC transformer 3.

Circuit power transformers

4a, 4b, 4c, 4d, and 4j are connected in series. This connection form is applied when the control circuit power transformer 4 is divided into

power supply circuits

17a, 17b, 17c, 17d and dispersed in series, and can be miniaturized in a distributed arrangement. Is assumed.

1: LLC resonant converter, 2: single-phase inverter, 3: LLC transformer, 4, 4a, 4b, 4c, 4d, 4j: power transformer for control circuit, 11: main circuit inverter, 12: multiple transformer, 13: for gate Power supply, 14: DC reactor, 16, 16a, 16b, 16c, 16d, 16j: Single-phase bridge rectifier circuit, 17: Power supply circuit, 17a, 17b, 17c, 17d: Gate power supply circuit, 17j: Control power supply circuit, 31: first reactor, 32: primary winding, 33: capacitor, 34: secondary winding, 41: primary winding, 42a, 42b, 42c, 42d, 42j: secondary winding, 100: main circuit, R: Rectifier part, C1, C2: Capacitor, In1, In2: Inverter part, Q1, Q2, Q3, Q4: Semiconductor element, Ti: Input terminal, To: Output terminal

Claims

A first inverter unit that obtains a DC input and gives a high-frequency output; an LLC transformer that converts a voltage of the high-frequency output of the first inverter unit; a rectifier unit that converts the output of the LLC transformer into a DC; A semiconductor element that constitutes the second inverter section through a rectifier circuit from a second inverter section that converts direct current output to alternating current and a control circuit power transformer that is connected in parallel to the secondary circuit of the LLC transformer And a power supply circuit that obtains the gate power of the semiconductor element and the power of the control circuit that supplies the gate signal of the semiconductor element, and the DC voltage of the rectifier circuit is divided and introduced into the control circuit as a DC output of the rectifier unit. Power converter.
The power conversion device according to claim 1,
The LLC transformer has a primary circuit in which a first reactor, a second reactor as a primary winding, and a capacitor are arranged in series.
The power conversion device according to claim 1 or 2, wherein
The control circuit power transformer includes one primary winding connected in parallel to the secondary circuit of the LLC transformer, and a plurality of secondary windings electromagnetically coupled to the primary winding. A power conversion apparatus characterized in that a gate power of each semiconductor element constituting the second inverter unit and a power of a control circuit that provides a gate signal of the semiconductor element are obtained from an output of a line.
The power conversion device according to claim 1 or 2, wherein
The control circuit power transformer includes a plurality of primary windings connected in series in parallel with the secondary circuit of the LLC transformer, and a plurality of secondary windings electromagnetically coupled to the respective primary windings, A power conversion apparatus characterized in that gate power of each semiconductor element constituting the second inverter unit and power of a control circuit that provides a gate signal of the semiconductor element are obtained from an output of each secondary winding.
The power conversion device according to claim 1 or 2, wherein
The control circuit power transformer includes a plurality of primary windings connected in parallel to the secondary circuit of the LLC transformer, and a plurality of secondary windings electromagnetically coupled to the respective primary windings, A power conversion apparatus characterized in that gate power of each semiconductor element constituting the second inverter unit and power of a control circuit that provides a gate signal of the semiconductor element are obtained from an output of each secondary winding.
The power conversion device according to claim 1 or 2, wherein
The control circuit power transformer includes a primary winding connected in parallel to the secondary circuit of the LLC transformer, and a plurality of secondary windings electromagnetically coupled to the primary winding. A first control circuit power transformer for providing gate power of each semiconductor element constituting the second inverter unit from an output; a primary winding connected in parallel to a secondary circuit of the LLC transformer; and the primary winding A second control circuit power transformer including a secondary winding electromagnetically coupled to the wire, and obtaining power of the control circuit for providing a gate signal of the semiconductor element from an output of the secondary winding;
The power converter according to claim 1, wherein a DC voltage of the rectifier circuit of the first control circuit power transformer is divided and introduced into the control circuit as a DC output of the rectifier unit.
The power conversion device according to claim 1 or 2, wherein
The control circuit power transformer includes a plurality of primary windings connected in series in parallel with the secondary circuit of the LLC transformer, and a plurality of secondary windings electromagnetically coupled to the respective primary windings, A power conversion apparatus characterized in that gate power of each semiconductor element constituting the second inverter unit and power of a control circuit that provides a gate signal of the semiconductor element are obtained from an output of each secondary winding.
A power supply device using the power conversion device according to any one of claims 1 to 7,
A power supply device comprising: a plurality of output circuits of the second inverter section connected in series for each phase to produce a multiphase AC output.
A power supply device using the power conversion device according to any one of claims 1 to 7,
The output circuit of the second inverter unit provides a multiphase AC output.
A power conversion device according to any one of claims 1 to 7, or a control method for a power supply device according to claim 8 or claim 9,
Regarding the resonance frequency in the LLC transformer and the drive frequency in the first inverter unit, an operation mode in which the drive frequency is equal to the resonance frequency, an operation mode in which the drive frequency is higher than the resonance frequency, and the drive frequency is A control method, wherein the first inverter unit is controlled by switching an operation mode lower than the resonance frequency.
The control method according to claim 10, comprising:
In the region where the input voltage of the first inverter unit is lower than the first voltage, the driving frequency is lower than the resonance frequency, and the input voltage of the first inverter unit is higher than the second voltage. In the region, the driving frequency is higher than the resonance frequency, and the input voltage of the first inverter unit is higher than the first voltage and lower than the second voltage, the driving frequency is set to the resonance frequency. The control method characterized by controlling to the driving | operation aspect made the same.