WO2017077599A1

WO2017077599A1 - Motor control device, vacuum cleaner, and hand dryer

Info

Publication number: WO2017077599A1
Application number: PCT/JP2015/081041
Authority: WO
Inventors: 崇山川; 裕次 ▲高▼山; 啓介植村; 篠本　洋介
Original assignee: 三菱電機株式会社
Priority date: 2015-11-04
Filing date: 2015-11-04
Publication date: 2017-05-11
Also published as: JPWO2017077599A1; JP6516863B2

Abstract

Provided is a motor control device that outputs a drive signal to a single-phase inverter for driving a single-phase motor, the motor control device performing PWM control on the single-phase inverter on the basis of a detected value of a motor current flowing to the single-phase motor. The motor control device executes synchronous PWM control in which a carrier is synchronized to the phase of a voltage command, and asynchronous PWM control in which the carrier is not synchronized to the phase of the voltage command, and the motor control device switches from asynchronous PWM control to synchronous PWM control according to an increase in the number of rotations of the motor. In the operation sections in which asynchronous PWM control and synchronous PWM control are executed, the carrier frequency used for asynchronous PWM control and synchronous PWM control is set to be lower than an upper limit frequency which is the upper limit of the carrier frequency.

Description

Motor control device, vacuum cleaner and hand dryer

The present invention relates to a motor control device that controls a single-phase inverter that drives a single-phase electric motor (hereinafter referred to as “single-phase motor”), a vacuum cleaner that uses the single-phase motor, and a hand dryer.

¡Motors used in vacuum cleaners and hand dryers are driven at a high speed of tens of thousands rpm (revolution per minute) or more by an inverter in a high-speed rotation range. In such a motor driven at a high speed, an inverter may be driven by one-pulse switching in a high-speed rotation range (see, for example, Patent Document 1 below).

Japanese Patent Laid-Open No. 2015-342

In a motor control device that performs switching control synchronized with a sensor cycle based on a signal from a position sensor, if one-pulse switching control disclosed in Patent Document 1 is performed, it may be difficult to ensure controllability of the motor. . Here, “securing the controllability of the motor” means maintaining a good performance index of the motor such as the rotation speed, noise or torque. Note that as the motor becomes smaller and the inertia of the motor becomes smaller, it becomes more difficult to ensure controllability of the motor, and it becomes difficult to realize stable motor control.

In general, in order to ensure the controllability of the motor, it is necessary to control the inverter at a carrier frequency that is about ten times the operating frequency, that is, the inverter output frequency (hereinafter referred to as “inverter frequency”). Become. The numerical value of 10 times the operating frequency is the same in the high-speed rotation range.

However, if the carrier frequency is set high in order to ensure controllability, the amount of heat generated by the switching element increases due to an increase in switching loss, and it is necessary to secure a heat dissipation performance structure, resulting in an increase in the size of the apparatus.

The present invention has been made in view of the above, and an object thereof is to provide a motor control device capable of realizing stable motor control while suppressing an increase in size of the device.

In order to solve the above-described problems and achieve the object, a motor control device according to the present invention PWM-controls the single-phase inverter based on a detected value of a motor current flowing in a single-phase motor driven by a single-phase inverter. A motor control device including a control unit, wherein synchronous PWM control in which a carrier is synchronized with a phase of a voltage command and asynchronous PWM control in which the carrier is not synchronized with a phase of the voltage command are executed, and In response to the increase, the asynchronous PWM control is switched to the synchronous PWM control, and the carrier frequency used for the asynchronous PWM control and the synchronous PWM control is set lower than the upper limit frequency that is the upper limit value of the carrier frequency.

According to the present invention, there is an effect that stable motor control can be realized while suppressing enlargement of the apparatus.

1 is a block diagram showing a configuration of a motor control system according to a first embodiment. The block diagram which shows the relationship between the input / output signal in the PWM control part of Embodiment 1, and an internal structure part 1 is a block diagram showing a detailed configuration of a PWM control unit according to Embodiment 1 Waveform diagram showing relationship between voltage command and carrier, PWM signal, and motor applied voltage in the first embodiment Block diagram showing an example of the configuration of an AD converter The figure which shows an example of transmission / reception of the signal between a processor and AD converter Timing chart for explaining timing of starting AD converter and reading digital data from AD converter Flow chart for determining the prohibited range Waveform diagram showing an example of variation of combination of carrier and voltage command in synchronous PWM control Waveform diagram showing motor applied voltage in comparison between asynchronous PWM control and synchronous PWM control The figure which compared and showed electric current THD in the case of asynchronous PWM control, and electric current THD in the case of synchronous PWM control Comparison chart comparing carrier frequency with generated noise and leakage current Timing chart for explaining timing of starting AD converter and reading digital data from AD converter when synchronous PWM control is applied (in case of synchronous 9 pulses) Timing chart for explaining timing of starting AD converter and reading digital data from AD converter when synchronous PWM control is applied (in case of synchronous 8 pulses) The figure for demonstrating the estimation method of the electric current value of the motor current in a prohibition period Timing chart for explaining timing of starting AD converter and reading digital data from AD converter when synchronous PWM control is applied (in case of synchronous 3 pulses) The figure which shows an example of a structure of the vacuum cleaner in Embodiment 2. The figure which shows an example of a structure of the hand dryer in Embodiment 2. The time chart which shows an example of the operation pattern which applied the control method which concerns on Embodiment 1 to the vacuum cleaner or the hand dryer The time chart which shows the other example of the operation pattern which applied the control method which concerns on Embodiment 1 to the vacuum cleaner or the hand dryer

Hereinafter, a motor control device, a vacuum cleaner, and a hand dryer according to an embodiment of the present invention will be described in detail based on the drawings. In addition, this invention is not limited by the following embodiment.

Embodiment 1 FIG.
FIG. 1 is a block diagram illustrating a configuration of a motor control system 1 according to the first embodiment. As shown in FIG. 1, the motor control system 1 includes a single-phase motor 12, a single-phase inverter 11 that is connected to the single-phase motor 12 and supplies AC power to the single-phase motor 12, and a direct current to the single-phase inverter 11. A power source 10 serving as a power source, a position sensor 21 that is provided in a stator (not shown) of the single-phase motor 12 and detects a rotor rotational position that is a rotational position of a rotor (not shown) of the single-phase motor 12, and an alternating current that flows through the single-phase motor 12 A current sensor 20 that is a current detection unit that detects a motor current that is a current; a voltage sensor 23 that detects a DC applied voltage that is a DC voltage that the power supply 10 applies to the single-phase inverter 11; a rotor rotational position, a motor current, and And a motor control device 2 that controls a single-phase inverter 11 that drives a single-phase motor 12 based on a DC applied voltage.

The single phase motor 12 is a brushless motor. That is, the rotor has a plurality of permanent magnets (not shown) arranged in the circumferential direction. The plurality of permanent magnets are arranged such that the magnetization direction is alternately reversed in the circumferential direction to form a plurality of magnetic poles of the rotor. A winding (not shown) is wound around the stator. The motor current is an alternating current flowing through the winding. In the following, the number of magnetic poles is four, but it may be other than this.

The position sensor 21 outputs a position sensor signal that is a digital signal to the motor control device 2. The position sensor signal is a signal for detecting the rotational position of the rotor, and shows a binary value of high and low according to the direction of the magnetic flux from the rotor. Therefore, the edge included in the position sensor signal corresponds to between the magnetic poles.

The single-phase inverter 11 that is a power converter includes switching elements 11a1 to 11a4 and diodes 11b1 to 11b4 that are free-wheeling diodes connected in antiparallel to the switching elements 11a1 to 11a4. In FIG. 1, bipolar transistors are illustrated as the switching elements 11a1 to 11a4. However, field effect transistors represented by MOSFET (Metal-Oxide-semiconductor Field Effect Transistor) may be used. In the case of using a MOSFET, parasitic diodes may be used as the diodes 11b1 to 11b4.

The current sensor 20 is connected between the single-phase motor 12 and the single-phase inverter 11 and detects the motor current. The detection value of the current sensor 20 is analog data. The voltage sensor 23 detects a DC applied voltage applied to the single-phase inverter 11 by the power supply 10. The detection value of the voltage sensor 23 is also analog data.

The motor control device 2 is detected by an analog-digital converter (hereinafter referred to as “AD converter”) 30 that converts analog data, which is a detected value of the motor current detected by the current sensor 20, into digital data, and a voltage sensor 23. AD converter 35 that converts analog data, which is a detected value of the DC applied voltage, into digital data, a motor current read from AD converter 30, a DC applied voltage read from AD converter 35, and position sensor 21 A control unit 25 that generates a PWM (Pulse Width Modulation) signal based on a position sensor signal from and a rotation speed command (rotation speed command value) (not shown), and a PWM signal output from the control unit 25 Based on this, drive signals for driving the switching elements 11a1 to 11a4 in the single-phase inverter 11 are generated. And a motion signal generating unit 32.

The control unit 25 includes a processor 31, a memory 34, and a PWM control unit 40. The control unit 25 generates a PWM signal described later by known PWM control. The drive signal generation unit 32 generates a drive signal for controlling on / off of the switching elements 11a1 to 11a4 in the single-phase inverter 11 based on the PWM signal from the control unit 25 and outputs the drive signal to the single-phase inverter 11. To do.

Next, the PWM control unit 40 will be described with reference to FIGS. 1 to 3. FIG. 2 is a block diagram illustrating a relationship between input / output signals and internal components in the PWM control unit 40 of the first embodiment. FIG. 3 is a block diagram illustrating a detailed configuration of the PWM control unit 40 according to the first embodiment.

The PWM control unit 40 includes a carrier generation unit 33 and a carrier comparison unit 38 as shown in FIG. 2, the voltage phase θv is input to the carrier generation unit 33, and the voltage phase θv, the DC applied voltage Vdc, and the voltage command Vm are input to the carrier comparison unit 38. The carrier generation unit 33 generates a carrier based on the voltage phase θv. The carrier comparison unit 38 generates the PWM signals Q1 to Q4 based on the carrier, the DC applied voltage Vdc, and the voltage command Vm.

The carrier generation unit 33 includes a carrier frequency setting unit 33a as shown in FIG. The carrier frequency setting unit 33 a sets a carrier frequency fc [Hz] that is a carrier frequency, generates a carrier synchronized with the voltage phase θv, and outputs the carrier to the carrier comparison unit 38 and the processor 31. As an example of the carrier waveform, a triangular wave carrier that goes up and down between “0” and “1” is shown at the tip of the arrow of the carrier frequency setting unit 33a. Note that the PWM control described in this embodiment includes synchronous PWM control and asynchronous PWM control. However, in the case of asynchronous PWM control, it is not necessary to synchronize with the voltage phase θv.

Further, as shown in FIG. 3, the carrier comparison unit 38 includes an absolute value calculation unit 38a, a division unit 38b,

multiplication units

38c, 38d, and 38f, an addition unit 38e,

comparison units

38g and 38h, and

output inversion units

38i and 38j. Have. The absolute value calculator 38a calculates the absolute value | Vm | of the voltage command Vm. In the dividing unit 38b, the absolute value | Vm | is divided by the DC applied voltage Vdc detected by the voltage sensor 23. In FIG. 1, the DC applied voltage Vdc is detected by the voltage sensor 23, but when the output voltage of the power supply 10 is stable, such as when the power supply 10 is connected to a commercial power supply, A value generated inside the PWM control unit 40 or a value transmitted from the processor 31 may be used without using the detection value of the voltage sensor 23.

The multiplication unit 38c calculates the sine value of the voltage phase θv and multiplies the output of the division unit 38b. In the division unit 38d, the output of the multiplication unit 38c is multiplied by 1/2. In the

adder

38e, 1/2 is added to the output result of the divider 38d. The multiplication unit 38f multiplies the output of the division unit 38b by -1. Here, the output of the adder 38e is input to the comparator 38g as a positive voltage command Vp for driving the switching elements 11a1 and 11a3 connected to the high potential side DC bus 11P among the switching elements 11a1 to 11a4. Then, the output of the multiplication unit 38f is input to the comparison unit 38h as a negative side voltage command Vn for driving the switching elements 11a2 and 11a4 connected to the DC bus 11N on the low potential side. The output of the comparison unit 38g becomes a PWM signal to the switching element 11a1, and the output of the output inversion unit 38i obtained by inverting the output of the comparison unit 38g becomes a PWM signal to the switching element 11a2. Similarly, the output of the comparison unit 38h becomes a PWM signal to the switching element 11a3, and the output of the output inversion unit 38j obtained by inverting the output of the comparison unit 38h becomes a PWM signal to the switching element 11a4. The presence of the output inverting unit 38i does not turn on the switching element 11a1 and the switching element 11a2 at the same time, and the presence of the output inverting unit 38j does not turn on the switching element 11a3 and the switching element 11a4 at the same time.

FIG. 4 is a waveform diagram showing the relationship between the voltage command and the carrier, the generated PWM signal, and the voltage applied to the single-phase motor 12 (hereinafter referred to as “motor applied voltage”) in the first embodiment. In FIG. 4, the horizontal axis represents the voltage phase θv, and the vertical axis represents the waveforms of the voltage commands Vp and Vn, the carrier, the PWM signals Q1 to Q4, and the motor applied voltage in order from the upper stage.

In FIG. 4, nine carrier peaks or troughs are included in one cycle of the voltage commands Vp and Vn, and the voltage commands Vp and Vn and the carrier are synchronized with respect to the voltage phase θv. That is, the waveform shown in FIG. 4 is a synchronous PWM waveform. The waveform shown in FIG. 4 is referred to as “synchronous 9 pulse waveform” or simply “synchronous 9 pulse” from the number of carriers included in one period of the voltage command.

In the case of synchronous PWM control, the carrier generation unit 33 generates a carrier synchronized with the voltage phase θv. At this time, the carrier comparison unit 38 compares the magnitudes of the carrier and the voltage commands Vp, Vn, and generates a “High” signal if the voltage commands Vp, Vn are larger than the carrier, so that the voltage commands Vp, Vn are carrier signals. If it is smaller than that, a "Low" signal is generated. The width of the “High” or “Low” interval varies depending on the voltage phase θv, and a series of “High” or “Low” signals are PWM signals. The switching elements 11a1 to 11a4 of the single-phase inverter 11 are driven by drive signals based on the PWM signals Q1 to Q4, and a motor voltage as shown in the lower part of FIG. 4 is applied to the single-phase motor 12, and the single-phase motor 12 rotates. To do.

Next, an example of the configuration of the AD converter 30 and the AD converter 35 will be described. The AD converter 30 and the AD converter 35 differ only in the input signals, and the basic operation is the same. Therefore, hereinafter, the AD converter 30 will be described, and the configuration and operation of the AD converter 35 will be omitted. Hereinafter, a case where the AD converter 30 is a successive approximation type will be described. However, the specific configuration of the AD converter 30 is not limited to the successive approximation type.

FIG. 5 is a block diagram showing an example of the configuration of the AD converter 30. As shown in FIG. 5, the AD converter 30 includes a control circuit 51, a comparator 52, and a DA (digital analog) converter 53. The details of the operation of the AD converter 30 shown in FIG. 5 are described in Japanese Patent Application Laid-Open No. 5-152960.

The control circuit 51 has a processor (not shown). The comparator 52 compares the comparison signal COM from the DA converter 53 with the analog input signal AIN and outputs the comparison result to the control circuit 51. The analog input signal AIN corresponds to the motor current. The control circuit 51 outputs a control signal CN approximating the analog input signal AIN to the DA converter 53 according to the comparison result. The DA converter 53 outputs a comparison signal COM corresponding to the control signal CN to the comparator 52. The control circuit 51 obtains digital data DOUT corresponding to the analog input signal AIN by executing a control that sequentially approximates the comparison signal COM to the analog input signal AIN. The control circuit 51 holds the digital data DOUT in a register (not shown).

FIG. 6 is a diagram illustrating an example of transmission / reception of signals between the processor 31 and the AD converter 30. First, the processor 31 outputs an activation signal S1 to the AD converter 30. The activation signal S1 is a signal that instructs the AD converter 30 to start AD conversion. The processor 31 activates the AD converter 30 by outputting the activation signal S1.

When the AD converter 30 receives the start signal S1, the AD converter 30 starts AD conversion processing for converting analog data into digital data. Specifically, when the control circuit 51 of the AD converter 30 receives the activation signal S1, the successive approximation process is started.

The AD converter 30 outputs a completion signal S2 indicating that the AD conversion is completed to the processor 31 after the AD conversion is completed. Specifically, the control circuit 51 outputs a completion signal S2 to the processor 31. After receiving the completion signal S <b> 2 from the AD converter 30, the processor 31 reads digital data from the AD converter 30. Specifically, the processor 31 reads digital data stored in a register in the control circuit 51.

As described above, the AD converter 30 starts the AD conversion process when the activation signal S1 is input from the processor 31, outputs the completion signal S2 to the processor 31 when the AD conversion process is completed, and stops the AD conversion process. To do.

Next, with reference to FIG. 7, the timing of starting the AD converter 30 and reading digital data from the AD converter 30 will be described. FIG. 7 is a timing chart for explaining the timing of starting the AD converter 30 and reading the digital data from the AD converter 30. The example of FIG. 7 is an example in which the motor current cycle T _I and the carrier cycle T _c are not in an integer multiple relationship, that is, in the case of asynchronous PWM control.

7, “position sensor signal” represents an output signal of the position sensor 21 input to the processor 31. The angle given immediately below the position sensor signal is the mechanical angle of the rotor. The position sensor signal includes edges at mechanical angles of 0 °, 90 °, 180 °, 270 °, 360 ° corresponding to a 4-pole rotor.

“Rotor rotation angle” represents the electrical angle of the rotor. That is, the electrical angle = mechanical angle × P / 2, where P is the number of magnetic poles. The processor 31 calculates the rotor rotation angle based on the position sensor signal. The angle given immediately below the rotor rotation angle is an electrical angle.

“Motor current” represents the motor current waveform. “Motor current” is shown for comparison with “rotor rotation angle”. As shown in FIG. 7, the edge of the position sensor signal is synchronized with the zero cross point of the motor current. Here, the zero cross point is a change point of polarity in the waveform of the signal, and is a point at which the polarity is switched from positive to negative or from negative to positive. In FIG. 7, the zero cross points A1 and A2 adjacent to each other are shown. The period from the zero cross point A1 to the zero cross point A2 is an electric half cycle of the motor current determined by the zero cross points A1 and A2. Thus, in the following, a case where control is performed in which the edge of the position sensor signal is synchronized with the zero cross point of the motor current will be described first. In this case, the rotor rotation angle gives phase information of the motor current.

Next, “AD converter operation timing” and “carrier” will be described. “AD converter operation timing” represents AD conversion processing, and “carrier” represents a carrier waveform. In the present embodiment, a certain phase angle range including the zero cross point is a prohibited range in which the start of the AD converter 30 is prohibited. Specifically, the total 2α phase angle range including α before and after the zero cross point is set as the prohibited range. The processor 31 does not output the start signal S1 to the AD converter 30 within the prohibited range, and the AD converter 30 to which the start signal S1 is not input does not execute the AD conversion process, indicating that the AD conversion process is completed. The signal S2 is not output.

The range other than the prohibited range is a permitted range in which activation of the AD converter 30 is permitted. Note that reading of digital data from the AD converter 30 is permitted regardless of the prohibited range or the permitted range. The permitted range is between adjacent prohibited ranges. A period corresponding to the prohibited range, that is, a time when the prohibited range is replaced with time is hereinafter referred to as a prohibited period. Similarly, a period corresponding to the permission range is hereinafter referred to as a permission period. The prohibited period is substantially the same as the prohibited range, and the permitted period is substantially the same as the permitted range.

With period T _I is an electrical cycle of the motor current, the length of the protection period is given by 2 × (α / 360) × T I, the permission period during a half cycle which is an electrical half-cycle of the motor current The length is given by T _I / 2-2 × (α / 360) × T _I.

Α is a predetermined angle greater than 0 and less than 90 °. In the illustrated example, α is 10 °. In this case, the prohibited range is -10 ° or more and 10 ° or less, 170 ° or more and 190 ° or less, 350 ° or more and 370 ° or less, 530 ° or more and 550 ° or less. The range is from 710 ° to 730 °. After calculating the rotor rotation angle, the processor 31 determines the prohibition range and the permission range based on the rotor rotation angle and a predetermined α.

In the example of FIG. 7, activation of the AD converter 30 and reading from the AD converter 30 by the processor 31 are performed at the timing of the carrier peak generated by the carrier generation unit 33. It may be performed at a timing other than the above, or may be performed at the timing of a valley point, or at the timing of both a mountain point and a valley point. Further, the timing may be determined without depending on the carrier. However, the AD conversion processing for one sampling data is executed in a time shorter than the carrier cycle _Tc .

Next, “AD converter operation timing” will be described in detail. In the following description, “period” is used, but it can be read as “range”. Based on the calculated rotor rotation angle, the processor 31 determines whether or not the time t0 that is the timing of the peak point is within the permission period. Since the time t0 is within the prohibition period, the processor 31 does not start the AD converter 30.

Next, based on the calculated rotor rotation angle, the processor 31 determines whether or not the time t1 that is the timing of the peak point following the time t0 is within the permission period. Since the time t1 is within the permission period, the processor 31 outputs the activation signal S1 to the AD converter 30 at the time t1. Note that the permission period in this case is included in the electrical half cycle from the zero cross point A1 to the zero cross point A2. When receiving the activation signal S1, the AD converter 30 executes AD conversion processing. In FIG. 7, the range during AD conversion is indicated by “AD conversion” with diagonal lines. When the AD converter 30 completes the AD conversion process, the AD converter 30 outputs a completion signal S2 to the processor 31, and the processor 31 receives the completion signal S2 from the AD converter 30.

Note that the operation waveform of FIG. 7 is an example in which two or more peak points of the carrier are included in one permission period. For this reason, time t2, which is the timing of the peak point following time t1, is within the permission period. Therefore, the processor 31 reads digital data from the AD converter 30 and outputs a start signal S1 to the AD converter 30 at time t2. The processor 31 uses this digital data for control. Upon receiving the activation signal S1, the AD converter 30 executes AD conversion processing and rewrites the register with the digital data after AD conversion. When the AD converter 30 completes the AD conversion process, the AD converter 30 outputs a completion signal S2 to the processor 31, and the processor 31 receives the completion signal S2 from the AD converter 30.

Next, the processor 31 determines whether or not the time t3 that is the timing of the peak point following the time t2 is within the permission period based on the calculated rotor rotation angle. Since the time t3 is within the permission period, the processor 31 reads the digital data from the AD converter 30 and outputs the activation signal S1 to the AD converter 30 at the time t3. The processor 31 uses this digital data for control. Upon receiving the activation signal S1, the AD converter 30 executes AD conversion processing and rewrites the register with the digital data after AD conversion. When the AD converter 30 completes the AD conversion process, the AD converter 30 outputs a completion signal S2 to the processor 31, and the processor 31 receives the completion signal S2 from the AD converter 30.

Next, the processor 31 determines whether or not the time t4 that is the timing of the peak point following the time t3 is within the permission period based on the calculated rotor rotation angle. Since the time t4 is within the permission period, the processor 31 reads the digital data from the AD converter 30 and outputs the activation signal S1 to the AD converter 30 at the time t4. The processor 31 uses this digital data for control. Upon receiving the activation signal S1, the AD converter 30 executes AD conversion processing and rewrites the register with the digital data after AD conversion. When the AD converter 30 completes the AD conversion process, the AD converter 30 outputs a completion signal S2 to the processor 31, and the processor 31 receives the completion signal S2 from the AD converter 30.

Further, based on the calculated rotor rotation angle, the processor 31 determines whether or not the time t5 that is the timing of the peak point following the time t4 is within the permission period. Since the time t5 is within the prohibited period, the activation signal S1 is not output to the AD converter 30. Further, the processor 31 determines whether or not the time t5 is in the electrical half cycle from the zero cross point A1 to the zero cross point A2. Since time t5 is in the electrical half cycle, the processor 31 reads digital data from the AD converter 30 at time t5. The processor 31 uses this digital data for control.

The processor 31 repeats the above operation during the operation of the single-phase motor 12.

FIG. 8 is a flowchart for determining the prohibited range. As shown in FIG. 8, the processor 31 calculates the rotor rotation angle from the position sensor signal (step S101), calculates the zero cross point of the motor current based on the calculated rotor rotation angle (step S102), and zero cross The range of front and rear α including the point is set as a prohibited range (step S103).

In the above description, it is assumed that the edge of the position sensor signal is synchronized with the zero cross point of the motor current. Here, the control when the edge of the position sensor signal and the zero cross point of the motor current are not synchronized will be briefly described.

The single-phase motor 12 is operated so that the edge of the position sensor signal and the zero cross point of the motor current are synchronized at the start of operation. However, as the rotational speed increases, the edge of the position sensor signal and the zero cross point of the motor current become out of synchronization. In particular, when the rotational speed reaches a so-called high rotational speed region such as 70,000 rpm or more, the edge of the position sensor signal and the zero cross point of the motor current become asynchronous. When the asynchronization occurs as described above, it may be necessary to reset the prohibited range set as shown in FIG.

The resetting of the prohibited range is performed as follows. The processor 31 monitors the digital data read from the AD converter 30 in time series, compares the previous value and the current value of the digital data within the same permitted range, and the polarity is switched between the previous value and the current value. Judge whether there is no. When the polarity is switched between the previous value and the current value, a zero-cross point is included in the permitted range. In this case, the prohibited range is reset based on the detected zero-cross point. The permitted range is also reset by resetting the prohibited range. Thereafter, the processor 31 activates the AD converter 30 and reads from the AD converter 30 only within the reset permission range.

The motor current can be detected even in asynchronous PWM control by the above method.

Next, a method for detecting the motor current when the synchronous PWM control is applied will be described. The synchronous PWM control is a method for controlling the carrier frequency to be an integral multiple of the electrical angle cycle of the motor as shown in the waveform shown in the upper part of FIG. When the synchronous PWM control is performed, if the electrical angular frequency is the same, the inverter output voltage during the electrical angular cycle remains the same voltage without changing depending on the cycle.

FIG. 9 is a waveform diagram showing an example of variation of combinations of carrier, voltage command, and synchronous PWM control. The horizontal axis represents voltage phase θv, and the vertical axis represents 3 synchronous pulses, 6 synchronous pulses from the bottom, The waveform of the carrier and the waveform of the voltage command Vp when controlling with 9 synchronous pulses are shown.

In the synchronous PWM control, control is performed so that the carrier frequency is, for example, 3 times, 6 times, or 9 times the frequency of the voltage command Vp (the same applies to the voltage command Vn). When the carrier frequency is changed to 3 times, 6 times, and 9 times, a PWM signal is generated in which the number of pulses included in the half cycle of the carrier is 3 pulses, 6 pulses, and 9 pulses, respectively. Since the carrier and the voltage phase θv are synchronized, these pulses are referred to as “synchronous 3 pulse”, “synchronous 6 pulse”, and “synchronous 9 pulse”. Although not shown in FIG. 9, the carrier frequency can be set to a frequency higher than nine times. However, since the number of pulses of the PWM signal increases with respect to one cycle of the voltage command Vp, the accuracy of the output voltage is improved, while the number of times of switching of the switching elements 11a1 to 11a4 is increased. That is, an increase in carrier frequency leads to an increase in switching loss, and an increase in carrier frequency and an increase in switching loss are in a trade-off relationship.

FIG. 10 is a waveform diagram showing a comparison between motor applied voltages in asynchronous PWM control and synchronous PWM control, and FIG. 11 is a graph showing total harmonic distortion (Total Harmonic Distortion: “Current” in the case of asynchronous PWM control). FIG. 6 is a diagram comparing and comparing current THD in the case of synchronous PWM control. In the case of asynchronous PWM control, as shown in the upper part of FIG. 10, the motor applied voltage on the positive voltage side and the motor applied voltage on the negative voltage side are at the center time t _m during one cycle when the positive voltage is switched to the negative voltage. On the other hand, the left and right are asymmetrical waveforms, and imbalance of the applied voltage occurs. On the other hand, in the case of synchronous PWM control, the left and right are symmetrical waveforms with respect to the center time t _{m in} one cycle, and the imbalance of the motor applied voltage is suppressed. For this reason, as shown in FIG. 11, the current THD in the case of synchronous PWM control is smaller than the current THD in the case of asynchronous PWM control. As a result, in the case of synchronous PWM control, torque pulsation generated due to current distortion can be suppressed, and generation of vibration and noise due to pulsation of the rotational speed of the single-phase motor 12 can be suppressed.

In the case of asynchronous PWM control, it is possible to suppress distortion of the output voltage when the carrier frequency is sufficiently high with respect to the voltage command, but when the carrier frequency is low with respect to the voltage command, the output voltage It is difficult to suppress the distortion. Therefore, in the case of asynchronous PWM control, it is preferable to set the carrier frequency to 10 times or more with respect to the voltage command.

On the other hand, in the case of synchronous PWM control, current pulsation can be suppressed even when the carrier frequency is lower than the voltage command. For example, a low carrier frequency such as 3 to 9 times the voltage command is selected. It becomes possible. For this reason, when the synchronous PWM control and the asynchronous PWM control are used together as in the present embodiment, the single-phase motor 12 can be operated even when the carrier frequency is reduced as compared with the conventional case where the control is performed only by the asynchronous PWM control. It becomes possible to drive stably.

FIG. 12 is a comparison diagram comparing the carrier frequency, generated noise, and leakage current with the conventional one. When the carrier frequency is reduced by using synchronous PWM control together, the number of switching times of the switching elements 11a1 to 11a4 is reduced. Therefore, noise generated in the single-phase inverter 11 and current leaked from the single-phase motor 12 are shown in FIG. As shown, it can be made lower than in the conventional case where only asynchronous PWM control is performed.

FIG. 13 is a timing chart for explaining the timing of starting the AD converter 30 and reading digital data from the AD converter 30 when the synchronous PWM control is applied. FIG. 13 shows an operation waveform in the case of synchronous 9 pulses, and the edge of the position sensor and the zero cross of the motor current are synchronized.

If synchronous PWM control is applied, motor control can be performed with an arbitrary number of carriers (9 in the figure) for one cycle of the rotor rotation angle, so the AD conversion timing should be the same for the current phase. Can do. In addition, since the prohibition period can be easily determined, it is easy to estimate the relationship between the current value obtained by AD conversion and the phase, and high-precision motor control is possible. Furthermore, since the carrier frequency is changed following the change of the rotor rotational frequency, the number of current data obtained in one cycle can be ensured regardless of the rotor rotational speed.

Further, from the relationship between the AD conversion prohibition period α and the number of synchronous PWM pulses, it is possible to estimate the number of times that AD conversion can be performed during one rotor rotation angle. Conversely, it is also possible to select the number of synchronous PWM pulses from the number of AD conversion implementations required in one cycle (that is, the number of motor current detections). The explanation here is for the case where the edge of the position sensor and the zero crossing of the motor current are synchronized. However, the number of AD conversion executions is obtained from the AD conversion inhibition period and the number of synchronous PWM pulses, and AD conversion is performed. The selection of the number of synchronous PWM pulses from the number of times can be performed even in an asynchronous case.

Next, the effect of this embodiment will be described. It is known that noise occurs at the zero cross point of the motor current. Specifically, since noise is generated when the switching elements 11a1 to 11a4 of the single-phase inverter 11 are turned on or off, noise caused by switching is included in the motor current at the zero cross point where the current polarity is switched. At the zero cross point of the motor current, a recovery current flows through the diodes 11b1 to 11b4 connected in antiparallel to the switching elements 11a1 to 11a4, and this recovery current also causes noise.

On the other hand, in the present embodiment, a certain range including each zero cross point of the motor current is set as a prohibition range in which the AD converter 30 is prohibited from starting, and the AD converter 30 is started between adjacent prohibition ranges. The permitted range is set to allow

The processor 31 outputs the start signal S1 to the AD converter 30 within the permitted range, and reads the digital data in the electrical half cycle determined by the adjacent zero cross points including the permitted range. Here, the permission period is a continuous period not including the zero cross point. As a result, the processor 31 can use the digital data converted by the AD converter 30 activated within the permitted range not including the zero-cross point for control, thereby suppressing the influence on noise control, Stable motor control can be realized.

In addition, since the influence of noise can be suppressed, the quality of the electric device provided with the motor control system 1 can be improved. Further, by suppressing the influence of noise, the filter constant can be reduced even when a filter for noise removal is provided in the motor control system 1, so that the filter can be miniaturized and the parts can be miniaturized. it can.

In general, the processor 31 activates the AD converter 30 within a continuous period that does not include the zero cross point in the electrical half cycle that is a period between adjacent zero cross points of the motor current, and the AD converter 30 is activated. The digital data obtained by starting the AD converter 30 is read during the same electrical half cycle. Thereby, the effect of this Embodiment mentioned above is acquired.

Here, a continuous period that does not include the zero cross point of the motor current is synonymous with the same permission period, and means that it does not consist of a plurality of periods across the zero cross point. In other words, in the present embodiment, the AD converter 30 is activated in the first permission period, and conversion processing is performed by the input of the activation signal S1 from the AD converter 30 in the second permission period after the first permission period. Processing such as reading digital data is eliminated.

As is clear from the above description, the continuous period not including the zero cross point of the motor current is shorter than a half cycle of the motor current and is set to a point between adjacent zero crosses.

In the present embodiment, the following effects can be obtained by using synchronous PWM control together.

First, AD conversion timing can be assumed in advance by synchronous PWM control, so current detection conditions (number of detections, phase, etc.) according to the number of rotations can be set optimally, and highly stable motor control Is possible.

Also, synchronous PWM control makes it possible to reduce the carrier frequency and reduce the switching frequency of the switching element, reduce the switching loss of the inverter, and realize a highly efficient drive system. Furthermore, by reducing the switching loss, it is possible to suppress the heat generation of the inverter board, and the heat dissipating fins can be reduced in size and reduced. Furthermore, by reducing the number of radiation fins, the system and products can be reduced in size and weight. Furthermore, since the degree of freedom is improved with respect to the heat dissipation structure, it is possible to reduce restrictions on the air path and arrangement for heat dissipation. In particular, in the case of air cooling, since it is not necessary to apply outside air to the substrate, it is possible to suppress deterioration of reliability such as short circuit, dielectric breakdown, element degradation, etc. due to dust, moisture, etc. in air. A reliable system can be realized.

Moreover, since the influence of the inverter short-circuit prevention time (dead time) can be suppressed by reducing the carrier frequency and the upper limit of the inverter output voltage can be suppressed, power is supplied from the power system such as a battery. In the case of a system (product) that does not exist, it is possible to reduce the size and weight of the power supply unit and the system and extend the period of use together with the improvement of efficiency.

In addition, the synchronous PWM control makes it possible to suppress the induced voltage distortion of the motor, and to reduce the noise caused by the iron loss and speed fluctuation caused by the voltage distortion. A system can be realized.

Further, as described with reference to FIG. 12, noise generated in the single-phase inverter 11 and leakage current from the single-phase motor 12 can be suppressed by reducing the carrier frequency. By suppressing noise and leakage current, the reliability of the device is improved and the cost for noise countermeasures can be reduced. For this reason, a thin material such as a PET film having a larger capacitance than that of a conventional insulating material may be wound around the slot of the stator as insulation, and the space factor of the winding can be improved. Further, even if the efficiency of the motor is improved by improving the space factor of the winding and the leakage current is increased, a countermeasure can be taken without adding an external circuit or the like.

FIG. 14 is a diagram illustrating an example of variations in the number of synchronous pulses in synchronous PWM control. Although FIG. 13 shows the reading timing in the case of synchronous 9 pulses, FIG. 14 shows the reading timing in the case of synchronous 8 pulses. As described above, in this embodiment, current detection is not performed before or after the edge of the position sensor signal synchronized with the zero cross point of the motor current, that is, near the switching of the position sensor signal. For this reason, when the number of synchronization pulses is odd, the relationship between the region where detection is not performed and the peak or valley of the carrier is asymmetric, but when the number of synchronization pulses is even, it is symmetric regardless of whether the current is positive or negative. It becomes. As a result, when the number of synchronization pulses is even, the number of AD conversions that can be performed is the same every time, so that the difference in controllability in phase is smaller and stable control than when the number of synchronization pulses is odd. Has the effect of becoming possible.

Next, motor control within the prohibited period will be described. In this embodiment, the processor 31 does not read from the AD converter 30 during the prohibition period, but estimates the current value of the motor current from the measured value of the motor current obtained in the permission period immediately before the prohibition period. The motor control may be performed using the estimated current value.

FIG. 15 is a diagram for explaining a method of estimating the current value of the motor current within the prohibited period. 15, “position sensor signal” and “motor current” are the same as those in FIG. The “detected current” consists of an actually measured “detected value” and an “estimated value” estimated within the prohibited period. The “detected value” is indicated by a black circle, and the “estimated value” is indicated by a white circle. Points N-3 to N-1 indicate three points actually measured in the permission period immediately before the prohibition period. Point N + 2 and point N + 3 indicate two points actually measured in the permission period immediately after the prohibition period. Point N and point N + 1 indicate two points estimated in the prohibition period. The “estimated value” can be obtained as follows. Since the zero cross point exists within the prohibition period, the motor current can be approximated by a straight line. Therefore, a straight line passing through the two points N-2 and N-1, which are the latest two points actually measured in the permission period immediately before the prohibition period, is obtained, and it is estimated that a current value exists on the straight line. N + 1 can be determined.

Note that the method of estimating the motor current within the prohibited period is not limited to the above example. For example, the point N and the point N + 1 may be estimated by polynomial approximation using a plurality of nearest points actually measured in the permission period immediately before the prohibition period.

In addition, in the motor control, when using vector control for controlling the motor current by dividing the motor current into two orthogonal dq axes, the motor current can be handled as a direct current amount, so that the above current value is accurately estimated. Is possible.

In this embodiment, α defining the prohibited range is set to 10 °, for example, but is not limited to this. However, if α is too large, the number of current values estimated within the prohibited range increases, and if α is too small, there is a possibility of being affected by noise generated at the zero cross point. Therefore, α may be selected from a range of 5 ° or more and 15 ° or less, for example. Further, the prohibited range may be asymmetric with respect to the zero cross point.

In this embodiment, the position sensor 21 is provided in the single-phase motor 12 and the rotor rotation angle is calculated based on the position sensor signal from the position sensor 21. May be estimated. The estimation of the rotational position in a so-called sensorless motor is described in, for example, Japanese Patent No. 5619195.

The switching elements 11a1 to 11a4 and the diodes 11b1 to 11b4 can be formed using a wide band gap semiconductor. The wide gap semiconductor is, for example, GaN (gallium nitride), SiC (silicon carbide), or diamond. By using wide band gap semiconductors for the switching elements 11a1 to 11a4, the voltage resistance and allowable current density of the switching elements 11a1 to 11a4 are increased. Therefore, the switching elements 11a1 to 11a4 can be downsized. The built-in semiconductor module can be downsized. In addition, since the wide band gap semiconductor has high heat resistance, the heat sink fins can be downsized.

7, 13, and 14 exemplify the case where the current detection is performed only with the carrier peak, the current detection may be performed only with the carrier valley. In addition, as shown in FIG. 16, current detection may be performed at both the peak and valley of the carrier. FIG. 16 shows the reading timing in the case of three synchronous pulses as a variation of the number of synchronous pulses in the synchronous PWM control. According to the example shown in FIG. 16, since current detection is performed at both the peak and valley of the carrier, it is possible to ensure the number of times of current detection even if the carrier frequency is reduced, and high efficiency and A highly reliable device can be realized.

In this embodiment, AD conversion is sequentially started in synchronization with the carrier. However, AD conversion processing is always performed in carrier synchronization so that only data reflected in the control is not used. There is no problem.

In general, single-phase motors are more susceptible to rotational speed fluctuations and torque fluctuations (also referred to as “torque pulsation”) than three-phase motors, and smaller motors have less inertia. It is easily affected and is difficult to drive stably under desired operating conditions.

Also, in the case of asynchronous PWM control, since the number of carriers in the electrical angle cycle is not an integer multiple relationship, there is a difference in the inverter output voltage for each electrical angle cycle, and optimal control is difficult. This is because the carrier phase is different from the electrical angle phase in each electrical angle cycle, and thus a control deviation occurs. This causes a change in torque and rotational speed in the motor control.

Therefore, by applying synchronous PWM control to the driving of a single-phase motor as in this embodiment, torque fluctuation and rotational speed fluctuation caused by control other than single-phase motor-specific torque pulsation are suppressed. Therefore, stable motor driving can be realized.

Also, in single-phase motor drive, torque pulsation occurs periodically, so synchronous PWM makes it easy to estimate the relationship between the carrier and motor phase, and inverter output control considering torque pulsation is relatively easy Therefore, stable motor driving can be realized.

Embodiment 2. FIG.
In the first embodiment, the motor control system 1 including the motor control device 2, the single-phase inverter 11, and the single-phase motor 12 has been described. In the present embodiment, an electric device including the motor control system 1 described in the first embodiment will be described. As the electric equipment, a vacuum cleaner and a hand dryer will be described in particular.

FIG. 17 is a diagram illustrating an example of the configuration of the electric vacuum cleaner 61. The vacuum cleaner 61 includes an extension pipe 62, a suction port 63, an electric blower 64, a dust collection chamber 65, an operation unit 66, a battery 67 and a sensor 68. The electric blower 64 includes the motor control system 1 described in the first embodiment. The vacuum cleaner 61 drives the electric blower 64 using the battery 67 as a power source, performs suction from the suction port body 63, and sucks dust into the dust collection chamber 65 through the extension pipe 62. In use, the operation unit 66 is held and the electric vacuum cleaner 61 is operated.

The operation unit 66 has a power switch and an acceleration switch (not shown). Here, the power switch is a switch for switching power supply from the battery 67 to a main circuit and a control circuit (not shown). The acceleration switch is a switch that switches to control for accelerating the electric blower 64 from low speed rotation to steady rotation.

In addition, low speed rotation means rotation of 1/10 or less of steady rotation speed. For example, when the steady rotation speed is 100,000 rotations, rotations of 10,000 rotations or less are low-speed rotations.

When the power switch is turned on and power supply from the battery 67 to the main circuit and the control circuit is started, the sensor 68 also starts detection at the same time.

Sensor 68 detects the movement of the vacuum cleaner 61 or the movement of a person. A low-speed activation of a motor (not shown) in the electric blower 64 is started, triggered by the signal from the sensor 68 that detects the movement of the electric vacuum cleaner 61 or the movement of a person being input into the electric blower 64.

The motor accelerates from the low speed rotation to the steady rotation speed by turning on the above acceleration switch after the start of the low speed start. If the acceleration switch is turned on before the power switch is turned on, the power switch is turned on to accelerate from the start up to the normal rotational speed and to perform normal operation.

Also, when only the acceleration switch is turned off from the state of rotating at the steady rotation speed, the motor continues to operate at a low speed without stopping. By continuing to operate at a low speed, it is possible to suppress the possibility that accumulated dust is discharged from the dust collection chamber 65 through the extension pipe 62 during the movement between cleanings. Also, idling loss during low-speed driving can be suppressed by performing synchronous PWM even at low-speed rotation.

Sensor 68 is a gyro sensor that detects the movement of the vacuum cleaner 61 or a human sensor that detects the movement of a person. In either case of starting up, it is possible to shorten the arrival time to the steady rotational speed. At this time, by applying the motor control system 1 described in the first embodiment to the electric vacuum cleaner 61, the detection accuracy of the analog signal that is the motor current or the motor voltage is improved. Can be stabilized.

The torque T generated when the motor rotates is determined by the product of the torque constant Kt and the motor current Ia as shown in the following equation.

T = Kt × Ia

Thus, since the torque T is proportional to the motor current Ia, it is necessary to generate a larger torque T in order to shorten the acceleration time, and it is also necessary to increase the motor current Ia. By flowing a large current Ia, the power consumption increases, the merit of shortening the operation time is reduced, and the reliability of components including the battery 67 is impaired.

In order to solve such a problem, it is common to control the acceleration rate. For example, by extending the acceleration time until the motor reaches the normal rotation speed, it is possible to extend the operation time and improve the reliability of the parts. At this time, by applying the motor control system 1 described in the first embodiment to the electric vacuum cleaner 61, the detection accuracy of the analog signal that is the motor current or the motor voltage is improved. It becomes possible to suppress the vibration of the rotation speed of the motor.

Furthermore, since the amount of heat generated by the component can be suppressed by suppressing the current that flows during startup, the reliability of the component is also improved.

Also, since the rotational speed increases gradually by slowing acceleration, vibration due to sudden acceleration can be suppressed. By suppressing vibration, it is possible to suppress discomfort to the human body and influence on peripheral devices. Further, by suppressing vibration, it is possible to suppress sound generated from the device.

In addition, when starting from a stationary state by the method as described above, since a larger force is required at the time of starting, more current is required. Therefore, in order to suppress the current peak, it is more effective to control the acceleration at the time of starting small. By applying the motor control system 1 described in the first embodiment to the vacuum cleaner 61, the detection accuracy of an analog signal that is a motor current or a motor voltage is improved, so that the acceleration can be finely controlled.

In addition, these acceleration methods may be provided with a changeover switch so that the user can switch, and the user can set it.

Here, the operation when the gyro sensor is used will be described. First, by manually turning on the power switch, the gyro sensor starts outputting a signal indicating the movement of the electric vacuum cleaner 61. When a signal that detects the movement of the electric vacuum cleaner 61 is output from the gyro sensor, the low speed rotation is started. By manually turning on the acceleration switch, the engine is accelerated from a low speed to a steady speed. When the cleaning is partially completed and moved to the next cleaning place, the low-speed rotation is resumed by manually turning off the acceleration switch. When cleaning again, the acceleration switch is manually turned on to accelerate to the normal rotational speed, and when cleaning is finished, the rotation is stopped by manually turning off the power switch.

The gyro sensor is attached to the vacuum cleaner 61 to detect the movement of the vacuum cleaner 61 that occurs when the vacuum cleaner 61 is used. The main body of the vacuum cleaner 61 always moves immediately before use. Therefore, by attaching a gyro sensor to the vacuum cleaner 61, the movement of the vacuum cleaner 61 can be detected and the vacuum cleaner 61 can be activated in advance. At this time, by applying the motor control system 1 described in the first embodiment to the electric vacuum cleaner 61, the detection accuracy of the analog signal that is the motor current or the motor voltage is improved, so that the acceleration is accelerated to the steady rotational speed more quickly. can do.

FIG. 18 is a diagram illustrating an example of the configuration of the hand dryer 70. The hand dryer 70 includes a casing 71, a hand detection sensor 72, a water receiver 73, a drain container 74, a cover 76, a sensor 77, and an intake port 78. Here, the sensor 77 is either a gyro sensor or a human sensor. The hand dryer 70 has an electric blower (not shown) in the casing 71. The electric blower has the motor control system 1 of the first embodiment. The hand dryer 70 has a structure in which water is blown off by blowing with an electric blower by inserting a hand into a hand insertion portion 79 at the top of the water receiving portion 73 and water is stored from the water receiving portion 73 into the drain container 74. ing.

The operation when the sensor 77 is a human sensor will be described. First, the sensor 77 detects that a person has come to the surroundings, and starts at a low speed. When a person holds his / her hand over the hand dryer 70 in order to dry his / her hands, the speed is accelerated to a steady rotational speed. When the drying is finished and the hand comes out from the hand insertion portion 79, the low speed operation is resumed. If the next person's hand is detected during low-speed driving, the vehicle is accelerated to the steady rotational speed again. If the surrounding people are not detected, the operation stop state is maintained.

Sensor 77 is a sensor that detects, for example, infrared rays, ultrasonic waves, or visible light. In addition, a temperature sensor or a sensor that detects a person by camera recognition may be used.

By attaching a human sensor to the hand dryer 70, it is possible to detect that the user has approached the hand dryer 70 and activate the hand dryer 70 in advance. At this time, by applying the motor control system 1 described in the first embodiment to the hand dryer 70, the detection accuracy of the analog signal, which is the motor current or the motor voltage, is improved. be able to.

Next, an example of an operation pattern when the control method according to the first embodiment is applied to the electric vacuum cleaner 61 or the hand dryer 70 according to the present embodiment will be described. FIG. 19 is a time chart illustrating an example of an operation pattern in which the control method according to the first embodiment is applied to the electric vacuum cleaner 61 or the hand dryer 70. In FIG. 19, (a) shows the change in the motor speed, (b) shows the change in the carrier frequency with respect to the motor speed, and (c) shows the generated noise and the leakage current.

In the example of the operation pattern shown in FIG. 19, asynchronous PWM control is performed in the low speed operation section (operation range (1) in the figure), which is the operation section from the start to the steady operation speed (steady rotation speed), and the steady operation speed. In the high-speed operation section that is an operation section higher than the steady operation speed, synchronous PWM control is performed in the order of synchronous 9 pulses, synchronous 6 pulses, and synchronous 3 pulses in accordance with the increase in the motor rotation speed (rotational speed). Here, two typical examples of the number of synchronization pulses, the motor rotation speed, and the carrier frequency in this operation pattern are shown below.

(First operation pattern)
Asynchronous ／ 9 pulses / 6 pulses / 3 pulses ・ Motor rotation speed: 40000rpm / ～ 50000rpm / ～ 70000rpm / ～ 120,000rpm
・ Carrier frequency: 9kHz / 9 ~ 12kHz / 8 ~ 12kHz / 6 ~ 12kHz

(Second operation pattern)
Asynchronous / 8 pulse / 6 pulse / 4 pulse Motor speed: 60000rpm / ~ 75000rpm / ~ 100,000rpm / ~ 150,000rpm
・ Carrier frequency: 18kHz / 18-20kHz / 15-20kHz / 13.3333-20kHz

Note that the first and second operation patterns shown above are examples, and switching from asynchronous to synchronous and switching of the number of synchronous pulses are not limited to these numerical values.

As shown in FIGS. 19A and 19B, asynchronous PWM control with a fixed carrier frequency is performed at the start-up time when the motor rotation speed is low and in the low-speed operation range (section (1) in the figure). . At that time, the motor rotational speed is increased to the steady operation speed, and then the steady operation speed is maintained. In the section (1), since the carrier frequency is fixed, as shown in FIG. 19C, the generated noise and the leakage current can be suppressed to an allowable value or less.

In addition, in the range of synchronous PWM control, that is, in the section (2) to section (4) in the figure, the asynchronous PWM control is switched to the synchronous PWM control, and the carrier frequency is increased or decreased so that the generated noise and the leakage current are less than the allowable values. However, the number of pulses is switched. In the section (2), the control is performed with 9 synchronous pulses, but the generated noise and the leakage current increase to near the allowable value as the rotational speed increases. Therefore, an upper limit frequency is set as the carrier frequency, and control is performed so that the carrier frequency does not exceed the upper limit frequency. The upper limit frequency of the carrier frequency can be determined based on the processing load of the microcomputer, generated noise, leakage current, and the like.

∙ Reduce the number of synchronization pulses before the carrier frequency reaches or reaches the upper limit frequency. In the section (3), the carrier frequency is reduced at the transition from the section (2) to the section (3), and the number of synchronization pulses is switched from 9 to 6, so that the generated noise and the leakage current do not exceed the allowable values. Is suppressed. Hereinafter, the same applies to the section (4), the carrier frequency is reduced at the transition from the section (3) to the section (4), the number of synchronization pulses is switched from 6 to 3, and the generated noise and leakage current are allowed. It is restrained not to exceed. As described above, the asynchronous PWM control and the synchronous PWM control are used in combination, and the number of synchronous pulses is reduced stepwise, thereby suppressing the generated noise and the leakage current, and the short time from the startup to the high speed operation range. Smooth acceleration is achieved.

FIG. 20 shows an example of an operation pattern different from that in FIG. FIG. 20C shows the characteristics of switching loss and module temperature.

In the operation pattern shown in FIG. 19, the range of the section (2) is narrowed and shifted to the section (3) due to the relationship between the generated noise and leakage current and their allowable values. When there is a margin, the operation pattern can be determined in consideration of the switching loss and the module temperature. In the operation pattern shown in FIG. 20, since the switching loss and the module temperature have a relatively large margin, the range of the section (2) can be widened. For this reason, synchronous PWM control with a margin compared to the operation pattern shown in FIG. 19 is possible.

In the above description, the operation switching of the starting 9 pulse, 6 pulse, and 3 pulse has been described. Needless to say, the number of pulses other than these may be controlled as shown in the second example. Control of the number of pulses exceeding 9 pulses (for example, 21 pulses, 15 pulses, 12 pulses) is also included in the gist of the present invention. It should be noted that the relationship of integer multiples may be momentarily or temporarily disrupted due to fluctuations, etc., but it is needless to say that such a case also has an integer multiple relationship. Further, in a condition where the motor rotation speed is low, the operation may be performed with a smaller number of synchronization pulses than in a condition where the motor rotation speed is high.

As described above, the motor control device according to the present embodiment performs control to switch from asynchronous PWM control to synchronous PWM control in accordance with an increase in the motor rotation speed. In that case, the carrier frequency used for asynchronous PWM control and synchronous PWM control is set lower than the upper limit frequency which is the upper limit value of the carrier frequency. Thereby, even if the motor rotation speed increases, generated noise and leakage current are suppressed.

In addition, the motor control device according to the present embodiment performs control so that the number of synchronous pulses decreases as the number of motor rotations increases when performing synchronous PWM control. As a result, a short time and smooth acceleration from the time of startup to the high-speed operation range is realized.

So far, the effect of applying the synchronous PWM control to the single-phase inverter that drives the single-phase motor has been described, but other effects will be described.

In a high-speed rotation application such as the electric vacuum cleaner 61 or the hand dryer 70, since a large current flows through the motor at the time of startup, the reliability of the battery and the elements used decreases as the number of startups increases. Therefore, it is desired to reduce the number of activations by continuing to rotate at a low speed without stopping the motor. By applying the motor control system 1 described in the first embodiment to such a high-speed rotation application, the detection accuracy of an analog signal that is a motor current or a motor voltage is improved. This can be realized and the reliability can be improved.

Furthermore, since the arrival time from the low speed startup to the steady rotation speed is significantly shortened, it is possible to reduce power consumption by frequently reducing the speed to low speed rotation. At this time, according to the motor control system 1 described in the first embodiment, the detection accuracy of the analog signal that is the motor current or the motor voltage is improved, so that vibration of the rotation speed of the motor can be suppressed and useless power consumption is achieved. Can be reduced.

Generally, in the vacuum cleaner 61, the operation unit 66 is turned on to reach the steady rotational speed from the start, but by providing a mode for operating at a low speed in advance, the operation unit 66 is turned on. It is possible to significantly reduce the time until actual use. At this time, by using the motor control system 1 described in the first embodiment, the detection accuracy of the analog signal that is the motor current or the motor voltage is improved, so that the vibration of the rotation speed of the motor can be suppressed, and the wasteful consumption. Electric power can be reduced.

For example, assuming that the time for rotating from the start of power supply to 2000 rpm is 1 s, and the time for rotating from 2000 rpm to 60000 rpm, which is the steady rotational speed, is 0.3 s, 1.3 s from the start to the steady rotational speed is reached. Necessary. Therefore, by performing a time-consuming start-up in advance, it is possible to realize the actual use from a switch-on to a steady rotational speed in only 0.3 s. At this time, by using the motor control system 1 described in the first embodiment, the detection accuracy of the analog signal that is the motor current or the motor voltage is improved, so that a shorter start-up time can be realized.

In addition, at the time of start-up in which a steep rise of the input current occurs, the reliability of the battery can be improved by setting a low acceleration rate and suppressing the steep rise. At this time, by using the motor control system 1 described in the first embodiment, the detection accuracy of an analog signal that is a motor current or a motor voltage is improved. Current interruption can be realized.

In addition, by reducing the acceleration rate until low speed operation, the current that flows to the motor at the time of startup also decreases, so suppressing the heat generation of the semiconductor element can suppress the heat generation of the component, leading to improved component reliability . At this time, by using the motor control system 1 described in the first embodiment, the detection accuracy of the analog signal that is the motor current or the motor voltage is improved, so that vibration at a low acceleration rate can be suppressed. Become.

In order to remove the heat generated from the semiconductor element, usually, a heat-dissipating fin having a good thermal conductivity is attached to the element surface, or a method of dispersing the heat to the mounting substrate using a surface-mounted element is used. In addition, there is a method of cooling the semiconductor element by providing a heat radiating fan, or cooling by water cooling, but these methods are suitable for small devices due to the cost of cooling and the increase in installation volume. Absent. However, if the electric device includes the electric blower described in the present embodiment, it is possible to heat the current configuration without providing additional parts by arranging these heating elements in the path of the wind generated by the electric blower. It is possible to escape.

Further, since no additional parts are required, it is possible to suppress an increase in cost, and it is possible to achieve further miniaturization because a space for providing additional parts is not necessary. Furthermore, it is possible to extend the operation time by allocating a space that can be reduced in size to the battery. At this time, by using the motor control system 1 described in the first embodiment, the detection accuracy of the analog signal that is the motor current or the motor voltage is improved, and therefore, unnecessary power that is unnecessary for the operation may be consumed. Since it decreases, it becomes possible to extend operating time more.

In addition, although the vacuum cleaner 61 and the hand dryer 70 were demonstrated in this Embodiment, the motor control system 1 of Embodiment 1 can be applied to the electric equipment with which a motor is mounted generally. Electric equipment equipped with motors include, for example, incinerators, crushers, dryers, dust collectors, printing machines, cleaning machines, confectionery machines, tea making machines, woodworking machines, plastic extruders, cardboard machines, packaging machines, hot air generators For object transportation, dust absorption, general air supply / exhaust, or OA equipment.

The configuration described in the above embodiment shows an example of the contents of the present invention, and can be combined with another known technique, and can be combined with other configurations without departing from the gist of the present invention. It is also possible to omit or change the part.

1 motor control system, 2 motor control device, 10 power supply, 11 single phase inverter, 11a1 to 11a4 switching element, 11b1 to 11b4 diode, 11P, 11N DC bus, 12 single phase motor, 20 current sensor, 21 position sensor, 23 voltage Sensor, 25 control unit, 30, 35 AD converter, 31 processor, 32 drive signal generation unit, 33 carrier generation unit, 33a carrier frequency setting unit, 34 memory, 38 carrier comparison unit, 38a absolute value calculation unit, 38b, 38d Division unit, 38c, 38d, 38f multiplication unit, 38e addition unit, 38g, 38h comparison unit, 38i, 38j output inversion unit, 40 PWM control unit, 51 control circuit, 52 comparator, 53 DA converter, 61 vacuum cleaner 62 extension pipe, 63 Port, 64 electric blower, 65 dust collection chamber, 66 operation section, 67 battery, 68, 77 sensor, 70 hand dryer, 71 casing, 72 hand detection sensor, 73 water receiving section, 74 drain container, 76 cover, 78 Air intake.

Claims

A motor control device comprising a control unit that PWM-controls the single-phase inverter based on a detected value of a motor current flowing in a single-phase motor driven by a single-phase inverter,
Synchronous PWM control in which the carrier is synchronized with the phase of the voltage command and asynchronous PWM control in which the carrier is not synchronized with the phase of the voltage command are executed,
The asynchronous PWM control is switched from the asynchronous PWM control to the synchronous PWM control in accordance with an increase in the motor speed, and the carrier frequency used for the asynchronous PWM control and the synchronous PWM control is set lower than the upper limit frequency that is the upper limit value of the carrier frequency. Motor control device.
The motor control device according to claim 1, wherein when performing the synchronous PWM control, the number of synchronous pulses changes in accordance with an increase in the motor rotational speed.
The motor control device according to claim 2, wherein the number of synchronization pulses is an even number.
The control unit includes an AD converter that converts analog data that is a detected value of the motor current into digital data;
The controller activates the AD converter in a continuous period not including the polarity change point in the electric half cycle determined by the adjacent polarity change points of the motor current, and is the same as the electric half cycle. The motor control device according to any one of claims 1 to 3, wherein the digital data is read during an electrical half cycle.
The control unit does not activate the AD converter within a prohibition period that is a fixed period including the polarity change point, and does not activate the AD converter within a permission period that is a period other than the prohibition period. The motor control device according to claim 4, wherein the digital data converted from the activation is read from the AD converter.
The control unit calculates a rotor rotation angle that is an electrical angle of the rotor based on a position sensor signal that detects a rotation position of the rotor, and determines the prohibition period and the permission period based on the rotor rotation angle. The motor control device according to claim 5, wherein the motor control device is determined.
The said control part estimates the said motor current in the said prohibition period using the said several digital data read from the said AD converter in the said permission period immediately before the said prohibition period. Motor control device.
The control unit includes a carrier generation unit that generates a carrier,
The timing at which the control unit activates the AD converter is the timing of the peak or valley point of the carrier,
8. The motor control device according to claim 5, wherein the timing at which the control unit reads the digital data from the AD converter is a timing of a peak point or a valley point of the carrier. 9.
9. The control unit according to claim 1, wherein the controller reads the digital data from the AD converter a plurality of times within a continuous period that does not include a change point of the polarity of the motor current. Motor control device.
The motor control device according to any one of claims 1 to 9, wherein the rotor has a plurality of permanent magnets.
The single-phase inverter has a plurality of switching elements,
11. The motor control device according to claim 1, wherein each of the plurality of switching elements is formed using a wide band gap semiconductor.
The motor control device according to claim 11, wherein the wide band gap semiconductor is silicon carbide, gallium nitride, or diamond.
A vacuum cleaner comprising the motor control device according to any one of claims 1 to 12, the single-phase inverter, and the single-phase motor.
A hand dryer comprising the motor control device according to any one of claims 1 to 12, the single-phase inverter, and the single-phase motor.