WO2011120299A1 - 电压变换装置、方法及供电系统 - Google Patents

电压变换装置、方法及供电系统 Download PDF

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Publication number
WO2011120299A1
WO2011120299A1 PCT/CN2010/078125 CN2010078125W WO2011120299A1 WO 2011120299 A1 WO2011120299 A1 WO 2011120299A1 CN 2010078125 W CN2010078125 W CN 2010078125W WO 2011120299 A1 WO2011120299 A1 WO 2011120299A1
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WIPO (PCT)
Prior art keywords
voltage
stage
output
isolated
transformation
Prior art date
Application number
PCT/CN2010/078125
Other languages
English (en)
French (fr)
Inventor
樊晓东
景遐明
刘志华
刘旭君
Original Assignee
华为技术有限公司
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
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Publication date
Application filed by 华为技术有限公司 filed Critical 华为技术有限公司
Priority to EP10790865.9A priority Critical patent/EP2393194A4/en
Priority to US12/980,047 priority patent/US20110241424A1/en
Publication of WO2011120299A1 publication Critical patent/WO2011120299A1/zh
Priority to US13/272,674 priority patent/US20120032509A1/en

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33561—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having more than one ouput with independent control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H02M1/007—Plural converter units in cascade
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33592—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present invention relates to the field of power supply, and in particular, to a voltage conversion device, method, and power supply system.
  • the multi-output voltage regulation technology mainly has power consumption benefits in the unequal power configuration scenario of the dual-emitting RF module, and improves the power supply efficiency.
  • MSR Multi-Service Routers
  • RAN Radio Access Network
  • wireless access network the emergence of successful cases of sharing, equivalent to the technical requirements of dual or multi-channel power supply voltage regulation, and the existing single output power supply can not be satisfied, using a single power supply simultaneously
  • an intermediate frequency power supply reliability problem Once the power supply has a problem, the entire module cannot work normally.
  • the intermediate frequency power supply can be combined by means of a combined circuit, and the abnormality of the RF module can be maintained.
  • FIG. 1 A multi-output circuit in the prior art is shown in FIG. 1.
  • the dashed box 12 is a fixed-duty half-bridge circuit
  • the dashed box 14 and the dashed box 16 are two BUCKs (step-down conversion).
  • the circuit uses synchronous trailing edge modulation.
  • the synchronization operation is required for the front-end stage operation, such as the synchronization signal synchronized with the control circuit 32 of the previous stage by the synchronization circuit 50 of FIG. 1, to control the control circuits 40 and 42 of the subsequent stage and the control circuit of the preceding stage.
  • 32 for synchronization the square wave voltage that needs to be synchronized.
  • Such a multi-output circuit in the prior art needs to pass a synchronization signal to operate normally during voltage conversion, and is susceptible to interference.
  • Embodiments of the present invention provide a voltage conversion device, method, and power supply system to reduce interference experienced in performing voltage conversion.
  • the embodiment of the invention provides a voltage conversion device, including:
  • a first transformer module for performing pre-stage voltage transformation on the input DC voltage, outputting an isolated DC voltage, wherein the pre-stage transformation includes converting a input DC voltage into a primary side transformation of the AC voltage to be changed, the original
  • the working cycle of the edge transformation includes an upper half cycle and a lower half cycle, and there is a dead zone time between the upper half cycle and the lower half cycle;
  • a capacitor filtering module configured to perform capacitance filtering compensation on the isolated DC voltage during the dead time, and output a stable intermediate DC voltage
  • the second transformer module is configured to perform at least two independent post-stage voltage transformations on the intermediate DC voltage, and output DC voltages required for at least two loads.
  • the embodiment of the invention provides a voltage conversion method, including:
  • the pre-stage transformation includes converting a input DC voltage into a primary side transformation of the AC voltage to be changed, and the working period of the primary side transformation includes a first half There is a dead time between the first half cycle and the second half cycle, and the isolated DC voltage is filtered and compensated at the dead time to output a stable intermediate DC voltage;
  • An embodiment of the present invention provides a power supply system including at least two loads, and further includes a voltage conversion device that supplies power to the at least two loads;
  • the voltage conversion device is configured to perform pre-stage voltage transformation on the input DC voltage, and output an isolated DC voltage, wherein the pre-stage transformation includes converting a input DC voltage into a primary side transformation of the AC voltage to be changed, the primary side
  • the working period of the transformation includes an upper half period and a lower half period, and there is a dead time between the upper half period and the second half period; the capacitive DC filter is compensated for the isolated DC voltage at the dead time, and the output is stable in the middle.
  • a DC voltage performing at least two independent post-stage voltage transformations on the intermediate DC voltage, and outputting a DC voltage required for the at least two loads.
  • the isolated DC voltage outputted after the pre-stage transformation is directly subjected to capacitance filtering compensation by the filter capacitor, and the stability can be obtained.
  • the intermediate DC voltage is used as the input voltage of the latter stage, so that the latter stage can be used. Synchronization with the previous stage to achieve multi-output, to achieve decoupling of the front-end mode of operation, reducing the degree of interference during voltage conversion.
  • Figure 1 is a structural diagram of a multi-output circuit of the prior art
  • FIG. 2 is a flow chart of a voltage conversion method according to an embodiment of the present invention.
  • FIG. 3 is a flow chart of a voltage conversion method according to an embodiment of the present invention.
  • FIG. 4 is a structural diagram of a voltage conversion device according to an embodiment of the present invention.
  • FIG. 5 is a structural diagram of a voltage conversion module according to an embodiment of the present invention.
  • Figure 6 is a structural diagram of a voltage conversion device according to an embodiment of the present invention.
  • Figure 7 is a structural diagram of a voltage conversion device according to an embodiment of the present invention.
  • FIG. 8 is a driving timing diagram of an embodiment of the present invention.
  • FIG. 9 is a driving timing diagram of an embodiment of the present invention.
  • FIG. 10 is a structural diagram of a power supply system according to an embodiment of the present invention.
  • an embodiment of the present invention provides a voltage conversion method, including:
  • the pre-stage transformation includes converting the input DC voltage into a primary side transformation of the AC voltage to be changed, and the primary side transformation
  • the duty cycle includes a first half cycle and a second half cycle, and there is a dead time between the first half cycle and the second half cycle.
  • the mutual switching between the upper half cycle and the off duty cycle may be: in one working cycle, the upper half cycle is switched to the lower half cycle; in another embodiment, it may be, the second half cycle of one duty cycle.
  • the switching to the first half of another working cycle is not specifically limited in the embodiment of the present invention.
  • the isolated DC voltage is directly subjected to capacitance filtering without an inductor, and the isolated DC voltage is subjected to capacitance filtering compensation to output a stable intermediate DC voltage.
  • the filter capacitor is used for filter compensation, which can filter the output voltage of the primary side to jump and output a stable intermediate DC voltage.
  • the filter capacitor is charged during the working time of the primary side conversion (non-dead time), and the energy obtained by the release of the charge is capacitively filtered to compensate the isolated DC voltage during the dead time of the primary side change, and the primary side can be filtered out. Transforms the voltage ripple generated during the dead time (this voltage ripple can cause the output voltage of the first half cycle to switch to the second half of the cycle to jump), and outputs a stable intermediate DC voltage.
  • the isolated DC voltage outputted after the pre-stage transformation is directly filtered by the filter capacitor, and the filter capacitor can be used.
  • the obtained stable intermediate DC voltage enables the latter stage to realize multi-output without synchronizing with the pre-stage, thereby realizing decoupling of the front-end stage operation mode, and reducing the degree of interference during voltage conversion.
  • an embodiment of the present invention provides a voltage conversion method, including:
  • the working period of the primary conversion includes an upper half period and a lower half period, and there is a dead time between the upper half period and the lower half period.
  • the mutual switching between the upper half cycle and the off duty cycle may be: in one working cycle, the upper half cycle is switched to the lower half cycle; in another embodiment, it may be, one duty cycle.
  • the second half of the cycle is switched to the first half of the other cycle, which is not specifically limited in the embodiment of the present invention.
  • the dead time causes the output voltage of the first half cycle and the output voltage of the second half cycle to be relatively hopping.
  • the preamplifier PWM Pulse Width
  • PWM Pulse Width
  • the signal drives a full-bridge circuit connected by a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) to perform primary-side conversion of the input DC voltage.
  • MOSFET Metal Oxide Semiconductor Field Effect Transistor
  • the duty ratio of the preceding PWM control signal can be controlled to be close to 50% by voltage feedforward frequency conversion, that is, the absolute value of the duty ratio of the pre-PWM signal and the 50% difference is not zero. And within the preset range. In practical applications, the duty ratio of the preceding PWM signal can be controlled to be 48%, 49% or 51%.
  • the input DC voltage can be primary transformed by driving a half-bridge circuit or a push-pull circuit with a fixed-duty pre-stage PWM signal.
  • the alternating voltage can be isolated and transformed by an isolation transformer to obtain a square wave voltage, which is an isolated alternating voltage.
  • the square wave voltage is synchronously rectified, the harmonic component is filtered out, and the isolated DC voltage is output; in one embodiment, the square wave voltage is synchronously rectified to filter out the harmonic component and output the isolated DC voltage;
  • the filter capacitor is used to filter and isolate the isolated DC voltage during the dead time of the primary full-bridge circuit to provide a certain amount of energy to maintain the DC voltage output and output a stable intermediate DC voltage.
  • the duty cycle of the pre-PWM control signal can be controlled by voltage feedforward frequency conversion to be close to 50%, that is, the duty ratio of the pre-stage PWM signal is absolutely different from 50%.
  • the value is not zero and is within the preset range.
  • the duty ratio of the above-mentioned pre-stage PWM signal can be controlled to be 48%, 49% or 51%.
  • the duty cycle is close to 50%, which can make the dead time smaller.
  • the isolated DC voltage of the primary-side rectified output of the pre-stage requires only a small-capacity filter capacitor for filtering, and no additional filter inductor is required.
  • the filter capacitor has a capacity below 1 Ouf.
  • the PWM voltage of the post-stage transformer can be dynamically adjusted according to the DC voltage output after the post-stage voltage transformation (for example, the duty ratio or phase or frequency of the PWM signal of the latter stage is dynamically adjusted according to the output voltage. ), to achieve interleaving control of two independent post-stage transformers.
  • the isolated DC voltage outputted after the pre-stage transformation is directly subjected to capacitance filtering compensation through the filter capacitor, and the filter capacitor is passed through the filter capacitor.
  • the stable intermediate DC voltage can be obtained, so that the latter stage can realize multi-output without synchronizing with the previous stage, and realize the decoupling of the front-end working mode, thereby reducing the degree of interference during voltage conversion.
  • the pre-stage PWM signal of the fixed duty ratio is controlled to operate at a fixed duty ratio of approximately 50%, and the dead time of the front stage is short, and the isolated DC voltage of the rectified output of the primary side of the front stage is only Filtering capacitors with small capacitance are required for filtering, eliminating the need for additional filter inductors and reducing circuit space.
  • an embodiment of the present invention provides a voltage conversion apparatus, including:
  • the first transformer module 310 is configured to perform pre-stage voltage transformation on the input DC voltage, and output an isolated DC voltage; the pre-stage transformer includes converting the input DC voltage into a primary side transformation of the AC voltage to be changed, and the primary side transformation
  • the duty cycle includes a first half cycle and a second half cycle, and there is a dead time between the first half cycle and the second half cycle.
  • the mutual switching between the upper half cycle and the off duty cycle may be: in one working cycle, the upper half cycle is switched to the lower half cycle; in another embodiment, it may be, the second half cycle of one duty cycle.
  • the switching to the first half of another working cycle is not specifically limited in the embodiment of the present invention.
  • the capacitor filtering module 320 is configured to perform capacitance filtering compensation on the isolated DC voltage in the dead time, and output an intermediate DC voltage;
  • the capacitor filter module 320 is used to filter and compensate the isolated DC voltage, which can provide a certain amount of energy, thereby maintaining the DC voltage output.
  • a stable intermediate DC voltage is output.
  • the second transformer module 330 is configured to perform at least two independent post-stage voltage transformations on the intermediate DC voltage, and output DC voltages required for at least two loads.
  • the second transformer module 330 can include at least two voltage conversion circuits.
  • any one of the at least two voltage conversion circuits is usually a buck conversion BUCK circuit.
  • the boost converter BOOST circuit can also be used.
  • the BUCK circuit has a small ripple and is relatively easy to control, it is widely used.
  • the BUCK circuit is taken as an example for description.
  • the isolated DC voltage outputted after the pre-stage transformation is directly subjected to capacitance filtering compensation through the filter capacitor, and the filter capacitor is passed through the filter capacitor.
  • the stable intermediate DC voltage can be obtained, so that the latter stage can realize multi-output without synchronizing with the previous stage, and realize the decoupling of the front-end working mode, thereby reducing the degree of interference during voltage conversion.
  • the first voltage conversion module 310 can include:
  • the primary side transform unit 311 is configured to perform primary side transform on the input DC voltage, and output an AC voltage to be changed.
  • the working period of the primary side transform unit includes an upper half period and a lower half period, and the upper half period and the lower half period are switched to each other. There is dead time between.
  • the primary side transform unit 311 can be a full bridge circuit; in one embodiment, the full-bridge circuit can be controlled by a fixed-cycle pre-stage PWM signal to perform pre-primary rectification of the input DC voltage. .
  • the duty cycle of the preceding PWM control signal can be controlled to be close to 50% by voltage feedforward frequency conversion. In practical applications, the duty cycle of the above PWM control signal can be controlled to be 48%, 49% or 51%.
  • the dead time can be made small (that is, the duty ratio of the first voltage conversion module is controlled to be close to 50%).
  • the capacitance filtering module 320 is composed of a small-capacity filter capacitor. For example, in one embodiment, the capacity of the filter capacitor is 10 uf or less.
  • the primary side transform unit 311 can also be a half bridge circuit; in one embodiment, the primary side transform unit 311 can also be a push-pull circuit.
  • the input DC voltage can be pre-primed by a fixed-duty pre-stage PWM signal to control the half-bridge or push-pull circuit.
  • the pre-PWM control signal can be controlled by voltage feedforward frequency conversion.
  • the duty cycle is close to 50%.
  • the duty cycle of the above PWM control signal can be controlled to be 48%, 49% or 51%.
  • the capacitance filtering module 320 is composed of a small-capacity filter capacitor.
  • the filter capacitor has a capacity of less than 10 uf.
  • the upper half period and the lower half period of the primary side transform unit 311 refer to the upper half period and the lower half period, the upper half period and the lower half period of the duty cycle of the full bridge circuit, the half bridge circuit, and the push pull circuit. There is a dead time between switching.
  • the transformer unit 312 is configured to perform isolation and voltage transformation on the alternating AC voltage, and output a square wave voltage, wherein the square wave voltage is an isolated alternating voltage;
  • the specific form of the transformer unit 312 can be an isolation transformer.
  • the rectifying unit 313 is used for synchronous rectification of the square wave voltage, filtering out harmonic components, and outputting an isolated DC voltage.
  • the counter wave voltage (ie, the isolated AC voltage) is synchronously rectified to filter out harmonic components and output an isolated DC voltage;
  • the isolated DC voltage outputted after the pre-stage transformation is directly subjected to capacitance filtering compensation through the filter capacitor, and the filter capacitor is passed through the filter capacitor.
  • the stable intermediate DC voltage can be obtained, so that the latter stage can realize multi-output without synchronizing with the previous stage, and realize the decoupling of the front-end working mode, thereby reducing the degree of interference during voltage conversion.
  • the pre-stage PWM signal of the fixed duty ratio is controlled to operate at a fixed duty ratio of approximately 50%, and the dead time of the front stage is short, and the isolated DC voltage of the rectified output of the primary side of the front stage is only Filtering capacitors with small capacitance are required for filtering, eliminating the need for additional filter inductors and reducing circuit space.
  • an embodiment of the present invention provides a voltage conversion apparatus, including:
  • Transformer Tl full bridge circuit Ql, rectifier circuit PI, filter capacitor Cl, and two independent BUCK transformer circuits J1 and J2. Specifically, according to Figure 6:
  • the full bridge circuit Q1 on the input side (ie, the primary side) of the transformer T1 is connected by four MOSFETs: Qlp Q2p and Q3p Q4.
  • Q2p and Q3p are a pair of bridge arms that are simultaneously turned on
  • Qlp and Q4p are another pair of bridge arms that are commonly turned on.
  • the full bridge circuit Q1 performs primary conversion on the input DC voltage to output an AC voltage to be changed; in one embodiment, the primary side of the full bridge circuit Q1 enters the input DC voltage.
  • the row switch is changed to change the input DC voltage to an AC voltage, that is, the AC voltage to be changed.
  • the working cycle of the full bridge circuit Q1 includes an upper half cycle and a lower half cycle.
  • the transformer T1 isolates and transforms the to-be-changed alternating voltage outputted by the full-bridge circuit Q1, and outputs a square wave voltage, which is an isolated alternating voltage; in one embodiment, the transformer T1 is an isolated transformer.
  • the rectifier circuit P1 synchronously rectifies the square wave voltage outputted by the transformer T1, filters out high frequency ripple (ie, harmonic component), and outputs an isolated DC voltage.
  • the rectifier circuit P1 is composed of two full bridge synchronous rectifiers: The first synchronous rectifier Q1SR and the second synchronous rectifier Q2SR are connected.
  • the first synchronous rectifier Q1 SR and the second synchronous rectifier Q2SR may be MOSFETs.
  • the gate of Q1 SR is connected to the drain of Q2SR to form a self-driving rectification circuit, and a synchronous rectification drive signal is generated by the transformer T1's own winding.
  • the self-driving timing in this embodiment is synchronized with the output signal of transformer T1.
  • Q2SR also similar to the self-driving connection, the gate of Q1 SR is connected to the drain of Q2SR.
  • Q1 SR and Q2SR can also adopt the driving mode, and the driving timing of the driving mode is the same as the self-driving timing.
  • the rectifier circuit P1 synchronously rectifies the opposite-wave voltage (that is, the isolated AC voltage), and outputs the isolated DC voltage.
  • the rectifying circuit P1 is synchronously rectified by a counter wave voltage (ie, an isolated AC voltage) for filtering out harmonic components and outputting an isolated DC voltage.
  • the isolated DC voltage is directly filtered by the filter capacitor C1 to output a stable intermediate DC voltage.
  • the square wave voltage outputted by the transformer T1 is synchronously rectified and directly connected to the filter capacitor C1 for filtering, and the filter capacitor C1 filters and compensates the isolated DC voltage to maintain the DC output and output. Intermediate DC voltage.
  • the MOS transistor of the full-bridge circuit In the dead time of the primary full-bridge circuit, the MOS transistor of the full-bridge circuit is not turned on, which will cause the output of the full-bridge circuit to jump, that is, the output voltage of the upper half cycle and the output voltage of the second half cycle. There is a jump, so that the rear BUCK transformer circuits J1 and J2 do not have enough stable input voltage.
  • the filter capacitor C1 isolates the DC voltage for filtering compensation, which can provide a certain amount of energy.
  • the filter capacitor C1 is charged by the output voltage of the rectifier circuit P1 during the non-dead time (eg, the upper half cycle or the second half cycle), at the dead time of the full-bridge circuit Q1, at which time the rectifier circuit P1
  • the output voltage isolated DC voltage
  • the filter capacitor will release the energy absorbed by the previous charge, and the isolated DC voltage will be filtered and compensated.
  • the output is stable. Intermediate DC voltage.
  • the intermediate DC voltage passes through two independent BUCK conversion circuits, such as the first BUCK conversion circuit.
  • MOSFET Q1B1 and MOSFET Q2B1 form a BUCK converter circuit, and inductor L1 and capacitor C2 are used to filter the output of the BUCK converter circuit.
  • MOSFET Q1B2 and MOSFET Q2B2 form another BUCK converter circuit, and inductor L2 and capacitor C3 are used to filter the output of the BUCK converter circuit.
  • the voltage conversion device further includes two PWM controllers, a first P WM controller 61 and a second P WM controller 62.
  • each BUCK converter circuit can independently regulate the output voltage and other protection functions.
  • MOSFET Q1B1 and MOSFET Q2B1 are connected to the OUT H and OUT 1 pins of the first PWM controller 61, respectively.
  • MOSFET Q1B2 and MOSFET Q2B2 are connected to the OUT H and OUT 1 pins of the second PWM controller 62, respectively.
  • each PWM controller is also connected to the output of each BUCK converter circuit, for example, the F/B terminal of the first PWM controller 61 and the output of the first BUCK converter circuit J1.
  • the voltage dividing resistor R1 is connected, and the F/B terminal and the second of the second PWM controller 62
  • the voltage dividing resistor R2 at the output of the BUCK converter circuit J2 is connected.
  • the PWM controller can dynamically adjust and control the PWM signal of each BUCK converter circuit according to the output voltage of each BUCK converter circuit (for example, dynamically adjusting the duty cycle or phase or frequency of the PWM signal according to the output voltage), thereby realizing Interleaving control of the BUCK converter circuit.
  • the PWM signal that controls the BUCK conversion circuit in one embodiment may employ trailing edge modulation; in one embodiment, the PWM signal that controls the BUCK conversion circuit may also employ leading edge modulation.
  • the PWM controller can also generate a pre-stage PWM signal with a fixed duty cycle (for example, a pre-stage PWM signal that provides a fixed duty cycle with a duty ratio of approximately 50%), and then the primary side can be driven by the isolation unit 60 and the drive unit 70.
  • the pre-stage PWM signal of the fixed duty cycle may also be generated by a separate pulse signal generating unit, or may be generated by the pre-stage PWM controller as in the above embodiment. .
  • the function of the isolation unit 60 is to transmit the generated pre-stage PWM signal of the prescribed duty ratio to the primary side.
  • the function of the driving unit 70 is to amplify the pre-stage PWM signal of the fixed duty ratio transmitted from the isolation unit 60.
  • the voltage conversion device may not include the isolation unit 60.
  • the filter capacitor C1 has a capacity of less than 10 uf.
  • the PWM controller can calculate the output voltage of the filter capacitor C1 by connecting the input voltage VIN terminal to the upper end of the filter capacitor C1, and then calculate the output voltage of the filter capacitor C1.
  • the primary side voltage can dynamically adjust the frequency of the pre-stage PWM signal, thereby adjusting the operating frequency of the primary side full bridge, controlling the alternating magnetic flux of the transformer T1, further reducing the core loss of the preceding stage, and expanding the pre-stage fixed occupation.
  • the input voltage operating range of the air ratio resonance topology can be used to calculate the output voltage of the filter capacitor C1 by connecting the input voltage VIN terminal to the upper end of the filter capacitor C1, and then calculate the output voltage of the filter capacitor C1.
  • the primary side voltage can dynamically adjust the frequency of the pre-stage PWM signal, thereby adjusting the operating frequency of the primary side full bridge, controlling the alternating magnetic flux of the transformer T1, further reducing the core loss of the preceding stage, and expanding the pre-stage fixed occupation.
  • the output voltage of the filter capacitor C1 can be detected according to the auxiliary power source or other manners, and the primary side voltage of the transformer T1 can be calculated according to the output voltage.
  • the primary voltage of the transformer T1 can dynamically adjust the frequency of the pre-PWM signal to adjust the operating frequency of the primary bridge.
  • the duty ratio of the pre-PWM signal of the fixed duty ratio may be close to 50%, that is, the absolute value of the duty ratio of the pre-PWM signal and the difference of 50% is not zero, and Within the scope of the setting.
  • the duty cycle of the PWM signal can be 49% fixed duty cycle, 48% fixed duty cycle or 53% fixed duty cycle.
  • the two PWM controllers described above may also be integrated into one chip.
  • the two outputs in FIG. 6 can be connected in parallel in parallel operation mode, in which case the two respective filter inductors L1 and L2 can be coupled inductors, when connected in parallel.
  • the two output voltages are the same.
  • this embodiment provides a driving timing diagram of Qlp ⁇ Q4p.
  • Qlp, Q2p, Q3, and Q4p in the primary full-bridge rectifier circuit operate at a fixed duty cycle close to 50% (the duty cycle is 49% in Figure 8), where Qlp and Q4p are the same drive timing.
  • Q2p and Q3p are the same driving timing, and Qlp and Q3p are complementary symmetrical control.
  • this embodiment provides a driving timing chart of Q1B1 and Q1B2.
  • the upper tubes Q1B1 and Q1B2 of the two-way BUCK converter circuit drive the phase shift 180 degrees, so that the two BUCKs can work alternately, reducing the input reflected current ripple at one time.
  • the load is three
  • a corresponding three-way BUCK converter circuit is required to output different DC voltages required for the three-way load.
  • the upper tube of the three-way BUCK conversion circuit can be controlled by the PWM signal to be phase-shifted by 120 degrees, so that the three-way BUCK conversion circuit can be alternately operated.
  • FIG. 6 and FIG. 7 shows the case when the load is two paths.
  • the load is multi-way (such as 3-way or 4-way)
  • the technical solution and two-way time are adopted. Similar, there is no substantial change, and will not be repeated here.
  • the primary side full-bridge circuit works at a fixed duty ratio close to 50% by the PWM controller, and the dead time is short.
  • the output only needs a small capacity.
  • the transformer can also obtain the primary side voltage VIN of the transformer T1 through CI or auxiliary power supply or other means, and dynamically adjust the operating frequency of the front-stage primary full-bridge according to the obtained input voltage VIN, and control the alternating magnetic flux of the transformer T1.
  • the core loss of the previous stage is further reduced, and the input voltage operating range of the pre-stage fixed duty cycle resonance topology is expanded.
  • an embodiment of the present invention provides a power supply system, including: a voltage conversion device 10 and at least two loads.
  • a voltage conversion device 10 shows that when the load is two paths (load 20 and load 30)
  • the structure diagram, when the load is multiplexed, does not affect the essence of the present invention.
  • the voltage conversion device 10 is configured to perform pre-stage voltage transformation on the input DC voltage, and output an isolated DC voltage voltage; the pre-stage transformation includes converting a input DC voltage into a primary side transformation of the AC voltage to be changed, and the primary side transformation
  • the duty cycle includes a first half cycle and a second half cycle, and there is a dead time between the first half cycle and the second half cycle; the isolated DC voltage of the output is subjected to capacitive filter compensation at the dead time, and the stable intermediate DC voltage is output.
  • the technical solution adopted is similar to that of the two-way, and there is no substantial change, and will not be described here.
  • the structure and function of the voltage conversion device 10 can be as described in any of the above embodiments, and will not be described again.
  • the isolated DC voltage outputted after the pre-stage transformation is directly filtered by the filter capacitor, and the filter capacitor can be used.
  • the obtained stable intermediate DC voltage enables the latter stage to realize multi-output without synchronizing with the pre-stage, thereby realizing decoupling of the front-end stage operation mode, and reducing the degree of interference during voltage conversion.

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Description

电压变换装置、 方法及供电系统 本申请要求于 2010年 4月 1日提交中国专利局， 申请号为 201010142326.X, 发明名称为"电压变换装置、 方法及供电系统"的中国专利申请的优先权， 其全 部内容通过引用结合在本申请中。 技术领域
本发明涉及供电领域， 特别涉及一种电压变换装置、 方法及供电系统。
背景技术
多路输出调压技术主要是在双发射射频模块不等功率配置场景下有功耗 收益， 提高供电效率， 随着 MSR ( Multiple Services Routers， 多业务开放路由 器） 多模协议和 RAN ( Radio Access Network, 无线接入网络）共享的成功案 例的出现， 等效的存在双路或多路电源调压此电路的技术需求， 而现有的单路 输出电源是无法满足的， 采用单路电源供电同时还存在中频供电可靠性问题， 一旦电源出现问题， 整个模块都不能正常工作， 而采用多路输出架构， 可以通 过合路给中频供电， 一路异常还可以维持射频模块继续工作。
现有技术中的一种多路输出电路如图 1所示， 在该电路中虚线框 12为固 定占空比的半桥电路，虚线框 14和虚线框 16为两个 BUCK (降压变换）电路， 采用同步后沿调制。 在该方案中前后级工作需要同步信号控制， 如通过图 1 中的同步电路 50产生与前级的控制电路 32同步的同步信号，来控制后级的控 制电路 40和 42与前级的控制电路 32进行同步， 需要同步的方波电压。
现有技术中的这种多路输出电路，在电压变换时需要通过同步信号才能正 常工作， 易受干扰。
发明内容
本发明实施例提供了一种电压变换装置、方法及供电系统， 以减少在进行 电压变换时受的干扰。 本发明实施例提供了一种电压变换装置， 包括：
第一变压模块，用于对输入的直流电压进行前级变压，输出隔离直流电压， 所述前级变压包括将输入的直流电压转换为待变交流电压的原边变换，所述原 边变换的工作周期包括上半周期和下半周期，上半周期与下半周期相互切换之 间存在死区时间；
电容滤波模块，用于在所述死区时间对所述隔离直流电压进行电容滤波补 偿， 输出稳定的中间直流电压；
第二变压模块， 用于对所述中间直流电压进行至少两路独立的后级变压， 输出至少两路负载所需的直流电压。
本发明实施例提供了一种电压变换方法， 包括：
对输入的直流电压进行前级变压，输出隔离直流电压， 所述前级变压包括 将输入的直流电压转换为待变交流电压的原边变换，所述原边变换的工作周期 包括上半周期与下半周期， 上半周期与下半周期相互切换之间存在死区时间； 在所述死区时间对所述隔离直流电压进行滤波补偿，输出稳定的中间直流 电压；
对所述中间直流电压进行至少两路独立的后级变压，输出至少两路负载所 需的直流电压。
本发明实施例提供了一种供电系统， 包括至少两路负载，还包括为所述至 少两路负载供电的电压变换装置；
所述电压变换装置用于对输入的直流电压进行前级变压，输出隔离直流电 压， 所述前级变压包括将输入的直流电压转换为待变交流电压的原边变换， 所 述原边变换的工作周期包括上半周期和下半周期，上半周期与下半周期相互切 换之间存在死区时间； 在所述死区时间对所述隔离直流电压进行电容滤波补 偿，输出稳定的中间直流电压； 对所述中间直流电压进行至少两路独立的后级 变压， 输出所述至少两路负载所需的直流电压。
本发明实施例通过以上技术方案，在前级变压过程中的原边变换的死区时 间，对经过前级变压后输出的隔离直流电压直接通过滤波电容进行电容滤波补 偿， 可以得到的稳定的中间直流电压作为后级的输入电压， 这样后级可以不用 和前级进行同步而实现多路输出， 实现前后级工作方式解耦， 减少了电压变换 时受干扰的程度。 附图说明
为了更清楚地说明本发明实施例中的技术方案，下面将对实施例描述中所 需要使用的附图作简单地介绍， 显而易见地， 下面描述中的附图仅仅是本发明 的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动性的前提 下， 还可以根据这些附图获得其他的附图。
图 1现有技术种的一种多路输出电路结构图；
图 2本发明实施例的一种电压变换方法的流程图；
图 3本发明实施例的一种电压变换方法的流程图；
图 4本发明实施例的一种电压变换装置的结构图；
图 5本发明实施例的一种电压变换模块的结构图；
图 6本发明实施例的一种电压变换装置的结构图；
图 7本发明实施例的一种电压变换装置的结构图；
图 8本发明实施例的一种驱动时序图；
图 9本发明实施例的一种驱动时序图；
图 10本发明实施例的一种供电系统的结构图。
具体实施方式
下面将结合本发明实施例中的附图，对本发明实施例中的技术方案进行清 楚、 完整地描述， 显然， 所描述的实施例仅仅是本发明一部分实施例， 而不是 全部的实施例。基于本发明中的实施例， 本领域普通技术人员在没有作出创造 性劳动前提下所获得的所有其他实施例， 都属于本发明保护的范围。
如图 2所示， 本发明实施例提供一种电压变换方法， 包括：
S101 , 对输入的 DC ( Direct Current, 直流） 电压进行前级变压， 输出隔 离直流压电压；前级变压包括将输入的直流电压转换为待变交流电压的原边变 换，原边变换的工作周期包括上半周期和下半周期，上半周期与下半周期相互 切换之间存在死区时间。 在一个实施例中， 上半周期与下班周期的相互切换可以是， 一个工作周期 中， 上半周期向下半周期切换； 在另一个实施例中， 还可以是， 一个工作周期 的下半周期向另一个工作周期的上半周期进行切换，本发明实施例不做特别的 限定。
S102,在上述死区时间对输出的隔离直流压电压进行电容滤波补偿，输出 稳定的中间 DC电压；
在一个实施例中， 在步骤 S101输出隔离直流电压后， 对隔离直流电压直 接进行不需要电感的电容滤波，对隔离直流电压进行电容滤波补偿，输出稳定 的中间 DC电压。
在原边变换的死区时间， 通过滤波电容进行滤波补偿， 可以滤除原边变换 的输出电压进行跳变）， 输出稳定的中间直流电压。
在一个实施例中， 滤波电容在原边变换的工作时间 （非死区时间）充电， 在原边变换的死区时间，通过释放充电得到的能量对隔离直流电压进行电容滤 波补偿， 可以滤除原边变换在死区时间时产生的电压紋波（此电压紋波可以导 致上半周期切换到下半周期的输出电压进行跳变），输出稳定的中间直流电压。
S103 ,对中间 DC电压进行至少两路独立的后级变压， 输出至少两路负载 所需的 DC电压。
本发明实施例通过以上技术方案，在前级变压过程中的原边变换的死区时 间， 对经过前级变压后输出的隔离直流电压直接通过滤波电容进行电容滤波， 通过上述滤波电容可以得到的稳定的中间 DC电压，使后级可以不用和前级进 行同步而实现多路输出， 实现前后级工作方式解耦， 减少了电压变换时受干扰 的程度。
如图 3所示， 本发明实施例提供一种电压变换方法， 包括：
S110, 对输入的 DC电压进行原边变换， 输出待变交流电压； 原边变换的 工作周期包括上半周期和下半周期，上半周期与下半周期相互切换之间存在死 区时间。
在一个实施例中， 上半周期与下班周期的相互切换可以是， 一个工作周期 中， 上半周期向下半周期切换； 在另一个实施例中， 还可以是， 一个工作周期 的下半周期向另一个工作周期的上半周期进行切换，本发明实施例不做特别的 限定。
在一个实施例中，死区时间会导致上半周期的输出电压和下半周期的输出 电压相对来说， 会出现跳变。
在一个实施例中， 可以通过固定占空比的前级 PWM ( Pulse Width
Modulation, 脉沖宽度调制 )信号驱动由 MOSFET ( Metal Oxide Semiconductor Field Effect Transistor,金属氧化物半导体场效应管）接成的全桥电路来对输入 的 DC电压进行原边变换。在一个实施例中可以通过电压前馈变频控制上述前 级 PWM控制信号的占空比接近 50 %， 也就是说，前级 PWM信号的占空比与 50 %的差的绝对值不为零， 且在预设的范围之内。 在实际应用中， 可以控制上 述前级 PWM信号的占空比为 48 %、 49 %或者 51 %等。通过控制上述前级 PWM 信号的占空比接近 50 %， 可以令死区时间较小。
在一个实施例中，可以通过固定占空比的前级 PWM信号驱动半桥电路或 者推挽电路等方式来对输入的 DC电压进行原边变换。
S120, 对待变交流电压进行隔离变压， 输出方波电压；
在一个实施例中， 可以通过隔离变压器对待变交流电压进行隔离变压，得 到方波电压， 该方波电压为隔离交流电压。
S130, 对方波电压进行同步整流， 滤除谐波分量， 输出隔离直流电压； 在一个实施例中， 对方波电压进行同步整流， 是为了滤除谐波分量， 输出 隔离直流电压；
S140,在上述死区时间对隔离直流电压进行电容滤波补偿，输出稳定的中 间 DC电压；
在一个实施例中， 在原边全桥电路的死区时间， 利用滤波电容对隔离直流 电压进行滤波补偿， 可以提供一定的能量， 从而维持直流电压输出， 输出稳定 的中间 DC电压。
在 S110 中提到， 在一个实施例中可以通过电压前馈变频控制前级 PWM 控制信号的占空比接近 50 %， 也就是说， 前级 PWM信号的占空比与 50 %的 差的绝对值不为零， 且在预设的范围之内。 在实际应用中， 可以控制上述前级 PWM信号的占空比为 48 %、 49 %或者 51 %等。通过控制上述前级 PWM信号 的占空比接近 50 %， 可以令死区时间较小。 这样， 对前级原边整流输出的隔 离直流电压只需要容量很小滤波电容进行滤波， 不需要额外的滤波电感。 例如 在一个实施例中， 滤波电容的容量在 1 Ouf以下。
S150,对中间 DC电压进行至少两路独立的后级变压， 输出至少两路负载 所需的 DC电压。
在一个实施例中可以根据后级变压后输出的 DC电压，动态调整控制后级 变压的后级 PWM信号（例如， 根据输出电压动态调整后级 PWM信号的占空 比或者相位或者频率等）， 实现对两路独立的后级变压进行交错控制。
本发明实施例通过以上技术方案，在前级变压过程中的原边变换的死区时 间，对经过前级变压后输出的隔离直流电压直接通过滤波电容进行电容滤波补 偿， 通过上述滤波电容可以得到的稳定的中间 DC电压， 使后级可以不用和前 级进行同步而实现多路输出， 实现前后级工作方式解耦， 减少了电压变换时受 干扰的程度。 进一步地， 通过固定占空比的前级 PWM信号控制前级原边变换 工作在接近 50 %的固定占空比， 前级死区时间很短， 对前级原边整流输出的 隔离直流电压只需要容量很小滤波电容进行滤波， 不需要额外的滤波电感， 缩 小了电路空间。
如图 4所示， 本发明实施例提供一种电压变换装置， 包括：
第一变压模块 310， 用于对输入的 DC电压进行前级变压， 输出隔离直流 压电压； 前级变压包括将输入的直流电压转换为待变交流电压的原边变换，原 边变换的工作周期包括上半周期与下半周期，上半周期与下半周期相互切换之 间存在死区时间。
在一个实施例中， 上半周期与下班周期的相互切换可以是， 一个工作周期 中， 上半周期向下半周期切换； 在另一个实施例中， 还可以是， 一个工作周期 的下半周期向另一个工作周期的上半周期进行切换，本发明实施例不做特别的 限定。
电容滤波模块 320， 用于在上述死区时间对上述隔离直流压电压进行电容 滤波补偿， 输出中间 DC电压；
在一个实施例中， 在原边全桥电路的死区时间， 利用电容滤波模块 320 对隔离直流电压进行滤波补偿，可以提供一定的能量，从而维持直流电压输出， 输出稳定的中间 DC电压。
第二变压模块 330， 用于对中间 DC电压进行至少两路独立的后级变压， 输出至少两路负载所需的 DC电压。
在一个实施例中， 第二变压模块 330可以包括至少两个电压变换电路， 在本实施例中，上述至少两个电压变换电路中的任一个变换电路通常为降 压变换 BUCK电路，在某些情况下也可以由升压变换 BOOST电路担任，但是 由于 BUCK电路的紋波较小， 也比较容易控制， 所以使用比较广泛， 在本实 施例中将重点以 BUCK电路为例进行描述。
本发明实施例通过以上技术方案，在前级变压过程中的原边变换的死区时 间，对经过前级变压后输出的隔离直流电压直接通过滤波电容进行电容滤波补 偿， 通过上述滤波电容可以得到的稳定的中间 DC电压， 使后级可以不用和前 级进行同步而实现多路输出， 实现前后级工作方式解耦， 减少了电压变换时受 干扰的程度。
如图 5所示， 在一个实施例中第一电压变换模块 310可以包括：
原边变换单元 311， 用于对输入的 DC电压进行原边变换， 输出待变交流 电压； 原边变换单元的工作周期包括上半周期和下半周期， 上半周期和下半周 期相互切换之间存在死区时间。
在一个实施例中， 原边变换单元 311可以为全桥电路； 在一个实施例中， 可以通过固定占空比的前级 PWM信号控制全桥电路来对输入的 DC电压进行 前级原边整流。在一个实施例中可以通过电压前馈变频控制上述前级 PWM控 制信号的占空比为接近 50 %。 在实际应用中， 可以控制上述 PWM控制信号 的占空比为 48 %、 49 %或者 51 %等。 通过控制上述前级 PWM信号的占空比 接近 50 %， 可以另死区时间较小 （也就是说， 此时第一电压变换模块工作的 占空比控制在接近 50 % )。 此时， 相应地， 所述电容滤波模块 320由小容量滤 波电容构成， 例如在一个实施例中， 滤波电容的容量在 10uf以下。
在一个实施例中，原边变换单元 311还可以为半桥电路；在一个实施例中， 原边变换单元 311还可以为推挽电路。 在一个实施例中， 可以通过固定占空比 的前级 PWM信号控制半桥电路或推挽电路来对输入的 DC电压进行前级原边 整流。在一个实施例中可以通过电压前馈变频控制上述前级 PWM控制信号的 占空比为接近 50 %。 在实际应用中， 可以控制上述 PWM控制信号的占空比 为 48 %、 49 %或者 51 %等。通过控制上述前级 PWM信号的占空比接近 50 %， 可以另死区时间较小， 此时， 相应地， 所述电容滤波模块 320由小容量滤波电 容构成。 例如在一个实施例中， 滤波电容的容量在 10uf以下。
需要说明的是， 原边变换单元 311的上半周期和下半周期是指全桥电路、 半桥电路和推挽电路的工作周期的上半周期和下半周期，上半周期和下半周期 相互切换之间存在死区时间。
变压单元 312， 用于对待变交流电压进行隔离变压， 输出方波电压， 该方 波电压为隔离交流电压；
在一个实施例中， 变压单元 312的具体形式可以为隔离变压器。
整流单元 313， 用于对方波电压进行同步整流， 滤除谐波分量， 输出隔离 直流电压。
在一个实施例中， 对方波电压（即， 隔离交流电压）进行同步整流， 是为 了滤除谐波分量， 输出隔离直流电压；
本发明实施例通过以上技术方案，在前级变压过程中的原边变换的死区时 间，对经过前级变压后输出的隔离直流电压直接通过滤波电容进行电容滤波补 偿， 通过上述滤波电容可以得到的稳定的中间 DC电压， 使后级可以不用和前 级进行同步而实现多路输出， 实现前后级工作方式解耦， 减少了电压变换时受 干扰的程度。 进一步地， 通过固定占空比的前级 PWM信号控制前级原边变换 工作在接近 50 %的固定占空比， 前级死区时间很短， 对前级原边整流输出的 隔离直流电压只需要容量很小滤波电容进行滤波， 不需要额外的滤波电感， 缩 小了电路空间。
如图 6所示， 本发明实施例提供一种电压变换装置， 包括：
变压器 Tl， 全桥电路 Ql， 整流电路 PI， 滤波电容 Cl， 以及两个独立的 BUCK变压电路 J1和 J2。 具体地， 根据图 6:
在变压器 T1的输入侧（即，原边）的全桥电路 Q1由 4个 MOSFET: Qlp Q2p , Q3p Q4 接成。其中， Q2p , Q3p 为一对同时导通的桥臂， Qlp , Q4p 为另一对共同导通的桥臂。 全桥电路 Q1对输入的 DC电压进行原边变换， 输 出待变交流电压； 在一个实施例中， 全桥电路 Q1的原边对输入的 DC电压进 行开关变换， 将输入的 DC电压变为交流电压， 即待变交流电压。 需要说明的 是， 全桥电路 Q1的工作周期包括上半周期和下半周期。 在上半周期， Qlp 和 Q4p导通， 在下半周期 Q2p 和 Q3p导通， 通过上半周期和半周期， 可以将输 入的直流电压变为待变交流电压。 需要说明的是， 在上半周期和下半周期相互 切换之间存在死区时间， 即， 在一个工作周期中， 从上半周期切换到下半周期 时， 并不是立刻切换的， 而是有一个死区时间， 在这个死区时间内， 全桥电路 的 MOS管并不导通。 或者， 在另一个实施例中， 从一个工作周期的下半周期 切换到另一个工作周期的上半周期时， 并不是立刻切换的， 而是有一个死区时 间， 在这个死区时间内， 全桥电路的 MOS管并不导通。
变压器 T1对全桥电路 Q1输出的待变交流电压进行隔离变压， 输出方波 电压， 该方波电压为隔离交流电压； 在一个实施例中， 变压器 T1为隔离变压 器。
整流电路 P1对变压器 T1输出的方波电压进行同步整流， 滤除高频紋波 (即， 谐波分量）， 输出隔离直流电压； 图 6中， 整流电路 P1由两个全桥同步 整流管： 第一同步整流管 Q1SR和第二同步整流管 Q2SR接成。 在一个实施 例中，第一同步整流管 Q1 SR和第二同步整流管 Q2SR可以为 MOSFET。在本 实施例中， Q1 SR的栅极连接到 Q2SR的漏极， 组成自驱动整流电路， 利用变 压器 T1 自身绕组产生同步整流驱动信号。 由于 Q2SR的栅极连接到了 Q1SR 的漏极（也就是变压器 T1的输出端），本实施例中的自驱动时序同步于变压器 T1的输出信号。 Q2SR， 同样为类似自驱动连接， Q1 SR的栅极连接到了 Q2SR 的漏极。 当然可以理解的是， 在另一个实施例中， Q1 SR和 Q2SR也可以采用 它驱方式， 采用它驱方式的时候驱动时序于自驱动时序保持相同。
整流电路 P1对方波电压 （即， 隔离交流电压）进行同步整流后， 输出隔 离直流电压。 在一个实施例中， 整流电路 P1对方波电压（即， 隔离交流电压） 进行同步整流， 是为了滤除谐波分量， 输出隔离直流电压。
利用滤波电容 C1对隔离直流电压进行直接滤波， 输出稳定的中间 DC电 压。
图 6中， 变压器 T1输出的方波电压经同步整流后直接连接滤波电容 C1 进行滤波， 滤波电容 C1对隔离直流电压进行滤波补偿， 维持直流输出， 输出 中间 DC电压。
在原边全桥电路的死区时间， 全桥电路的 MOS管并不导通， 此时就会造 成全桥电路输出跳变的情况，即上半周期的输出电压和下半周期的输出电压相 比来说会有一个跳变，这样后级 BUCK变压电路 J1和 J2就没有足够稳定输入 电压， 而本实施例中， 通过滤波电容 C1隔离直流电压进行滤波补偿， 可以提 供一定的能量， 从而维持直流电压输出， 在原边全桥电路工作存在死区时间， 通过滤波电容进行滤波补偿，可以滤除原边全桥电路工作在死区时间时产生的 电压紋波（此电压紋波可以导致上半周期和下半周期进行相互切换时， 输出电 压进行跳变）， 输出稳定的中间直流电压。
在本实施例中， 滤波电容 C1在非死区时间（如， 上半周期或者下半周期） 利用整流电路 P1的输出电压进行充电， 在全桥电路 Q1的死区时间， 这时整 流电路 P1的输出电压（隔离直流电压）就会发生跳变， 此时， 由于电压不稳 定（跳变引起的）， 滤波电容就会释放之前充电吸收的能量， 对隔离直流电压 进行滤波补偿， 输出稳定的中间直流电压。
中间 DC电压经过两个独立的 BUCK变换电路， 如第一 BUCK变换电路
J1和第二 BUCK变换电路 J2， 输出两路负载所需的不同 DC电压。 图 6中， MOSFET Q1B1和 MOSFET Q2B1组成一个 BUCK变换电路， 电感 L1、 电容 C2 用于对该 BUCK 变换电路的输出进行滤波。 同样， MOSFET Q1B2 和 MOSFET Q2B2组成另一个 BUCK变换电路，电感 L2、电容 C3用于对该 BUCK 变换电路的输出进行滤波。
根据图 6， 在一个实施例中， 该电压变换装置还包括两个 PWM控制器， 第一 P WM控制器 61和第二 P WM控制器 62。由于有两个控制环路，每个 BUCK 变换电路可以独立调节输出电压及其它保护功能。 根据图 6， MOSFET Q1B1 和 MOSFET Q2B1分别接到第一 PWM控制器 61的 OUT H和 OUT 1引脚。 MOSFET Q1B2和 MOSFET Q2B2分别接到第二 PWM控制器 62的 OUT H和 OUT 1引脚。 每个 PWM控制器的反馈输入端（ F/B端）还与每个 BUCK变换 电路的输出端相连， 例如， 第一 PWM控制器 61的 F/B端和第一 BUCK变换 电路 J1的输出端的分压电阻 R1相连， 第二 PWM控制器 62的 F/B端和第二 BUCK变换电路 J2的输出端的分压电阻 R2相连。这样 PWM控制器就可已根 据每个 BUCK 变换电路的输出电压动态调整控制每个 BUCK 变换电路的 PWM信号 （例如， 根据输出电压动态调整 PWM信号的占空比或者相位或者 频率等），从而实现对 BUCK变换电路的交错控制。在一个实施例中控制 BUCK 变换电路的 PWM信号可以采用后沿调制； 在一个实施例中控制 BUCK变换 电路的 PWM信号还可以采用前沿调制。
PWM控制器也可以产生固定占空比的前级 PWM信号（例如提供占空比 为接近 50 %的固定占空比的前级 PWM信号）， 然后可以通过隔离单元 60和 驱动单元 70驱动原边的 MOSFET: Qlp, Q2p , Q3p, Q4p的栅极控制各栅极 导通， 使原边全桥电路 Q1工作在固定占空比（如， 接近 50 %的固定占空比）。 当然艮好理解的是， 在一个实施例中， 固定占空比的前级 PWM信号也可以由 单独的脉沖信号产生单元产生，也可以像上述本实施例中由前级 PWM控制器 一并产生。 在本实施例中， 因为 PWM控制器放在副边， 因此隔离单元 60的 作用就是将产生的规定占空比的前级 PWM信号传递到原边。 驱动单元 70的 作用是将隔离单元 60传递过来的固定占空比的前级 PWM信号进行放大。 当 然， 在另一个实施例中， 该电压变换装置也可以不包括隔离单元 60。
而且， 由于原边全桥电路工作在接近 50%的固定占空比， 死区时间很短， 这样依靠变压器 T1 自身的漏感， 就只需要容量很小的滤波电容 C1对隔离直 流电压进行滤波补偿， 而不需要额外的滤波电感， 缩小了电路的空间。 例如在 一个实施例中， 滤波电容 C1的容量在 10uf以下。
如图中的虚线示， 上述 PWM控制器可以通过将输入电压 VIN端连接到 滤波电容 C1的上端， 通过测量得到滤波电容 C1的输出电压， 再才艮据此输出 电压， 可以计算出变压器 T1的原边电压， 从而可以动态的调节前级 PWM信 号的频率， 从而调节原边全桥的工作频率， 控制变压器 T1的交变磁通， 进一 步减小前级的磁芯损耗， 扩大前级固定占空比谐振拓朴的输入电压工作范围。
当然， 可以理解的是， 在另一个实施例中， 还可以根据辅助电源或则其他 方式检测得到滤波电容 C1的输出电压，再根据此输出电压，计算出变压器 T1 的原边电压才艮据。 变压器 T1的原边电压可以动态的调节前级 PWM信号的频 率， 从而调节原边全桥的工作频率。 在一个实施例中，上述固定占空比的前级 PWM信号的占空比可以为接近 50 % , 即前级 PWM信号的占空比与 50 %的差的绝对值不为零， 且在预设的 范围之内。 在实际应用前级 PWM信号的占空比可以为 49 %的固定占空比、 48 %的固定占空比或者 53 %的固定占空比等。
当然可以理解的是， 在一个实施例中， 还可以将上述两个 PWM控制器集 成在一个芯片里面。
另外如图 7所示， 在另一个实施例中， 图 6中的两路输出可以并联在一起 并联工作模式下， 这时两路各自的滤波电感 L1和 L2可以为耦合电感， 当并 联在一起时， 两路输出电压相同。
如图 8所示， 本实施例提供一种 Qlp ~ Q4p的驱动时序图。 根据图 8， 原 边全桥整流电路中的 Qlp、 Q2p、 Q3 和 Q4p工作在接近 50 %的固定占空比 (图 8中的占空比为 49 % )， 其中 Qlp和 Q4p为同一驱动时序， Q2p与 Q3p 为同一驱动时序， Qlp与 Q3p 为互补对称控制。
如图 9所示，本实施例提供一种 Q1B1和 Q1B2的驱动时序图。根据图 8， 两路 BUCK变换电路的上管 Q1B1和 Q1B2驱动移相 180度， 从而能使两路 BUCK实现交替工作， 减小一次测输入反射电流紋波。
可以理解的是， 在一个实施例中， 如果负载为 3路， 相应的就需要 3路 BUCK变换电路来输出 3路负载所需的不同 DC电压。 这时， 可以通过 PWM 信号控制 3路 BUCK变换电路的上管驱动互相移相 120度， 从而能使 3路 BUCK变换电路实现交替工作。
当然很好理解的是，图 6和图 7对应的实施例给出了负载为两路时的情况， 当负载为多路（如 3路或者 4路）时， 采用的技术方案和两路时类似， 并没有 实质的改变， 在此不再赘述。
本发明实施例通过以上技术方案，通过 PWM控制器使原边全桥电路工作 在接近 50%的固定占空比， 死区时间很短， 依靠变压器 T1 自身的漏感， 输出 只需要容量很小滤波电容， 不需要额外的滤波电感， 缩小了电路空间； 电路中 有了输出电容 C1后， 在死区时间， 通过 C1上得到的稳定的中间 DC电压， 后级可以实现多路输出， 而不用和前级进行同步， 可以实现前后级工作方式解 耦。 通过多个 PWM控制器实现交错 PWM后沿调制工作方式； 该 PWM控制 器还可以通过 CI或者辅助电源或则其他方式得到变压器 T1得原边电压 VIN， 根据得到的输入电压 VIN，动态的调节前级原边全桥的工作频率，控制变压器 T1 的交变磁通， 进一步减小前级的磁芯损耗， 扩大前级固定占空比谐振拓朴 的输入电压工作范围。
如图 10所示， 本发明实施例提供一种供电系统， 包括： 电压变换装置 10 和至少两路负载， 为方便描述， 图 10 中给出了负载为两路（负载 20和负载 30 ) 时的结构图， 当负载为多路时， 不影响本发明的实质。
电压变换装置 10， 用于对输入的 DC 电压进行前级变压， 输出隔离直流 压电压； 前级变压包括将输入的直流电压转换为待变交流电压的原边变换， 所 述原边变换的工作周期包括上半周期和下半周期，上半周期与下半周期相互切 换之间存在死区时间； 在死区时间对输出的隔离直流压电压进行电容滤波补 偿，输出稳定的中间 DC电压；对中间 DC电压进行至少两路独立的后级变压， 输出两路负载（负载 20和负载 30 )所需的 DC电压。
当负载为多路（如 3路或者 4路）时， 采用的技术方案和两路时类似， 并 没有实质的改变， 在此不再赘述。
电压变换装置 10的结构和功能可以如上述任一实施例所述， 在此不再赘 述。
本发明实施例通过以上技术方案，在前级变压过程中的原边变换的死区时 间， 对经过前级变压后输出的隔离直流电压直接通过滤波电容进行电容滤波， 通过上述滤波电容可以得到的稳定的中间 DC电压，使后级可以不用和前级进 行同步而实现多路输出， 实现前后级工作方式解耦， 减少了电压变换时受干扰 的程度。
以上所述仅为本发明的几个实施例，本领域的技术人员依据申请文件公开 的可以对本发明进行各种改动或变型而不脱离本发明的精神和范围。

Claims

权 利 要 求
1、 一种电压变换装置， 其特征在于， 包括：
第一变压模块，用于对输入的直流电压进行前级变压，输出隔离直流电压， 所述前级变压包括将输入的直流电压转换为待变交流电压的原边变换，所述原 边变换的工作周期包括上半周期和下半周期，上半周期与下半周期相互切换之 间存在死区时间；
电容滤波模块，用于在所述死区时间对所述隔离直流电压进行电容滤波补 偿， 输出稳定的中间直流电压；
第二变压模块， 用于对所述中间直流电压进行至少两路独立的后级变压， 输出至少两路负载所需的直流电压。
2、 如权利要求 1所述的电压变换装置， 其特征在于， 所述第一变压模块 包括：
原边变换单元，用于对输入的直流电压进行原边变换，输出待变交流电压； 变压单元， 用于对所述待变交流电压进行隔离变压， 输出方波电压， 所述 方波电压为隔离交流电压；
整流单元， 用于对所述方波电压进行同步整流， 输出隔离直流电压。
3、 如权利要求 2所述的电压变换装置， 其特征在于， 所述原边变换单元 为全桥电路、半桥电路或者推挽电路的一种， 所述原边变换过程中上半周期与 下半周期指全桥电路、 半桥电路或推挽电路工作周期的上半周期与下半周期。
4、如权利要求 3所述的电压变换装置，其特征在于，所述装置还包括 PWM 控制器， 所述 PWM控制器用于产生驱动所述全桥电路、 半桥电路或者推挽电 路的固定占空比的前级 PWM信号， 所述前级 PWM控制信号的占空比控制在 接近 50%， 所述电容滤波模块由小容量滤波电容构成。
5、 如权利要求 4所述的电压变换装置， 其特征在于， 所述第二变压模块 包括至少两个电压变换电路，所述至少两个电压变换电路中的任一个变换电路 为降压变换电路。 6、 如权利要求 5所述的电压变换装置， 其特征在于， 所述 PWM控制器 还用于产生驱动所述至少两个电压变换电路的后级 PWM信号， 所述 PWM控 制器根据所述至少两个电压变换电路的输出电压动态调整所述后级 PWM信 号。
7、 如权利要求 2所述的电压变换装置， 其特征在于， 所述整流单元包括： 第一同步整流管和第二同步整流管，所述第一同步整流管的栅极连接到所述第 二同步整流管的漏极，所述第二同步整流管的栅极连接到所述第一同步整流管 的漏极。
8、 一种电压变换方法， 其特征在于， 包括：
对输入的直流电压进行前级变压，输出隔离直流电压， 所述前级变压包括 将输入的直流电压转换为待变交流电压的原边变换，所述原边变换的工作周期 包括上半周期与下半周期， 上半周期与下半周期相互切换之间存在死区时间； 在所述死区时间对所述隔离直流电压进行滤波补偿，输出稳定的中间直流 电压；
对所述中间直流电压进行至少两路独立的后级变压，输出至少两路负载所 需的直流电压。
9、 如权利要求 8所述的电压变换方法， 其特征在于， 所述对输入的直流 电压进行前级变压， 输出隔离直流电压， 包括：
对输入的直流电压进行原边变换， 输出待变交流电压；
对所述待变交流电压进行隔离变压， 输出方波电压， 所述方波电压为隔离 交流电压；
对所述方波电压进行同步整流， 输出隔离直流电压。
10、 如权利要求 8或 9所述的电压变换方法， 其特征在于， 所述原边变换 的驱动信号为固定占空比的前级脉沖宽度调制 PWM信号。
11、 如权利要求 8或 9所述的电压变换方法， 其特征在于， 所述对输入的 直流电压进行原边变换是通过全桥电路、半桥电路或推挽电路将输入的直流电 压变为待变交流电压， 所述原边变换过程中上半周期与下半周期指全桥电路、 半桥电路或推挽电路工作周期的上半周期与下半周期。
12一种供电系统， 包括至少两路负载， 其特征在于， 还包括为所述至少 两路负载供电的电压变换装置；
所述电压变换装置用于对输入的直流电压进行前级变压，输出隔离直流电 压， 所述前级变压包括将输入的直流电压转换为待变交流电压的原边变换， 所 述原边变换的工作周期包括上半周期和下半周期，上半周期与下半周期相互切 换之间存在死区时间； 在所述死区时间对所述隔离直流电压进行电容滤波补 偿，输出稳定的中间直流电压； 对所述中间直流电压进行至少两路独立的后级 变压， 输出所述至少两路负载所需的直流电压。
13、 如权利要求 12所述的供电系统， 其特征在于， 所述电压变换装置具 体用于：
第一变压模块，用于对输入的直流电压进行前级变压，输出隔离直流电压， 所述前级变压包括将输入的直流电压转换为待变交流电压的原边变换，所述原 边变换的工作周期包括上半周期和下半周期，上半周期与下半周期相互切换之 间存在死区时间；
电容滤波模块，用于在所述死区时间对所述隔离直流电压进行电容滤波补 偿， 输出稳定的中间直流电压；
第二变压模块， 用于对所述中间直流电压进行至少两路独立的后级变压， 输出至少两路负载所需的直流电压。
14、 如权利要求 12所述的供电系统， 其特征在于， 所述第一变压模块工作 的占空比控制在接近 50%， 所述电容滤波模块由小容量滤波电容构成。
PCT/CN2010/078125 2010-04-01 2010-10-26 电压变换装置、方法及供电系统 WO2011120299A1 (zh)

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