US6950639B2 - Local oscillator for generating an HF signal for direct mixing by using avalanche photodiodes - Google Patents

Local oscillator for generating an HF signal for direct mixing by using avalanche photodiodes Download PDF

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US6950639B2
US6950639B2 US09/911,272 US91127201A US6950639B2 US 6950639 B2 US6950639 B2 US 6950639B2 US 91127201 A US91127201 A US 91127201A US 6950639 B2 US6950639 B2 US 6950639B2
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apd
local oscillator
resonant circuit
frequency
capacitance
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US20020032014A1 (en
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Torsten Gogolla
Andreas Winter
Helmut Seifert
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Hilti AG
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Hilti AG
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D9/00—Demodulation or transference of modulation of modulated electromagnetic waves
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/48—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S17/00
    • G01S7/491—Details of non-pulse systems
    • G01S7/4912—Receivers
    • G01S7/4913—Circuits for detection, sampling, integration or read-out
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
    • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
    • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
    • H03B5/1237—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator
    • H03B5/124—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator the means comprising a voltage dependent capacitance
    • H03B5/1243—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator the means comprising a voltage dependent capacitance the means comprising voltage variable capacitance diodes
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03L—AUTOMATIC CONTROL, STARTING, SYNCHRONISATION OR STABILISATION OF GENERATORS OF ELECTRONIC OSCILLATIONS OR PULSES
    • H03L7/00—Automatic control of frequency or phase; Synchronisation
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03L—AUTOMATIC CONTROL, STARTING, SYNCHRONISATION OR STABILISATION OF GENERATORS OF ELECTRONIC OSCILLATIONS OR PULSES
    • H03L7/00—Automatic control of frequency or phase; Synchronisation
    • H03L7/06—Automatic control of frequency or phase; Synchronisation using a reference signal applied to a frequency- or phase-locked loop
    • H03L7/08—Details of the phase-locked loop
    • H03L7/099—Details of the phase-locked loop concerning mainly the controlled oscillator of the loop
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03L—AUTOMATIC CONTROL, STARTING, SYNCHRONISATION OR STABILISATION OF GENERATORS OF ELECTRONIC OSCILLATIONS OR PULSES
    • H03L7/00—Automatic control of frequency or phase; Synchronisation
    • H03L7/06—Automatic control of frequency or phase; Synchronisation using a reference signal applied to a frequency- or phase-locked loop
    • H03L7/16—Indirect frequency synthesis, i.e. generating a desired one of a number of predetermined frequencies using a frequency- or phase-locked loop
    • H03L7/18—Indirect frequency synthesis, i.e. generating a desired one of a number of predetermined frequencies using a frequency- or phase-locked loop using a frequency divider or counter in the loop
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S17/00—Systems using the reflection or reradiation of electromagnetic waves other than radio waves, e.g. lidar systems
    • G01S17/02—Systems using the reflection of electromagnetic waves other than radio waves
    • G01S17/06—Systems determining position data of a target
    • G01S17/08—Systems determining position data of a target for measuring distance only
    • G01S17/32—Systems determining position data of a target for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated
    • G01S17/36—Systems determining position data of a target for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated with phase comparison between the received signal and the contemporaneously transmitted signal
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/48—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S17/00
    • G01S7/491—Details of non-pulse systems

Definitions

  • the invention is directed to a local oscillator for generating a narrow-band high-frequency signal (HF signal) for direct signal mixing with a reception signal generated by a reverse-biased avalanche photodiode (APD) from a light signal impinging on the APD.
  • HF signal narrow-band high-frequency signal
  • APD reverse-biased avalanche photodiode
  • Avalanche photodiodes are used in sensitive optoelectronic measurement systems, among others.
  • optoelectronic measurement of distances up to 100 m with accuracies of a few millimeters has great importance for numerous applications, particularly in the construction industry and in plants engineering.
  • Distance measuring systems of this kind should be as highly dynamic as possible so that very weak, as well as very strong, light reception signals can be processed. This renders superfluous the use of defined target marks on the object whose distance from a reference point is to be determined.
  • the possibility of direct distance measurement at determined surfaces, i.e., without the use of target marks makes it possible, particularly in the technical fields and industries mentioned above, to reduce manufacturing time and to cut costs while simultaneously reducing manufacturing tolerances.
  • a preferably sinusoidal intensity-modulated beam of a light source is directed to a measurement object (EP 0 701 702 B1, DE 196 43 287 A1, U.S. Pat. No. 4,403,857).
  • the intensity-modulated light which is reflected or backscattered from the measurement object is detected by a photodiode.
  • the distance to be measured is given by the phase displacement—relative to the emitted light intensity of the light source—of the light intensity which is modulated in sinusoidal manner and backsctattered from the measurement object.
  • a high voltage is required in the blocking direction in order to generate the high electric field strengths in the avalanche zone of the APD.
  • this voltage ranges from 40 V to 500 V.
  • Typical gain factors of the photocurrents are between 10 and 200. They are highly dependent on the semiconductor material (e.g., Si, InGaAs), the construction of the photodiode, the blocking voltage and the temperature.
  • a method of direct signal mixing may be applied in the receiver as is known, for example, from U.S. Pat. No. 4,503,857, cited above, and from a scientific paper by K. Seta, T. Oh'Ishi, “Distance Measurement Using a Pulse Train Emitted from a Laser Diode”, Japanese J. of Appl. Physics, Volume 26, No. 10, pages L1690-L1692, October 1987, and as also has been suggested in an especially advantageous modification in German Patent Application 100 06 493.0, also cited above.
  • the (preferably) sinusoidal signal of frequency f LO of a local oscillator and an amplitude of >1 V are superimposed on the APD blocking voltage, so that along with the blocking voltage the gain factor M of the APD, i.e., its internal current source, is also modulated.
  • M(t) is the modulated APD gain depending on time t
  • i FOTO (t) is the internal photocurrent generated by the light incidence.
  • a mix product results, i.e., an intermediate frequency signal (IF signal), which oscillates with the frequency difference between the frequency f LO of the local oscillator and the frequency f MESS of the modulated detected light output.
  • IF signal intermediate frequency signal
  • the output signal of the APD i.e., the IF signal, is (comparatively) low-frequency and can accordingly be further processed without difficulty. Since the mixing process takes place within the chip of the APD, the structural sizes of the mixing arrangement are typically smaller than the utilized modulation wavelength by three or four orders of magnitude.
  • the IF signal arising from the direct mixing is, as a rule, comparatively very low-frequency; e.g., 10 kHz to 100 kHz. No interference is to be expected in this frequency range. Also, parasitic characteristics of other electronic components are negligible at these low frequencies. Since the output signal of the APD is in the IF range, no other high-frequency components are needed in the reception part aside from the local oscillator. Accordingly, production costs as well as power consumption of an optoelectronic distance measuring system outfitted with APDs of this type can also be drastically reduced.
  • the present invention solves the above discussed problems by providing a local oscillator for generating a narrow-band HF signal for direct mixing with a (first) reception signal which is generated from an incident light signal by a reverse biased avalanche photodiode (APD), wherein the depletion layer capacitance of the APD is incorporated in a controllable HF resonant circuit whose resonant frequency forming the HF signal is determined by selective adjustments of a control unit which is acted upon by the reference oscillator.
  • APD reverse biased avalanche photodiode
  • the invention proceeds from the idea that the depletion layer capacitance of the APD which is used simultaneously as direct mixer is incorporated, as an element which (co-)determines oscillation, in an HF oscillator, especially a PLL-stabilized HF oscillator, wherein the LC oscillator is of primary relevance. Not only does this PLL-stabilized operation ensure the required narrow band of the local oscillator signal which is required for mixing with the reception signal used as measurement signal, but a comparatively high voltage amplitude is also achieved with low power consumption.
  • the adjustable divider ratio of the PLL-circuit together with the frequency of the reference oscillator determines the resonant frequency of the HF resonant circuit.
  • the HF resonant circuit is constructed as an LC resonant circuit whose resonant frequency is adjusted by means of a capacitance, in particular a capacitance diode, which is connected in parallel or in series with the depletion layer capacitance and which is controllable by means of the PLL-circuit.
  • a capacitance in particular a capacitance diode
  • the HF resonant circuit in which the depletion layer capacitance of the APD is incorporated can also be realized by means of a resonator which is controllable either directly or by the additional use of a controllable capacitance and/or a controllable inductance.
  • Resonators of this type can be, for example, crystals, dielectric ceramic resonators, cavity resonators or line resonators whose equivalent circuits correspond to an LC circuit as is known by one skilled in the art.
  • FIG. 1 shows a first embodiment of a local oscillator according to the invention which is realized as a PLL-stabilized LC parallel resonant circuit with a capacitance diode connected in parallel with the depletion layer capacitance of an APD;
  • FIG. 2 shows a modification of the first embodiment in which the PLL-circuit of FIG. 1 is replaced by a digital control loop or regulator circuit;
  • FIG. 3 shows a second embodiment of a local oscillator according to the invention based on an LC parallel resonant circuit with a capacitance diode connected in series with the depletion layer capacitance of the APD;
  • FIG. 4 shows a third embodiment of the local oscillator according to the invention based on an LC series resonant circuit
  • FIG. 5 shows a fourth embodiment of a PLL-stabilized LC oscillator according to the invention in which there are two APDs which are high-frequency connected in parallel and which are incorporated in an LC parallel resonant circuit;
  • FIG. 6 shows a fifth embodiment of a PLL-stabilized LC oscillator in which two APDs are connected in parallel with respect to high frequency and DC voltage;
  • FIG. 7 shows a sixth embodiment of a PLL-stabilized LC oscillator in which two APDs are connected in parallel with respect to high frequency and are incorporated in an LC series resonant circuit;
  • FIG. 8 shows a seventh embodiment of a PLL-stabilized LC oscillator in which two APDs are connected in series with respect to high frequency and are incorporated in an LC series resonant circuit.
  • a high voltage is supplied in the blocking direction, via an ohmic resistance R, to an APD 2 which is used, for example, as a measurement signal receiver.
  • the capacitance C of a capacitance diode 3 and an inductance L are connected in parallel with respect to the high-frequency signal with the depletion layer capacitance C APD of the APD 2 .
  • This circuit arrangement forms an LC parallel resonant circuit which is operated in idle at its resonant frequency.
  • the coupling capacitances C K in the circuit between the cathode-side base points of the APD 2 and the capacitance diode 3 and the inductance L and between the inductance L and the output of an HF amplifier 4 constitute a short circuit for the signal with the resonant frequency. As is known by the person skilled in the art, they serve only to decouple the different DC voltages to be applied to the APD 2 , capacitance diode 3 and inductance L.
  • the mixing product which is formed directly in the APD 2 from the reception signal of the APD 2 and the IF signal obtained from the narrow-band HF signal generated by the LC oscillator according to the invention can be tapped on the APD 2 via a low-pass 7 .
  • the low-pass 7 can be a capacitive short circuit to ground, for example, for the HF signals, i.e., for the reception signal of the PD 2 and for the local oscillator signal proceeding from the LC oscillator; however, the IF signal can pass unimpeded. Further, the anode of the APD 2 is grounded through the low-pass 7 via a relatively high-impedance resistance, resulting in a DC voltage component of almost 0 volts. Together with a feedback network 5 , which will not be described in more detail herein because it is known to the person skilled in the art, the HF amplifier 4 , e.g., in the form of a high-frequency transistor, ensures that the LC oscillator is stimulated to oscillate.
  • the HF amplifier 4 e.g., in the form of a high-frequency transistor
  • the capacitance C of the capacitance diode 3 connected in the blocking direction depends on its applied blocking voltage.
  • a small proportion of the LC oscillator signal is supplied, along with the output signal of the amplifier 4 , to a PLL-circuit 1 in which the frequency of the LC oscillator signal is reduced, e.g., by a frequency divider, and subsequently compared with respect to phase position to a phase-locked reference oscillator signal generated by a reference oscillator 6 .
  • the PLL-circuit 1 regulates the diode capacitance C and therefore the resonant frequency of the LC oscillator, so that there is phase equality between the frequency-reduced LC oscillator signal and the reference oscillator signal.
  • the resonant frequency of the LC oscillator adopts a reference value which is dependent on the adjustable divider ratio of the frequency divider of the PLL-circuit 1 and on the frequency of the reference oscillator 6 .
  • the reference oscillator 6 is the common master oscillator of the entire system, that is, particularly and by way of example, the optoelectronic distance measurement arrangement.
  • the resonant frequency of the LC oscillator is only dependent on the freely selectable divider ratio of the PLL-circuit 1 and on the frequency of the reference oscillator 6 .
  • it does not depend on the depletion layer capacitance C APD of the APD 2 which can involve substantial specimen scatter and is, moreover, dependent on temperature, voltage and reception power. Fluctuations in APD capacitance C APD are compensated completely by the controllable capacitance C of the capacitance diode 3 .
  • phase behavior of the local oscillator signal and accordingly also the phase behavior of the reception signal, particularly the measurement signal, which is obtained via the APD 2 is influenced in the intermediate frequency range by the capacitance variations of the APD 2 which are dependent on temperature and reception power, any distance measurement error caused by this is eliminated at the same time. Accordingly, a defined system state is always adjusted automatically independent from interfering ambient conditions. Other interfering parasitic circuit capacitance is also compensated.
  • the defined narrow-band system states described above can be realized at optional frequencies in the frequency lock-in range of the PLL-circuit 1 with the programmable divider ratio of the PLL circuit 1 .
  • the PLL-circuit can be replaced by a digital control loop or regulating circuit according to FIG. 2 .
  • the actual value of the frequency of the local oscillator signal proceeding from the LC oscillator is constantly measured by a frequency counter 20 using the reference oscillator signal and is subsequently supplied to a control unit 21 .
  • the control unit 21 is preferably formed by a digital computer, that is, for example, by a microcontroller or a signal processor.
  • the control unit 21 Based on the constantly measured actual frequency value, the control unit 21 generates suitable control voltages via a regulating algorithm by a digital-to-analog converter 22 , these control voltages being applied to the capacitance diode 3 , so that the resonant frequency adopts a predetermined, stabilized reference value. Fluctuations in the LC resonator values, depletion layer capacitance C APD and inductance L are accordingly compensated by a capacitance C of the capacitance diode 3 which is changed by the regulating algorithm via the control voltage.
  • the resonant frequency is roughly adjusted via the control voltage of the capacitance diode 3 which is generated by the control unit 21 with the digital-to-analog converter 22 ; this can be based on tabular values, for example. Variations in the resonant frequency are not compensated, i.e., no regulating algorithms are used. Often, the adjusted frequency value need only correspond approximately to a predetermined reference value, i.e., small, constant deviations from the reference value, but also time variations, are allowed. However, the instantaneous value of the resonant frequency and local oscillator frequency at a defined point in time must often be known exactly for subsequent calculations. Therefore, the actual frequency value is measured and supplied to the control unit 21 .
  • FIG. 3 shows a variant of the LC oscillator.
  • the capacitance C of the capacitance diode 3 is connected in series with the depletion layer capacitance of the APD 2 .
  • this series connection forms a parallel resonant circuit.
  • the anode of the capacitance diode 3 is connected to 0 volts, i.e., to ground, with respect to DC voltage via the inductance L.
  • the positive control voltage of the PLL-circuit 1 is again supplied to the cathode of the capacitance diode 3 .
  • FIG. 4 shows another series resonant circuit comprising inductance L, capacitance C of the capacitance diode and the APD depletion layer capacitance C APD .
  • the anode of the capacitance diode 3 is connected to ground potential with respect to DC voltage via a choke Dr.
  • the advantage of the arrangements according to FIGS. 3 and 4 consists in that with the necessary, relatively large diode capacitances C (e.g., 10-times greater than C APD ) the voltage amplitude across the capacitance diode 3 is lower than in the circuit according to FIG. 1 or FIG. 2 .
  • the capacitance diode 3 in a half-wave of the local oscillator signal is connected in the forward conducting direction. In this way, the amplitude is limited and strong harmonic wave components occur in the local oscillator signal. Therefore, in order to reduce the amplitude at the capacitance diode in the embodiment example in FIG. 1 , another capacitance (not shown) must generally be connected in series with the diode capacitance C in addition. This corresponds to a capacitive voltage divider. However, as a result of this step, the effective capacitance of this series connection is small, so that the tunable frequency range of the resonant circuit is also limited.
  • variable range of the effective resonator capacitance i.e., the series connection of C and C APD
  • the variable range of the effective resonator capacitance can be very small as in the circuit according to FIG. 1 due to the series connection of the variable capacitance C of the capacitance diode 3 and the depletion layer capacitance C APD of the APD 2 because of the small depletion layer capacitance of, e.g., 2 pF.
  • the frequency tuning range of the resonant circuit is accordingly limited.
  • the amplitude of the voltage at the APD 2 is dependent on the control voltage of the PLL due to the capacitive voltage divider comprising C and C APD due to the variable capacitance C.
  • FIG. 5 shows another supplemented variant of the basic circuit arrangement according to FIG. 1 .
  • the described PLL-stabilized LC oscillator is formed by an HF-signal parallel connection of a first depletion layer capacitance C APD,R of the APD 2 which in this case serves as a reference receiver of an optoelectronic distance measurement device, the depletion layer capacitance C APD,M of an APD 10 used as a measurement signal receiver, the capacitance C of the capacitance diode 3 and the inductance L.
  • the APD 2 of the reference receiver receives the sinusoidal intensity-modulated light emitted by a laser diode transmitter (not shown) directly after traversing a reference path with a known optical path length.
  • the APD 10 of the measurement receiver detects the intensity-modulated light which is backscattered from the remote measurement object and which is likewise sinusoidal.
  • the two electric signals generated in the APDs 2 , 10 are converted by direct mixing with the local oscillator signal proceeding from the LC oscillator in the (comparatively) low-frequency intermediate frequency range.
  • the IF reference signal IF R and the IF measurement signal IF M are extracted via associated low-pass filters 7 and 11 , respectively.
  • the distance to be measured can be determined. Phase errors generated by the laser diode transmitter are eliminated by this subtraction.
  • reference is had to the optoelectronic distance measurement device which was already mentioned and which is described in DE Patent Application 100 06 493.0.
  • the two APDs according to the embodiment example in FIG. 6 can be connected in parallel with respect to the HF signal as well as with respect to DC voltage.
  • the common voltage signal of the local oscillator is transmitted to the respective depletion layers of the APDs 2 , 10 in phase in that the APD 10 of the measurement branch and the APD 2 of the reference branch are connected in parallel with respect to HF signal in case the bulk resistances of the APDs are negligible. Since the bulk resistances in silicon APDs are generally less than 0.1 ⁇ , this precondition is also given in practice.
  • the phase of the local oscillator signal is influenced only by the sum capacitance of the parallel circuit C
  • the mixing process takes place in the internal power source of the respective APD, so that the high-frequency reception signal is also not influenced by the depletion layer capacitance of the respective APD and other external parasitic capacitances.
  • the frequency of the frequency-converted IF signal is so low that the depletion layer capacitance which is dependent on environmental conditions and specimen scatter plays no part in this respect. Since, in this case, the APD 10 of the measurement branch and the APD 2 of the reference branch are acted upon by the same local oscillator and variations in the sum capacitance have the same effect for both branches, error compensation can be achieved within certain limits by subtracting the phases of the respective IF signals.
  • the inductance L for the IF signal constitutes a good short circuit, so that crosstalk of the IF signals from the reference signal branch to the measurement signal branch is sharply reduced.
  • compensation of capacitances is only significant for generating the constant resonant frequency.
  • the depletion layer capacitances C APD,R and C APD,M of the avalanche photodiodes 2 and 10 are incorporated in an LC series resonant circuit.
  • the LC circuit is formed by the inductance L through the capacitance C of the capacitance diode 3 connected in series with L and by the sum capacitance C APD,R +C APD,M of the parallel-connected APDs 2 , 10 which is connected in series with L.
  • the coupling capacitances C K are again HF-signal short circuits.
  • the low-pass filters 7 and 11 serve to decouple the IF signals of the measurement channel and reference channel and generate a DC voltage component of virtually 0 volts at the anodes of the APDs 2 , 10 in that a connection to ground is produced via a relatively high-impedance resistance.
  • the low-pass filters 7 , 10 do not pass the HF signals, i.e., the reception signals of the APDs 2 , 10 , or the local oscillator signal proceeding from the LC oscillator.
  • the amplifier 4 together with the feedback network 5 in turn leads to a stable operation of the LC oscillator with constant amplitude.
  • the resonant frequency is also regulated in this example in the manner described above by the PLL-circuit 1 by influencing the blocking voltage at the capacitance diode 3 .
  • the anode of the capacitance diode is connected to earth, i.e., grounded, with respect to DC voltage via the choke Dr.
  • the choke Dr does not pass the HF signals and constitutes a short circuit for the low-frequency IF signals, so that crosstalk of IF signals from the reference signal branch to the measurement signal branch is also sharply reduced in this case.
  • the coupling capacitances C K are high impedance for the low-frequency IF signals.
  • the two APDs 2 , 10 can be connected in parallel with respect to DC voltage similar to FIG. 6 in that the coupling capacitances at the cathodes of the APDs 2 , 10 are replaced in FIG. 7 by conducting connections.
  • the depletion layer capacitances C APD,R and C APD,M of the avalanche photodiodes 2 and 10 are incorporated in an LC series resonant circuit.
  • This circuit comprises inductance L, the capacitance C of the capacitance diode 3 which is connected in series with L, and the capacitances C APD,R and C APD,M of the APDs 2 , 10 which are connected in series with L.
  • the coupling capacitances C K are again short circuits for the HF signals.
  • the low-pass filters 7 and 11 serve to decouple the IF signals of the measurement channel and reference channel. They do not pass the HF signals, i.e., the reception signals of the APDs 2 , 10 , or the local oscillator signal proceeding from the LC oscillator.
  • the amplifier 4 together with the feedback network 5 , brings about a stable operation of the oscillator with constant amplitude.
  • the resonant frequency is also regulated in this example by the PLL-circuit 1 by influencing the blocking voltage at the capacitance diode 3 .
  • the anode of the capacitance diode 3 and the anodes of APDs 2 and 10 are connected to earth, i.e., grounded, with respect to DC voltage via the chokes Dr 1 and Dr 2 .
  • the chokes do not pass the HF signals and, for the low-frequency IF signals, represent a short circuit, so that crosstalk of IF signals from the reference signal branch to the measurement signal branch is also sharply reduced.
  • the coupling capacitances C K have a high impedance for the low-frequency IF signals. If it is not necessary for the gain factors of the APD 10 of the measurement receiver and of the APD 2 of the reference receiver to be adjustable separately by their depletion layer voltages, the cathodes of the two APDs 2 , 10 can also be connected with respect to DC voltage in that the coupling capacitance between them in FIG. 8 is replaced by a conducting connection.
  • variable range of the effective resonator capacitance can be small due to the series connection of the capacitance diode 3 with the depletion layer capacitances of the APDs 2 , 10 because of the small APD capacitance of, e.g., 2 pF, so that the frequency tuning range of the resonant circuit is limited.
  • the amplitude of the APD voltage is dependent on the control voltage of the PLL-circuit 1 due to the capacitive voltage divider and the variable capacitance C of the capacitance diode 3 . Further, care should be taken in the variant in FIG.
  • the local oscillator signal is transmitted asymmetrically to the respective APDs 2 , 10 in case of different depletion layer capacitances C APD,R and C APD,M . Accordingly, specimen scatter and environmental influences are not compensated to the same extent as in the other examples.
  • the voltage amplitudes at the respective APDs 2 , 10 are halved by the series connection of the APD depletion layer capacitances due to the capacitive voltage divider proceeding from the latter.
  • the amplitude of the LO signal should be as high as possible for highly efficient direct mixing.
  • connection combinations for producing an LC resonant circuit For example, a parallel resonant circuit comprising an inductance and the series connection of a capacitance diode with series-connected APD depletion layer capacitances can be used. Also, a parallel resonant circuit comprising an inductance and the series connection of a capacitance diode with parallel-connected APD depletion layer capacitances would also be possible. Depending on the application, the APDs can be connected with respect to high-frequency signals or DC voltage. Naturally, in all embodiment examples, the digital control unit according to FIG. 2 can also be used instead of the PLL-regulated local oscillators, wherein a regulated or unregulated system is possible according to the description of FIG. 2 .
  • An APD is a current source supplying an internally generated photocurrent.
  • the depletion layer capacitance of the APD is connected in parallel with this current source. At high frequencies, the signal is short-circuited by this depletion layer capacitance, so that a low-pass behavior results. Also, the local oscillator signal “sees” this depletion layer capacitance and is likewise short-circuited to ground by it. However, this is only the case when the local oscillator signal is supplied, e.g., via an amplifier.
  • the capacitive influence, and accordingly also the low-pass behavior originating from the respective depletion layer capacitance is eliminated, i.e., tuned out, by the parallel-connected or series-connected inductance L.
  • the depletion layer capacitance is integrated in the resonant circuit and, therefore, can no longer generate any low-pass behavior.
  • the low-pass behavior is now only formed by the drift times of the generated charge carriers within the semiconductor.

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US09/911,272 2000-07-31 2001-07-23 Local oscillator for generating an HF signal for direct mixing by using avalanche photodiodes Expired - Lifetime US6950639B2 (en)

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DE10065353A DE10065353B4 (de) 2000-07-31 2000-12-27 Lokaloszillator zur Erzeugung eines HF-Signals zur Direktmischung mittels Avalanche-Fotodioden

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US20050174199A1 (en) * 2003-12-29 2005-08-11 Stmicroelectronics S.A. Integrable acoustic resonator and method for integrating such resonator
US20050189997A1 (en) * 2003-12-29 2005-09-01 Stmicroelectronics S.A. Integrable amplitude-locked loop including an acoustic resonator
US20050189999A1 (en) * 2003-12-29 2005-09-01 Stmicroelectronics S.A. Integrated electronic circuit comprising a tunable resonator
US20050189998A1 (en) * 2003-12-29 2005-09-01 Stmicroelectronics S.A. Integrable phase-locked loop including an acoustic resonator
US20070030099A1 (en) * 2005-07-01 2007-02-08 Stmicroelectronics S.A. Band pass filtering circuit fitted with acoustic resonators
US20070035365A1 (en) * 2005-03-18 2007-02-15 Stmicroelectronics S.A. Integrable tunable filter circuit comprising a set of baw resonators
US20080024244A1 (en) * 2006-07-28 2008-01-31 Stmicroelectronics Sa Filtering circuit fitted with acoustic resonators
US20090072925A1 (en) * 2007-09-03 2009-03-19 Stmicroelectronics Sa Frequency tuning circuit for lattice filter
US20100235128A1 (en) * 2006-01-13 2010-09-16 Peter Wolf Measuring device, in particular distance measuring device
US10066993B2 (en) 2012-06-19 2018-09-04 Klein Medical Limited Spectroscopic analysis

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US9052065B2 (en) * 2010-12-01 2015-06-09 Gp Strategies Corporation Liquid dispenser
US10128398B1 (en) * 2014-05-23 2018-11-13 Stc.Unm Resonance avalanche photodiodes for dynamic biasing
US10707905B2 (en) * 2015-06-23 2020-07-07 Skyworks Solutions, Inc. Wideband multiplexer for radio-frequency applications
US11181626B2 (en) * 2016-03-14 2021-11-23 Plethron Inc. Injecting an RF local oscillator signal into an avalanche photo diode using photons emitted from a light emitting diode
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Cited By (23)

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US7274274B2 (en) 2003-12-29 2007-09-25 Stmicroelectronics S.A. Integrable acoustic resonator and method for integrating such resonator
US7218181B2 (en) * 2003-12-29 2007-05-15 Stmicroelectronics S.A. Integrable amplitude-locked loop including an acoustic resonator
US20050189997A1 (en) * 2003-12-29 2005-09-01 Stmicroelectronics S.A. Integrable amplitude-locked loop including an acoustic resonator
US20050189999A1 (en) * 2003-12-29 2005-09-01 Stmicroelectronics S.A. Integrated electronic circuit comprising a tunable resonator
US20050189998A1 (en) * 2003-12-29 2005-09-01 Stmicroelectronics S.A. Integrable phase-locked loop including an acoustic resonator
US7423502B2 (en) 2003-12-29 2008-09-09 Stmicroelectronics Sa Electronic circuit comprising a resonator to be integrated into a semiconductor product
US20050174199A1 (en) * 2003-12-29 2005-08-11 Stmicroelectronics S.A. Integrable acoustic resonator and method for integrating such resonator
US7345554B2 (en) * 2003-12-29 2008-03-18 Stmicroelectronics S.A. Integrable phase-locked loop including an acoustic resonator
US7187240B2 (en) 2003-12-29 2007-03-06 Stmicroelectronics S.A. Integrated electronic circuit comprising a tunable resonator
US20050174198A1 (en) * 2003-12-29 2005-08-11 Stmicroelectronics S.A. Electronic circuit comprising a resonator to be integrated into a semiconductor product
US20090174503A1 (en) * 2005-03-18 2009-07-09 Stmicroelectronics Sa Integrable tunable filter circuit comprising a set of baw resonators
US20070035365A1 (en) * 2005-03-18 2007-02-15 Stmicroelectronics S.A. Integrable tunable filter circuit comprising a set of baw resonators
US7825748B2 (en) 2005-03-18 2010-11-02 Stmicroelectronics Sa Integrable tunable filter circuit comprising a set of BAW resonators
US7492242B2 (en) 2005-03-18 2009-02-17 Stmicroelectronics S.A. Integrable tunable filter circuit comprising a set of BAW resonators
US7525400B2 (en) 2005-07-01 2009-04-28 Stmicroelectronics S.A. Band pass filtering circuit fitted with acoustic resonators
US20070030099A1 (en) * 2005-07-01 2007-02-08 Stmicroelectronics S.A. Band pass filtering circuit fitted with acoustic resonators
US20100235128A1 (en) * 2006-01-13 2010-09-16 Peter Wolf Measuring device, in particular distance measuring device
US20080024244A1 (en) * 2006-07-28 2008-01-31 Stmicroelectronics Sa Filtering circuit fitted with acoustic resonators
US7696844B2 (en) 2006-07-28 2010-04-13 Stmicroelectronics Sa Filtering circuit fitted with acoustic resonators
US20090072925A1 (en) * 2007-09-03 2009-03-19 Stmicroelectronics Sa Frequency tuning circuit for lattice filter
US7920036B2 (en) 2007-09-03 2011-04-05 Stmicroelectronics S.A. Frequency tuning circuit for lattice filter
US10066993B2 (en) 2012-06-19 2018-09-04 Klein Medical Limited Spectroscopic analysis
US10352768B2 (en) 2012-06-19 2019-07-16 Klein Medical Limited Spectroscopic analysis

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EP1191697A3 (de) 2002-08-14
US20020032014A1 (en) 2002-03-14
EP1191697B1 (de) 2011-12-21
JP4719383B2 (ja) 2011-07-06
EP1191697A2 (de) 2002-03-27
JP2002124827A (ja) 2002-04-26

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