TWI569565B - Staggered high boost DC converter - Google Patents

Description

Interleaved high boost DC converter

The present invention relates to a boost converter, and more particularly to an interleaved high boost DC converter.

Referring to FIG. 1, a conventional boost converter, when the parasitic resistance RL on the inductor L is not considered, its voltage gain M is as shown in Equation 1, where the parameters V _O , V _in , and D are output voltage and input voltage, respectively. , the responsibility of the switch to conduct ratio.

Formula 1

Considering the influence of the parasitic resistance R _L on the voltage gain M, the voltage gain M is as shown in Equation 2, wherein the parameter R _O is the resistance of the load, and referring to FIG. 2, is a voltage gain pair of a conventional boost converter. The relationship between the conduction ratios.

Formula 2

The conversion efficiency is turned on as shown in Equation 3.

...three

However, the disadvantages of conventional boost converters are:

1. Although increasing the turn-on ratio can increase the voltage gain, the higher the turn-on ratio is, the more likely it is to generate a large input current chopping, which will result in a decrease in the service life of the fuel cell providing the input current, and it can be seen from FIG. When the conduction ratio exceeds 0.9 or more, the voltage gain is not increased and decreased by the influence of the parasitic resistance, and in order to avoid large input current ripple, the voltage conversion ratio is limited to about 5 times or less, and the high voltage gain is not satisfied. Demand.

2. This circuit architecture will have a reverse recovery current generated by the diode D0, which will cause considerable power loss and reduce the conversion power, which is not suitable for high power applications.

Accordingly, it is an object of the present invention to provide an interleaved high step-up DC converter that solves the above problems.

Thus, the interleaved high-boost DC converter of the present invention comprises a first coupled inductor and a second coupled inductor, a first switch, a second switch, a first diode, a second diode, An energy storage component, a third diode, a secondary side capacitor, and a rectified output stage.

Each of the coupled inductors has a primary side winding and a secondary side winding, each side winding having a first end and a second end, wherein the first ends of the primary side windings of the first and second coupled inductors are electrically Connected together to receive a DC input voltage, the second end of the secondary side winding of the first coupled inductor is electrically coupled to the second end of the secondary side winding of the second coupled inductor.

The first switch has a first end electrically connected to the first end of the primary side winding of the first coupled inductor, and a second end, and the first switch is controlled to switch between a conducting state and a non-conducting state, When the first switch is turned on, the primary side winding of the first coupled inductor receives a first current charge to generate a first primary side voltage, and when the first switch is not turned on, the secondary side of the first coupled inductor The winding is magnetically induced to produce a first secondary side voltage proportional to the first primary side voltage.

The second switch has a first end electrically connected to the first end of the primary side winding of the second coupled inductor, and a second end, and the second switch is controlled to switch between the conducting state and the non-conducting state, When the second switch is turned on, the primary side winding of the second coupled inductor receives a second current charge to generate a second primary side voltage, and when the first switch is not turned on, the second coupled inductor secondary The side winding is magnetically induced to generate a second secondary side voltage proportional to the second primary side voltage, wherein a phase difference exists between the first switch and the second switch in an on state.

The first diode has a cathode electrically connected to the second end of the first switch and an anode. The second diode has an anode electrically connected to the second end of the second switch and a cathode electrically connected to the first end of the secondary side winding of the second coupled inductor.

The energy storage component is electrically connected between the first end of the first switch, the anode of the first diode, and the cathode of the second diode. When the first diode is turned on, the energy storage component is used. A stored voltage is generated based on the discharge from the primary side windings of the first and second coupled inductors, the stored voltage being proportional to the sum of the first and second primary side voltages.

The third diode has an anode electrically connected to the cathode of the first diode, and a cathode.

The secondary side capacitor has a first end electrically connected to the first end of the secondary side winding of the first coupled inductor, and a second end of the cathode electrically connected to the third diode, the secondary side capacitor is used A capacitance voltage proportional to the first secondary side voltage is generated in accordance with a discharge from the secondary side winding of the first coupled inductor.

a rectifying output stage electrically connected to the second end of the secondary side capacitor for generating a DC according to a discharge from the secondary side winding of the second coupled inductor, a discharge of the energy storage element, and a discharge of the secondary side capacitor The output voltage is proportional to the sum of the storage voltage, the capacitance voltage, and the second secondary side voltage.

The invention has the advantages of complying with high voltage gain and high power applications at the same time, and reducing component cost.

Before the present invention is described in detail, it should be noted that in the following description, similar elements are denoted by the same reference numerals.

Referring to FIG. 3, an embodiment of the interleaved high-boost DC converter of the present invention includes a first coupled inductor 1 and a second coupled inductor 2, a first switch S1, a second switch S2, and a first two. The pole body D1, a second diode D2, a third diode D3, an energy storage element 3, a secondary side capacitor C3, a rectifying output stage 4 and a control unit 5.

The first and second coupled inductors 1 have a primary side winding Np1 and a secondary side winding Ns1, and the second coupled inductor 2 has a primary side winding Np2 and a secondary side winding Ns2, each side winding Np1, Np2, Ns1 The Ns2 has a first end and a second end, wherein the first ends of the primary side windings Np1, Np2 of the first and second coupled inductors 1, 2 are electrically connected together to receive a DC input voltage Vin, The second end of the secondary side winding Ns1 of the first coupled inductor 1 is electrically connected to the second end of the secondary side winding Ns2 of the second coupled inductor 2. The first end of each of the primary side windings Np1, Np2 is a striking end, and the second end of each of the primary side windings Np1, Np2 is a non-tapping end. The first end of each of the secondary side windings Ns1, Ns2 is a striking end, and the second end of each of the secondary side windings Ns1, Ns2 is a non-tapping end. In this embodiment, each of the coupled inductors 1 and 2 is a magnetic core and has a turns ratio n, wherein the magnetizing inductance and the leakage inductance of the two coupled inductors 1 and 2 in a non-ideal effect analysis will be Further explanation is given later.

The first switch S1 has a first end electrically connected to the first end of the primary side winding Np1 of the first coupled inductor 1, and a second end, and the first switch S1 is controlled to switch to the conducting state and not During the on state, when the first switch S1 is turned on, the primary side winding Np1 of the first coupled inductor 1 receives a first current charge to generate a first primary side voltage, and when the first switch S1 is not turned on, The secondary side winding Ns1 of the first coupled inductor 1 is magnetically induced to generate a first secondary side voltage proportional to the first primary side voltage. The first switch S1 is an N-type power semiconductor transistor, and the first end of the first switch S1 is a drain, and the second end of the first switch S1 is a source.

The second switch S2 has a first end electrically connected to the first end of the primary side winding Np1 of the second coupled inductor 2, and a second end, and the second switch S2 is controlled to switch to the conducting state and not During the on state, when the second switch S2 is turned on, the primary side winding Np2 of the second coupled inductor 2 receives a second current charge to generate a second primary side voltage. When the first switch S1 is not turned on, The secondary side winding Ns2 of the second coupled inductor 2 is magnetically induced to generate a second secondary side voltage proportional to the second primary side voltage. The second switch S2 is an N-type power semiconductor transistor, and the first end of the first switch S1 is a drain, and the second end of the first switch S1 is a source. There is a phase difference between the first switch S1 and the second switch S2 in the on state. And the switching of the first and second switches S1, S2 is zero current switching, which will be further explained below in conjunction with the mode operation of the present invention.

The first diode D1 has a cathode electrically connected to the second end of the first switch S1 and an anode. The second diode D2 has an anode electrically connected to the second end of the second switch S1 and a cathode electrically connected to the first end of the secondary side winding Ns2 of the second coupled inductor 2. The third diode D3 has an anode electrically connected to the cathode of the first diode D1, and a cathode.

The energy storage device 3 is electrically connected between the first end of the first switch S1, the anode of the first diode D1 and the cathode of the second diode D2. When the first diode D1 is turned on, the The energy storage component 3 is configured to generate a storage voltage according to discharges from the primary side windings Np1, Np2 of the first and second coupled inductors 1, 2, the storage voltage being proportional to the first and second primary sides The sum of the voltages. The energy storage component 3 includes a first capacitor C1 and a second capacitor C2. The first capacitor C1 has a first end electrically connected to the first end of the first switch S1, and a second end electrically connected to the anode of the first diode D1. The second capacitor C2 has a first end electrically connected to the first end of the first switch S1, and a second end electrically connected to the cathode of the second diode D2.

The secondary side capacitor C3 has a first end electrically connected to the first end of the secondary side winding Ns1 of the first coupled inductor 1, and a second end electrically connected to the cathode of the third diode D3. The side capacitor C3 is for generating a capacitance voltage proportional to the first secondary side voltage according to the discharge from the secondary side winding Ns1 of the first coupled inductor 1.

The rectified output stage 4 is electrically connected to the second end of the secondary side capacitor C3 for discharging according to the secondary side winding Ns2 from the second coupled inductor 2, discharging of the energy storage element 3, and the secondary side capacitor C3 The discharge produces a DC output voltage Vo that is proportional to the sum of the stored voltage, the capacitor voltage, and the second secondary side voltage. The rectified output stage 4 includes a fourth diode D4 and an output capacitor C4.

The fourth diode D4 has an anode electrically connected to the second end of the secondary side capacitor C3, and a cathode. The output capacitor C4 is electrically connected between the cathode of the fourth diode D4 and the anode of the first diode D1 for providing the output voltage Vo.

The control unit 5 generates a first pulse modulation signal for switching the first switch S1 and a second pulse modulation signal for switching the second switch S2, the first pulse modulation signal and the second pulse wave The modulated signals have the same cycle time, and the phase differences of the first and second pulse modulated signals are one-half of the cycle time. The switching timing diagram of the switches S1, S2 will be further explained below in eight stages.

Referring to FIG. 4, an equivalent circuit diagram of the embodiment is used to illustrate the magnetizing inductances Lm1 and Lm2 and the leakage inductances Lk1 and Lk2 in the non-ideal equivalent circuit of the two coupled inductors. Wherein, the parameters v _D1 , v _D2 , v _D3 , and v _D4 represent the crossover voltages of the first to fourth diodes D1 to D4, respectively, and the parameters V _C1 and V _C2 represent the cross of the first capacitor C1 and the second capacitor C2, respectively. The parameters i _LK1 and i _LK2 represent the leakage inductance current flowing through the two coupled inductors 1, 2, respectively, the parameter i _in represents the input current, and the parameters i _D1 ～ i _D4 represent the flow through the first to fourth diodes D1, respectively. ~D4 current, parameter i _O represents total output current

Referring to FIG. 5, it is an operation timing diagram of the embodiment, wherein the parameters v _gs1 and v _gs2 respectively represent voltages of the first and second pulse modulation signals that control whether the first and second switches S1 and S2 are turned on. The parameters v _ds1 and v _ds2 respectively represent the two-terminal voltage across the first and second switches S1 and S2, and the parameter T _S is the cycle time of the first pulse modulation signal.

The following is a circuit diagram of the eight-stage operation of the present embodiment, in which the conductive elements are indicated by solid lines, and the non-conducting elements are indicated by broken lines, which are described below for each stage.

The first stage (time:):

Referring to FIG. 5 and FIG. 6, the first switch S1 is turned from non-conducting to conducting, and the second switch S2 is not turned on, the first diode D1 is not turned on, the second diode D2 is not turned on, and the third diode D3 is turned on. When turned on, the fourth diode D4 is not turned on.

The first phase begins when the first switch S1 is switched to be on, and the presence of its leakage inductance Lk1 provides a current that rises linearly from zero, while the first switch S1 has a soft switching performance with zero current switching, reducing switching losses. As the leakage inductance current i _LK1 rises, when Magnetizing inductance of the first coupled inductor 1 The stored energy is still transmitted to the secondary side group Ns1 by the primary side winding Np1, so only the third diode D3 remains in the conducting state, and the first diode D1, the second diode D2 and the first The quadrupole D4 is reverse biased and not turned on, due to the leakage inductance of the first and second coupled inductors 1, 2 with The falling rate of the current of the third diode D3 is controlled, and the reverse current of the third diode D3 is prevented from being switched rapidly, so as to alleviate the reverse recovery problem of the third diode D3. When, the current of the third diode D3 When the voltage drops to 0 and the third diode D3 turns into non-conduction, the first phase ends.

second stage( ):

Referring to FIG. 5 and FIG. 7, the first switch S1 is turned on, and the second switch S2 is turned on, the first diode D1 is not turned on, the second diode D2 is not turned on, and the third diode D3 is turned on to be non-conductive. The fourth diode D4 is not conductive.

The second phase begins with the third diode turning into a non-conducting state, all of which are reverse biased and not conducting. The input voltage Vin crosses the primary side of the two coupled inductors 1, 2, that is, across its magnetizing inductances Lm1, Lm2 and the leakage inductances Lk1, Lk2, so that the leakage inductor current , Linear rise, slope is From an energy point of view, the input voltage Vin charges the primary side windings Np1, Np2 of the first and second coupled inductors 1, 2, that is, the primary side windings Np1 of the first and second coupled inductors 1, 2 Np2 stores energy in the second stage. When the second switch S2 is switched to be non-conducting, the second phase ends.

The third stage( ):

Referring to FIG. 5 and FIG. 8, the first switch S1 is turned on, and the second switch S2 is turned into non-conducting, the first diode D1 is not turned on, and the second diode D2 is turned from non-conducting to conducting, the third two The polar body D3 is not turned on, and the fourth diode D4 is turned from non-conducting to conducting.

The third phase begins with the second switch S2 switched to non-conducting, and the leakage inductance current of the second coupled inductor 2 Continuity of the second diode D2 is turned on, leakage inductance current The second capacitor C2 is charged by flowing through the second diode D2, the second capacitor C2, and the first switch S1. Magnetizing inductance of the primary side winding Np2 of the second coupled inductor 2 Transmitting energy to the secondary side winding Ns2 causes the fourth diode D4 to be turned on, and the fourth diode current Charging the fourth capacitor C4, the first switch S1 remains conductive, and the leakage current is current It decreases linearly. When the energy stored in the leakage inductance Lk2 is completely released, When the second diode D2 is turned into non-conduction, the third phase ends. Since the current i _D2 flowing through the second diode _D2 first drops to 0, the second diode D2 is turned into a non-conducting state, so the second diode D2 has no problem of reverse recovery loss.

Fourth stage ):

Referring to FIG. 5 and FIG. 9, the first switch S1 is turned on, and the second switch S2 is not turned on, the first diode D1 is not turned on, the second diode D2 is turned on to be non-conductive, and the third diode D3 is not. Turned on, the fourth diode D4 is turned on.

The fourth phase begins when the energy of the leakage inductance is completely released to the second capacitor C2, and the second diode D2 is turned into a non-conduction. Magnetizing inductor current of the second coupled inductor 2 Fully reflected by the primary side winding Np2 to the secondary side winding group Ns2, thus , the fourth diode current Charging the fourth capacitor C4, the current flowing through the first switch S1 is equal to the sum of the currents of the magnetizing inductances Lm1, Lm2 of the first and second coupled inductors 1, 2, that is, . When the second switch S2 is switched to be on, the fourth phase ends.

The fifth stage ( ):

Referring to FIG. 5 and FIG. 10, the first switch S1 is turned on, and the second switch S2 is turned from non-conducting to conducting, the first diode D1 is not turned on, the second diode D2 is not turned on, and the third diode D3 is not. Turned on, the fourth diode D4 is turned on.

The fifth phase begins with the second switch S2 switched to be turned on. Due to the presence of the leakage inductance Lk2 of the second coupled inductor 2, the second switch S2 has a soft switching performance of zero current switching, reducing switching loss. Leakage inductor current Rise, when The energy storage of the magnetizing inductance Lm2 of the second coupled inductor 2 is still magnetically induced to transmit the secondary side winding Ns2 by the primary side winding Np2, and the fourth diode D4 remains in the fourth stage of conduction state, flowing through the fourth two Polar body D4 current Falling, the first diode D1, the second diode D2, and the third diode D3 are reverse biased and are not turned on. The leakage inductances Lk1, Lk2 of the first and second coupled inductors 1, 2 control the fourth diode current The rate of descent can therefore alleviate the problem of reverse recovery of the fourth diode D4. When, the fourth diode current Down to 0, the fourth diode D4 turns into a non-conducting state, and the fifth phase ends.

The sixth stage ( ):

Referring to FIG. 5 and FIG. 11, the first switch S1 is turned on, and the second switch S2 is turned on, the first diode D1 is not turned on, the second diode D2 is not turned on, and the third diode D3 is not turned on, the fourth two The pole body D4 is turned from conduction to non-conduction.

The sixth phase begins with the fourth diode D4 transitioning to non-conducting, all the diodes are reverse biased and not conducting, and the first switch S1 and the second switch S2 are both turned on. The input voltage Vin spans the primary side windings Np1, Np2 of the first and second coupled inductors 1, 2, that is, across its magnetizing inductances Lm1, Lm2 and the leakage inductances Lk1, Lk2, and the leakage inductance current , Linear rise, slope is From the energy point of view, the primary side windings Np1, Np2 of the first and second coupled inductors 1, 2 store energy in the sixth stage. When the first switch S1 is switched to be non-conducting, the sixth phase ends.

The seventh stage ( ):

Referring to FIG. 5 and FIG. 12, the first switch S1 is turned on and turned off, and the second switch S2 is turned on, the first diode D1 is not turned on, the second diode D2 is not turned on, and the third diode D3 is turned on. The fourth diode D4 is not conductive.

The seventh phase begins with the first switch S1 switching to non-conducting. Leakage inductor current of the first coupled inductor 1 Continuity causes the first diode D1 to transition to a conduction leakage current Flowing through the first capacitor C1 and the first diode D1, charging the first capacitor C1, the magnetizing inductance Lm1 of the first coupling inductor 1 transfers energy from the primary side winding Np1 to the secondary side winding Ns1 such that the third diode D3 transition is conductive, third diode current Charging the third capacitor C3, the second switch S2 is kept conductive, and the leakage current is current It decreases linearly. When the energy stored in the leakage inductance Lk1 is completely released, When the first diode D1 transitions to non-conducting, the seventh phase ends. Since the current i _D1 flowing through the first diode _D1 first drops to zero, the first diode D1 is turned into non-conduction, so the first diode D1 has no problem of reverse recovery loss.

The eighth stage ( ):

Referring to FIG. 5 and FIG. 13 , the first switch S1 is not turned on, and the second switch S2 is turned on, the first diode D1 is turned on and turned off, the second diode D2 is not turned on, and the third diode D3 is turned on. The fourth diode D4 is not conductive.

The eighth phase begins when the energy of the leakage inductance is completely released to the first capacitor C1, and the first diode D1 is turned into a non-conducting state. The magnetizing inductance Lm1 of the first coupled inductor 1 is completely reflected by its primary side winding Np1 to the secondary side winding Ns1, Therefore, the third diode current Charging the third capacitor C3, the current flowing through the second switch S2 is equal to the sum of the currents of the magnetizing inductors Lm1, Lm2, that is, . When the first switch S1 is switched to be on, the eighth phase ends and the next switching cycle is entered.

Voltage gain analysis:

Since the voltages of the first and second capacitors C1 and C2 can be regarded as the output voltage of the conventional boost converter, the magnetization inductances Lm1 and Lm2 of the first and second coupled inductors 1 and 2 satisfy the volt-second equilibrium theorem, and can be derived. Obtaining first and second capacitor voltages , As in the fourth.

...four

Third and fourth capacitor voltages , It can be obtained by deriving the primary side voltage of the first and second coupled inductors 1, 2 to the secondary measured voltage. In the seventh stage, the first switch S2 is not turned on, the second switch S2 is turned on, and the third diode D3 is turned on, and the third capacitor voltage can be derived. As in the fifth.

Five

The parameters v _Ns1 and v _Ns2 are the first secondary side voltage on the secondary side winding Ns1 of the first coupled inductor 1 and the second secondary side voltage on the secondary side winding Ns2 of the second coupled inductor 2, respectively. The parameter n is the turns ratio of the secondary side windings Np1, Np2 and the primary side windings Ns1, Ns2, the parameter k is the coupling coefficient of the first and second coupled inductors 1, 2, the parameter Vin is the input voltage, and the parameter D is the first And the conduction ratio of the second switches S1, S2.

In the third stage, the first switch S1 is turned on, the second switch S2 is not turned on, the fourth diode D4 is turned on, and the fourth capacitor voltage is turned on. As in the sixth.

Equation six

Output voltage Equivalent to the fourth capacitor voltage As in the seventh.

Equation seven

Therefore, the voltage gain of the embodiment can be derived. As in the eighth.

Equation eight

See Figure 14 for different coupling factors (k =1, 0.95, 0.9) and voltage gain Relationship graph, when When the figure shows the coupling coefficient The effect on voltage gain is very small. Coupling coefficient , the voltage gain is as shown in Equation 9.

Formula nine

It can be seen from the above equation that the voltage gain of the present embodiment has two design degrees of freedom of the coupled inductor turns ratio and the turn-on ratio, and the high boost ratio can be achieved by appropriately designing the turns ratio of the coupled inductor, and does not have to be operated in an extremely large conduction. ratio. Referring to FIG. 15 , a voltage gain graph corresponding to the coupled inductor turns ratio and the turn-on ratio is shown. FIG. 15 shows the turn-on ratio. , When the voltage gain is 10 times; , The voltage gain is 20 times.

It can be seen from the seventh stage and the third stage of the embodiment that the voltage stress of the first switch S1 and the second switch S2 is as shown in Equation 10 and Equation 11, the first diode D1, the second diode D2, and the third diode. The voltage stress of the body D3 is as shown in Equation 12, Equation 13, and Equation 14.

Ten

Equation eleven

Twelve

Thirteen

Fourteen

Since the power switching voltage stress of the conventional interleaved boost converter is the output voltage The switching voltage stress of this embodiment is only the output voltage. of Therefore, the first and second switches S1 and S2 can be realized by using a metal oxide half-effective transistor having a low rated withstand voltage and a low on-resistance, which can reduce the switch conduction loss. The first to third diodes D1 to D3 with lower voltage stress can use Schottky diodes, because the typical forward voltage drop of the Schottky diode is 0.3 V, which is higher than the general power diode. The conduction voltage is reduced, and the conduction loss is further reduced.

Experimental simulation:

Regarding the steady-state characteristic analysis of the present embodiment, it can be assumed that the first and second switches S1 and S2, the first to third diodes D1 to D3 have a conduction voltage drop of zero and a transient phase with a very short time, including the first In the first, fourth, fifth and eighth stages, only the second, third, sixth and seventh stages are considered. The first to fourth capacitors are large enough to ignore voltage chopping so that the capacitor voltage is considered constant during a switching period. Refer to Figure 16 for the drive signals of the switches S1 and S2, the input voltage Vin and the output voltage Vo waveform. , , , the theoretical value conduction ratio is approximately The simulation result conforms to the voltage gain of Equation 9.

As shown in FIG. 17, which is a diagram of the switch drive signal and the switch cross-voltage signal, it is verified that when the first switch S1 or the second switch S2 is not conducting, the voltage across the switch is verified. or All about , only the output voltage Vo= The first and second switches S1, S2 of the present embodiment have a lower value than the analysis result of Equation 10 and Equation 11, compared with the switching voltage stress of the conventional boost converter. The advantage of voltage stress.

As shown in FIG. 18, the leakage inductance currents of the first and second coupled inductors 1, 2 , And input current Waveform diagram, as can be seen from the figure , The chopping current is about the same size And input current The chopping current is only about the same size It was verified that the interleaved operation of the present embodiment has a reduced chopping current utility.

Figure 19 shows the magnetizing inductor current waveforms of the first and second coupled inductors 1, 2, verifying that the present embodiment operates in a continuous conduction mode (CCM).

20 is a waveform diagram of current and voltage of the first and second diodes D1 and D2, as shown in the figure. with There is no reverse recovery problem, so there is no reverse recovery loss. And the first diode D1 voltage stress is , only one quarter of the output voltage, the voltage stress of the second diode D2 is approximately Only one-half of the output voltage Vo is verified, and this embodiment is verified to conform to the analysis results of Equations 12 and 13.

21 is a current and voltage waveform diagram of the third and fourth diodes D3 and D4. It can be seen from the figure that the voltage stresses of the third and fourth diodes D3 and D4 are both 200V, and it is verified that the present embodiment conforms to the fourteenth embodiment. Analysis results. The reverse recovery current of the currents of the third and fourth diodes D3 and D4 is small because the leakage inductances of the first and second coupled inductors 1, 2 with The existence eases the reverse recovery problem.

As shown in FIG. 22, the first and second switches S1 and S2 have zero current switching waveform diagrams. It can be seen that the first and second switches S1 and S2 are first reduced to zero across the voltage, and current flows through the switch. Therefore, the soft-cutting effect of the zero current switching (ZCS) is achieved, the switching loss can be reduced, and the power switching zero current switching function of the present embodiment is verified.

In summary, the above embodiment has the following advantages:

1. With higher voltage gain, it can be known from Equation 9 that the first and second capacitors C1 and C2 are connected in series, and the third side capacitors C3 are connected in series with the secondary side windings Ns1 and Ns2 of the first and second coupled inductors 1, 2 The circuit architecture, and more adjustable the first and second coupled inductors 1, 2 turns ratio amplification voltage gain, meets the requirements of high voltage gain without the need to operate at a very large conduction ratio.

2. With low voltage stress, since the voltage stress of the first and second switches S1, S2 is only one quarter of the output voltage Vo, a low-rated piezoelectric crystal with a small on-resistance can be used, and Equation 14 shows that the first to fourth diodes D1 to D4 also have low voltage stress, and therefore, the conduction loss can be reduced.

3. With zero current switching efficiency, the first and second switches S1, S2 have the soft cutting performance of zero current switching, and can reduce the switching loss.

4. There is no reverse current power consumption. Before the first and second diodes D1 and D2 are turned into non-conducting, the current flowing through them is first reduced to zero, so the first and second diodes D1 and D2 are not. Reverse recovery power loss problem.

5. The problem of voltage surge is avoided. When the first and second switches S1 and S2 are switched to be non-conducting, the leakage inductance energy of the first and second coupled inductors 1, 2 can be transmitted from the primary side to the secondary side. To avoid the problem of voltage surge.

6. High conversion efficiency, due to lowering the conduction losses of the first and second switches S1, S2, the first to fourth diodes D1 to D4, and the zero current switching effect, the first to fourth diodes D1 to D4 No reverse current consumption and avoiding voltage surges, which can effectively improve conversion efficiency.

7. Decrease the input current chopping because the first ends of the primary side windings Np1, Np2 of the first and second coupled inductors 1, 2 are connected in parallel to an input voltage source (such as a solar cell module output) that provides the input current i _in End, fuel cell), and the first and second switches S1, S2 are separated by an interval of 1/2 switching period, so that the current flows through the primary side windings Np1, Np2 of the first and second coupled inductors 1, 2, respectively The chopping wave can cancel, reduce the chopping size of the input current i _in , and can reduce the number of electrolytic capacitors at the output end of the solar cell module and prolong the service life of the fuel cell, thereby reducing the overall cost of the system.

8. The number of magnetic cores in this case is only two. It is not only easy to design the first and second coupling inductors 1, 2, but also reduce the component cost, and the number of other components is also small. For example, the number of switches is only two, and the number of the diodes is only two. The number is only four and the number of capacitors is only four. Therefore, the object of the present invention can be achieved.

However, the above is only the embodiment of the present invention, and the scope of the invention is not limited thereto, and all the equivalent equivalent changes and modifications according to the scope of the patent application and the patent specification of the present invention are still The scope of the invention is covered.

1‧‧‧First coupled inductor

Np1‧‧‧ primary side winding

Ns1‧‧‧ secondary side winding

2‧‧‧Second coupled inductor

Np2‧‧‧ primary side winding

Ns2‧‧‧ secondary side winding

S1‧‧‧ first switch

S2‧‧‧ second switch

D1‧‧‧First Diode

D2‧‧‧ second diode

D3‧‧‧ third diode

3‧‧‧ Energy storage components

C1‧‧‧first capacitor

C2‧‧‧second capacitor

C3‧‧‧Secondary side capacitor

4‧‧‧Rectified output stage

D4‧‧‧ fourth diode

C4‧‧‧fourth capacitor

5‧‧‧Control unit

Lm1‧‧‧ Magnetized Inductance

Lm2‧‧‧ Magnetized Inductance

Lk1‧‧‧ leakage inductance

Lk2‧‧‧ leakage inductance

Vin‧‧‧Input voltage

Vo‧‧‧ output voltage

i _in ‧‧‧Input current

i _{Lk1 ‧‧‧Leakage} inductor current

i _{Lk2 ‧‧‧Draining} inductor current

i _D1 ‧‧‧current flowing through the first diode

i _D2 ‧‧‧current flowing through the second diode

i _D3 ‧‧‧current flowing through the third diode

i _D4 ‧‧‧current flowing through the fourth diode

v _D1 ‧‧‧The cross-pressure of the first diode

v _D2 ‧‧‧Secondary voltage of the second diode

v _D3 ‧‧‧Transverse pressure of the third diode

v _D4 ‧‧‧Transverse pressure of the fourth diode

v _C1 ‧‧‧cross pressure of the first capacitor

v _C2 ‧‧‧cross-voltage of the second capacitor

v _C3 ‧‧‧cross-voltage of the third capacitor

v _C4 ‧‧‧cross voltage of the fourth capacitor

Other features and effects of the present invention will be apparent from the following description of the drawings, wherein: Figure 1 is a circuit diagram of a conventional boost converter; Figure 2 is a conventional boost converter voltage Figure 3 is a circuit diagram of an embodiment of the interleaved high-boost DC converter of the present invention; Figure 4 is an equivalent circuit diagram of the embodiment; Figure 5 is an equivalent circuit diagram of the embodiment Figure 6 is a circuit diagram of the operation of the first stage of the embodiment; Figure 7 is a circuit diagram of the operation of the second stage of the embodiment; Figure 8 is a circuit diagram of the operation of the embodiment in the third stage; Figure 9 is a circuit diagram of the operation of the embodiment in the fourth stage; Figure 10 is a circuit diagram of the operation of the embodiment in the fifth stage; Figure 11 is a circuit diagram of the operation of the embodiment in the sixth stage; Figure 12 is a circuit diagram of the embodiment For example, a circuit diagram of the seventh stage is operated; FIG. 13 is a circuit diagram of the eighth stage of operation of the embodiment; FIG. 14 is a relationship diagram of different coupling coefficients and voltage gain; FIG. 15 is a coupled inductor turns ratio and conduction. Than one Gain graph; Figure 16 is a waveform diagram of the drive signal, input voltage and output voltage of the switch; Figure 17 is a waveform diagram of the switch drive signal and the switch across the voltage signal; Figure 18 is the leakage of the first and second coupled inductors FIG. 19 is a waveform diagram of magnetizing inductor currents of first and second coupled inductors; FIG. 20 is a waveform diagram of current and voltage of first and second diodes; 21 is a waveform diagram of current and voltage of the third and fourth diodes; and FIG. 22 is a waveform diagram of zero current switching of the first and second switches.

1‧‧‧First coupled inductor

Np1‧‧‧ primary side winding

Ns1‧‧‧ secondary side winding

2‧‧‧Second coupled inductor

Np2‧‧‧ primary side winding

Ns2‧‧‧ secondary side winding

S1‧‧‧ first switch

S2‧‧‧ second switch

D1‧‧‧First Diode

D2‧‧‧ second diode

D3‧‧‧ third diode

3‧‧‧ Energy storage components

C1‧‧‧first capacitor

C2‧‧‧second capacitor

C3‧‧‧Secondary side capacitor

4‧‧‧Rectified output stage

D4‧‧‧ fourth diode

C4‧‧‧fourth capacitor

5‧‧‧Control unit

Vin‧‧‧Input voltage

Vo‧‧‧ output voltage

Claims (10)

An interleaved high-boost DC converter comprising: a first coupled inductor and a second coupled inductor, each coupled inductor having a primary side winding and a secondary side winding, each side winding having a first end and a second end, wherein the first ends of the primary side windings of the first and second coupled inductors are electrically connected together to receive a DC input voltage, and the second end of the secondary side winding of the first coupled inductor is electrically a second end of the secondary side winding of the second coupled inductor; a first switch having a first end electrically coupled to the first end of the primary side winding of the first coupled inductor, and a second end, And the first switch is controlled to switch between the conductive state and the non-conductive state. When the first switch is turned on, the primary side winding of the first coupled inductor receives a first current charge to generate a first primary side voltage. When the first switch is not conducting, the secondary side winding of the first coupled inductor is magnetically induced to generate a first secondary side voltage proportional to the first primary side voltage; a second switch having an electrical connection a first end of the first end of the primary side winding of the second coupled inductor, and a second end, and the second switch is controlled to switch between a conducting state and a non-conducting state, when the second switch is turned on, The primary side winding of the second coupled inductor receives a second current charge to generate a second primary side voltage. When the first switch is non-conductive, the secondary side winding of the second coupled inductor is magnetically induced to produce a proportional a second secondary side voltage of the second primary side voltage, wherein a phase difference exists between a time when the first switch and the second switch are in an on state; a first diode having an electrical connection to the first switch a cathode of the second end and an anode; a second diode having an anode electrically connected to the second end of the second switch and a cathode electrically connected to the first end of the secondary side winding of the second coupled inductor; An energy storage component electrically connected between the first end of the first switch, the anode of the first diode, and the cathode of the second diode, when the first diode is turned on, the energy storage component Used to derive from the first and the first Discharging the primary side winding of the inductor to generate a stored voltage that is proportional to the sum of the first and second primary side voltages; a third diode having an electrical connection to the first two An anode of the cathode of the polar body, and a cathode; a secondary side capacitor having a first end electrically connected to the first end of the secondary side winding of the first coupled inductor, and an electrical connection to the third diode a second end of the cathode, the secondary side capacitor is configured to generate a capacitor voltage proportional to the first secondary side voltage according to the discharge from the secondary side winding of the first coupled inductor; and a rectified output stage, Connecting a second end of the secondary side capacitor for generating a DC output voltage according to a discharge from the secondary side winding of the second coupled inductor, a discharge of the energy storage element, and a discharge of the secondary side capacitor, The output voltage is proportional to the sum of the storage voltage, the capacitance voltage, and the second secondary side voltage. The interleaved high-boost DC converter of claim 1, wherein the rectified output stage comprises: a fourth diode having an anode electrically connected to the second end of the secondary side capacitor, and a cathode; And an output capacitor electrically connected between the cathode of the fourth diode and the anode of the first diode to provide the output voltage. The interleaved high-boost DC converter of claim 1, wherein the energy storage component comprises: a first capacitor having a first end electrically connected to the first end of the first switch, and an electrical connection a second end of the anode of the first diode; and a second capacitor having a first end electrically connected to the first end of the first switch, and a second electrically connected to the cathode of the second diode Two ends. The interleaved high-boost DC converter of claim 1, wherein the first end of each primary side winding is a striking end and the second end of each primary side winding is a non-tapping end. The interleaved high-boost DC converter of claim 1, wherein the first end of each secondary side winding is a striking end and the second end of each secondary side winding is a non-tapping end. The interleaved high-boost DC converter of claim 1, wherein the first switch is an N-type power semiconductor transistor, and the first end of the first switch is a drain, and the first switch The second end is the source. The interleaved high-boost DC converter of claim 1, wherein the second switch is an N-type power semiconductor transistor, and the first end of the first switch is a drain, and the first switch The second end is the source. The interleaved high-boost DC converter of claim 1, wherein the switching of the first and second switches is a zero current switching. The interleaved high-boost DC converter according to claim 1, further comprising a control unit, wherein the control unit generates a first pulse modulation signal for switching the first switch and a second switch for switching the second switch a pulse wave modulation signal, the first pulse wave modulation signal having the same cycle time as the second pulse wave modulation signal. The interleaved high-boost DC converter of claim 9, wherein the phase difference of the first and second pulse-modulated signals is one-half of a cycle time.