RU2450418C1 - Broadband frequency synthesiser - Google Patents

Abstract

FIELD: radio engineering.

SUBSTANCE: device comprises a frequency-tuned signal source, a generator of control commands and serially connected units of frequency tuning range expansion, every of which comprises a divider with a switched division ratio, a mixer and a band-pass filter.

EFFECT: increased width of frequency tuning range with provision of high purity of output signal spectrum and high speed of frequency tuning.

4 cl, 10 dwg

Description

The invention relates to radio engineering and can be used in radar, in measuring equipment, in communication technology, namely in devices where fast frequency tuning over a wide range and high purity of the output signal spectrum are required.

Known frequency synthesizers [Appendix No. 1. Shapiro D.N., Pain A.A. Fundamentals of the theory of frequency synthesis. Radio and Communications, 1981, pp. 102-111] based on a reference generator that generates several values of fixed reference frequencies, a device for switching signals of reference frequencies and series-connected blocks. Each of the blocks contains a mixer, a bandpass filter (PF) and a frequency divider with a fixed division ratio. Frequency tuning is achieved by independent switching of the frequencies of the reference signals in each block.

The first drawback of these synthesizers is a narrow output frequency tuning band, since in order to obtain a low level of discrete components, it is necessary to ensure such a ratio between the converted frequencies of the signals received at the input of the local oscillator and the input of the mixer IF to prevent spurious spectral components of the mixer output signal from falling into the PF passband . In practice, this ratio should be at least 10. The second drawback is the need for a large number of blocks to obtain a small frequency tuning step, which makes the design of the synthesizer quite complicated.

Known frequency synthesizer (copyright certificate SU No. 1695488), containing a reference oscillator, blocks of the formation of the frequency grid and the shaper of the control signals. Each block contains a mixer, PF and a divider with a switchable division ratio (DPKD). The division ratio is switched using a shaper of control signals. In each block, the signal from the mixer output of the previous block is input to the DPKD input, the reference frequency signal is input to the mixer local oscillator input, and the signal from the DPKD output is input to the intermediate frequency (IF) input. For the first block, the reference frequency signal is supplied simultaneously to the DPKD and the input of the mixer local oscillator. Frequency tuning is achieved by independently switching the division coefficients of the DPKD in each block. The number of synthesized frequencies is determined by the number of combinations of the values of the division coefficients of the DPKD D _i and increases exponentially with an increase in the number of frequency grid blocks. This device provides suppression of the combination components that occur during frequency conversion in all blocks except the last due to the subsequent division of the frequency.

, где f_{вых max} - максимально возможное значение частоты выходного сигнала, D_min - минимальное значение коэффициента деления ДПКД в последнем блоке. С учетом необходимости обеспечения низкого уровня комбинационных составляющих, возникающих за счет нелинейности смесителя и попадающих в полосу полезного сигнала, соотношение между частотами сигналов, поступающих на вход гетеродина и вход ПЧ, должно быть не менее 10. В результате ширина полосы перестройки частоты выходного сигнала составит несколько процентов от частоты опорного сигнала.The disadvantage of this device in the case of a low level of spurious components and a small step of frequency tuning is the narrow frequency tuning range. The larger the minimum value of the division coefficient of the DPKD D _{i, min} , the lower the level of spurious spectral components in the output signal of the synthesizer will be, but, at the same time, the smaller will be the tuning bandwidth at the output frequency, theoretically no more

where f _{o max} - the maximum possible value of the frequency of the output signal, D _min - the minimum value of the division coefficient of the DPKD in the last block. Taking into account the need to ensure a low level of combinational components that arise due to the nonlinearity of the mixer and fall into the useful signal band, the ratio between the frequencies of the signals fed to the local oscillator input and the IF input should be at least 10. As a result, the frequency tuning bandwidth of the output signal will be several percent of the frequency of the reference signal.

A frequency synthesizer is also known (US patent No. 5495202), which consists of a reference generator, a digital direct synthesis synthesizer (DSPC), a frequency conversion and division unit, and a frequency conversion unit up. The frequency conversion and division unit consists of a mixer, a band-pass filter, a divider with a fixed division coefficient and a low-pass filter. The up-conversion unit consists of a mixer and a band-pass filter. By converting the frequency of the DSPC signal into a high frequency range and then dividing the frequency, a reduction in the level of spurious discrete components in the spectrum of the output signal is achieved to the desired level while maintaining a high frequency tuning rate. The frequency conversion unit up transfers the output frequency of the conversion and frequency division unit to the required frequency range.

The disadvantage of this synthesizer is the narrow frequency tuning range, since the resulting output frequency tuning band is D times smaller than the DSP frequency tuning range, where D is the divisor division factor.

The closest analogue to the claimed device is a synthesizer (US patent No. 4791377), which consists of four reference signal shapers based on phase-locked oscillators (PLL), a digital signal converter, a command shaper, frequency conversion and division blocks, and a frequency conversion block up . Each frequency conversion and division unit consists of a mixer, a band-pass filter and a divider with a fixed division coefficient. The up-conversion unit consists of a mixer and a band-pass filter. This synthesizer provides a high speed tuning frequency. Another advantage is the wide band of tuning the frequency of the output signal: the ratio of the maximum value of the frequency of the output signal to the minimum is 1.453.

However, while maintaining a wide frequency tuning range, this device does not allow to obtain a high level of suppression of spurious spectral components due to the small value of the division coefficient of the frequency dividers and the structure of the conversion and frequency division blocks, in which the signals with non-multiple frequencies are fed to the inputs of the local oscillator and the inverter of the mixers. At the same time, the signal frequency at the input of the inverter of the mixers is only 3 times less than the frequency at the input of the mixer local oscillator, which leads to the combination of low-order combination components with a high level in the passband of the bandpass filter at the output of the mixer.

The task of the invention is to increase the width of the frequency tuning range of the synthesizer while ensuring high purity of the output signal spectrum and high frequency tuning speed.

The essence of the claimed invention lies in the fact that in a broadband frequency synthesizer containing a frequency-tunable signal source, a command shaper and blocks for expanding the frequency tuning range, the input of the first block being connected to the output of the signal source and each of the blocks containing a mixer, a bandpass filter, and a divider ; in each unit for expanding the frequency tuning range, the divider is made with a switchable division factor and has a high-frequency input and a control input, while in each block the high-frequency input of the divider and the input of the mixer local oscillator are connected to the unit input, the divider control input is connected to the output of the control command generator, the divider output connected to the input of the intermediate frequency of the mixer, the output of the latter is connected through a band-pass filter to the output of the block, and the blocks are connected so that the input of each subsequent block with SINGLE output directly from the previous one.

In addition, a broadband frequency synthesizer is claimed in which an additional frequency tuning range expansion unit is additionally connected to the output of the last frequency tuning range extension unit by means of two switches that can be controlled, the first of which has one input connected to the output of the last unit and two outputs, the first of which is connected to the first input of the second switch, which has two inputs and one output, the second to the input of the added unit, the output of which is connected to the second input cerned switch.

In addition, a broadband frequency synthesizer is claimed, which additionally includes a divider with a switchable division coefficient, a high-frequency input of which is connected to the output of the second switch, and a control input - with the output of the control command generator.

In addition, a broadband frequency synthesizer is claimed in which the frequency-tunable signal source consists of a reference signal generator, a local oscillator signal conditioning unit comprising a mixer, a switchable dividing factor divider, a low-pass filter and a bandpass filter, and a digital meshing unit containing a digital direct synthesis synthesizer, filter, mixer, band-pass filter, switchable dividing factor divider and low-pass filter, and up-frequency converting unit, containing a mixer and a band-pass filter, while in the local oscillator signal generating unit, the input of which is connected to the reference frequency driver, the mixer divider and the local oscillator input are connected to the input of the local oscillator signal generator, the output of the divider through the low-pass filter is connected to the intermediate frequency input of the mixer, the output the latter is connected to a band-pass filter, the output of which is the output of the local oscillator signal generating unit, connected to the frequency grid generating unit, in which the digital input a direct synthesis synthesizer is connected to a reference frequency driver, the output is through a filter with an intermediate frequency input of the mixer, the local oscillator input is connected to the output of the local oscillator signal generation unit, the mixer output is connected through a bandpass filter with a divider, the output of which is connected through the low-pass filter to the output of the node forming a frequency grid connected to the frequency conversion unit up, in which the input of the intermediate frequency of the mixer is connected to the output of the node forming the frequency grid, input heterodyne on the mixer it is connected to the driver of the reference frequency signals, the output of the mixer is connected through a band-pass filter to the output of the frequency conversion unit up, which is the output of the frequency-tunable signal source.

The technical result of the claimed invention is to solve the problem by using the proposed connection of the units for expanding the range of the frequency adjustment based on the divider with a switchable division ratio. The serial connection of the blocks of the claimed structure allows you to get a wide frequency tuning band at the output of the synthesizer - about 35% (the ratio of the maximum value of the frequency of the output signal to the minimum is 1.35). The inventive synthesizer provides a frequency tuning step less accurate than the frequency setting and a continuous frequency tuning band, which allows you to get any frequency of the output signal with the required accuracy. The inventive design of the blocks, in which the input of the local oscillator and the input of the inverter of the mixer receives signals with multiple frequencies, allows you to suppress all spurious components in the output signal of the mixer to the desired level using a band-pass filter, ensuring high purity of the spectrum of the output signal. In addition, the claimed design of the blocks does not reduce the frequency switching speed, i.e. the speed of tuning the frequency of the output signal remains equal to the speed of tuning the frequency of the signal source due to the use in the design of blocks only broadband filters.

In addition, the use at the output of the cascade of blocks of an additional block for expanding the frequency tuning range connected through two switches allows us to further expand the frequency tuning range of the output signal and to obtain the total synthesized frequency bandwidth equal to an octave at the output.

The use of an additional divider connected to the output of the second switch with a switchable division factor provides an additional extension of the frequency tuning range of the output signal and allows any frequency value within a few octaves to be obtained at the output of the inventive device.

The use of a frequency-tunable source with a step less than the required accuracy of setting the output frequency of the synthesizer allows us to solve the problem of uneven step of the frequency grid that occurs when converting frequencies in cascades of the frequency converter and to provide the possibility of obtaining any frequency from the tuning range. The proposed scheme tunable in frequency of the source allows you to get the frequency tuning range of the source is 5-10 times larger than in the device according to US patent No. 5495202, with the same level of suppression of spurious spectral components of the DSPC and maintaining a high frequency tuning speed.

The invention is illustrated using Fig.1-5, which depict:

figure 1 is a basic diagram of a broadband frequency synthesizer;

figure 2 - diagram of a broadband frequency synthesizer with an additional unit

extend the frequency tuning range (BRDP) connected with

using two switches, and an additional divider with

switchable division factor (DPKD);

figure 3 is a diagram of the tuning of the frequency of the output signal BRDP;

figure 4 is a diagram of the tuning of the frequency of the output signal when using an additional LDP;

5 is a diagram of a signal source;

In figure 1, 2 and 5, positions 1-26 are indicated:

1 - signal source;

2.i - i-th block of the expansion of the range of the frequency adjustment (BRDP), where i = 1,2,3, ..., N;

3 - control command shaper (PKU);

4.i - divider with switchable division coefficient (DPKD) of the i-th BRDP;

5.i - low-pass filter (LPF) of the i-th BRDP;

6.i - mixer of the i-th BRDP;

7.i - band-pass filter (PF) of the i-th BRDP;

8 - the first switch;

9 - the second switch;

10 - DPKD;

11 - a shaper of signals of reference frequencies (FSOCH);

12 - site for the formation of local oscillator signals (UFSG);

13 - node forming a frequency grid (UFSC);

14 - node frequency conversion up (UPCHV);

15 - DPKD in UFSG;

16 - low-pass filter in UFSH;

17 - mixer in UFSG;

18 - PF in UFSG;

19 - digital direct synthesis synthesizer (DSPC);

20 - filter in the UFSC;

21 - mixer in UFSK;

22 - PF in UFSK;

23 - DPKD in UFSH;

24 - low-pass filter in UFSC;

25 - mixer in UPCHV;

26 - PF in the HRE.

Broadband frequency synthesizer contains a signal source 1, tunable in frequency, N blocks for expanding the range of the frequency adjustment (BRDP) 2 and the shaper control commands (PKU) 3. Each BRDP 2.i consists of a divider with a switchable division ratio (DLC) 4.i, low-pass filter (LPF) 5.i, mixer 6.i and band-pass filter (PF) 7.i. The input BRDP 2.i is connected through a branch with the RF input of the DPKD 4.i and the input of the local oscillator of the mixer 6.i. The output of the DPKD 4.i through the low-pass filter 5.i is connected to the input of the inverter of the mixer 6.i, the output of which is connected to the PF 7.i. The output of PF 7.i is the output of the BJF 2.i. The control inputs of all DPKD 4.i are connected to the output of the PKU 3.

To the output of the last unit of the BRDP 2.N, an additional BRDP 2.N + 1 is connected using the first 8 and second 9 switches (Figure 2). The first switch 8 has one input and two outputs. The second switch 9 has two inputs and one output. The input of the first switch 8 is connected to the output of the BRDP 2.N. The first output of the first switch 8 is connected to the first input of the second switch 9. The second output of the first switch 8 is connected to the input of the PDU 2.N + 1. The output of the PDU 2.N + 1 is connected to the second input of the second switch 9. The first 8 and second 9 switches are made with the possibility of control through PKU 3.

The broadband synthesizer may further comprise a DPKD 10, the RF input of which is connected to the output of the second switch 9 (Figure 2). The control input DPKD 10 is connected to the output of the PKU 3.

The signal source 1 can be made in accordance with the circuit depicted in Fig.5. It consists of a connected signal shaper of reference frequencies (FSOCH) 11, a node for generating local oscillator signals (UFSG) 12, a node for forming a frequency grid (UFSC) 13 and a unit for frequency conversion up (UPCHV) 14. UFSG 12 consists of a DPKD 15, low-pass filter 16, mixer 17 and PF 18. The input UFSG 12 connected to one of the outputs of FSOCH 11 is connected via branching to the RF input of DPKD 15 and the input of the local oscillator of mixer 17. The output of DPKD 15 through LPF 16 is connected to the input of the IF of mixer 17, the output of which is connected to PF 18. The output of PF 18 is the output of UFSG 12. UFSK 13 consists of a digital synthesizer direct synthesis (DSPC) 19, filter 20, mixer 21, PF 22, DPKD 23 and low-pass filter 24. The reference input of the DSPC 19 is connected to one of the outputs of the FSOC 11. The output of the DSPC 19 is connected through the filter 20 to the input of the inverter of the mixer 21, the local oscillator input which is connected to the output of the UFSG 12. The output of the mixer 21 is connected through the PF 22 to the RF input of the DPKD 23. The output of the DPKD 23 is connected to the low-pass filter 24, the output of which is the output of the UFSC 12. The up-conversion unit (UPCHV) 14 consists of a mixer 25 and PF 26 The input of the inverter of the mixer 25 is connected to the output of the FSSC 13. The input of the local oscillator of the mixer 25 is connected to one of the outputs of the FSOC 11. You the mixer 25 is connected to the PF 26, the output of which is the output of the UPCHV 14. The output of the latter is the output of the signal source 1. The control inputs DPKD 15, DPKD 23 and TsSPS 19 are connected to the output of the PKU 3.

The device operates as follows.

Using a signal source 1, tunable in frequency in a narrow range (of the order of 1%), a signal is formed with a frequency f _in . This signal is fed to the input of the first BRDP 2.1, in which, branching, it is fed to the RF input of the DPKD 4.1 and the input of the local oscillator of the mixer 6.1. The division coefficient D ₁ DPKD 4.1 set by PKU 3 in the range from D _1min to D _1max . From the output of the DPKD 4.1 through the low-pass filter 5.1, a signal with a frequency of f _I / D _{1 is} fed to the input of the inverter of the mixer 6.1. At the output of mixer 6.1, the useful signal has a frequency:

, где f_вх - частота сигнала, формируемого источником сигнала 1, D₁ - значение коэффициента деления ДПКД 4.1.

where f _I - the frequency of the signal generated by the signal source 1, D ₁ - the value of the division coefficient of the DPKD 4.1.

The parasitic spectral components of the signal are suppressed using PF 7.1, which can be tuned to the total (f _in + f _vx / D ₁ ) or differential (f _in -f _vx / D ₁ ) component of the signal frequency from the output of the mixer 6.1. Further, for definiteness, we will take into account that the filters are tuned to the total signal component at the output of the mixers. This choice does not affect the operation of the claimed device.

The diagram of the tuning of the output frequency BRDP 2.1 for various division ratios of the DPKD 4.1 and the implementation of the tuning of the input frequency is shown in Fig.3. At the output of the BRDP 2.1, the formation of any frequency in the range from f _{o min} to f _{o max max is provided} . For this, it is necessary that, with a division coefficient D ₁ and a minimum required frequency of the signal source 1 f _{in min, the} value of the frequency of the output signal is less than or equal to the value of the output frequency with a division coefficient increased by unity, i.e. at D ₁ +1, and the maximum frequency of the signal source is 1 f _{in max} . For docking of the tuning sections at D _1min and D _1min +1, the largest tuning range of the signal source frequency 1 f _in is required. Mathematically, the condition of continuous frequency tuning is determined by the expression:

f _1out (D _1min , f _{in min} ) ≤f _{1 out} (D _1min +1, f _{in max} ).

Choosing an equal sign in the expression for continuous frequency tuning, you can get the minimum and maximum necessary values for tuning the input frequency of the BRDP 2.1 while ensuring continuous tuning in frequency of the output signal of the BRDP 2.1. The ratio of the maximum to the minimum value of the input frequency BRDP 2.1 is determined using the formula:

This formula is valid for all SLDS 2.i. In this case, as the value of the division coefficient, the coefficient D _{i min} corresponding to the selected block is used, f _{in max} and f _{in min} correspond to the minimum and maximum values of the frequency of the output signal of the previous block of the PDLU 2i-1.

The ratio of the maximum value of the frequency tuning range at the output of the BRDP 2.1 to the minimum is calculated by the formula:

The greater the maximum value of the division coefficient, the greater the tuning band at the output of the PDL 2.1, but the higher the requirements for the selectivity of PF 7.1 at the output of the mixer 6.1. The maximum value of the division coefficient is selected from the feasibility condition PF 7.1.

The efficiency of expanding the frequency tuning range using one BRDP2.1 is estimated using the expansion coefficient K _exp , expressed as the ratio of the tuning bandwidth at the output to the tuning bandwidth at the input:

где f_{1 вых max(min)} - максимальное (минимальное) значение частоты выходного сигнала БРДП2.1, f_{1 вых max(min)} - (минимальное) значение частоты входного сигнала БРДП 2.1.

where f _{1 o output max (min)} is the maximum (minimum) value of the frequency of the output signal BRDP 2.1, f ₁ o _{output max (min)} is the (minimum) value of the frequency of the input signal BRDP 2.1.

The magnitude of the expansion coefficient K _exp can be expressed in terms of the minimum D _1min and maximum D _1max dividing factor _DPKD 4.1:

.

.

If we assume that the ratio of the maximum to the minimum division coefficient of the DPKD 4.1 D _max / D _min is 2, then the expression for the expansion coefficient takes the form:

.

.

With a division coefficient D _1min equal to 2, the expansion coefficient K _rash will be 3.5, with D _1min equal to 16, K _ras will be 10.0625. However, despite the fact that with an increase in D _1min, the expansion coefficient K _exp also increases, the absolute value of the tuning band will be small, since for large values of the division coefficient D _1min a narrow frequency tuning band of the input signal is required. For example, for D _1min equal to 16, the bandwidth of the frequency tuning of the input signal must satisfy the expression:

.

.

Accordingly, the frequency bandwidth of the output signal will be equal to:

f _{1out max} -f _{1out min} ≈0,035 · f _{in min} .

The use of the serial connection of several BJC 2.i, which have an identical design and differ in specified dividing coefficients and filter passbands, allows to obtain a wide tuning band.

поступает на вход следующего БРДП 2.2. Минимальный коэффициент деления ДПКД 4.2 в БРДП 2.2 выбирают исходя из диапазона перестройки выходной частоты БРДП 2.1, используя выражение (*) с учетом замены D_1min на D_2min. При правильном D_2min рассчитанное по формуле (*) значение

должно быть меньше или равно

. С учетом преобразования частоты в БРДП 2.2 значение частоты выходного сигнала рассчитывается по формуле:In this case, from the output of the PDLP 2.1 a useful signal with a frequency

arrives at the input of the next BRDP 2.2. The minimum division coefficient of the DPKD 4.2 in the PDLF 2.2 is selected based on the tuning range of the output frequency of the PDLP 2.1 using the expression (*) taking into account the replacement of D _1min by D _2min . With the correct D _{2min, the value} calculated by the formula (*)

must be less than or equal

. Taking into account the conversion of frequency into the PDL 2.2, the value of the output signal frequency is calculated by the formula:

, где D₂ - коэффициент деления второго ДПКД 4.2. Для N последовательно соединенных БРДП 2.i выходная частота может быть рассчитана по формуле:

where D ₂ is the division coefficient of the second DPKD 4.2. For N series-connected PDLF 2.i, the output frequency can be calculated by the formula:

, где D_N - коэффициент деления ДПКД 4.N в БРДП 2.N.

, where D _N is the division coefficient of the DPKD 4.N in the NLR 2.N.

, т.е не более 35% от минимального значения выходной частоты. При этом на выходе предпоследней ячейки БРДП 2.N-1 обеспечивают отношение максимального значения частоты к минимальному, равное 9/8.Let us evaluate the maximum possible tuning range of the output frequency of the signal at the output N of the PDL 2.i. The minimum value of the division coefficient of the DPKD 4.i cannot be less than 2. The maximum value is chosen equal to 4, so that the ratio D _imax / D _imin does not exceed the value 2. The maximum possible ratio of the output frequency of the signal to the input in the N-th MTPL 2.N in accordance with the formula is:

, i.e. not more than 35% of the minimum value of the output frequency. At the same time, at the output of the penultimate cell of the BJB 2.N-1, the ratio of the maximum frequency to the minimum value is equal to 9/8.

, где f - частота на входе гетеродина смесителя, K - целое число, δf - отстройка значения реальной частоты от ближайшего кратного значения. В общем случае на выходе смесителя образуются спектральные составляющие, частоты которых соответствуют формуле:A useful feature of the BRID 2.i circuit is the ability to suppress all spurious spectral components of the output signal arising due to the nonlinear properties of the mixer to any desired level due to the fact that signals with multiple frequencies are involved in the frequency conversion. Explain this. When converting frequency, it is traditional to use multiple frequencies at the inputs of the local oscillator and the inverter of the mixer. The value of the frequency at the input of the inverter of the mixer can be represented as

where f is the frequency at the input of the mixer local oscillator, K is an integer, δf is the detuning of the real frequency from the nearest multiple value. In the general case, at the mixer output, spectral components are formed whose frequencies correspond to the formula:

, где m и n - целые числа.

where m and n are integers.

The amplitude of the spectral components formed at the mixer output is the higher, the smaller the sum of n and m, which are the order of nonlinearity and lead to the formation of the corresponding components in the spectrum of the output signal. The frequency of one of the closest to the frequency of the useful signal combination components of the output signal of the mixer can be represented by the formula:

.

.

The frequency of the useful signal will be:

.

.

, которая соответствует К+1 гармонике сигнала, поступающего на вход ПЧ смесителя. Для преодоления данной проблемы значение отношения частот сигналов, участвующих в преобразовании, должно быть величиной больше 10(K≥10), что приводит к ограничению диапазона перестройки выходной частоты в синтезаторах прямого синтеза, построенных по известным схемам (Приложение №2. А.Ченакин. Частотный синтез: текущие решения и новые тенденции. ЭЛЕКТРОНИКА: Наука, Технология, Бизнес, январь 2008, стр.92-97).For small δf, the parasitic component of the spectrum will be located in close proximity to the useful signal and cannot be filtered by the FS, and for small K it will have a large amplitude, since is a product of low order nonlinearity. A similar problem arises with a parasitic component of the form

, which corresponds to the K + 1 harmonic of the signal supplied to the input of the inverter of the mixer. To overcome this problem, the value of the frequency ratio of the signals involved in the conversion should be greater than 10 (K≥10), which leads to a limitation of the tuning range of the output frequency in direct synthesis synthesizers constructed according to well-known schemes (Appendix No. 2. A. Chenakin. Frequency synthesis: current solutions and new trends. ELECTRONICS: Science, Technology, Business, January 2008, pp. 92-97).

The frequency conversion circuit in the inventive device is devoid of the disadvantage described above, since when the frequency multiplicity at the input of the local oscillator and the input of the inverter of the mixer 6.i all possible combination components of its output signal can be described by the formula:

f _{CMout m, n} = f _in | m ± n / D _i |.

. ПФ 7.i с соответствующими полосами пропускания и заграждения позволяет подавить все паразитные спектральные компоненты до требуемого уровня. Выбор полосы пропускания и полосы заграждения ПФ 7.i зависит от отношения коэффициентов деления ДПКД 4.i D_1max/D_1min. Если данное отношение меньше 1,5, на выходе смесителя 6.i достаточно одного ПФ 7.i. Если значения отношения D_max/D_min находятся в интервале от 1,5 до 2, то для подавления паразитных спектральных компонент на выходе смесителя необходимо использовать два переключаемых полосовых фильтра. При этом верхняя граница полосы пропускания более низкочастотного фильтра должна совпадать с нижней границей полосы пропускания более высокочастотного фильтра. При еще больших значениях D_1max/D_1min количество необходимых фильтров возрастает.It can be seen from the formula that the smallest detuning of the nearest spurious spectral component from the useful one is

. PF 7.i with the appropriate bandwidths and barriers allows you to suppress all spurious spectral components to the desired level. The choice of the passband and the fence bandwidth of PF 7.i depends on the ratio of the division coefficients of the DPKD 4.i D _1max / D _1min . If this ratio is less than 1.5, at the output of the mixer 6.i one PF 7.i is enough. If the values of the ratio D _max / D _min are in the range from 1.5 to 2, then to suppress spurious spectral components at the output of the mixer, two switched band-pass filters must be used. In this case, the upper limit of the passband of the lower-pass filter should coincide with the lower limit of the passband of the higher-pass filter. At even greater values of D _1max / D _{1min, the} number of necessary filters increases.

The inventive scheme of a broadband synthesizer has another advantage - a high frequency tuning speed, regardless of the tuning step. When converting the signal frequency to BRDP 2.1, the frequency tuning time increases slightly, and its increase is associated with the passage of PF 7.1, the band of which is several times wider than the tuning band of signal source 1. The transition time when the input frequency is changed is determined by the duration of the pulse characteristic of the filter, since the signal at the output of the bandpass filter is a convolution of the input signal with the impulse response of the bandpass filter. For filters, the duration of the impulse response is inversely proportional to the bandwidth. Therefore, the duration of the impulse response of PF 7.1 will be several times less than the duration of the impulse response of the filters used in signal source 1 and which are more narrow-band. The duration of the impulse response of the filters in the remaining BRJ 2.i is even shorter, therefore the main contribution to the frequency tuning time will be made by the narrowest filter in the BRJ cascade 2. Ultimately, the speed of the output frequency of the synthesizer will be determined by the speed of the signal at the output of signal source 1 .

Connecting the PDU 2.N + 1 to the output of the N-th PDU 2.N through the first 8 and second 9 switches, as shown in Figure 2, allows you to further expand the frequency tuning range of the synthesizer output signal.

With one position of the switches of the switches 8 and 9, the output from the second switch 9 receives a signal from the output of the PDU 2.N without conversion. In a different position, the output of the second switch 9 receives a signal, the tuning band of which is further expanded with the help of an additional BRD 2.N + 1. This scheme makes it possible to use in aggregate the tuning range of the frequency of the output signal of the last BRDP 2.N and the range obtained at the output of the BRDP 2.N + 1. The resulting frequency bands, complementing each other, form a total tuning range greater than the maximum possible tuning range of the output signal of the 2.N.

Figure 4 clearly illustrates the mechanism of continuous tuning of the signal frequency at the output of the second switch 9. The abscissa axis shows the frequency of the signal at the output of the N-th PDL 2.N normalized to its minimum value, along the ordinate axis, the signal frequency at the output of the second switch 9 , also normalized to the minimum value of the frequency of the output signal of the N-th BRDP 2.N. The diagram is built for the values of division factors 2 and 4 of the DPKD 4.N + 1 (division coefficient 3 is not used). The “transit” line corresponds to the position of the switches of the switches 8 and 9, at which the signal bypasses the PDL 2.N + 1, i.e. tuning the frequency of the signal in the circuit N of the series-connected BRD 2.i, providing a tuning range of the output frequency in the range from 1 to 4/3.

, где f_вых - частота сигнала с выхода БРДП 2.N+1, f_вх - частота входного сигнала БРДП 2.N+1.To obtain frequency tuning at the output of the synthesizer from 4/3 to 3/2, the circuit from N BRDP 2.i is tuned so that at its output a tuning range of 16/15 to 6/5 is obtained, and the division coefficient of the DPKD is 4.N + 1 set using PKU 3 equal to 4, providing frequency conversion in accordance with the expression:

, where f _o is the frequency of the signal from the output of the BRDP 2.N + 1, f _in - the frequency of the input signal of the BRDP 2.N + 1.

.To get the output frequency tuning range from 3/2 to 2, the circuit from N PDL 2.i is set up in such a way as to obtain the frequency tuning range from 1 to 4/3 at its output. The division coefficient of the DPKD 4.N + 1 is set using PKU 3 equal to 2, providing frequency conversion in accordance with the expression:

.

Together, the three frequency tuning sections of FIG. 4 form a continuous tuning band of the output frequency with an octave width.

, а верхнее будет равно 1 и будет совпадать с минимальным значением при коэффициенте деления равном 1, обеспечивая непрерывность диапазона перестройки выходной частоты в полосе от

до 2. Аналогично будет обеспечена стыковка участков перестройки выходной частоты при больших значениях коэффициента деления, равных 2^p. В результате общий диапазон возможных значений выходной частоты составит от

до 2.To build a frequency synthesizer with a frequency tuning range of more than an octave, in accordance with FIG. 3, the output of the second switch 9 is connected to the DPKD 10 with the fission factor 3 switched by PKU 3 equal to 2 ^p at p = 0,1,2 ... p _max . The division coefficient of the DPKD 10, equal to unity at p = 0, corresponds to the direct passage of the signal from the output of the second switch 9 without conversion. When setting the division ratio equal to 2, the normalized value of the lower cutoff frequency of the tuning range will be equal to

, and the upper one will be equal to 1 and will coincide with the minimum value with a division coefficient equal to 1, ensuring continuity of the tuning range of the output frequency in the band from

up to 2. Similarly, the docking of sections of the tuning of the output frequency with large values of the division coefficient equal to 2 ^p will be provided. As a result, the total range of possible values of the output frequency will be from

up to 2.

The serial connection of the PDU 2.i provides the conversion of the frequency of the signal source 1, equivalent to multiplying the frequency by a fractional number of times. When switching the division factor, the frequency tuning step at the output of the synthesizer becomes uneven.

раз.To ensure an equidistant step of tuning the output frequency, tuning of the input frequency must be carried out with an uneven pitch, depending on the division factors D _i DPKD 4.i. Thus, for practical implementation, it is necessary to use an input signal tunable to a frequency with a step smaller than the required accuracy of setting the frequency in

time.

This requirement can be satisfied when executing the signal source 1, for example, in accordance with the scheme in figure 5.

The reference frequency signal generator (FSOC) 11 provides the formation of three coherent signals of a fixed frequency.

.One of the signals of FSOS 11 with a frequency f _{OP2 is} fed to the input of the FSD 12, the circuit of which is similar to the BRPS 2.L circuit since the frequency f _{OP2 is} fixed, then the number of generated frequencies at the output of the FSD 12 will correspond to the number of division coefficients D1 of the DPDC 15. Frequency values by the output of the UFSG 12 is determined by the formula f _OP2 (1 + 1 / D1). The frequency tuning step will be the largest for the values D1 = D1 _min and D1 = D1 _min +1 and can be calculated by the formula

.

From the output of the UFSG 12, the signal is fed to the input of the local oscillator of the mixer 21 of the frequency grid forming unit (UFSM) 13, to the inverter of which the signal generated by the DSPC 19 is supplied through the filter 20 in accordance with the formula:

,

,

where S is the capacity of the accumulating adder TsPSS 19; R is the value of the code that controls the frequency of the DSPC 19 supplied to the control input of the DSPC 19 from the output of the FCU 3, f _OP3 is the clock frequency of the DSPC 19 supplied to the reference input of the DSPC 19 from the output of the FSOC 11. The signals from the output of the mixer 21 will have the frequency:

f _CM2 = f _OD2 (1 + 1 / D1) ± f _DSSR .

Using PF 22 select one of the signals from the output of the mixer 21, suppressing all undesirable combination components. In the future, we assume that this signal is a signal with a frequency f _OP2 (1 + 1 / D1) ± f _DSPC .

If the tuning bandwidth of the DSPC 19 is Δf _DSPC , then at the output of the mixer 21 it will be possible to generate a signal with any frequency value in the band:

.

.

When providing conditions Δf _TSSPS ≥Δf _UFSG frequency tuning at the output of mixer 21 is continuous.

The frequency bandwidth of this signal will be wider than the tuning band of the DSPC 19, due to the possibility of changing the division coefficient D1 of the DPKD 15 in the UFSG 12. Due to the peculiarities of the operation of the DSPC 19, its output signal contains spurious components near the frequency of the useful signal (Appendix No. 3. Makarenko V. Components for building wireless communication devices, part 7. Direct digital synthesis frequency synthesizers (Exi, No. 1, January 2010, Fig. 5), which will also be contained in the signal at the output of PF 22. Their level is reduced from using division often you. Explain this.

Low-level spurious components can be represented in the form of components formed due to spurious amplitude and angular modulations. Spurious amplitude modulation can be suppressed by limiting the amplitude of the signal at the RF input of the DPKD. When dividing the signal frequency, the phase fluctuation due to spurious angular modulation is retained in absolute value, however, the period of the output signal increases by D times (where D is the division coefficient), which leads to a decrease in the spurious angular modulation index, the value of which determines the level of spurious spectral components. When dividing the frequency of the output signal of the mixer 21 using the DPKD 23, the division coefficient of which is D2, the level of spurious components falling into the passband of the PF 22 will decrease by 20 logD2, however, the tuning bandwidth will also decrease by D2 times.

To expand the range of tuning the frequency of the output signal, a divider with a switchable division ratio is used, providing frequency tuning at the RF input of the DPKD 23 for each value of the division coefficient. Chart tuning the frequency of the output signal DPKD 23 will be similar to the diagram shown in Fig.3. To ensure overlap of the sections of the adjustment at the output of the DPKD 23 with the values of the division factors D2 _min and D2 _min +1, the largest adjustment range is necessary. The required tuning bandwidth at which continuous frequency tuning is possible is determined from the condition:

,

,

where f _{CM2 min (max)} is the minimum (maximum) possible value of the output frequency

PF 22; D2 _min is the smallest division coefficient of the DPKD 23.

In this case, the width of the adjustment band at the output of the DPKD 23 will be equal to:

.

.

The full frequency tuning range, taking into account the possibility of switching the division coefficient D2 of the DPKD 23, can be calculated by the formula:

,

,

where f _{DPKD min (max)} - the minimum (maximum) possible value of the frequency at the output of the DPKD 23; D2 _{min (max)} is the smallest (largest) division coefficient of the DPKD 23;

f _{CM2 min} - the value of the frequency of the output signal of the mixer 21.

.To compensate for the reduction in frequency during dividing by the DPCD 23 and to provide the required boundaries for tuning the output frequency, a signal with a frequency f of the _DPCD through the low-pass filter 24 is fed to the input of the inverter of the mixer 25 in the IFA 14. At the input of the local oscillator of the mixer 25, a signal of a fixed frequency f _{OP1 is received} from the FSOCH 11. at the mixer output signal 25 is generated with frequency f _O = f ± f _{DPKD OP1.} Using PF 26 at the output of mixer 25, the total or difference component of the signal is selected, which is the useful frequency of signal source 1. Since the frequency of the output signal of the DSPC 19 is divided, the maximum step of the frequency tuning at the output of signal source 1 will be no more

.

Compared with the technical solution described in US patent No. 5495202, the proposed signal source circuit 1 allows to obtain a wider frequency tuning band by expanding the frequency tuning range of the DSPC 19 when it is converted with different values of the mixer oscillator frequency 25 and using several values of the division coefficient in DPKD 23. As a result, the frequency tuning range of the signal source 1 is 5-10 times larger than in the device of US Pat. No. 5,495,202, with the same level of suppression of spurious spectra components of the DSPC 19 and the same value of the frequency tuning speed.

Example.

The practical implementation of the inventive device is illustrated using Fig.6-10, which depict:

figure 6 is a structural diagram of a practical implementation of the synthesizer;

7 is a diagram of the frequency tuning of the output signal of the BRDP 2.1;

on Fig is a diagram of the tuning frequency of the output signal BRDP 2.2;

figure 9 is a diagram of the frequency tuning of the output signal of the BRDP 2.3;

figure 10 is a diagram of the tuning of the frequency of the output signal BRDP 2.4.

In Fig.6 additionally noted:

27 - divider with a fixed division ratio of 4;

28-30 - switches.

The scheme of the broadband frequency synthesizer shown in Fig.6, allows to provide the following output signal parameters:

1) the range of output frequencies from 0.75 GHz to 12 GHz;

2) frequency tuning step less than 0.025 Hz;

3) the relative level of spurious discrete components in the spectrum of the output signal is not more than -90 dB;

4) the frequency tuning time (for any two values of the frequencies of the output frequency range) is less than 200 ns;

5) the relative level of the spectral density of the frequency fluctuations at a frequency of 12 GHz is not more than -135 dB _ns / Hz with a detuning of 50 kHz from the carrier;

6) the instability of the frequency of the output signal is equal to the instability of the frequency of the reference oscillator.

As FSOCH 11 use a reference generator with one value of the output frequency. With it, a signal with a frequency of 4000 MHz with a low level of phase noise is formed.

Using UFSG 12, a local oscillator signal is generated from the signal of the reference generator 11 for the mixer 21, which can have seven frequency values depending on the division coefficient D1 of the DPKD 15. The division coefficient D1 can take integer values from 11 to 17. The possible frequencies of the output signal of the mixer 17 MHz: 4363.64; 4,333.33; 4,307.69; 4,285.71; 4,266.67; 4250; 4235.29. To suppress all unwanted spectral components, the low-pass filter 16 must have a cutoff frequency of at least 364 MHz and the start frequency of the obstacle band 470 MHz, the PF 18 bandpass filter must have a passband from 4285 MHz to 4364 MHz and suppress signals with a frequency below 4000 MHz and above 4470 MHz . Fundamentally, the low-pass filter 16 may be absent in the circuit. However, in this case, at the output of mixer 17 there will be an increased level of combinational components from the signal supplied to the IF input, which complicates the implementation of PF 18. At the output of PF 18, all undesirable components should be suppressed to the level of -70 dB. This value of the suppression value should be no more than the level of spurious components at the output of the DSP 19, falling into the passband of the PF 20. The value of -70 dB corresponds to a typical level of spurious components near the carrier frequency of the signal for DSP chips. The occurrence of these spurious components is associated with a limitation of the bit depth of the high-speed digital-to-analog converter (DAC) included in the DSPC. The frequency values of the reference signal of the DSPC 19 and the band of output frequencies generated by the DSPC 19 are selected so that the combination band below the fifth order does not fall into the passband of the PF 20.

DSPC 19 generates an output signal in the frequency band from 357 MHz to 388 MHz. Since the values of the output frequency are 0.357-0.388 of the sampling frequency of the DAC in the DSPC 19, the combination components are formed both above and below the band of useful frequencies. Therefore, at the output of the DSPC 19 in this circuit, a band-pass filter is used instead of a low-pass filter. To suppress the combination components below the fifth order, it is necessary that the PF 20 has a passband corresponding to the output signal frequency band and suppresses frequencies below 285 MHz and above 429 MHz.

From the output of the PF 20, the signal is fed to the input of the inverter of the mixer 21, to the input of the local oscillator of which the signal from the output of the FSB 12 has seven different frequency values. In the spectrum of the output signal of the mixer 21 using PF 22 allocate the total frequency. As a result, a signal is formed whose frequency is tuned in the range from 4602 MHz to 4752 MHz. To suppress the parasitic components arising in the mixer 21, PF 22 must suppress frequencies less than 4364 MHz and more than 4970 MHz to the level of -70 dB.

PF 18 and PF 22 with the above parameters can be performed by standard microstrip technology. In this case, the cascade connection of two PFs may be required. As a lower frequency PF 20, it is advisable to use a filter on lumped parameters. Since the combination components at the output of the DSPC 19 usually have a level of less than -45 dB with respect to the useful signal, the suppression requirements for the PF 20 are much lower - it is enough to provide attenuation in the obstacle band of about -30 dB.

The signal from the output of the PF 22 is fed to the RF input of the DPKD 23, the minimum value of the dividing coefficient of which is chosen so as to ensure continuity of the tuning of the output frequency when changing the division ratio and when tuning the frequency at its input in the range from 4692 MHz to 4752 MHz. Changing the division coefficient D2 from 31 to 38 provides continuous tuning of the signal frequency at the output from 121.21 MHz to 153.27 MHz. A low-pass filter 24 with a cut-off frequency of 154 MHz and a blocking frequency of 242 MHz provides suppression of harmonic components in the output signal of the DPKD 23. When dividing the frequency, the level of parasitic components decreases by at least 20 lg (D2) in decibels. With D2 equal to 31, the suppression is minimal and amounts to 29.8 dB. Thus, if the relative level of spurious components at the PF 22 output is not more than -70 dB, then the level of spurious components will be no more than -100 dB. Using a mixer 25, to the input of the local oscillator of which a reference frequency signal of 4000 MHz is supplied, the frequency band of the signal at the output of the DPKD 23 is transferred to the range of the required input frequencies of the BRDP 2.1.

The required values of the boundary frequencies of the tuning band at the input of the PDL 2.1 are determined as follows. First, find the value of the upper cutoff frequency in accordance with the formula:

,

,

where f _{o, max} is the value of the frequency of the output signal of the synthesizer, Di _min is the smallest value of the division coefficient of the DPKD 4.1-4.4, i = 3, 4, 5, 6.

и по известному значению f_{вх 2-1,max} определяют f_{вх 2-1, max}. С учетом значений коэффициента деления, используемых в схеме на Фиг.6: f_{вх 2-1, min}=3846,8 МГц, f_{вх 2-1, max}=3878,8 МГц, поэтому в смесителе 25 в качестве полезного сигнала используют разностную составляющую.Then, using the formula (*) and substituting D1 _min into it, determine

and from the known value of f _{in 2-1, max} determine f _{in 2-1, max} . Taking into account the values of the division coefficient used in the scheme of FIG. 6: f _{in 2-1, min} = 3846.8 MHz, f _{in 2-1, max} = 3878.8 MHz, therefore, in the mixer 25, a differential signal is used as a useful signal component.

When implementing the PDLP 2.1, the values of the division coefficients of the DPKD 7.1 are used from 10 to 16. The frequency tuning diagram of the output signal of the PDLP 2.1 is shown in Fig.7. The requirements for the low-pass filter 5.1 are as follows: the cutoff frequency of the passband is 388 MHz, the cutoff frequency of the suppression band is 482 MHz. LPF 5.1 can be implemented on lumped elements. PF 7.1 should pass a signal in the band from 4096 MHz to 4266.7 MHz, suppress frequencies below 3847 MHz and above 4337 MHz and provide suppression of spurious components to a level of -100 dB relative to the level of the useful signal. To implement this requirement, it is necessary to use a cascade connection of two identical filters. PF7.1 can be performed using standard microstrip technology.

Because of the wide range of variation of the division coefficient of the DPKD 4.2, it is not possible to suppress all parasitic components using one PF 7.2 connected to the output of the mixer 6.2, because of the wide limits of variation of the division coefficient of DPKD 4.2, therefore, the switching filters PF 7.2.1 and PF 7.2.2 are used in the circuit LPF 5.2.1 and LPF 5.2.2. A similar circuit solution is used in the BRDP 2.3 and the BRDP 2.4. In order to ensure suppression of spurious components to a level of from -90 to -100 dB, it is also advisable to use a cascade connection of filters.

In the PDLM 2.2, for the DPKD 4.2, division factors from 4 to 7 are used. The diagram of the tuning of the output frequency of the PDLF 2.2 is presented in Fig. 8. The low-pass filter 5.2.1 should pass signals with a frequency below 711.1 MHz and suppress above 1185.2 MHz. The low-pass filter 5.2.2 should pass signals with a frequency below 1066.6 MHz and suppress above 1659.2 MHz. PF 7.2.1 should suppress signals with a frequency of less than 4266.7 MHz and more than 5333.3 MHz. PF 7.2.2 should suppress signals with a frequency of less than 4266.7 MHz and more than 5807.3 MHz. PF 7.2.1, PF 7.2.2 and low-pass filter 5.3.1, low-pass filter 5.3.2 switch at the same time when changing the division coefficient of the DPKD 4.2 from 6 to 5.

In the PDLD 2.3, for the PDKD 4.3, the values of the division coefficients equal to 2, 3, and 4 are used. The diagram of the tuning of the output frequency of the PDLD 2.3 is shown in Fig. 9. The low-pass filter 5.3.1 must pass signals with a frequency below 1700 MHz and suppress above 2400 MHz. The low-pass filter 5.3.2 must pass signals with a frequency below 2666.7 MHz and suppress above 3400 MHz. PF 7.3.1 should suppress signals with a frequency of less than 5333.3 MHz and more than 6800 MHz. PF 7.3.2 should suppress signals with a frequency of less than 5333.3 MHz and more than 8475.5 MHz. PF 7.3.1, PF 7.3.2 and low-pass filter 5.3.1, low-pass filter 5.3.2 at the frequency tuning switch simultaneously. When the division coefficient of the DPKD 4.3 is 4, for all values of the input frequency of the block use PF 7.3.1 and low-pass filter 5.3.2. These filters are also used with a division ratio of 3 and an input frequency of less than 5100 MHz. In other cases, use PF 7.3.2 and low-pass filter 5.3.1. At the output of the BRDP 2.3, frequency tuning in the band from 6 to 8 GHz is provided.

In the BRDP 2.4 for the DPKD 4.4, the values of the division coefficients 2 and 4 are used. The diagram of the tuning of the output frequency of the BRDP 2.4 is presented in FIG. 10. LPF 5.4.1 should pass signals with a frequency of less than 4000 MHz and suppress more than 6200 MHz. LPF 5.4.2 should pass signals with a frequency of less than 1860 MHz and suppress more than 3200 MHz. PF 7.4.1 shall suppress signals with a frequency of less than 7440 MHz and more than 9600 MHz. PF 7.4.2 should suppress signals with a frequency of less than 8000 MHz and more than 12400 MHz. LPF 5.4.1, LPF 5.4.2 and PF 7.4.1, PF 7.4.2 switch simultaneously with changing the division coefficient of the DPKD 4.4 from 4 to 2. This ensures the formation of output frequencies in the range from 8 to 12 GHz.

Tuning in the band from 6 to 8 GHz is ensured by using the output signal of the PDLD 2.3 by switching the switches 8 and 9 to the position corresponding to the transit passage of the signal.

At the output of the BRDP 2.4, frequency tuning in the range from 6 to 12 GHz is provided. The extension of the frequency tuning range to the low-frequency region without reducing the requirements on the parameters of the frequency synthesizer can be obtained by introducing the DPKD 10 connected to the output of the second switch 9. For the range from 6 to 12 GHz, the signal through the switches 29 and 30 goes around the DPKD 10. For receiving tuning in the range from 3 to 6 GHz, the signal is fed to the RF input DPKD 10, the coefficient of which is set equal to 2. Using DPKD 10 with coefficients 2, 4, 8, you can get the minimum frequency value at the output of the synthesizer 750 MHz at maximum SG value of the output signal frequency of 12 GHz.

, максимальное значение которого 3,09 для схемы на Фиг.6. При увеличении частоты в 3,09 раза уровень фазового шума увеличится на 20lg(3,09)=9,8 дБ. Однако, это справедливо для уровня шума вблизи несущей при отстройках менее 10 кГц. При отстройках более 100 кГц шум дополнительно увеличится ориентировочно на 6-7 дБ из-за добавки собственных шумов ДПКД 4.1-4.4 и 10. В результате уровень спектральной плотности фазовых шумов при отстройке от несущей более 100 кГц для синтезатора в соответствии с Фиг.6 составит около -135 дБ_нес/Гц.To achieve the declared level of spectral power density of phase noise, a low-noise reference generator is required. When comparing the absolute values of the phase noise of the reference generator 11 and the DPKD 23, the noise of the reference generator 11 will be higher than the noise of the DPKD 23 for detuning from the carrier of less than 10 kHz. When the detunings are more than 10 kHz, the phase noise of the DPKD 23 will be greater than the phase noise of the reference oscillator 11. Therefore, the noise at the output of signal source 1 when the detuning from the carrier is less than 10 kHz will correspond to the phase noise level of the reference oscillator, and when the detuning is more than 100 kHz, it will be of the order of -153 dB _ns / Hz. The frequency conversion of the signal, produced in the BJP 2.1-2.4, is equivalent to multiplying the frequency by a fractional number equal to

, the maximum value of which is 3.09 for the circuit in Fig.6. With a frequency increase of 3.09 times, the phase noise level will increase by 20lg (3.09) = 9.8 dB. However, this is true for the noise level near the carrier with offsets less than 10 kHz. With detunings of more than 100 kHz, the noise will additionally increase by approximately 6-7 dB due to the addition of intrinsic noise of DPKD 4.1-4.4 and 10. As a result, the level of spectral density of phase noise when detuning from a carrier of more than 100 kHz for the synthesizer in accordance with FIG. 6 will be about -135 dB _bore / Hz.

When using a high-frequency crystal oscillator with a frequency of 100 MHz and multiplying the frequency by 40 times as the reference oscillator 11, it is possible to obtain the phase noise level for the output signal with detunings of more than 100 kHz at about -135 dB _ns / Hz, which is much higher than the noise level introduced by the DPCD. With the increase of 3.09 times the phase noise at synthesizer output will be -135+ _9,8≈-bore 125 dB / Hz. To provide a phase noise less than -135 dB _bore / Hz is necessary to use a reference oscillator 11, constructed according to a more complex scheme, which uses high-frequency oscillator synchronization stabilized high-Q SAW resonator or dielectric resonator, the high frequency signal of the crystal oscillator with a frequency of 100 MHz using a narrow-band phase locked loop (PLL). The PLL loop band is selected to be approximately equal to the detuning frequency from the carrier, at which the spectral density of the phase noise of the crystal oscillator, taking into account the multiplication factor, is 40 times equal to the spectral density of the phase noise of the synchronized high-frequency generator. For example, a low-noise generator stabilized by a high-quality SAW resonator with an output frequency of 500 MHz has the following spectral density values of the power of spectral fluctuations: when detuning from a carrier at 100 Hz, about -105 dB _ns / Hz, when detuning from a carrier at 1 kHz, about -135 dB _nsec / Hz, when tune away from the carrier more than 50 kHz less than -170 dB _nsec / Hz. When multiplying its frequency by 8 times for detunings of more than 50 kHz, the level of the spectral power density of the frequency fluctuations will be less than -152 dB _nsec / Hz, which coincides with the phase noise level of digital frequency dividers. The optimal PLL loop bandwidth will be approximately 1.5 kHz.

To ensure the declared switching frequency of the output frequency, it is necessary to use a high-speed PKU 3. In the case when a broadband synthesizer is used as a master oscillator in communication systems or in radar stations, a relatively small (from 10 to 10,000) number of operating frequencies is usually required. Moreover, all the values of the division coefficients of the DPKD 4.1-4.4, 10, 15, and 23 and the control codes of the DSPC 19 can be calculated in advance and stored in the memory of the PCU 3. With this construction, it is easy enough to provide a frequency update rate close to the frequency switching speed (200 ns). When using the inventive synthesizer as a signal generator, the required values in the general case must be calculated for each frequency value and it is very difficult to ensure a frequency update rate of the order of 200 ns due to the finite speed of computing devices. To simplify the computational algorithms, it is advisable to preliminarily calculate the boundary frequencies of the tuner band of the output frequency of the synthesizer for all used combinations of division coefficients of the DPKD 4.1-4.4, 10, 15, and 23 and write them to the memory of the PKU 3. Moreover, for each specific frequency value specified with an accuracy of 0.025 Hz, the values of all the division coefficients of the DPKD 4.1-4.4, 10, 15, and 23 are determined by comparing with the calculated boundary frequencies without performing calculations, and only the value of the frequency code of the DSPC 19 is calculated.

Claims (4)

1. A broadband frequency synthesizer comprising a frequency-tunable signal source, a control command generator and frequency range extension blocks, wherein the input of the first block is connected to the output of the signal source and each of the blocks contains a mixer, a bandpass filter and a divider, characterized in that each unit for expanding the frequency tuning range, the divider is made with a switchable division factor and has a high-frequency input and a control input, while in each block a high-frequency input divides of the mixer and the input of the mixer local oscillator are connected to the input of the unit, the control input of the divider is connected to the output of the control command generator, the output of the divider is connected to the intermediate frequency input of the mixer, the output of the latter is connected through the bandpass filter to the output of the unit, and the blocks are connected so that the input of each subsequent block is connected directly with the release of the previous one. 2. The broadband frequency synthesizer according to claim 1, characterized in that to the output of the last frequency extension range extension unit, an additional frequency adjustment range extension unit is connected by means of two switches that can be controlled, the first of which has one input connected to the output of the last unit, and two outputs, the first of which is connected to the first input of the second switch, which has two inputs and one output, the second to the input of the added unit, the output of which is connected to the second input of the second about the switch. 3. The broadband frequency synthesizer according to claim 2, characterized in that it includes an additional divider with a switchable division ratio, the high-frequency input of which is connected to the output of the second switch, the control input - with the output of the control command generator. 4. The broadband frequency synthesizer according to claim 1, characterized in that the frequency-tunable signal source consists of a signal generator of the reference frequencies, a local oscillator signal generating unit comprising a mixer, a divider with a switchable division factor, a low-pass filter and a bandpass filter, node forming a frequency grid containing a digital direct synthesis synthesizer, a filter, a mixer, a band-pass filter, a switchable dividing factor divider and a low-pass filter, and a frequency conversion unit up containing a mixer and a band-pass filter, while in the local oscillator signal generating unit, the input of which is connected to the reference frequency driver, the mixer divider and the local oscillator input are connected to the input of the local oscillator signal generator, the output of the divider through the low-pass filter is connected to the intermediate frequency input of the mixer, the output of the latter is connected to a band-pass filter, the output of which is the output of the local oscillator signal generation unit, connected to the frequency grid generation unit, in which the digital input a direct synthesis synthesizer is connected to a reference frequency driver, the output is through a filter with an intermediate frequency input of the mixer, the local oscillator input is connected to the output of the local oscillator signal generation unit, the mixer output is connected through a bandpass filter with a divider, the output of which is connected through the low-pass filter to the output of the unit forming a frequency grid connected to the frequency conversion unit up, in which the input of the intermediate frequency of the mixer is connected to the output of the frequency forming unit, the input is hetero ina mixer coupled to the reference frequency signal generator, the output of the mixer is coupled through a bandpass filter with the output frequency converting unit up, which is the output of a tunable source.

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