RU2323536C1 - Method for receiving noise-like frequency-manipulated signals with continuous phase - Google Patents

Abstract

FIELD: radio engineering, possible use in radio navigation and radio communication systems for receiving noise-like frequency-manipulated signals with continuous phase.

SUBSTANCE: in accordance to the method, input signal is decomposed onto quadrature components separately for each one of two symbolic frequencies, quadrature components of signal are decoded, cophased and quadrature components of signal are combined in pairs by two frequency channels, cophased and quadrature components of mutual correlation function of received and synchronous supporting noise-like signals are composed by integrating results of conjugation of cophased and quadrature components of signal in two frequency channels on spectrum which is equal to duration of noise-like signal, with following extraction of information symbols on basis of detection of sign of cophased component of mutual correlation function.

EFFECT: increased interference resistance when receiving noise-like frequency-manipulated signals with continuous phase.

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Description

The invention relates to the field of radio engineering and can be used in radio navigation and radio communication systems for receiving noise-like frequency-manipulated signals with a continuous phase.

There is a method of receiving opposite noise-like signals, which consists in multiplying the received signal with an exact copy of the useful signal for a “single” message, integrating the result of multiplication over an interval equal to the duration of the noise-like signal, and deciding on the value of the information symbol based on the sign of the correlation integral [1].

The method provides a potentially achievable noise immunity of reception, i.e. it is optimal in case of white noise, however, it cannot be implemented in practice, since it requires the formation of an exact copy of a noise-like signal that matches the received signal in time, frequency and phase.

A known method of receiving opposite noise-like signals, which consists in dividing the input signal into quadrature components by multiplying it by two quadrature copies of the noise-like signal shifted by π / 2 phase angle, isolating the quadrature low-frequency components using low-pass filters, forming a reference harmonic signal with a frequency, equal to the average frequency of the received signal, using a phase locked loop, forming two quadrature copies of the received noise a signal by multiplying a medium-frequency signal with a reference code sequence generated by a self-tuning ring in time, and the reference noise-like signal for a quadrature channel is distinguished by a phase shift of π / 2 relative to the reference signal for a synchronous channel, from the output of which a demodulated signal is removed [2].

This method allows to achieve noise immunity close to potential, with high accuracy of synchronization in time and phase of the carrier.

The disadvantage of this method is the difficulty of implementation in the case of receiving noise-like frequency-manipulated signals with a continuous phase due to the difficulties associated with the formation of quadrature copies of the received noise-like signal.

A known method of receiving orthogonal frequency-manipulated signals, which consists in parallel processing of the received signal through the channel "logical units" and the channel "logical zero", each of which contains in turn two quadrature channel, multiplying the received signal with the reference signals of the frequency of the sending unit or “zero” shifted relative to each other by a phase angle π / 2, integrating the results of multiplication in each channel on an interval equal to the signal duration, deciding on the value of information symbol based on the result of comparing the squares of the modules of the output signals of the receiving channels of the “one” and “zero” packages [3].

The disadvantage of this method is the low noise immunity due to the incoherent method of receiving and using orthogonal signals instead of the opposite ones, as well as the complexity of the implementation of the receiving device in the case of noise-like frequency-manipulated signals with a continuous phase.

The present invention is intended to solve the problem of increasing the noise immunity of receiving noise-like frequency-manipulated signals with a continuous phase with the simplicity of the hardware implementation of the reception method.

The problem is solved in that in a method for receiving noise-like frequency-manipulated signals with a continuous phase, comprising decomposing the input signal into quadrature components separately for each of the two symbol frequencies by multiplying the input signal with reference harmonic signals of each of the symbol frequencies shifted relative to each other by π / 2 phase angle, switching and decoding the quadrature components of the signal separately for each of the symbol frequencies by multiplying by modi encoded code sequences, pairwise combination of in-phase and quadrature components of the signal over two frequency channels, the formation of reference coherent harmonic orthogonal signals of each of the two symbol frequencies using the phase error signal obtained by multiplying the quadrature component of the mutual correlation function of the received and synchronous reference noise-like signals with an information symbol, obtained by determining the sign of the in-phase component of the mutual correl function, the formation of reference code sequences coinciding in time with the received signal, using the time error signal obtained by subtracting the module of the mutual correlation function of the received signal and the delayed reference signal from the module of the mutual correlation function of the received signal and the leading reference signal, the formation of in-phase and quadrature components the mutual correlation function of the received and synchronous reference noise-like signals by integrating I am the results of combining common-mode and quadrature components of a signal over two frequency channels on an interval equal to the duration of a noise-like signal, with the subsequent selection of information symbols based on determining the sign of the common-mode component of the mutual correlation function.

The essence of the proposed method for receiving noise-like frequency-manipulated signals with a continuous phase is illustrated by a description of a functional diagram of a device that implements the proposed method, including a correlation receiver (Fig. 1), a code synchronization block (Fig. 2), as well as timing charts (Fig. 3).

The correlation receiver (figure 1) contains a block 1 for generating quadrature signals, including the first, second, third and fourth multipliers 2 ₁ , 2 ₂ , 2 ₃ and 2 ₄ , the signal inputs of which are combined and are the input of the correlation receiver, and the outputs are connected to the signal the inputs of the fifth, sixth, seventh and eighth multipliers 2 ₅ , 2 ₆ , 2 ₇ and 2 ₈ , respectively, while the outputs of the multipliers 2 ₅ , 2 ₇ and 2 ₆ , 2 _{8 are} combined through the first and second adders 3 ₁ and 3 _3, respectively. The first and second integrators 4 ₁ and 4 ₂ with a reset, respectively connected to the outputs of the adders 3 ₁ and 3 ₂ , a decision block 5, the input of which is connected to the output of the integrator 4 ₁ with a reset, and the output is connected to one of the inputs of the multiplier 2 ₉ connected on the other input to the output of the integrator 4 ₂ with reset. The output of the multiplier 2 _{9 is} connected to the first loop filter 6, connected in series with the tunable generator 7 of the carrier frequency and the synthesizer 8 of the reference signals, and the first, second, third and fourth outputs of the latter are connected respectively to the reference inputs of the multipliers 2 ₁ , 2 ₂ , 2 ₃ and 2 ₄ . A code synchronization unit 9 is connected to the outputs of the multipliers 2 ₁ , 2 ₂ , 2 ₃ , 2 ₄ , the first and second outputs of which are connected to the pairwise connected reference inputs of the multipliers 2 ₅ , 2 ₆ and 2 ₇ , 2 _8, respectively, and the third output combined synchronizing inputs of integrators 4 ₁ and 4 ₂ with a reset and a decision block 5, the output of which is the output of the demodulator.

Block 9 code synchronization (figure 2) contains two channels 10 ₁ and 10 ₂ temporary discriminators, each of which includes the tenth, eleventh, twelfth, thirteenth multipliers 2 ₁₀ , 2 ₁₁ , 2 ₁₂ , 2 ₁₃ , the signal inputs of which are connected to the outputs multipliers 2 ₁ , 2 ₂ , 2 ₃ and 2 _4, respectively, of block 1 for generating quadrature signals, the outputs of multipliers 2 ₁₀ and 2 ₁₂ , 2 ₁₁ and 2 _{13 are} combined respectively through the third and fourth adders 3 ₃ and 3 ₄ , the outputs of the latter are respectively connected to inputs third and fourth integrators April ₃ and April ₄ with Sat. osom, integrators outputs 4 ₃ and 4 ₄ through the fourteenth and fifteenth multipliers February ₁₄ and 2 _15, respectively connected to the inputs of the fifth adder March _5, which is connected to the output device 11 taking the square root. Output device 11 taking the square root of each channel portion 10 ₁ and 10 ₂ is connected to a subtractor 12 which is connected to the series-connected to the second loop filter 13 controlled by the timing generator 14, the generator 15, the code sequence to the output of which is connected to the inverter 16 code sequences and the generator 17 pulses . The first and second outputs of the code sequence converter 16 are connected to the combined reference inputs of the multipliers 2 ₅ , 2 ₆ and 2 ₇ , 2 _8, respectively, the third and fourth, fifth and sixth outputs of the code sequence converter 16 are connected, respectively, to the combined reference inputs of the multipliers 2 ₁₀ , 2 ₁₁ and 2 ₁₂ , 2 ₁₃ channels 10 ₁ and 10 _{2 of the} temporary discriminator, the output of the pulse shaper 17 is connected to the synchronizing inputs of the integrators 4 ₁ , 4 ₂ , 4 ₃ and 4 ₄ with a reset. The control input of the code sequence generator 15 is connected through the delay control unit 18 to the output of the search unit 19, the first, second, third and fourth inputs of the search unit 19 are connected to the outputs of the multipliers 2 ₁ , 2 ₂ , 2 ₃ and 2 _4, respectively, of the quadrature signal generating unit 1 .

The method of receiving noise-like frequency-manipulated signals with a continuous phase is as follows. The correlation receiver performs the functions of a demodulator of the received signal, as well as a synchronization device, and can operate in two modes: search by the delay time (initial mode) and tracking the delay and phase of the received signal. In search mode, code synchronization of the received and reference noise-like signals (SHPS) is carried out with an accuracy of up to half the duration of the ShSS element. In the tracking mode, accurate synchronization is performed by the delay time and the phase of the carrier frequency, and digital information is extracted (demodulation).

The input noise-like frequency-shift keyed signal (FM-SHPS) with a continuous phase can be represented as

where D is an information symbol taking values ± 1; f ₀ is the average frequency; φ is the initial phase (the amplitude is assumed to be unity); T _P - repetition period equal to the duration of the information symbol; θ (t) is a function that determines the law of frequency manipulation (Fig.3, a, b):

,

,

where d ₁ - element of the code sequence d ₀ , d ₁ , ..., d _N-1 , taking values ± 1; N is the length of the code sequence; rect (t) is a rectangular pulse of unit amplitude and duration T.

Input multipliers 2 ₁ , 2 ₂ , 2 ₃ , 2 ₄ blocks 1 of the formation of quadrature signals multiply the signal (1) with coherent harmonic orthogonal signals of symbol frequencies f ₁ = f ₀ + 1 / 4T and f ₂ = f ₀ -1 / 4T (manipulation index m _f = 0.5) generated by the synthesizer 8 reference signals.

At the outputs of the multipliers 2 ₁ , 2 ₂ , 2 ₃ and 2 _4, respectively, quadrature components of the input signal are formed separately for each of the symbol frequencies f ₁ and f ₂ (Fig. 3, c-c):

When recording (3), it was taken into account that the components of the total frequencies f _j + f _k (j = 1, 2; k = 1, 2) are completely suppressed by the subsequent signal processing path.

и

в (3) описывают изменение фазы опорных сигналов частот f₁ и f₂ соответственно относительно фазы сигнала средней частоты f₀. При совпадении частоты i-го элемента ШПС и частоты опорного сигнала квадратурные низкочастотные составляющие (3) имеют вид прямоугольных импульсов длительности Т и амплитудой, равной

(полярность импульсов определяется структурой кодовой последовательности). При несовпадении частот элемента ШПС и опорного сигнала на выходе перемножителя формируется колебание в виде полупериода частоты f_T/2=1/2T, начальная фаза которого равна либо φ, либо φ+π в соответствии со структурой кодовой последовательности.Functions

and

in (3) describe the phase change of the reference signals of frequencies f ₁ and f _2, respectively, relative to the phase of the signal of the middle frequency f ₀ . With the coincidence of the frequency of the i-th element of the BBA and the frequency of the reference signal, the quadrature low-frequency components (3) have the form of rectangular pulses of duration T and amplitude equal to

(the polarity of the pulses is determined by the structure of the code sequence). If the frequencies of the BSS element and the reference signal do not coincide, an oscillation is formed at the output of the multiplier in the form of a half-period of frequency f _T / 2 = 1 / 2T, the initial phase of which is equal to φ or φ + π in accordance with the structure of the code sequence.

The quadrature components (3) of the signal are fed to the inputs of multipliers 2 ₅ , 2 ₆ , 2 ₇ , 2 ₈ , where they are switched and decoded by multiplying by modified code sequences (Fig. 3, l, m):

respectively, for channels of frequency f ₁ and frequency f ₂ . Elements of sequences (4) are associated with elements of the original code sequence (2), which determines the law of frequency manipulation, by the following relationships:

where l and m are respectively the number of characters d _j = -1 and d _j = 1 in the sequence d ₀ , d ₁ , ..., d _i-1 . Elements of modified code sequences (5) can take values 0, ± 1, and if element a _i = ± 1, then element b _i = 0 and vice versa, if b _i = ± 1, then a _i = 0.

The decoding of the quadrature signals (3) is carried out by multiplying the ith element by -1 if the number of packages of the alternative frequency at the previous positions (from 0 to i-1) is odd, as a result of which the phase difference θ (t) -θ _j (t) = π (j = 1,2; t∈ [(i-1) T, iT]).

The formation of code sequences (4) is performed by block 9 code synchronization. The output signals of the multipliers 2 ₅ , 2 ₆ and 2 ₇ , 2 _8, respectively, are combined in pairs in the adders 3 ₁ and 3 ₂ , forming the full quadrature components of the signal (Fig.3, n, o):

Integrators 4 ₁ and 4 ₂ with reset, gated by pulses generated by the code synchronization block 9, calculate the correlation integrals of the form

, sign(x) - знаковая функция.The output value z _{1 of the} in-phase channel is supplied to the decision block 5, which forms an estimate of the information symbol

, sign (x) is a sign function.

The output value z _{2 of the} quadrature channel is used to generate a phase mismatch signal in accordance with the algorithm

implemented multiplier 2 ₉ . The loop filter 6 smooths out fluctuations in the error signal (7) caused by the influence of noise at the input of the receiver and generates a control signal for a tunable generator 7 that generates a carrier frequency signal f ₀ in phase with the received signal (1). The carrier frequency signal is used to generate coherent reference signals of symbol frequencies f ₁ and f ₂ (block 8).

Code synchronization of the correlation receiver is carried out by block 9 code synchronization (figure 2). The signal processing algorithms in the linear part of each channel 10 ₁ and 10 _{2 of the} temporary discriminator of block 9 (up to the inputs of the multipliers 2 ₁₄ and 2 _{15 are} similar to the signal processing algorithms in the correlation receiver (Fig. 1). The only difference is that the reference code sequences in channels 10 ₁ and 10 ₂ are shifted relative to synchronous code sequences (4) by + T / 2 and -T / 2, respectively.

The quadrature components of the signal at the outputs of the adders 3 ₃ and 3 ₄ channels 10 ₁ and 10 ₂ respectively have the form

I ₊ (t) = a (t + T / 2) I ₁ (t) + b (t + T / 2) I ₂ (t),

Q ₊ (t) = a (t + T / 2) Q ₁ (t) + b (t + T / 2) Q ₂ (t),

I _- (t) = a (tT / 2) I ₁ (t) + b (tT / 2) I ₂ (t),

Q _- (t) = a (tT / 2) Q ₁ (t) + b (tT / 2) Q ₂ (t),

where a (t ± T / 2) and b (t ± T / 2) are the reference code sequences (the “+” sign corresponds to channel 10 ₁ , and “-” to channel 10 _{2 of the} temporary discriminator).

The output values of the integrators 4 ₃ and 4 ₄ channels 10 ₁ and 10 ₂ are formed as

where R (τ) is the modulus of the normalized autocorrelation function of the FM-SHPS (1); τ - synchronization error (time alignment error of the complex envelopes of the received and reference signals).

The time mismatch signal is generated by the subtractor 12 by comparing the output signal modules of the channels 10 ₁ and 10 _{2 of the} discriminator:

The delay of the reference code sequences is controlled by supplying a control signal from the output of the loop filter 13 to the controlled clock 14. The code sequence generator 15 generates a periodic pseudorandom sequence of length N, which is then converted into synchronous reference code sequences intended for the correlation receiver, as well as leading and delayed reference sequences for channels 10 ₁ and 10 _{2 of the} temporary discriminator.

In the search mode, the code sequence generator 15 is controlled by supplying a control signal from the search unit 19 to the delay control unit 18.

The search unit 19 searches for the LNB by the delay time and the initial setting of the code sequence generator 15. Examples of the implementation of a search device in the form of a multichannel correlation receiver that calculates the cross-correlation function (VKF) of the complex envelope of the input signal with all its possible copies for discrete values of the delay time and the choice of the reference signal delay value in the channel with the maximum VKF value as an estimate are given in [4 ]. The signals to the input of the search unit 19 come from the outputs of the multipliers 2 ₁ , 2 ₂ , 2 ₃ , 2 _{4 of the} block 1 of the formation of quadrature signals. Each channel of the M-channel search block 19 (M≤N) is made in the same way (Fig. 2) as the channels 10 ₁ , 10 _{2 of the} temporary discriminator (the delay discrete reference code sequences are T).

The proposed method for receiving FM-ShPS with a continuous phase provides noise immunity close to potentially achievable.

Immunity demodulator characterized probability P _err erroneous reception, which in case of perfect synchronization (optimal coherent detection) is defined as [5]

P _OSH = 1-F (q),

- отношение сигнал/шум на выходе синфазного канала (выходная величина z₁ в (6) при фазовой ошибке φ=0 равна DE);

- энергия сигнала (1); N₀/2 - спектральная плотность мощности белого шума.where Φ (q) is the probability integral,

- the signal-to-noise ratio at the output of the common-mode channel (the output value z ₁ in (6) with a phase error φ = 0 is equal to DE);

- signal energy (1); N _0/2 is the spectral power density of white noise.

и

установившихся флуктуационных ошибок (соответственно по задержке и фазе).The accuracy of synchronization can be characterized by the variance

and

steady-state fluctuation errors (delay and phase, respectively).

With high accuracy of synchronization, error dispersion can be defined as [6]

where N _τ and N _φ are the spectral density of the equivalent time and phase fluctuations, respectively, reduced to the input of the discriminator (time and phase for blocks of code and phase synchronization, respectively); F _Шτ , F _Шφ is the noise band of delay and phase tracking systems (code and phase synchronization blocks, respectively).

It can be shown that the variances of equivalent fluctuations are determined by the expressions

With high accuracy of synchronization (q >> 1), formulas (8) coincide with the known results for the variances of maximum likelihood estimates for the delay and signal phase [7]:

where F _e = 1 / 4T is the effective width of the spectrum of the signal (1).

Thus, the proposed method for receiving noise-like frequency-manipulated signals with a continuous phase allows to achieve noise immunity of the reception, which is close to potentially achievable, with low hardware costs for its implementation. This is the technical and economic effect in comparison with the known methods of quadrature reception of frequency-manipulated signals.

Information sources

1. G.I. Tuzov. Statistical theory of the reception of complex signals. - M .: Owls. Radio 1977, p. 301 (Fig. 6.11).

2. Radio engineering systems / Yu.P. Grishin, V. P. Ipatov, Yu. M. Kazarinov and others. Ed. Yu.M. Kazarinova. - M .: Higher. school, 1990, p. 314 (Fig. 14.10).

3. B.R. Levin. Theoretical foundations of statistical radio engineering. Book 2. - M .: Owls. Radio, 1968, p. 394 (Fig. 29).

4. Radio engineering systems / Yu.P. Grishin, V. P. Ipatov, Yu. M. Kazarinov and others. Ed. Yu.M. Kazarinova. - M .: Higher. school, 1990, p. 64 (Fig. 3.16).

5. Radio engineering systems / Yu.P. Grishin, V. P. Ipatov, Yu. M. Kazarinov et al. Ed. Yu.M. Kazarinova. - M .: Higher. school, 1990, p. 63 (file 3.43).

6. Radio engineering systems / Yu.P. Grishin, V. P. Ipatov, Yu. M. Kazarinov et al. Ed. Yu.M. Kazarinova. - M .: Higher. school., 1990, p.332 (f-ly 15.17, 15.18).

7. Radio engineering systems / Yu.P. Grishin, V. P. Ipatov, Yu. M. Kazarinov et al. Ed. Yu.M. Kazarinova. - M .: Higher. school, 1990, p.127 (file 5.2), p.139 (file 5.14).

Claims (1)

A method of receiving noise-like frequency-manipulated signals with a continuous phase, which consists in decomposing the input signal into quadrature components separately for each of the two symbol frequencies by multiplying the input signal with reference harmonic signals of each of the symbol frequencies shifted relative to each other by the phase angle π / 2, characterized in that the quadrature components of the signal are decoded separately for each of the symbol frequencies by multiplying by the corresponding modified code sequences whose elements take values (0,1, -1) and are uniquely associated with elements of a binary code sequence that determines the law of frequency manipulation of the received noise-like signal, pairwise combination of in-phase and quadrature components of the signal over two frequency channels, signal generation with a frequency equal to the carrier frequency of the received noise-like signal using the phase-mismatch signal obtained by multiplying the quadrature cross-correlation component the functions of the received and synchronous reference noise-like signals with an information symbol obtained by determining the sign of the in-phase component of the mutual correlation function, generating reference harmonic signals of each of the symbol frequencies shifted relative to each other by a phase angle π / 2, forming reference code sequences matching in time with the received signal using the time mismatch signal obtained by subtracting the module of the mutual correlation function of the received s channel and the delayed reference signal from the module of the mutual correlation function of the received signal and the leading reference signal, the formation of in-phase and quadrature components of the mutual correlation function of the received and synchronous reference noise-like signals by integrating the results of combining the in-phase and quadrature components of the signal over two frequency channels on an interval equal to the duration of the noise-like signal, followed by the selection of information symbols based on the definition of the sign in phase th component of the mutual correlation function.

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