RU2307474C1 - Method for receipt of noise-like signals with minimal frequency manipulation - Google Patents

Abstract

FIELD: radio engineering, possible use in radio navigation and radio communication systems for receiving noise-like signals with minimal frequency manipulation.

SUBSTANCE: in the method for receiving noise-like signals in search mode, code synchronization is set between received and supporting noise-like signals with precision up to duration of the element, and then in tracking mode precise synchronization is set based on delay time and phase of bearing frequency and demodulation. The method for receiving noise-like signals includes division of input signal onto quadrature components by means of multiplication of input signal and supporting harmonic signals of bearing frequency, integration of multiplication results in each quadrature channel separately on two intervals, equal to double duration of noise-like signal element and shifted from each other for element duration, creating correlations z_1ci, z_1si, z_1ci and z_2si at each interval, decoding and coherent accumulation of element-wise processing in four channels on interval, which is equal to the duration of the noise-like signal.

EFFECT: increased interference resistance of noise-like signal receipt with minimal frequency manipulation.

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Description

The invention relates to the field of radio engineering and can be used in radio navigation and radio communication systems for receiving noise-like signals with minimal frequency manipulation.

There is a method of correlation reception of phase telegraphy signals, which consists in multiplying the received signal with an exact copy of the useful signal to send a “unit”, integrating the result of multiplication by the duration of the information symbol and deciding on the value of the information symbol based on the sign of the correlation integral [1]. The method allows to realize the potential noise immunity in any form of signal (both for simple and complex signals) with perfect synchronization of the received and reference signals in time, frequency and phase. The practical implementation of the correlation method as applied to noise-like phase-manipulated signals is possible using either a coherent synchronization system [2] or an incoherent system [3]. In the first case, practically potential noise immunity of the reception is achieved if the synchronization errors in the delay and phase are negligibly small. The incoherent synchronization system loses in noise immunity to the coherent system, but it allows you to save the capture state in the carrier tracking circuit under conditions of strong interference, in search mode, etc.

However, this method cannot be used to receive noise-like frequency-manipulated signals, since the reference signals for phase detection in the quadrature channels of the phase discriminator cannot be generated by multiplying the carrier signal by a code sequence. In addition, the use of low-pass filters instead of integrators in the quadrature channels of the phase discriminator leads to losses in noise immunity.

A device for receiving complex phase-shifted signals is known, containing the first and second multipliers, the signal inputs of which are combined, and the outputs are connected to an incoherent processing unit for orthogonal signals and an adder, the output of which is connected in series to a bandpass filter, a third multiplier and a delay tracking unit, the output of which is connected to the synchronizing input of the block of incoherent processing of orthogonal signals, to the output of the adder connected in series to the evaluation unit frequencies and amplitudes of the signal, a tunable generator and fourth and fifth multipliers combined at the input, the other inputs of which are connected respectively to the additional outputs of the delay tracking unit, and the outputs are connected to the reference inputs of the first and second multipliers, respectively, while the amplitude output of the frequency and amplitude estimation unit the signal is connected to an additional input of the delay tracking unit, and the other input of the third multiplier is connected to the output of the bandpass filter [4].

However, the known device has low noise immunity and a long synchronization time.

A known method of quadrature reception of frequency-manipulated signals with a minimum shift, which consists in dividing the input signal into quadrature components by multiplying the input signal with reference signals shifted relative to each other by π / 2, highlighting the low-frequency quadrature components; multiplying low-frequency quadrature components, forming from the result of multiplication and the output demodulated voltage signal a half-cycle frequency; the formation of the results of multiplying the quadrature components and the voltage of the half-cycle frequency of the voltage mismatch between the carrier signal and the reference signals, the formation using the voltage mismatch of the reference signals coinciding in frequency with the carrier of the frequency-manipulated signals and shifted relative to each other by π / 2, differentiation of the quadrature components, calculating the difference of the differentiated quadrature components, calculating the sum of the quadrature components, multiplied and the differences of the differentiated quadrature components by the sum of the quadrature components; subtracting the half-cycle frequency voltage from the result of multiplying the difference of the differentiated quadrature components by the sum of the quadrature components and then filtering the resulting voltage to obtain an output demodulated signal [5].

The disadvantage of this method is the low noise immunity of receiving noise-like signals with minimal frequency manipulation due to the use of low-pass filters in quadrature channels with a bandwidth equal to half the spectral width of the noise-like signal.

The present invention is intended to solve the problem of improving the noise immunity of receiving noise-like signals with minimal frequency manipulation. The problem is solved in that in a method for receiving noise-like signals with minimal frequency manipulation, including dividing the input signal into quadrature components by multiplying the input signal with reference harmonic carrier signals shifted relative to each other by π / 2, according to the invention, the results of multiplication are integrated into each quadrature calal separately at two intervals equal to twice the duration of the noise-like signal element and offset by each flax other on element length, forming on each interval correlation z _1ci, z _2ci, z _1si and z _2si, decoding and coherent accumulation piecemeal processing results in four channels at an interval equal to the duration of a noise-like signal to form correlation z _1c, z _2c, z _1s and z _2s , the formation of reference harmonic orthogonal signals with a frequency equal to the carrier frequency of a noise-like signal, using the phase mismatch signal obtained by multiplying the quadrature component of the mutual correlation function n the received and synchronous reference signals with an information symbol obtained by determining the sign of the in-phase component of the mutual correlation function, the formation of the reference clock pulses used for gating integrators performing integration, and code sequences coinciding in delay time with the received signal used in decoding using the time mismatch signal obtained by subtracting the squared module of the mutual correlation function of the received signal and the delayed reference signal from the square of the module of the mutual correlation function of the received signal and the leading reference signal, the formation of the in-phase z ₁ = z _1c -z _2s and quadrature z ₂ = z _2c + z _1s components of the mutual correlation function of the received and synchronous reference noise-like signals with subsequent the selection of information symbols based on determining the sign of the in-phase component of the mutual correlation function.

Figure 1 and 2 shows a diagram of the correlation receiver and the code synchronization block used to implement the proposed method, and figure 3 is a timing diagram explaining the operation of these devices.

The correlation receiver (Fig. 1) contains a block 1 for elementwise processing of a noise-like signal, including the first and second multipliers 2 ₁ and 2 ₂ , the signal inputs of which are combined, the first, second, third, fourth integrators 3 ₁ , 3 ₂ , 3 ₃ , 3 ₄ moreover, the inputs of the first and third integrators 3 ₁ and 3 ₃ , the second and fourth integrators 3 ₂ and 3 _{4 are} combined and connected respectively to the output of the first and second multipliers 2 ₁ and 2 ₂ , the filter 4 compresses a noise-like signal, including the third, fourth, fifth sixth multipliers February _3, 2 _4, 2 _5, 2 ₆ whose inputs are x are connected to the outputs of integrators March _1, 3 _2, 3 _3, 3 _4, respectively, first, second, third, fourth accumulators January ₅ 5 _2, 5 _3, 5 ₄ inputs of which are connected to the outputs of the multipliers February _3, February ₄ , 2 ₅ , 2 ₆ , respectively, the outputs of the accumulating adders 5 ₁ and 5 ₄ , 5 ₂ and 5 _{3 are} combined, respectively, through the first subtractor 6 ₁ and the first adder 7 ₁ , the output of which is connected to the first input of the seventh multiplier 2 ₇ , the decision block 8 whose input is connected to the output of the subtractor 6 ₁ , and the output connected to the second input of the multiplier 2 ₇ is the output ohm of the demodulator, a first loop filter 9 connected in series to the output of the multiplier 2 ₇ and a tunable generator 10, the first and second outputs of which are connected to the reference inputs of the multipliers 2 ₁ and 2 _2, respectively, the code synchronization unit 11, the first, second, third and fourth signal inputs of which are connected to outputs of integrators March _1, 3 _2, 3 ₃ and 3 _4, respectively, and a control input connected to the first output of generator 10 is being adjusted, the first and second outputs of block 11 are combined with synchronize of integrators inputs 3 _1, 3 ₂ and 3 _3, 3 _4, respectively, third and fourth outputs of block 11 are connected with joint reference inputs of multipliers February _3, 2 ₄ and 2 _5, 2 _6, respectively, and a fifth output 11 coupled to the block with the combined sync inputs of accumulating adders 5 ₁ , 5 ₂ , 5 ₃ , 5 ₄ .

Block 11 code synchronization (figure 2) contains the first channel 12 of the temporary discriminator, including the eighth, ninth, tenth, eleventh multipliers 2 ₈ , 2 ₉ , 2 ₁₀ , 2 ₁₁ , the signal inputs of which are connected to the outputs of the integrators 3 ₁ , 3 ₂ , 3 ₃ and 3 _4, respectively, of block 1 of the bitwise processing, the fifth, sixth, seventh, eighth accumulating adders 5 ₅ , 5 ₆ , 5 ₇ , 5 ₈ , the signal inputs of which are connected to the outputs of the multipliers 2 ₈ , 2 ₉ , 2 ₁₀ and 2 ₁₁ accordingly, the outputs of the accumulating adders 5 ₅ and 5 ₈ , 5 ₆ and 5 _{7 are} combined through the second subtractor 6 ₂ and the second adder 7 _2, respectively, the outputs of the latter through the twelfth and thirteenth multipliers 2 ₁₂ and 2 ₁₃ respectively, are connected to the first and second inputs of the third adder 7 ₃ , the second channel 13 of the temporary discriminator, including the fourteenth, fifteenth, sixteenth, seventeenth multipliers 2 ₁₄ , 2 ₁₅ , 2 ₁₆ , 2 ₁₇ , the signal inputs of which are connected to the outputs of the integrators 3 ₁ , 3 ₂ , 3 ₃ and 3 _4, respectively, the ninth, tenth, eleventh, twelfth accumulating adders 5 ₉ , 5 ₁₀ , 5 ₁₁ , 5 ₁₂ , the signal inputs which are connected to the outputs multipliers 2 ₁₄ , 2 ₁₅ , 2 ₁₆ and 2 ₁₇ respectively, the outputs of accumulating adders 5 ₉ and 5 ₁₂ , 5 ₁₀ and 5 _{11 are} combined through the third subtractor 6 ₃ and the fourth adder 7 _4, respectively, the outputs of the latter through the eighteenth and nineteenth multipliers 2 ₁₈ and 2 ₁₉ , respectively, are connected to the first and second inputs of the fifth adder 7 ₅ , the fourth subtractor 6 ₄ , the summing input of which is connected to the output of the adder 7 _{3 of the} first channel 12 of the temporary discriminator, and the subtracting input to the output of the adder 7 _{5 of the} second channel 13 of the temporary discriminator the second loop filter 14, the input of which is connected to the output of the subtractor 6 ₄ , the phase shifter 15, the pulse shaper 16, the first and second additional outputs of which are connected to the combined synchronization inputs of the integrators 3 ₁ , 3 ₂ and 3 ₃ , 3 ₄ , respectively, respectively, block 1 by bit processing, a frequency divider 17, the input of which is connected to the first output of the adjustable oscillator 10, and the output is connected to the control input of the phase shifter 15, the search block 18, the delay control unit 19, the code sequence generator 20 Ostey, decoder 21, while the first, second, third and fourth inputs of the search unit 18 are connected to the signal inputs of the multipliers 2 ₈ and 2 ₁₄ , 2 ₉ and 2 ₁₅ , 2 ₁₀ and 2 ₁₆ , 2 ₁₁ and 2 ₁₇ respectively, the synchronizing input the code sequence generator 20 is connected through the delay control unit 19 to the output of the search unit 18, the control input of the block 19 is connected to the output of the clock pulse generator 16, the first and second outputs of the code sequence generator 20 are connected to the combined reference inputs of the multipliers 2 ₈ , 2 ₉ and 2 ₁₀ , 2 ₁₁ respectively Thus, the third and fourth outputs of the generator 20 are connected to the combined reference inputs of the multipliers 2 ₁₄ , 2 ₁₅ and 2 ₁₆ , 2 _17, respectively, and the fifth and sixth outputs of the generator 20 are connected to the combined reference inputs of the multipliers 2 ₃ , 2 ₄ and 2 ₅ , 2 ₆ , corresponding to the filter 4 for compressing a noise-like signal, the input of the decoder 21 is connected to the additional output of the code sequence generator 20, and the output of the decoder 21 is connected to the combined clock inputs of the accumulating adders 5 ₅ , ..., 5 ₁₂ of the code synchronization block 11 and 5 ₁ ,. ..,5 ₄ filters 4 compression noise-like signal.

The method of receiving noise-like signals with minimal frequency manipulation is as follows. The correlation receiver can operate in two modes: search by the delay time (initial mode) and tracking the delay and phase of the received signal. In the search mode, code synchronization of the received and reference noise-like signals (SHPS) is established up to the duration of the signal element. In tracking mode, accurate synchronization is established by the delay time and the phase of the carrier frequency, the allocation of digital information (demodulation).

The input of the correlation receiver (Fig. 1) receives a noise-like signal with minimal frequency shift keying (MFM-SHPS) of the form

where f _о is the carrier frequency, φ is the initial phase (the amplitude is assumed to be unity); D is an information symbol; I (t) = cosθ (t) and Q (t) = sinθ (t) are the envelopes of the quadrature phase-stimulated (FM) components of the BSS, the elements of which are defined as

- функция, определяющая закон угловой модуляции; d(t) - псевдослучайная последовательность (см. фиг.3, б, в); С_i и S_i - элементы кодовых последовательностей, определяющих законы ФМ квадратурных компонентов сигнала (1), Т - длительность элемента МЧМ-ШПС.Here

- a function that determines the law of angular modulation; d (t) is a pseudo-random sequence (see Fig. 3, b, c); C _i and S _i are the elements of code sequences defining the laws of FM quadrature components of the signal (1), T is the duration of the element MFM-SHPS.

Timing diagrams (Fig. 3) are given for steady-state operation under the assumption that there is no noise, and the errors of code and phase synchronization are negligible (for example, the length of the pseudo-random sequence N = 7).

The input multipliers 2 ₁ and 2 ₂ multiply the received signal (1) with reference harmonic signals сs2πf ₀ t and sin2πf ₀ t of the carrier frequency generated by the tunable generator 10.

At the outputs of the multipliers 2 ₁ and 2 ₂ are formed of low-frequency components, respectively

(see figure 3, g, d), as well as the components of the doubled frequency 2f ₀ , which are filtered out by the subsequent processing path.

In block 1 of the element-by-element processing of SHPS, correlations are formed (see Fig. 3, e, g) by integrating the results of multiplication at intervals corresponding to the ith element of low-frequency signals (2):

The gating of integrators 3 ₁ , ... 3 _{4 of the} block 1 of the bitwise processing of the NPS is performed by pulses generated by the code synchronization block 11. The results (3) of the element-by-element processing of SHPS (see Fig. 3, h, and) are supplied to the signal inputs of multipliers 2 ₃ , ... 2 ₆ , filter 4 of compression SHPS, where phase manipulation is removed by multiplying by the elements of code sequences C ₀ , C ₁ , ... C _M-1 and S ₀ , S ₁ , ... S _M-1 (see FIG. 3, k, l) generated by the code sequence generator 20 of the code synchronization block 11. The accumulating adders 5 ₁ , ... 5 ₄ are used to coherently accumulate the results of (3) bitwise processing of the NPS on the observation interval [0, T _s ] defined by the pulses of the decoder 21 of the code synchronization block 11.

The output values of accumulating adders, respectively, 5 ₁ , 5 ₂ , 5 ₃ and 5 ₄ , taking into account the above, can be represented as

where M = (N + 1) / 2 is the number of elements of quadrature FM-SHPS on the observation interval t∈ [0, T _s ], N is the length of the pseudo-random sequence d ₀ , d ₁ , ... d _N.

информационного символа, sign(x) - знаковая функция. Составляющая z₂ поступает на сигнальный вход перемножителя 2₇ фазового дискриминатора, на опорный вход которого подается оценка

для снятия цифровой модуляции в составляющей z₂. Сигнал фазового рассогласования с выхода перемножителя 2₇ поступает на петлевой фильтр 9, формирующий сигнал управления частотой и фазой подстраиваемого генератора 10.Values (4) are combined in pairs in the adder 7 ₁ and subtracter 6 ₁ , forming, respectively, the quadrature component z ₂ = z _2C + z _1S and the in-phase component z ₁ = z _1C -z _{2S of the} output value of the filter of the NPS compression 4 (see Fig. 3 , m, n). Component z ₁ is input to the decision block 8 of the demodulator, which generates an estimate

information symbol, sign (x) is a sign function. Component z _{2 is} fed to the signal input of the multiplier 2 ₇ phase discriminator, the reference input of which is evaluated

to remove digital modulation in component z ₂ . The phase mismatch signal from the output of the multiplier 2 _{7 is} fed to a loop filter 9, which generates a frequency and phase control signal for the adjustable oscillator 10.

Block 11 code synchronization (figure 2) works as follows. The signal inputs of the multipliers 2 ₈ , ... 2 _{11 of the} first channel 12 of the temporary discriminator with the leading reference signal, as well as the multipliers 2 ₁₄ , ... 2 _{17 of the} second channel 13 of the discriminator with the delayed reference signal, receive the results (3) of the element-by-element SHPS processing with corresponding outputs of integrators 3 ₁ , ... 3 _{4 of} block 1 of the correlation receiver (Fig. 1). The reference inputs of the multipliers 2 ₁₀ , 2 ₁₁ and 2 ₈ , 2 _{9 of the} first channel 12 are respectively supplied with a code sequence C ₀ , C ₁ , ..., C _M-1 and a cyclic shift S _M-1 , S ₀ , ... S _M-2 sequences S ₀ , S ₁ , ... S _M-1 , a the code sequence S ₀ , S ₁ , .. respectively are applied to the reference inputs of the multipliers 2 ₁₄ , 2 ₁₅ and 2 ₁₆ , 2 _{17 of the} second channel 13, respectively. .S _M-1 and cyclic shift C ₁ , C ₂ , ..., C _{0 of the} code sequence C ₀ , C ₁ , ..., C _M-1 . These sequences are generated by the generator 20 code sequences.

The accumulating adders 5 ₅ , ... 5 ₈ and 5 ₉ , ... 5 _12, respectively, of the first and second channels 12 and 13 carry out coherent accumulation of decoded results (3) of processing elements of quadrature FM-SHPS on the observation interval in accordance with the algorithm

Argument “1” in correlations (5) corresponds to the first channel 12 leading by T, and the argument “-1” to the second channel 13 with the reference signal delayed by T relative to the reference signal of the correlation receiver (Fig. 1).

и

на выходах сумматоров 7₃ и 7₅.The output values (5) of the accumulating adders 5 ₆ , 5 ₇ and 5 ₅ , 5 ₈ , as well as 5 ₁₀ , 5 ₁₁ and 5 ₉ , 5 ₁₂ are combined in pairs in the adders 7 ₂ , 7 ₄ and subtractors 6 ₂ , 6 ₃ , respectively forming the quadrature components z ₂ (1) = z _2C (1) + z _1S (1), z ₂ (-1) = z _2C (-1) + z _1S (-1) and in-phase components z ₁ (1) = z _1C (1) -z _2S (1), z ₁ (-1) = z _1C (-1) -z _2S (-1) of the output values of the channels 12 and 13 of the temporary discriminator. In multipliers 2 ₁₂ , 2 ₁₃ and 2 ₁₈ , 2 _19, these values are squared, and then combined, forming squares of modules

and

at the outputs of adders 7 ₃ and 7 ₅ .

The time mismatch signal is generated by the subtractor 6 _{4 of the} temporary discriminator and fed to the loop filter 14. Clock pulses for the code sequence generator 20 are generated by dividing the carrier frequency in the frequency divider 17. Control of the delay of the clock pulses is carried out by applying a control signal from the output of the loop filter 14 to the controlled phase shifter 15, the output of which is connected to the shaper 16 clock pulses. The latter also generates pulses for gating integrators 3 ₁ , ... 3 ₄ block 1 bitwise processing (figure 1). Generator 20 generates synchronous reference code sequences for multipliers 2 ₃ , ... 2 ₆ of correlation receiver unit 4, as well as reference code sequences ahead of T and delayed by T for multipliers 2 ₈ , ... 2 ₁₁ and 2 ₁₄ , .. .2 ₁₇ channels 12 and 13 of the discriminator, respectively. In the search mode, the code sequence generator 20 is controlled by supplying a control signal from the search unit 18 to the delay control unit 19.

The search unit 18 carries out a parallel search of the SHPS by the delay time and the initial installation of the code sequence generator 20. Examples of the implementation of the search unit in the form of a multichannel correlation receiver that calculates the cross-correlation function (VKF) of the input signal with all its possible copies for discrete values of the delay time and select, as an estimate of the delay, the delay time of the reference signal in the channel with the maximum value of the VKF, are given in [6]. In the proposed device, each channel of the search unit 18 is implemented according to the scheme of one channel (12 or 13) of the temporary discriminator (figure 2).

The proposed method for receiving noise-like signals with minimal frequency manipulation provides noise immunity close to potentially achievable.

The noise immunity of the correlation receiver (Fig. 1) is characterized by the probability P _{Osh of} erroneous reception of information symbols, which in the case of perfect synchronization (optimal coherent reception) is defined as [7]

where Ф (q) is the probability integral, q is the signal-to-noise ratio at the output of the in-phase channel of the correlation receiver.

It can be shown that the average values of the in-phase and quadrature components z ₁ and z _{2 are} determined by the expressions

,

и

,

- средние значения корреляций (3) соответственно для "косинусного" канала и "синусного" канала квадратурного преобразователя (черта сверху означает статистическое усреднение);

- энергия сигнала (1); R(τ) - нормированная ВКФ комплексных огибающих сигнала (1) и опорного сигнала, соответствующего замене в (2) косинусоидальной весовой функции прямоугольным импульсом единичной амплитуды и длительности 2Т; τ и φ - ошибки кодовой и фазовой синхронизации соответственно.Where

,

and

,

- average values of correlations (3), respectively, for the "cosine" channel and the "sine" channel of the quadrature transducer (the bar above means statistical averaging);

- signal energy (1); R (τ) is the normalized VKF of the complex envelopes of the signal (1) and the reference signal, corresponding to the replacement in (2) of the cosine weight function by a rectangular pulse of unit amplitude and duration 2T; τ and φ are the errors of code and phase synchronization, respectively.

и

для средних значений (7) можно записатьWith high accuracy of synchronization, when the errors τ and φ can be neglected, setting

and

for average values (7), we can write

Dispersion of each of the output values of the quadrature channels of the correlation receiver

- дисперсия величин (3), не зависящая от номера i элемента ШПС;Where

- the variance of the quantities (3), not depending on the number i of the NPS element;

N _0/2 is the spectral density of the noise power; E ₀ - energy of a harmonic reference signal of duration 2T.

The signal-to-noise ratio in this case, taking into account (8), (9), is equal to

- отношение сигнал/шум при оптимальном корреляционном способе приема (опорный сигнал - точная копия принятого сигнала);

- множитель, характеризующий потери в помехоустойчивости предлагаемого способа по сравнению с оптимальным способом приема.Where

- signal-to-noise ratio with optimal correlation method of reception (reference signal is an exact copy of the received signal);

- a factor characterizing the loss in noise immunity of the proposed method compared to the optimal method of reception.

и

установившихся флуктуационных ошибок (соответственно по времени и фазе), которые при высокой точности синхронизации можно определить как [8]The accuracy of synchronization can be characterized by the variance

and

steady-state fluctuation errors (respectively in time and phase), which with high accuracy of synchronization can be defined as [8]

,

- дисперсии эквивалентных временных и фазовых флуктуаций, приведенных ко входу дискриминатора (соответственно временного (ВД) и фазового (ФД);

,

- дисперсии флуктуаций на выходе дискриминатора (временного и фазового); k_ВД, k_ФД - крутизна дискриминационной характеристики (соответственно ВД и ФД);

- шумовая полоса систем слежения за задержкой и фазой (систем кодовой и фазовой синхронизации соответственно).Where

,

- variances of equivalent temporal and phase fluctuations reduced to the input of the discriminator (respectively, temporary (VD) and phase (FD);

,

- variance of fluctuations at the output of the discriminator (time and phase); k _VD , k _FD - steepness of the discriminatory characteristics (respectively VD and FD);

- noise band of delay and phase tracking systems (code and phase synchronization systems, respectively).

It can be shown that the variances of equivalent fluctuations are determined by the expressions

where η ² = 8 / π ² (-0.9 dB) is a parameter characterizing the energy loss of the proposed method of reception compared with the optimal correlation method. For η = 1 (q = q _opt ≫1), formulas (12) coincide with the known results for the variances of the maximum likelihood estimates for the delay and signal phase [9]:

where F _e is the effective width of the spectrum of the signal (for signal (1) F _e = 1 / 4T).

The noise immunity of the search block is characterized by the probability Р _{Ош of the} erroneous search termination, which with the length of the pseudo-random sequence N≫1 can be defined as the probability of recognition error M of orthogonal signals [10]:

where r is the VKF value during a temporary detuning | τ | ≤T (in the worst case, at τ = ± T, r = 0.5). To ensure a given error probability (13), the proposed method requires a 1 / η = q _opt / q times greater signal-to-noise ratio than the optimal search method (M-channel correlation receiver with quadrature channels).

Thus, the proposed method for receiving noise-like signals with minimal frequency manipulation makes it possible to achieve reception noise immunity close to potentially achievable (energy loss less than 1 dB), with little hardware and computational costs for the implementation of the proposed method for receiving SHPS with minimal frequency manipulation. This is the technical and economic effect in comparison with the known methods of quadrature reception of frequency-manipulated signals.

Information sources

1. G.I. Tuzov. Statistical theory of the reception of complex signals. - M .: Owls. Radio 1977, p. 301 (Fig. 6.11).

2. Radio engineering systems / Yu.P. Grishin, V. P. Ipatov, Yu. M. Kazarinov et al. / Ed. Yu.M. Kazarinova. - M .: Higher. school, 1990, p. 314 (Fig. 14.10).

3. Radio engineering systems / Yu.P. Grishin, V. P. Ipatov, Yu. M. Kazarinov et al. / Ed. Yu.M. Kazarinova. - M .: Higher. school, 1990, p. 318 (Fig. 14.12), p. 328 (Fig. 15.3).

4. A.S. 1046943 (USSR). Correlation receiver of complex phase-manipulated signals / V.I.Zhuravlev, N.P. Trusevich. Publ. 10/07/83. Bull. VNIIIPI No. 37.

5. Patent 2192101 (RF). The method of quadrature reception of frequency-manipulated signals with a minimum shift / A.M. Karlov, E.V. Volkhonskaya. E.N. Avdeev. Publ. 10/27/2002.

6. Radio engineering systems / Yu.P. Grishin, V. P. Ipatov, Yu. M. Kazarinov et al. / Ed. Yu.M. Kazarinova. - M .: Higher. school, 1990, p. 64 (Fig. 3.16).

7. Radio engineering systems / Yu.P. Grishin, V. P. Ipatov, Yu. M. Kazarinov et al. / Ed. Yu.M. Kazarinova. - M .: Higher. school, 1990, p. 63 (file 3.43).

8. Radio engineering systems / Yu.P. Grishin, V. P. Ipatov, Yu. M. Kazarinov et al. / Ed. Yu.M. Kazarinova. - M .: Higher. school., 1990, p.332 (f-ly 15.17, 15.18).

9. Radio engineering systems / Yu.P. Grishin, V. P. Ipatov, Yu. M. Kazarinov et al. / Ed. Yu.M. Kazarinova. - M .: Higher. school, 1990, p.127 (file 5.2).

10. G.I. Tuzov. Statistical theory of the reception of complex signals. - M .: Owls. Radio, 1977, p. 301 (Fig. 6.11), p. 46 (file 1.98).

Claims (1)

The method of receiving noise-like signals with minimal frequency manipulation, which consists in dividing the input signal into quadrature components by multiplying the input signal with reference harmonic carrier signals shifted relative to each other by π / 2, characterized in that the multiplication results are integrated in each quadrature channel separately at two intervals equal to twice the duration of the noise-like signal element and shifted relative to each other by the duration of the element Forming on each correlation interval z _1ci, z _2ci, z _1si and z _2si, decoding and coherent accumulation piecemeal processing results in four channels at an interval equal to the duration of a noise-like signal, with the formation of the correlations of the form _{z 1c, z 2c, z 1s} z 2s , the formation of reference harmonic orthogonal signals with a frequency equal to the carrier frequency of a noise-like signal, using the phase mismatch signal obtained by multiplying the quadrature component of the mutual correlation function of the received and synchronous reference of signals with an information symbol obtained by determining the sign of the in-phase component of the mutual correlation function, generating reference clock pulses used for gating integrators integrating, and code sequences coinciding in delay time with the received signal used in decoding using a time mismatch signal obtained by subtracting the squared module of the mutual correlation function of the received signal and the delayed reference signal from the squared module of the mutual correlation function of the received signal and the leading reference signal, the formation of the in-phase z ₁ = z _1c- z _2s and quadrature z ₂ = z _2c + z _1s components of the mutual correlation function of the received and synchronous reference noise-like signals with subsequent allocation of information symbols to the basis of determining the sign of the in-phase component of the mutual correlation function.

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