RU2286005C2 - Broadband amplifier - Google Patents

Abstract

FIELD: radio engineering and communications, possible use as high frequency amplification cascades in different microelectronic devices.

SUBSTANCE: broadband differential amplifier (dwg.1) contains input parallel-balance cascade 1 on input transistors 2,3 with a source of supporting current 4, bases are connected to first and second inputs of broadband differential amplifier, and collectors - to first and second current outputs 7,8, while source of supporting current 4 has parasitic capacitance 9. Introduced to circuit is first (second) auxiliary transistor 10, 11, base of which is connected to base of transistor 1,2, emitter is connected to first (second) additional supporting current source 11,12, having parasitic capacitance 13,14, collector of transistor 10,11 is connected to second (first) current output 8,7 through first (second) current divider 15,16.

EFFECT: increased upper limit frequency.

3 cl, 8 dwg

Description

The invention relates to the field of radio engineering and communications and can be used in the structure of broadband analog microcircuits for various functional purposes.

Known broadband amplifiers based on differential stages, which became the basis for the construction of modern analog microcircuits of the radio frequency range [1, 2]. The problem of expanding their bandwidth is one of the urgent problems of modern analog microcircuitry.

The closest prototype of the claimed device is a broadband amplifier (ШУ) [US Patent No. 4517525, Н 03 F 3/45], containing an input parallel-balanced stage at the first and second input transistors, the emitters of which are connected to the first reference current source, the bases are connected to the first and the second inputs of the differential amplifier, and the collectors with the corresponding first and second current outputs, and the first current source has a stray capacitance.

A significant disadvantage of the known SH is that it has relatively low values of the upper cutoff frequency of the steepness of the conversion of the input voltage to the output current.

The main objective of the invention is to increase the cutoff frequency of the steepness of the conversion of the input voltage to the output current.

This goal is achieved by the fact that in a broadband amplifier containing an input parallel-balanced cascade on the first and second input transistors, the emitters of which are connected to the first reference current source, the bases are connected to the first and second inputs of the differential amplifier, and the collectors to the corresponding first and second current outputs, and the first current source has a stray capacitance, new elements are introduced and the connections between them are the first auxiliary transistor, the base of which is connected to the base of the first input trans stories, whose emitter is connected to a first reference current source further having a parasitic capacitance, the collector of the first auxiliary transistor is connected to the second current output through the first current divider.

The proposed broadband amplifier (Fig. 1) contains an input parallel-balanced stage 1 at the first 2 and second 3 input transistors, the emitters of which are connected to a reference current source 4, the bases are connected to the first 5 and second 6 inputs of the differential amplifier, and the collectors to the corresponding the first 7 and second 8 current outputs, and the first reference current source 4 has a stray capacitance 9. To this part of the drawing of FIG. 1 there corresponds a prototype amplifier. The first 10 and second 11 auxiliary transistors are introduced into the circuit, the bases of which are connected to the corresponding bases of the first 2 and second 3 input transistors, the emitters are connected to the first and second additional reference current sources 11 and 12 having spurious capacitances 13 and 14. The collector of the first auxiliary transistor 10 is connected to the second current output 8 through the first current divider 15, and the collector of the second auxiliary transistor 11 is connected to the first current input 7 through the second current divider 16.

When identical elements are used in the circuit, when the stray capacitances 9, 13 and 14 are the same, with the same static currents of all transistors 2, 3, 10, 11, as well as with the same capacitance, the collector-base of transistors 10, 11 and 2, 3 have the optimal transmission coefficients current dividers 15 and 16 in accordance with claim 3 of the claims are selected in the range of 0.3-0.8.

Figure 2 shows an example of constructing a current divider. Due to the appropriate choice of emitter areas of the used transistors, it is possible to provide a current transfer coefficient in the range of 0.3 ÷ 0.7.

Figure 3 shows a diagram explaining the operation of the inventive device in the high frequency range.

Figure 4 shows the schematic diagram of the control circuit (for the case when the signal is removed in the collector circuit Q ₁₃ ), implemented on the basis of the base matrix crystal of NPO Integral (Minsk). It was investigated by the authors in the environment "PSpice" (figure 5). The axis "Y" in the drawing of figure 5 corresponds to the value of the slope of the conversion of the input voltage of the amplifier into its output current.

The frequency dependence of the module coefficient of conversion of the input voltage to the output current (steepness SHU) is shown in the drawing of figure 5.

The dependence of the upper cut-off frequency F _c (level - 3 dB) of the steepness of the ШУ of Fig. 4 on the current transfer coefficient K _{i of the} divider 15 (16) is shown in Fig. 6. From this graph, as well as other experiments, it follows that the extremum of the function F _c = φ (K _i ) lies in the range 0.5–0.8.

Analysis of the prototype amplifier. An equivalent circuit of the prototype ШУ for high frequencies is shown in the drawing of Fig. 3, where it is indicated: r _e2 , r _e3 are the differential resistances of the emitter junctions of transistor 2 and 3. The current complex of the emitter of transistor 3 is connected with the parameters of the circuit elements of Fig.3 with obvious relations

Where

- постоянная времени конденсатора 9;

- the time constant of the capacitor 9;

ω = 2πf is the circular frequency of the input signal.

Therefore, the collector current of transistor 3

Where

- комплексный коэффициент передачи по току эмиттера транзистора 3,

- complex current transfer coefficient of the emitter of the transistor 3,

ω_α - верхняя граничная частота транзистора 3 в схеме с общей базой.

ω _α is the upper cutoff frequency of the transistor 3 in the circuit with a common base.

Equation (2) can be reduced to the form

Where

- эквивалентная постоянная времени высоких частот.

- equivalent time constant of high frequencies.

The numerical values of ω _α for modern integrated transistors lie in the range of hundreds of megahertz - units of gigahertz, while the time constant τ _2.3 is in many cases lower frequency, especially when the transistors 2 and 3 are in micro mode. Indeed, at I _e = 10 μA, C ₉ = 5 pF, we obtain the following numerical values of the parameters:

r _e2 = r _e3 = φ _t / I _e = 2.5 kOhm, τ _2.3 = 6 · 10 ^-9 s,

where φ _m ≈25 mV is the temperature potential,

I _e - the static current of the emitter of the transistor.

Thus, in the considered example, starting from the frequency f _2.3 = 25 MHz, the steepness of the transmission of the prototype amplifier begins to decrease. This is its significant drawback, which is not eliminated in the framework of well-known circuitry solutions. The capacitance of the capacitor 9, which in real circuits consists of the capacitance on the substrate With _p = 3 ÷ 5 pF and the collector capacitance of the transistor With _K = 0.1 ÷ 0.3 pF, which runs the reference current source 4, limits the passband of the known amplifier from above the level of tens of hundreds of megahertz.

приведены на чертеже фиг.3. Из рассмотрения составляющих выходного тока (тока выхода 8) следует, что токи

,

, с учетом их фазовых сдвигов, могут взаимно компенсировать друг друга при определенных значениях K_i. Причем составляющая

определяется величиной выходной емкости 13 источника тока 13 (емкости на подложку). Найдем условия этой компенсации.Analysis of the claimed amplifier. Directions of alternating currents in the circuit of figure 1 when applying a positive half-wave of the input voltage

shown in the drawing of figure 3. From the consideration of the components of the output current (output current 8) it follows that the currents

,

, taking into account their phase shifts, can mutually cancel each other at certain values of K _i . Moreover, the component

determined by the value of the output capacitance 13 of the current source 13 (capacitance to the substrate). We find the conditions for this compensation.

In view of (1), at τ _α ≪τ _2.3 , the collector current of transistor 3

where τ _2.3 = τ _2.3 · C ₉ .

On the other hand, the collector current of transistor 10, due to a capacity of 13 C ₁₃ (at C _K10 ≪C ₁₃ ):

where r _e10 is the resistance of the emitter junction of the transistor 10,

- постоянная времени конденсатора С₁₃.

- the time constant of the capacitor C ₁₃ .

Therefore, in the high-frequency range, the total output current 8

or subject to (5) and (6)

After transformations of formula (8), we can find

Where

, τ₁₃=(r_э2+r_э3)С₁₃,

, τ ₁₃ = (r _e2 + r _e3 ) C ₁₃ ,

) последнее уравнение можно привести к видуAfter additional transformations (at

) the last equation can be reduced to

Where

- выходной ток усилителя в диапазоне низких частот.

- the output current of the amplifier in the low frequency range.

From formula (10), you can determine the formula for the normalized frequency response of the amplifier of figure 3

The last equation allows you to compare the frequency characteristics of the claimed 0 <K _i <1 and the known (K _i = 0) differential amplifiers, as well as an amplifier with an arbitrary value of K _i .

Equation (11) explains the extremum of the function M = φ (ω) for K _i = 0.5 - in its numerator there is a coefficient τ ₁₃ -K _i τ ₁₃ , which for r _e10 = r _e2 = r _e3 assumes a zero value if K _i = 0.5.

If we put K _i = 0, then from (11) we get the normalized frequency response of the prototype amplifier:

To identify a positive effect, the authors performed a numerical calculation of the coefficient M according to formula (11) in a MAPL medium with C ₁₃ = C ₉ = 5 pF, r _e2 = r _e3 = r _e10 = 25 kOhm (graph of Fig. 7) and C ₁₃ = With ₉ = 5 pF, r _e2 = r _e3 = r _e10 = 2.5 kOhm (graph of Fig. 8) at various current transfer coefficients K _i = 0 ÷ 1. An analysis of these graphs shows that there is some optimal value of K _i ≈0.5, at which the upper boundary frequency of the coefficient M improves significantly. These data are confirmed by other experimental results performed by the authors in the environment of PSpice (figure 5, 6). Moreover, a certain shift of the extremum of the function F _c = φ (K _i ) from the level K _i = 0.5 is explained by the small influence of the capacitances of the collector junctions of the transistors. In general, the gain in the upper cutoff frequency reaches 20–30 times.

BIBLIOGRAPHY

1. Alekseenko A.G. Fundamentals of microcircuitry. Elements of the morphology of microelectronic equipment / A.G. Alekseenko. - Vol. 2, revised and ext. - M., "Sov. Radio", 1977. - 408 p.

2. Design and use of operational amplifiers / Ed. J. Graham, J. Toby, L. Huhlsman // Per. from English V.I. Levina and I.M. Kheifets. Under. ed. Ph.D. I.N. Teplyuk. - M.: Mir Publishing House, 1974.

3. US patent No. 4517525, H 03 F 3/45 (prototype).

4. US patent No. 5311144, H 03 F 3/45.

5. US patent No. 427479, H 03 F 3/45.

6. US patent No. 5115205, H 03 F 3/45.

7. US patent No. 3883816, H 03 F 3/45.

8. Patent US 2004/0145414 A1, H 03 F 3/45.

9. US patent No. 4586000, H 03 F 3/45.

10. US patent No. 6060956 (figure 1), H 03 F 3/45.

11. US patent No. 3693108, H 03 F 3/45.

Claims (3)

1. A broadband differential amplifier containing an input parallel-balanced cascade on the first and second input transistors, the emitters of which are connected to a reference current source, the bases are connected to the first and second inputs of the differential amplifier, and the collectors to the corresponding first and second current outputs, and the reference source current has a parasitic capacitance, characterized in that the first auxiliary transistor is introduced into the circuit, the base of which is connected to the base of the first input transistor, the emitter is connected to the first an additional reference current source having a stray capacitance, the collector of the first auxiliary transistor is connected to the second current output through the first current divider. 2. The device according to claim 1, characterized in that the second auxiliary transistor is introduced into the circuit, the base of which is connected to the base of the second input transistor, the emitter is connected to a second additional reference current source having a stray capacitance, the collector of the second auxiliary transistor is connected to the first current output through the second current divider. 3. The device according to claim 2, characterized in that for the same values of stray capacitances of the first and additional current sources and the same values of the capacitance of the collector junctions of the input and additional transistors, the optimal current transfer coefficient of the first and second current dividers lies in the range of 0.3-0 ,8.

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