KR20150031501A - Compound loop antenna - Google Patents

Abstract

Embodiments of the present invention relate to planar (double-sided) and printed (cross-sectional) composite electric field antennas. These embodiments relate to improvements and, in particular, are not exclusively limited to achieving performance gains in coplanar field radiators and high bandwidth (low Q), high radiation intensity / power / gain, and high efficiency To a composite loop antenna having magnetic loops having an electric field orthogonal to the magnetic fields. Embodiments also relate to a self-contained counterpoise compound field antenna comprising a transition formed on the magnetic loop and having a transition width greater than the width of the magnetic loop. The transition portion generally isolates the counterpoise formed on the magnetic loop opposite or adjacent to the electric field radiator.

Description

[0001] COMPOUND LOOP ANTENNA [0002]

Preliminary notes on related uses

This application claims priority from U.S. Provisional Application No. 61 / 303,594, filed February 11, 2010, and is a continuation-in-part of U.S. Application Nos. 12 / 878,016, 12 / 878,018, filed on September 8, 2010, 12 / 878,020.

Embodiments of the present invention relate to planar (double-sided) and printed (cross-sectional) composite electric field antennas, and in particular, but not exclusively, provide coplanar field radiators and performance gains at high bandwidths (low Q) A large radiation intensity / power / gain, and magnetic loops having an electric field orthogonal to the magnetic fields to achieve high efficiency. Embodiments also relate to a self-contained counterpoise compound field antenna comprising a transition portion formed on the magnetic loop and having a transition width greater than the width of the magnetic loop. The transition portion substantially insulates the counterpoise formed on the magnetic loop opposite or adjacent to the electric field radiator.

The ever-decreasing size of modern telecommunication devices poses a need for improved antenna design. Known antennas in devices such as mobile / cellular telephones represent one of the severe limitations in performance and are included in almost any way in almost all cases.

In particular, the efficiency of an antenna can seriously affect the performance of the device. The more efficient the antenna is, the more energy is emitted from the transmitter. Similarly, due to the inherent compatibility of the antenna, the more efficient the antenna, the more received signals are converted to electrical energy for processing with the receiver.

In order to establish maximum energy transfer (in both receive and transmit modes) between the transceiver (which acts as both a transmitter and a receiver) and the antenna, the impedances of both must scale to each other. If there is any misalignment between them, in the case of transmission, the performance of the lane where energy is reflected from the antenna into the transmitter is exerted. When operating as a receiver, the performance of the lane of the antenna results in a lower receiving power than other possible cases.

Known simple loop antennas are generally current supply devices, which primarily produce a magnetic field. As such they are not generally suitable as transmitters. This is particularly pronounced in the case of small loop antennas (ie, smaller ones with diameters smaller than one wave). Conversely, voltage supply antennas such as dipoles produce both electrical (E) and magnetic (H) fields, and can be used in both receive and transmit modes.

The amount of energy received by the loop antenna, or transmitted from the loop antenna, is determined in part by its area. In general, when the area of the loop is halved each time, the amount of energy that can be received / transmitted is reduced by about 3 dB based on application parameters, such as initial size, frequency, and so on. This physical constraint is presumed to mean that very small loop antennas can not actually be used.

Composite antennas can be used in both transverse-electromagnetic (TM) and transverse electric (TE) modes for high performance gains such as high bandwidth (low Q), high radiation intensity / power / gain, Are the antennas that are energized to achieve.

In the late 1940s, Wheeler and Chu first tested the characteristics of an electrical short-range (ELS) antenna. Through his work, some numerical formulas have been formed that account for the limitations of antennas as the physical size decreases. One particularly important limitation of the ELS antennas mentioned by the wheeler and weight is that they have large radiation quality factors (Q) and store more energy in time averages than they emit. According to Wheeler and Chul, ELS antennas have a high radiological coefficient (Q), resulting in minimal resistance loss to the antenna or matching network and also very low radiation efficiency, typically between 1 and 50% . As a result, since the 1940s, the fact that ELS antennas have narrow bandwidth and poor radiation efficiency has been accepted by Science World. Most accomplishments in wireless communication systems using modern ELS antennas have resulted from thorough experimentation and optimization of modulation schemes and on air protocols, but today commercially available ELS antennas It still reflects the narrow bandwidth, low efficiency characteristics that the wheeler and the additional established.

In the early 1990s, Grassdale et al. (Dale M. Grimes and Craig A. Grimes) found mathematically that certain combinations of the TM and TE modes work together in ELS antennas that exceed the low radiation Q limits established by the theory of Wheeler and Chou. . They described their work in a publication entitled " Q and Bandwidth of Antennas Emitting TE and TM Mode ", published in May 1995, IEEE Transactions on Electromagnetic Compatibility, This argument has sparked many controversies and led to the concept of a "complex electric field antenna" in which both the TE and TM modes are excited, which is very different from the "simple electric field antenna" in which the TM or TE mode is excited singly. The gain of the composite electric field antenna is lower than the wheeler and the further limited Q, increased radiant intensity, and directivity (gain). Radiation, and radiation efficiency, U.S. Pat. It has been mathematically proven by renowned RF experts, including the group employed by the Naval Air Warfare Center Weapons Division [P. L. Overfelft, D. R. Bowling, and D. J. White, "Magnetic Loops, Electron Dipole Array Antennas (Preliminary Results)," Interim rept. September 1994].

Composite field antennas have proven to be complex and difficult to implement physically due to the difficulties associated with undesirable device coupling effects and the design of low loss passive networks for combining electrical and magnetic radiators.

There are many examples for a two-dimensional non-composite antenna consisting of a generally printed metal strip on a circuit board. However, such antennas are supplied with voltage. An example of such an antenna is a planar inverted-F antenna (PIFA). Most similar antenna designs also consist primarily of quarter wavelengths (or a multiple of a quarter wavelength), supply voltage, and dipole antennas.

Planar antennas are also known in the art. For example, U.S. Patent No. 5,061,938 to Zahn et al. Requires expensive Teflon substrates or similar materials to operate the antenna. U.S. Patent No. 5,376,942 to Shiga discloses a planar antenna capable of receiving but not transmitting microwave signals. The antenna of the above-mentioned market also requires an expensive semiconductor substrate. U.S. Patent No. 6,677,901 to Nalbandian relates to a planar antenna requiring a substrate having a permittivity to permeability ratio of 1: 1 to 1: 3, which can be used only in the HF and VHF frequency ranges 3 to 30 MHz and 30 to 300 MHz). Techniques for printing rather low frequency devices on expensive glass-reinforced epoxy laminate sheets such as FR-4, commonly used for conventional printed circuit boards, are known, but the dielectric loss in FR-4 is too high It is also contemplated to have a dielectric constant that can not be sufficiently tightly controlled for such substrates to be used for microwave frequencies. For these reasons, alumina substrates are generally used more. Also, none of these planar antennas is a composite loop antenna.

In terms of bandwidth, efficiency, gain and radiation intensity, the basis for increased performance of composite electric field antennas is formed from energy effects stored in the proximity field of the antenna. In RF antenna design, it is desirable to transmit as much radiant energy as possible to the antenna. The energy stored in the antenna proximity field has historically been referred to as a reaction force and also acts to limit the amount of power that can be emitted. When discussing complex power, there are real and imaginary (sometimes referred to as "reactive") portions. The actual power will never return from leaving the source while the virtual or reactive power has a tendency to oscillate and interact with the source (within half wavelength) of the source, thereby affecting the operation of the antenna . The presence of real power is added directly from multiple sources, but multiple sources of virtual power can be attached or subtracted (erased). The gain of the composite antenna is driven by both the TM and TE sources so that engineers can obtain a design that utilizes reactive power cancellation that was previously not available in a simple magnetic field antenna and thus improves the actual power transmission characteristics of the antenna It is in point.

In order to be able to cancel the reactive power in the composite antenna, the electric field and the magnetic field must operate at right angles to each other. Many arrangements of the electric field radiator (s) required to emit the electric field, and the magnetic loop required to generate the magnetic field have been proposed, but all such designs have been unchanged based on three-dimensional antennas. For example, U.S. Patent No. 7,215,292 issued to McLean requires a pair of magnetic loops in parallel planes having electrical dipoles on a third parallel plane located between a pair of magnetic loops . U.S. Patent No. 6,437,750 to Grimes et al. Requires two pairs of magnetic loops and electrical dipoles to be physically aligned at right angles to each other. US application US 2007/0080878, filed by McLean, discloses an arrangement in which the magnetic dipoles and electrical dipoles are located in right-angled planes.

BRIEF DESCRIPTION OF THE DRAWINGS Fig.
Fig. 2 shows a circuit layout of an embodiment incorporating four separate antenna elements; Fig.
FIG. 3A is a detailed view of one of the antenna elements of FIG. 2 including a phase tracker. FIG.
Figure 3b is a detail view of one of the antenna elements of Figure 2 that does not include a phase tracker.
4A is a view showing an embodiment of a compact cross-section type composite antenna.
Figure 4b shows an embodiment of a compact cross-sectional composite antenna with a magnetic loop in which the corners are cut at an angle of about 45 degrees.
Figure 4c illustrates an embodiment of a compact cross-sectional composite antenna with a magnetic loop having two symmetrical wide-narrow-wide transitions.
5 is a view for explaining an embodiment of a compact double faced composite antenna.
6 is a view for explaining an embodiment of a large composite antenna array composed of four composite antenna elements.
Figure 7 shows how the dimension of the phase tracker affects inductance and capacitance.
8 illustrates a ground plane of the antenna embodiment of Fig. 6; Fig.
9A is a view for explaining an embodiment of a self-contained type counterpoise antenna having a balun;
9B is a view for explaining another embodiment of the antenna according to FIG. 9A in which the balun is removed; FIG.
Figure 10A illustrates an embodiment of a self-contained counterpoise antenna having a curved trace between an array of electric field radiators and electric field radiators.
Figure 10b illustrates an embodiment of a self-contained counterpoise antenna having an array of electric field radiators but without the curved traces.
Figures 11a-11c are schematic diagrams illustrating 2D radiation patterns for the antenna according to Figure 9;
Figures 12A-12C are schematic diagrams illustrating 2D radiation patterns for the antenna according to Figure 10A;
13A is a schematic diagram illustrating a plot of voltage standing wave ratio for the antenna according to Fig. 9. Fig.
Figure 13b is a schematic diagram illustrating a chart of measured return loss for the antenna according to Figure 9;
14A is a schematic view for explaining a plot of voltage standing wave ratio for an antenna according to FIG.
Figure 14b is a schematic diagram illustrating a chart of measured return loss for the antenna according to Figure 10;
15 is a schematic diagram illustrating an embodiment of a self-contained counterpoise antenna having a tapered transition;

Embodiments provide an improved planar composite loop (CPL) antenna that can operate in both transmit and receive modes and can provide much better performance than known loop antennas. The two main elements of the CPL antenna are the magnetic loop that generates the magnetic field (field H) and the electric field radiator that emits the electric field (field E).

The electric field radiator may be physically located inside the loop or outside the loop. For example, Figure 1 shows an embodiment of a single CPL antenna element with an electric field radiator positioned on the inside of the loop bound by an electrical trace, while Figures 3a and 3b show an embodiment of a single CPL antenna element 2 shows two embodiments of a single CPL antenna element with an electric field radiator. Figure 3A includes a phase tracker for broadband use, as further described below, while Figure 3B does not include a phase tracker and is also more suitable for less broadband applications. Figures 4a, 4b and 4c illustrate other embodiments of a small cross-sectional antenna in which an electric field radiator (s) is located within the magnetic loop. Embodiments of antennas formed using any of these techniques may be readily assembled into a portable portable device, such as, for example, a telephone, PDA, laptop, and may also be assembled as a separate antenna. Figure 2 and other figures illustrate an embodiment of a CPL antenna array using microstrip forming techniques. Such a printing technique makes the antenna to be designed or manufactured to be compact and consistent.

The antenna 100 shown in Fig. 1 is arranged and printed on a section of the printed circuit board 101. Fig. The antenna comprises a magetic loop 110, in which case the loop is necessarily rectangular and generally has an open base. The two ends of the open base portion are fed from the coaxial cable 130 to the drive point in a generally known manner.

An electric field radiator or series of resonant circuits 120 is located within the loop 110. The series of resonant circuits 120 includes a J-type trace (not shown) on the circuit board 101 bound to the loop 100 by a serpentine trace 124 that acts as an inductor, which means having inductance or inductive reactance. (122). The J-type trace 122 necessarily has capacitive reactance characteristics indicated by its dimensions and the materials used for the antenna. The traces 122 together with the trapezoidal traces 124 serve as a series of resonant circuits.

Here, the antenna 100 is shown so as to be easily understood. The actual embodiment may not physically resemble the depicted antenna. One end of the loop 132 is connected to the center conductor of the cable 130 while the other end of the loop 134 is connected to the center conductor of the cable 130. In this case, Lt; RTI ID = 0.0 > 130 < / RTI > The loop antenna 100 differs from known loop antennas in that the series of resonant circuits 120 are bound to the loop 134 halfway around the loop. Such a binding position serves as an important part in the operation of the antenna, as described below.

The series of resonant circuits 120 and the traces 124 are carefully positioned relative to the magnetic loop 110 so that the electric field and the magnetic fields generated / received by the antenna 100 are transmitted to the magnetic field- But may be formed to be orthogonal to each other without having to be arranged physically orthogonal to the loop 110. [ This orthogonal relationship has the effect of enabling the electromagnetic waves radiated by the antenna 100 to effectively propagate through the space. In order to achieve such an effect, the series of resonant circuits 120 and the trapezoidal traces 124 are located at an electrical position of about 90 degrees or about 270 degrees along the magnetic loop 110. In an alternative embodiment, the trapezoidal trace 124 may be located at a point along the magnetic loop 110 where the current flowing through the magnetic loop is a reflective minimum. Thus, the trapezoidal trace 124 may or may not be located at about 90 degrees or 270 degrees electrical point. The point of the magnetic loop 110 where current is a reflection minimum depends on the geometry of the magnetic loop 110. For example, the point at which the current is the minimum of reflection may be recognized as the first area of the magnetic loop initially. After adding or removing a metal to the magnetic loop to achieve impedance matching, the point at which the current is at the minimum of reflection may change from the first region to the second region.

The magnetic loop 110 may have many different electrical and physical lengths; However, with respect to the predetermined frequency band (s), electrical lengths that are wavelengths, quarter wavelengths and multiples of 1/8 wavelength are provided for more efficient operation of the antenna. By adding inductance to the magnetic loop, the electrical length of the magnetic loop increases. By adding capacitance to the magnetic loop, the electrical length of the magnetic loop is reduced.

The vertical relationship between the magnetic field and the electric field is such that the physical location changes based on the frequency of the signals transmitted / received by the antenna to a physical location that is 90 degrees or 270 degrees from the drive point around the magnetic loop, (120) and the trapezoidal traces (124). As mentioned, such a position may be at 90 degrees or 270 degrees from the driving point (s) of the magnetic loop 110, determined by the ends 132 and 134, respectively. Thus, if the end portion 132 is connected to the center conductor of the cable 130, the trapezoidal trace 124 may be located at a 90 degree position as shown in FIG. 1, or at a 270 degree position (Not shown in FIG. 1).

The vertical relationship between the magnetic field and the electric field is also determined by locating the series of resonant circuits 120 and the trapezoidal traces 124 at a physical location around the magnetic loop where the current flowing through the magnetic loop is located at the reflective minimum, . As described above, the position at which the current reaches the reflection minimum depends on the geometry of the magnetic loop 110.

The vertical relationship between the electric field and the magnetic field, which causes the antenna 100 to function more effectively as both the receive and transmit antennas by arranging the circuit elements in such a way that the phase relationship between the elements is 90 degrees, . The magnetic field is generated solely (or necessarily solely) by the magnetic loop 110 while the electric field is provided in a form suitable for transmission over a much greater distance of energy transmitted from the antenna, Is emitted by circuit 120.

The series of resonant circuits 120 may include an inductive (L) element and a capacitive (C) element (s) so as to resonate at the operating frequency of the antenna 100 and to have a value selected to match the inductive reactance with the capacitive reactance . This is because resonance produces the best efficiency when the reactance of the capacitive element is equal to the reactance of the inductive element, i.e., X _L = X _C. Thus, the values of L and C may be selected to provide a predetermined operating range. For example, other types of resonant circuits using a crystal oscillator may be used to provide different operating characteristics. If a crystal oscillator is used, the Q-value of such a circuit is much larger than that of the simple LC circuit shown, thus limiting the bandwidth characteristics of the antenna.

As described above, the series of resonant circuits 120 effectively acts as an electric field radiator (which also means as an electric field receiver due to the reciprocity inherent in the antenna). As shown, the series of resonant circuits 120 are quarter wave antennas, but the series resonant circuits may also be multiples of full wave antennas, multiples of a quarter wavelength antenna or multiples of a quarter wavelength antenna Can operate. If a particular constraint interferes with a predetermined wavelength of the material to be used as the trace 122, a propagation delay (" propagation delay ") may be used to achieve a series of resonant circuits 120, , It is possible to use the trapezoidal trace 124. [0050] It is also to that, but may be possible in theory, in practice, instead of simply physically connected to the loop current flowing through the magnetic loop in FIG point or 90/270 positioned at minimum reflection point of the series resonant circuit _L X = X _C , it is not a common practice to use a rod antenna of a predetermined wavelength.

As noted above, the position of the series resonant circuit 120 is important: it is important that the phase difference between the electric field and the magnetic field is 90 degrees or 270 degrees, or that the current flowing through the magnetic loop Loop and can be bound. A point where the series of resonant circuits 120 are bound to the magnetic loop 110 is referred to as a "connection point ", and a connection point at 90 degrees or 270 degrees electrical point along the magnetic loop is & 270 connection point ", and the connection point where the current is at the reflection minimum is referred to as the" reflection minimum connection point ".

The position variation of the connection point depends to some extent on the intended use of the antenna and on the magnetic loop geometry. For example, an optimal connection point can be found by comparing the performance of an antenna using a 90/270 connection point versus the performance of an antenna using a reflection minimum connection point. Then a connection point may be selected which causes the highest efficiency for the intended use of the antenna. The 90/270 connection point may not be different from the reflection minimum connection point. For example, an embodiment of an antenna may have current at a 90/270 degree point or a reflection minimum approaching 90/270 degree point. If a 90/270 degree connection point is used, the variation from the correct 90/270 degrees is somewhat based on the intended use of the antenna, but is generally located close to 90/270 degrees, It gets better. The magnitudes of the electric and magnetic fields should also ideally be substantially the same or similar.

Indeed, the point at which the series of resonant elements 120 are bound to the loop 110 can be found empirically through the use of electric field and magnetic field probes that define a position at which the current is located at the reflection minimum or 90/270 degrees position have. The point at which the trapezoidal trace 124 can be bound to the loop 110 can be determined by moving the trace 124 until a predetermined 90/270 degree difference is observed. Another way to determine the reflex minimum connection point and the 90/270 connection point along the loop 110 is to determine the best connection point along the loop 110 as the area (s) of the minimum surface current magnitude There is a method of visualizing the surface current of an electromagnetic software simulation program to be visualized.

Thus, the degree of empirical measurement, trial and error, requires ensuring the sinful performance of the antenna, even if the principles underlying the arrangement of the elements are well understood. This is simply due to the nature of the printed circuits, often requiring a degree of " tuning " before certain performance is achieved.

While known simple loop antennas offer a very wide bandwidth, typically one octave, known antennas, such as dipoles, have very narrow bandwidths - generally a much smaller fraction of the operating frequency (such as 20% of the operating center frequency) Respectively.

Printed circuit technology is well known and is not described in detail here. It is sufficient that the copper traces are arranged and printed on a suitable substrate (usually by etching or laser trimming) with a particular dielectric effect. With careful selection of materials and dimensions, certain values of capacitance and inductance can be achieved without requiring separate, separate elements. However, as will be discussed further below, the design of this embodiment relaxes the substrate limitations of previous high frequency planar antennas.

As discussed above, these embodiments are arranged and fabricated using known microstrip techniques, in which the final design is reached as a result of a certain amount of manual calibration and the physical traces on the substrate are thereby adjusted. Indeed, calibrated capacitance sticks are used and include metal elements having known capacitance elements of, for example, 2 picoFarad. For example, a capacitance stick may be placed in contact with various locations of the antenna trace, wherein the performance of the antenna is measured.

Under the care of skilled technicians or designers, such techniques indicate the fact that traces constituting the antenna can be scaled, such as by adjusting the capacitance and / or inductance. After a number of iterations, an antenna having a predetermined performance can be achieved.

The connection point between the series of resonant elements and the loop is again determined empirically using electric field and magnetic field probes. Note that once the proper connection position is determined, the L and C values and / or fine adjustment to the connection can be achieved by laser trimming the in situ trace, taking into account the fact that minimal intervention from the test equipment can have a large real effect at the frequencies discussed. Can be performed. Once the final design is formed, it can be reproduced with the product repeatedly. Alternatively, the point of connection between the series of resonant elements and the loop may be determined by using an electromagnetic software simulation program to visualize the surface current, and also by selecting the area or regions where the surface current is located at the minimum.

An antenna formed according to the above-described embodiment provides an improvement in the efficiency of the antenna, which is generally known, over a similar volume.

In a further embodiment, a plurality of discrete antenna elements may be combined to provide greater performance than would be achieved by the use of a single element.

Fig. 2 shows an antenna 200 arranged and printed on a section of circuit board 205 in a known manner. Although the circuit board 205 has been described in plan, there is a certain amount of thickness on the substrate constituting the circuit board, and a ground plane (not shown) is formed on the ground plane region 624 On the back side of the circuit board 205 in a similar manner. In Figure 2, the antenna 200 comprises four functionally identical separate antenna elements 210 arranged in two sets, each driven in parallel.

The effect of providing many instances of the basic antenna element 210 is that it improves the overall performance of the antenna 200. In the absence of any loss associated with the configuration of the antenna, it may be possible to construct an antenna that theoretically includes a very large number of individual instances of the basic antenna elements 210, each having a number of elements adding 3 dB of gain to the antenna . However, in practice, losses, especially dielectric heating effects, mean that it is not possible to add extra devices indefinitely. In Fig. 2, an example shown with a four-element antenna is suffi- ciently a physically possible range and adds a gain of 6dB (less dielectric losses) to an antenna consisting of a single element.

2 is suitable for use in articles of micro-cellular base stations or other fixed wireless infrastructure, while single device 210 may be a cellular or mobile handset, a wireless pager, a PDA or a laptop It is suitable for use in mobile devices such as computers. The only real decision issue is size. The components and operation of the elements 210 are further described and shown in Figures 3A and 3B for antennas 310 and 370, respectively.

3A is a diagrammatic representation of the phase tracking antenna element 300 through the inclusion of the phase tracking antenna element 300 specially adapted to provide a greater operating bandwidth (optical bandwidth) than the narrow bandwidth antenna 100 of FIG. 1, A single antenna 310 (one embodiment of the elements 210 of FIG. 2) capable of achieving a large bandwidth up to a half octave is described. In particular, such a wide bandwidth is achieved by combining phase tracker 330 with rectangular electric field radiator 320 and loop element 350. The rectangular electric field radiator 320 replaces the series of resonant circuits 120 shown in FIG. However, the operating bandwidth of the rectangular electric field radiator 320 is also wider than that of the tuning circuit 120 due to the operation of the phase tracker 330 as will be described below.

For an alternative embodiment to antenna 310, by having the same rectangular electric field radiator 320, loop element 350, and driving or feed point 340 but not the phase tracker 330 Is illustrated in FIG. 3B as an antenna 370, such as antenna 310 of FIG. 3A, having a narrower operating bandwidth than antenna 310. In FIG. Another way to incorporate broadband operation is illustrated by the CPL antenna element of Figure 4A incorporating multiple electric field radiators 404 and 408 as further described below.

In the case of the tuning circuit 120, the connection point between the tuning circuit and the loop was important in determining the overall performance of the antenna 100. In the case of the electric field radiator 320 at the antennas 310 and 370 from Figs. 3A and 3B located on the outside of the loop 350, although the connection point is still generally at the center frequency, 350, or because the connection point is effectively distributed along the length of one side of the electric field radiator, the exact position as described above will be Less important. As such, the end points at which the edges of the electric field radiator 320 meet the loop 350 determine the operating frequency range of the antennas 310, 370 along with the dimensions of the loop.

The dimensions of the loop 350 are also important in determining the operating frequency of the antennas 310, 370. In particular, the overall length of the loop 350 is a key dimension as described above. To allow for a wide operating frequency range, a triangular phase tracker element 330 is provided directly across the electric field radiator 320 (located in one of two possible positions as shown in FIG. 2). The phase tracker 330 effectively acts as an automatic variable length tracking device that lengthens or shortens the electrical length of the loop 350 based on the frequency of the RF signal fed into it at the feed or drive point 340.

The phase tracker 330 is like a near-infinite series of L-C elements, only a portion of it resonates at a given frequency, thus automatically changing the effective length of the loop. In this way, a wider operating bandwidth can be achieved than with a simple loop that does not have any such phase tracking elements.

The phase trackers 330 shown in FIG. 2 have two different possible positions. These locations are selected for each antenna element 210 in the group of antenna elements 210 shown in FIG. 2 to minimize mutual interference between adjacent antenna elements 210. From an electrical point of view, the two configurations are functionally identical.

Since the magnetic loop 350 is a completely short signal current, a large bandwidth (1 ½ or less octave) of the antennas 310 and 370 is possible. As shown in FIGS. 3A and 3B, the magnetic loop is completely short because it is a half short wave, but it can also be completely short in quarter wave open and full short wave. The phase of the antenna is determined by the dimension 360. The dimension 360 extends between the length of the electric field radiator 320 and the length of the left side of the magnetic loop 350. The signal is shortened at a point where the signal is located at 180 degrees from the phase. A magnetic field with the largest magnitude is generated by the magnetic loop, and a small magnitude magnetic field is generated by the electric field radiator. In addition, the magnetic loop may change from an RF short having a very low actual impedance to a RF open with a very high real impedance. The largest magnitude electric field is diverted by one or more electric field radiator elements. However, the magnetic loop also produces a small electric field that is smaller in magnitude than the electric field emitted by the electric field radiator and opposite to the magnetic field.

The efficiency of the antenna is achieved by minimizing the current in the magnetic loop to produce the highest possible magnetic field. This is accomplished by designing the antenna so that the current travels into the electric field radiator and is reflected in the opposite direction, as further described below in FIG. As more current can be used for transmission purposes, the maximum magnetic field is projected in all directions that maximizes the efficiency of the antenna from the antenna. The maximum field energy that can be generated occurs when the magnetic loop is a complete RF short or when the magnetic loop has a very low actual impedance. However, under normal circumstances, RF shorts are undesirable because they cause the transmitter driving the antenna to fail. The transmitter produces an energy set at the set impedance. By using the impedance matching characteristic of the electric field, it becomes possible to have a near RF short loop without causing the transmitter to malfunction.

The current flowing through the magnetic loop flows into the electric field radiator. The current is then reflected into the magnetic loop by an electric field radiator along the opposite direction so that the electric field reflects into the magnetic field to form a short of the electric field radiator and forms an orthogonal electric and magnetic field.

The dimension 365 is composed of the width of the electric field radiator 320. The dimension 365 does not affect the efficiency of the antenna, but its width determines whether the antenna is narrowband or broadband. The dimension 365 has a large width only to widen the band of the antenna 310 described in FIG. 3A.

For example, all of the trace elements of the magnetic loop described in Fig. 3A can be made very thin without affecting the performance or efficiency of the antenna. However, by making these loop trace elements thicker, it is possible to accommodate a larger input power, or to have a smaller size, such as may be required by many other portable devices, such as mobile phones, The physical size of the antenna can be changed so as to match the space of the antenna.

It will be clear to those skilled in the art that any form of electric field radiator can be used in the multi-element configuration shown in FIGS. 2, 3A and 3B, together with a rectangular electric field radiator 320 constructed solely as an example. Likewise, a single element embodiment may use a rectangular electric field radiator, a tuning circuit, or any other suitable type of antenna. The multi-element version shown in FIG. 2 uses four separate antennas 210, but it does not include the available space with some limitation on the upper extent of the elements 210 as described below, Can be changed up and down based on the requirements.

Embodiments of the present disclosure have superior performance characteristics, are operable over much larger bandwidths, and permit the use of single or multi-element antennas, in comparison to known antennas of similar size. It also does not require any complicated components and creates low cost devices that are applicable to a wide range of RF devices. The embodiments of the present disclosure find particular use in mobile communication devices, but can be used in any device for which an efficient antenna is desired.

One embodiment comprises a compact, cross-sectional composite antenna ("cross-sectional antenna" or "print antenna"). "Cross-sectional shape" means that the antenna elements are positioned or printed on a single layer or plane, if desired. As used, the phrase "print antenna" is intended to encompass all of the elements of the printed antenna, whether or not the elements of the printed antenna are printed or etched, laminated, sputtered, or some other way of providing a metal layer on the surface, Regardless of whether or not it is formed of a cross-sectional antenna. The multiple layers of the cross-sectional antennas can be combined into a single device to provide wider bandwidth operation at smaller physical volumes, but each of the devices is still of a cross-sectional shape. The above-described cross-sectional antenna is necessarily a shortened device that does not have any ground plane on the rear side or bottom plane and represents a new concept in antenna designs on its own. The cross-sectional antenna is balanced, but can be driven with a balanced line or an unbalanced line if an important ground plane is present in the device of the intended application. The physical size of such an antenna can vary greatly based on the performance characteristics of the antenna, but the antenna 400 illustrated in FIG. 4A is about 2 cm x 3 cm. Similar or larger implementations are possible.

The cross-sectional antenna 400 is composed of two electric field radiators physically located inside the magnetic loop. 4A, the cross-sectional antenna 400 is comprised of a magnetic loop 402, wherein the first electric field radiator 404 includes a magnetic loop 402 having a first electrical trace 406, And the second electric field radiator 408 is coupled to a magnetic loop 402 having a second electrical trace 410. [ The electrical traces 406 and 410 connect the electric field radiators 404 and 408 to the magnetic loop 402 at corresponding 90/270 degree electrical positions for a feed or drive point. Alternatively, the electrical traces 406 and 410 may couple the electric field radiators 404 and 408 to the magnetic loop in a region where a current flowing through the magnetic loop is located at the reflection minimum. As described above, for other frequencies, the connection or connection point of the traces 406, 410 changes, which is why at one frequency the radiator 404 is positioned at a different point from the radiator 408, To the loop 402 will be described. At low frequencies, it takes a long time for the waves to reach the 90/270 degree point; Consequently, the physical position of the 90/270 degree point will be higher along the magnetic loop compared to the high frequency. At high frequencies, the time to reach the 90/270 degree position is less, resulting in a physical position of 90/279 degrees relative to the low frequency being lowered along the magnetic loop. Likewise, the points along the magnetic loop where the current is located at the reflection minimum may also depend on the frequency of the electric field radiator. Finally, alternate embodiments of the antenna 400 may be composed of one or more electric field radiators directly coupled to the magnetic loop 402 without electrical traces.

The electric field radiator 404 also has a different magnitude than the electric field radiator 408 because each electric field radiator emits waves at a different frequency. The small electric field radiator 404 may have a short wavelength and thus a high frequency. The large electric field radiator 408 may have a long wavelength and a low frequency.

The physical arrangement of the electric field radiator (s) physically located inside the magnetic loop reduces the size of the entire antenna as compared to other embodiments in which the physical location of the electric field radiator (s) and the magnetic loop are external to each other And provides a broadband device at the same time. Alternative embodiments may have different numbers of electric field radiators, each arranged at different positions around the loop. For example, the first embodiment may have only one electric field radiator located inside the magnetic loop, whereas the second embodiment with two electric field radiators may have one electric field radiator on the inside of the magnetic loop, And a second electric field radiator on the outer side of the magnetic loop. Alternatively, more than two electric field radiators may physically be located inside the magnetic loop. As with the other antennas described above, the cross-sectional antenna 400 acts as a transducer due to the electrical and magnetic fields.

As described above, the use of multiple electric field radiators is allowed for broadband functionality. Each electric field radiator can be configured to emit waves at different frequencies, with the result that the electric field radiators cover a wide band range. For example, the cross-sectional antenna 400 may be configured to cover a standard IEEE 802.11b / g radio frequency range with the use of two electric field radiators configured in two frequency ranges. The first electric field radiator 404 may be configured to cover, for example, a frequency of 2.41 GHz, while the second electric field radiator 408 may be configured to cover, for example, a frequency of 2.485 GHz. This allows the sectioned antenna 400 to cover the frequency band 2.41 GHz to 2.485 GHz, which corresponds to the IEEE 802.11b / g standard. The use of two or more electric field radiators generates a wideband operation without the use of a phase tracker (as shown in Figures 2 and 3), as described in connection with the physically large antenna embodiments described above. In an alternative embodiment, a wideband antenna can also be achieved by tapering multiple electric field radiators using a log scale similar to a YAGI antenna.

In general, the length of the electric field radiator determines the frequency they cover. The frequency is inversely proportional to the wavelength. Therefore, a small electric field radiator has a small wavelength and causes high frequency. On the other hand, large electric field radiators have long wavelengths and result in low frequencies. However, this generalization is also an operational characterization.

For optimal efficiency, the electric field radiator may have an electrical length of about one wavelength, one quarter wavelength, or one-eighth the wavelength of the generated frequency. If, as described above, the electrical length of the electric field radiator is limited so that the amount of available physical space is less than a predetermined wavelength, a slab trace may be used to electrically elongate the electric field radiator and add propagation delay .

4A and 4B, the electrical traces 406 and 410 are inductors, and their individual lengths versus their shape or other characteristics determine their inductance. For optimal efficiency, the inductive reactance of the electrical trace can be matched to the capacitive reactance of the corresponding electric field radiator. The electrical traces 406 and 410 are bent to reduce the overall size of the antenna. For example, the curves of the electrical traces 406 may be formed in close proximity to the magnetic loop 402 instead of being closely formed with the electric field radiator 404, or the curve of the traces 406 may be formed It may be lowered instead of ascending similarly to the first embodiment. The electrical traces are shaped to extend their length, but not because the shape has any particular significance apart from the context. For example, instead of having a linear electrical trace, a curve may be added to the electrical trace to increase its length and correspondingly increase its inductive reactance. However, the sharp corners on the electrical traces and the sinusoidal form of the electrical traces may adversely affect the efficiency of the antenna. In particular, electrical traces having a sinusoidal shape cause the electrical traces to emit a small electric field that partially outphases the electric field radiator, reducing the efficiency of the antenna. Thus, the efficiency of the antenna can be improved by using soft and elegant curves or by using electrical traces molded into as few bends as possible.

The spacing between the elements in the cross-sectional antenna 400 adds capacitance to the entire antenna. For example, the distance between the top of the electric field radiator 404 and the magnetic loop 402, the distance between the two electric field radiators 404 and 408, the left side of the electric field radiators 404 and 408, The gap between the right side of the electric field radiators 404 and 408 and the magnetron loop 402 and the gap between the bottom of the electric field radiator 408 and the magnetic loop 402, All affect the capacitance of the antenna 400. As described above, in order to resonate the antenna 400 with optimum efficiency, the inductive reactance and the capacitive reactance of the entire antenna are matched in a predetermined frequency band (s). Once the inductive reactance is determined, the distance between the various elements can be determined based on the capacitive reactance value needed to match the inductive reactance value for the antenna.

Formulas are given to find the space between the elements and the associated edge capacitance, and the optimal spacing between the elements can be determined using multi-objective optimization. The optimal spacing between elements or between any two adjacent antenna elements can be optimized using linear programming. Alternatively, nonlinear programming, such as a genetic algorithm, can be used to optimize the spatial values.

As discussed above, the size of the cross-sectional antenna 400 is based on a number of factors including a predetermined operating frequency, narrow band versus wide band function, and tuning of capacitance and inductance.

In the case of the antenna element 400 of FIG. 4A, the length of the magnetic loop 402 is one wavelength (360 degrees) designed with optimal efficiency, although a multiple of other wavelengths may also be used. When designed for optimum efficiency, a portion of the magnetic loop also acts as an electric field radiator, and the electric field radiator also generates a small magnetic field in addition to the directivity and efficiency of the antenna. The length of the magnetic loop may be arbitrary, or it may be a multiple of one wavelength, one quarter wavelength or one eighth wavelength, which is a specific length that can increase efficiency over other cases. One wavelength is an open circuit for voltage and a short circuit for current. Alternatively, the length of the magnetic loop 402 may be physically less than one wavelength, but extra inductance may be added to electrically lengthen the loop by increasing the propagation delay. The width of the magnetic loop 402 is based primarily on the capacitance as well as the desired effect on the inductance of the magnetic loop 402. For example, by physically shortening the magnetic loop 402, the wavelength is made small, resulting in a high frequency. For a design for optimal efficiency of the magnetic loop 402, the inductance and capacitance must satisfy the equation w = l / sqrt (LC), where w is the wavelength of the loop 402. Thus, the magnetic loop 402 can be tuned by varying the inductance and capacitance affecting the electrical length. By reducing the width of the magnetic loop, it also adds inductance. In thinner magnetic loops, more electrons must be pressed through small areas, additional delay.

The upper portion 412 of the magnetic loop 402 is thinner than any other portion of the magnetic loop 402. Thereby allowing the size of the magnetic loop to be adjusted. The upper portion 412 can be reduced because it has a minimal impact on the 90/270 degree connection point. The upper portion 412 of the magnetic loop 402 may also be trimmed to increase the electrical length of the magnetic loop 402 and increase the inductance so that the inductive reactance matches the overall capacitive reactance of the antenna Can help. Alternatively, the height of the upper portion 412 may be increased to increase the capacitance (or equivalently, to reduce the inductance). As described above, the reflection minimum connection point depends on the geometry of the magnetic loop. Thus, by geometrically changing the loop, either by smoothing the upper portion 412, increasing the upper portion 412, or by altering any other aspect of the magnetic loop, the loop geometry is modified So that the current is required to be located at the reflection minimum.

The magnetic loop 402 should not be square as illustrated in FIG. 4A. In an embodiment, the magnetic loop 402 may be in a rectangular or peculiar form and two electric field radiators 404 and 408 may be located at corresponding 90/270 degree connection points or reflection minimum connection points . For optimal efficiency, the electrical length of the odd shaped loop may be a multiple of about one wavelength or a multiple of about 1/4 or 1/8 wavelength in a given frequency band (s). The electric field radiators may be located on the inside or outside of the magnetic loop of the specific type. The key is to identify the connection point along the magnetic loop that maximizes the efficiency of the antenna. The connection point may be a 90/270 degree electric point along the magnetic loop or a point where the current flowing through the magnetic loop is located at the reflection minimum.

For example, in a smartphone, the singular shape of the antenna design can be matched in a space of a usable specific type, such as a back cover of a mobile device. Instead of having a square shape, the magnetic loop may have a rectangular shape, a circular shape, an elliptical shape, a generally E-shaped shape, a generally S-shaped shape, and the like. Likewise, an antenna of the miniature singular shape can be matched in a non-uniform space on a laptop computer or other portable electronic device.

As noted above, the position of the electrical traces may be such that the electric field radiated by the electric field radiator is perpendicular to the magnetic field generated by the magnetic loop, such that the electrical trace is approximately 90/270 degrees electrical point along the magnetic loop, Lt; / RTI > The 90/270 degree connection point and the reflection minimum connection point are important because these points prevent the reactive power (virtual power) from being transmitted off the antenna and returned. Reactive power is generally generated and stored around the proximity electric field of the antenna. The reactive power oscillates around the fixed position near the source and affects the operation of the antenna.

Referring to FIG. 4A, the dashed line 414 indicates that the most important regions of the edge capacitance phenomenon occur. Two-piece metal in the antenna, such as magnetic loop and electric field radiators, at a certain distance away can form a level of edge capacitance. Through the use of edge capacitance, embodiments of cross-sectional antennas allow all elements of an antenna to be printed on one side of suitable substrate materials of almost any type, including inexpensive dielectric materials. An example of an inexpensive dielectric material that can be used as a substrate includes a glass-reinforced epoxy laminate (FR-4) having a dielectric constant of about 4.7 +/- 0.2. For example, in the cross-sectional antenna 400, there is no need for a rear surface or a ground surface. Rather, lead is connected to each end of the magnetic loop, and one of the leads is grounded. As noted above, this full wave antenna design means a shorted composite loop antenna with optimal efficiency. Indeed, the cross-sectional antenna will be most optimally performed in the presence of a counterpoise ground plane, as is common in buried antenna designs where the counterpoise is provided by the object on which the antenna is mounted.

The 2D design of the embodiment of the cross-sectional antenna has several advantages. With the use of a suitable substrate or dielectric base that can be very thin, the traces of the antenna are actually sprayed or printed on the surface and still function as a composite loop antenna. In addition, the 2D design allows the use of antenna materials, such as very cheap substrates, which are not generally seen as suitable for microwave devices. Another advantage is that the antenna can be placed on surfaces of a singular shape, such as the back of the cell phone case cover, the edges of the lamp tower, and the like. Embodiments of the cross-sectional antenna may be printed on the dielectric surface with an adhesive located on the back side of the antenna. Next, the antenna can be attached to various computing devices with leads connected to the antenna to provide the required power and ground. For example, as described above, with this design, an IEEE 802.11b / g wireless antenna can be printed on a surface with an approximate postage stamp size. The antenna may be attached to a cover of a laptop, a case of a desktop computer, or a back cover of a cell phone or other portable electronic device.

A variety of dielectric materials may be used as embodiments of the cross-sectional antenna. The advantage of FR-4 as a substrate on other dielectric materials such as polytetrafluoroethylene (PTFE) is that it has low cost. Generally, dielectrics used for high frequency antenna designs have much lower loss characteristics than FR-4, but they are generally more costly than FR-4.

Embodiments of the cross-sectional antenna may also be used for narrow band applications. The narrowband is associated with a channel in which the bandwidth of the message does not exceed the channel coherence bandwidth. For a wide band, the message bandwidth greatly exceeds the channel correlation bandwidth. Examples of narrow-band antenna applications include Wi-Fi and point-to-point long-range microwave links. According to the embodiments described above, for example, an array of narrow band antennas can be printed on a sticker and then used for Wi-Fi access with good signal strength and greater distance compared to standard Wi-Fi antennas May be located on a laptop.

4B illustrates another embodiment of a cross-sectional antenna 420 with a magnetic loop 422 having a corner that is cut at an angle of about 45 degrees. By cutting the corners of the magnetic loop 422 at a certain angle, the efficiency of the antenna is improved. By having a magnetic loop having corners formed at an angle of about 90 degrees, it affects the flow of current flowing through the magnetic loop. When the current flowing through the magnetic loop strikes a 90 degree angle corner, it causes the current to recoil, and the reflected current flows against the main current flow or forms an eddy pool. Energy losses as a result of the 90 degree corners may adversely affect the performance of the antenna, particularly the small antenna embodiment. And improves the current flow around the corners of the magnetic loop when cutting the coaters of the magnetic loop at an angle of about 45 degrees. Thus, the angled corners make the electrons in the current less imperceptible as they flow through the magnetic loop. Although it is suitable to cut the corners at a 45 degree angle, alternate embodiments of cutting at a different angle than the 45 degree are also possible.

4C illustrates an alternate embodiment of a cross-sectional antenna 440 that uses transitions of various widths in the magnetic loop 442 to add inductance or add capacitance to the magnetic loop 442. FIG. The corners of the magnetic loop 442 were cut at an angle of about 45 degrees to improve the current flow as the current flowed around the corners of the magnetic loop 442, thereby increasing the efficiency of the antenna. A single electric field radiator 444 is physically located inside the magnetic loop 442. The electric field radiator 444 is connected to the magnetic loop 442 with electrical traces 446 having a soft curve shape. As described above, the efficiency of the antenna is improved by having electrical traces 446 that are not sinusoidal and also have soft curves that minimize the number of bends in the trace.

The term transition is used to refer to changes in the width of the magnetic loop. 4C, the magnetic loop 442 takes a generally rectangular shape and includes a first transition on the left side and a second transition on the right side. In the embodiment shown in Figure 4c, the first transition is symmetrical to the second transition. The transition on both the left and right sides of the magnetic loop 442 is thinner than the remaining portion of the magnetic loop 442 and between the first wide section 450 and the second wide section 452, Wherein the first wide section (450) and the second wide section (452) comprise a narrow section (448) or a medium narrow section located adjacent to both of the narrow section (448) and the narrow section It has a large width. In particular, the magnetic loop transitions from the first wide section 450 to the middle narrow section 448 and the middle narrow section 448 transitions to the second wide section 452. The wide-narrow-wide transition in the magnetic loop produces pure inductance, thus increasing the electrical length of the magnetic loop. Thus, the use of the wide-narrow-wide transition in the magnetic loop increases the electrical length of the magnetic loop 442 by adding inductance to the magnetic loop 442. The length of the intermediate narrow section 448 may also be increased or decreased depending on the need to add a predetermined inductance to the magnetic loop. For example, in FIG. 4C, the middle narrow section 448 spans about one-quarter of the right and left sides of the magnetic loop 442. However, the middle narrow section 448 can be increased to about a half of a different ratio of the right and left sides of the magnetic loop 442, so that the inductance of the magnetic loop 442 increases do.

Transitions are not limited to sections or segments that have a smaller width than the remaining portion of the magnetic loop 442. [ Alternate transition portions may include intermediate wide sections or intermediate wide segments that are wider than the remaining portion of the magnetic loop 442 and are located between and adjacent to the first narrowing section and the second narrowing section, The first narrowing section and the second narrowing section have a smaller width than the wide section. Particularly, in such an alternative embodiment, the magnetic loop is transitioned from the first narrowing section to the middle narrow section, and the intermediate wide section later shifts to the second narrow section. The narrow-wide-narrow transition in the magnetic loop produces a capacitance, so that the electrical length of the magnetic loop is shortened. The length of the intermediate width can be increased or decreased by adding capacitance to the magnetic loop.

By using a transition in the magnetic loop, i. E. By changing the width of the magnetic loop to more than one or more sections or segments of the magnetic loop. Transitions that change widths in the magnetic loop may also be tapered to further add inductance or capacitance to ensure that the reactive inductance and reactive capacitance of all elements of the antenna are matched. For example, in a wide-narrow-wide transition, the first wide section may taper to a small width of the middle narrow section at its large width. Likewise, the intermediate narrow section may be tapered from its narrow width to a large width of the first wide section or the second wide section or both. The sections in the narrow-wide-narrow-width transition and the wide-narrow-wide transition can be tapered independently of each other. For example, in the first narrow-wide-narrow transition, only the intermediate wide section may be tapered, whereas in the second narrow-wide-narrow transition, only the first narrow section is tapered have. The tapering may be linear, stepped or curved.

The actual difference in width between the portions of the magnetic loop depends on the amount of inductance or capacitance required to reliably match the overall reactive capacitance of the antenna to the overall reactive inductance of the antenna. The embodiment illustrated in Figure 4c illustrates two wide-narrow-wide transitions positioned symmetrically opposite each other. Alternative embodiments, however, may have transitions only on one side of the magnetic loop 442. Also, if more than one transition is used in the magnetic loop, these transitions need not be symmetrical. For example, the magnetic loop of the singular shape may have two transitions, the transitions having different lengths and widths. Other types of transitions can also be used on a single magnetic loop. For example, a magnetic loop may have one or more narrow-wide-narrow transitions or one or more wide-narrow-wide transitions.

Fig. 5 illustrates an embodiment of a compact double-sided antenna or a planar antenna 500. Fig. The planar antenna 500 includes a tunable patch shown as a dashed line 502 that generates a capacitive reactance to match the inductive reactance of the magnetic loop 504 for a particular frequency The second plane is used. The variable patch 502 is a generally square metal piece having a fixed position relative to other elements of the antenna 500. In one embodiment, the variable patch 502 is positioned at a point away from a 90/270 degree electrical point along the magnetic loop, as shown in FIG. 5, or at a point at which a current is reflected by the upper left corner of the antenna 500 And may be located at a position remote from the region located at the minimum. The electric field radiator 506 is positioned inside the magnetic loop 504 to reduce the overall size of the double-sided antenna 500. For optimal efficiency, the electric field radiator 506 may have an electrical length corresponding to approximately a quarter wavelength at its corresponding operating frequency. If the electric field radiator is small, it causes a smaller wavelength at higher frequencies. The electric field radiator 506 is generally bent in the shape of J to match the entire length of the inside of the magnetic loop 504. Alternatively, the electric field radiator 506 may be strained to be positioned in a straight line or bent into another shape, rather than being bent into the J shape. Such an embodiment is contemplated herein, but it makes the antenna wider and also increases the overall size of the antenna.

The electrical trace 508 connects the electric field radiator 506 to the magnetic loop 504 at the 90/270 degree connection point or at the minimum reflection connection point. The upper portion 510 of the magnetic loop 504 is smaller than the other sides of the magnetic loop 504. This serves to lengthen the electron length of the magnetic loop 504 and to increase the inductance. The increase in inductance also causes the inductive reactance to match the overall capacitive reactance of the antenna 500, as in the case of the small cross-sectional antenna 400, and can be adjusted as described above.

The variable patch 502 may also be positioned anywhere along the upper portion 510 of the magnetic loop 504. However, when the variable patch 502 is located away from the point where the magnetic loop 504 is connected to the electric field radiator 506, the variable attenuator 502 exhibits the best performance. The size of the variable patch 502 can also be increased by varying its depth, length, and height. By increasing the depth of the variable patch 502, an antenna design that can take up more space is caused. Alternatively, the variable patch 502 can be made very thin, but its length and height can be correspondingly adjusted. Instead of having the variable patch 502 covering the upper left corner of the antenna 500, the length and height may be increased to cover the left half of the antenna 500. Alternatively, the length of the variable patch 502 can be increased and the upper half of the antenna 500 is allowed to expand. Similarly, the height of the variable patch 502 can be increased, and the left side of the antenna 500 is allowed to expand. The variable patch can also be made smaller.

Similar to the cross-sectional antenna, various dielectric materials may be used in embodiments of the double-sided antenna 500. Available dielectric materials include FR-4, PTFE, cross-linked polystyrene, and the like.

FIG. 6 illustrates an embodiment of a large antenna 600, which consists of an array of four antenna elements 602 and has a bandwidth of one or one-half octave. Each antenna element 602 may be a TE mode transverse electric radiator or a magnetic field (H field) radiator or a magnetic loop dipole 604 (shown schematically in dashed lines and having a magnetic loop 604) and a TM mode (transverse) radiator, or an electric field (E) radiator, or an electric field dipole 606 (referred to as a rectangle shaped area and also referred to as an electric field radiator 606). The magnetic loop 604 should be one electrical wavelength that causes a short circuit. On the other hand, the magnetic loop 604 may be physically smaller than one wavelength, while electrically looping the magnetic loop 604 by adding extra inductance as discussed below. The physical width of the magnetic loop 604 can also be adjusted to obtain the appropriate impedance / capacitance of the magnetic loop 604 so that it can resonate at a predetermined frequency. The physical parameters of the magnetic loop 604 do not depend on the amount of dielectric material used for the antenna elements 602, as will be described below.

As discussed above, the magnetic loop 604 is fully shorted to maximize the amount of current in the magnetic loop and also to generate the highest magnetic field. At the same time, the impedance is matched to the load from the transmitter to prevent the transmitter from becoming unavailable as a result of the short circuit. Current flows from the magnetic loop 604 into the electric field radiator 606 in the direction of the arrow 607 and from the electric field radiator 606 in the direction of the arrow 609 into the magnetic loop 604 in the opposite direction, Is returned.

In an embodiment, each of the antenna elements 602 has a height of about 4.45 cm wide x about 2.54 cm, as described in FIG. However, as discussed above, the size of all components is determined by the operating frequency or other characteristics. For example, the traces of the magnetic loop 604 may be made very thick to increase the gain of the antenna element 602 and also to increase the physical size of the inter-antenna element 602 and thus the size of the antenna 600 It is possible to change to match any given physical space and still be in a resonance state, to sustain a portion of the gain that is equally increased and to maintain a similar level of efficiency, It is not supplied at all. As long as the altered design continues (1) a magnetic loop with successive closed surface currents, (2) energy reflection from the electric field radiator into the magnetic loop, and (3) matched impedance of the components, It can be adjusted to any size. Similar levels of efficiency can be achieved, although the gain varies based on the particular size and shape selected for the antenna.

The phase tracker 608 (represented by the triangular shaped area) makes the antenna 600 broadband and can also be eliminated for narrow band design. The tip of the phase tracker 608 is ideally located in the 90/270 degrees electrical position along the magnetic loop 604. [ However, in an alternative embodiment, the tip of the phase tracker may be located at the minimum reflection junction. The dimension 610 of the electric field radiator 606 is not a real issue for the overall operation of the antenna element 602. [ The dimensions 610 only have a width to make the antenna element 602 broader and the dimensions 610 can be reduced if the antenna element 602 has a tendency to become a narrow band device. As described, antenna element 602 has a tendency to become broadband because it includes phase tracker 608. Dimension 612 is determined by the center operating frequency and also determines the phase of antenna element 602. [ The dimension 612 is located over the length of the electric field radiator 606 and the length of the left side of the magnetic loop 604. [ Dimension 612 can generally be a quarter wavelength and is slightly adjusted for the dielectric material used as the substrate. The electric field radiator 606 has a length representing a quarter wavelength at the frequency of interest. The length of the electric field radiator 606 may also be sized to be a multiple of a quarter wavelength at the frequency of interest, but such a change may reduce the effectiveness of the antenna.

The width of the upper portion 614 of the magnetic loop 604 is intended to be smaller than any other portion of the magnetic loop 604, but such a difference may not appear in the figure of FIG. This difference in size is similar to the small antenna embodiments already described, and the upper portion 614 can be removed to increase the electrical length and add inductance. The upper portion 614 of the magnetic loop 604 can be removed because it has minimal effect on the electrical position of 90/270 degrees. When the inductance is added by removing the upper portion 614, the magnetic loop 604 exhibits an electrically long lifetime.

The dimensions 616, 617, and 618 of the magnetic loop 604 are all determined by the wavelength dimension. Dimension 616 comprises the width of the magnetic loop 604. The dimension 617 consists of the length of the left side portion of the bottom side of the magnetic loop 604. That is, the dimension 617 is composed of the length of the bottom portion of the magnetic loop 604 with respect to the left side of the magnetic loop opening 619. The dimension 618 consists of the entire length of the magnetic loop 604. The best antenna performance is achieved when the dimension 616 is the same size as the dimension 618, resulting in a rectangular loop. However, a magnetic loop 604 having a rectangular or irregular shape may also be used.

As described above, the phase tracker 608 is included for the broadband operation of the antenna 600, and the removal of the phase tracker 608 causes the antenna 600 to be less broadband. The antenna 600 may optionally be formed in a narrow band by reducing the physical vertical dimensions of the phase tracker 608 and the dimensions of the electric field radiator 606. The support of broadband operation in the phase tracker 608 and antenna has the potential to reduce the total number of antennas used in various devices such as cell phones. The dimensions of the phase tracker 608 also affect its inductance and capacitance as described in FIG. The range of the capacitance and inductance of the phase tracker 608 may be tuned by adjusting the physical dimensions of the phase tracker 608. The inductance L of the phase tracker 608 is based on the height of the phase tracker 608. The capacitance (C) of the phase tracker (608) is based on the width of the phase tracker (608).

The pairs of antenna elements 602 and antenna elements 602 have a set of gaps formed therebetween. The two antenna elements 602 located on the left side of the antenna 600 constitute a first pair of antenna elements 602 while the two antenna elements 602 located on the right side of the antenna 600, The elements 602 constitute a second pair of antenna elements 602. There is a first gap 620 between each pair of antenna elements 602 and there is a second gap 622 between each pair of antenna elements 602. A first gap 620 between each pair of elements 602 and a second gap 622 between each pair of antenna elements 602 is defined as the distance between the far- Are designed to align the far field radiation patterns generated by the antenna elements 602 in the most effective manner so that they are added to the antenna elements. Well-known phase antenna array techniques can be used to determine the optimal spacing between multiple CPL antenna elements 602, resulting in the far-field radiation pattern of each of the elements being added.

In an embodiment, the far field radiation patterns can be modeled on a computer based on the relationship of the other components of the antenna elements 602. [ For example, the size of the antenna elements 602, the spacing between the pairs of antenna elements 602 or between the pairs of antenna elements 602, and the relationship of the components may be determined by additional orientations of the long- And till the tack is accomplished. Alternatively, the far-field emission patterns can be measured using electrical equipment having a relationship of components that are adjusted on such a base.

6, the antenna elements 602 are supplied by a microstrip feed line indicated by a broken line 624. The supply line in the dashed line 624 is matched with the network to drive the impedance and is based on the dielectric material used. The symmetry of the feed lines is also important to avoid unnecessary phase delay resulting in far-field radiation patterns generated by the antenna elements being removed instead of the additive.

With reference to FIG. 6, the embodiment uses a common synthesizer / separator 626 to split the incoming signal into two to feed two sets of antenna elements and combine the return signals. The second and third synthesizer / separator 628 then feeds each pair of antenna elements 602 and separates the return signals into two to combine the return signals. Separators 626 and 628 are preferred because they prevent nearly all of the impedance matching that occurs along the supply lines over a wide frequency range and that power is returned along the supply line, Do.

8 illustrates a bottom layer 800 of an antenna 600 including elements 802, 812, 814 and 816, each of which includes a trapezoidal element 804, a choke joint ) Region 806 and a laser 808. In this embodiment, The elements 802,812,814 and 816 are configured such that even if the elements 812 and 814 also set the phase angle of the antenna 600 by reflecting the signal or RF energy to the bottom of the bridge element 820 , And act as capacitors. The distance 826 from the bottom of the trapezoidal elements 804 to the bottom of the bridge element 820 may be reduced by a factor of 1/4 if a spherical shape for the derivative pattern generated by the antenna 600 is desired, It can not be larger than the wavelength. By varying the distance 826 for each of the elements 802, 812, 814, 816, other types of radiation patterns can be formed. Finally, cutting elements 822 and 824 are provided at locations where trace materials are removed from the bottom left corner and bottom right corner of the bridge element 820 to prevent reflection of the elements 802 and 816, And consequently changes the phase angle set by the elements 812, 814.

The trapezoidal elements 804 maintain the magnetic loop 604 of each corresponding antenna element 602 that is matched due to the fact that each trapezoidal element 804 is dimensionally log-driven. The inclination of each trapezoidal element 804, in particular the slope of the upper side of the trapezoidal element 804, adds inductance and capacitance, which is varied to match the inductive reactance to the capacitive reactance of the antenna 600 Lt; / RTI > By adding capacitance through the trapezoidal elements 804, the electrical length of each corresponding magnetic loop 604 on the other side of the antenna 600 can be adjusted. The trapezoidal elements 804 are aligned with the upper traces 614 of the magnetic loop 604 on the other side of the antenna 600. The choke connectors 806 act to isolate the trapezoidal elements 804 from ground, thus preventing leakage of the derived signal. The sides 809 and 810 of the trapezoidal elements 804 are counterpoised to the electric field radiators 606 on the other side of the antenna 600 that need to be grounded to establish polarization. The side 809 is composed of the right side of the trapezoidal elements 804 and the upper right side of the laser 808 overlying the choke connector 806. That is, side 809 comprises the right side of each element 802, 812, 814, 816 overlying choke connector 806. The side surface 810 is constituted by the left side surface of the trapezoidal elements 804 and the left side surface of the laser 808. That is, side 810 comprises the left side of each element 802, 812, 814, 816 overlying ground plane element 828. The counter poises 809 and 810 increase the transmission / reception efficiency of the antenna 600. The ground plane element 828 is a standard for microstrip antenna design where, for example, a 50 ohm (.OMEGA.) Trace on a 4.7 dielectric is about 100 mils wide.

As described above, the trapezoidal elements 804 can be fine tuned to change the inductance of the corresponding magnetic loop or to change the capacitance. The fine tuning process includes a process of reducing or enlarging sections of the trapezoidal elements 804. For example, it can determine if additional capacitive reactance is needed to match the inductive reactance of the magnetic loop. Thus, the trapezoidal elements 804 can be enlarged to increase the capacitance. An alternative fine tuning step is to vary the slope of the trapezoidal elements 804. For example, the slope may be varied from 15 degrees to 30 degrees. Alternatively, if the magnetic loop 604 is modified by increasing its area or by reducing the width of the top trace 614 of the magnetic loop 604, The metal on the ground plane must also be adjusted accordingly. For example, the upper side of the trapezoidal element 804 or the overall length of the trapezoidal element 804 may be reduced or increased based on whether the upper trace 614 of the magnetic loop 604 is reduced or increased .

The simultaneous energization of the TM and TE radiators results in a zero reactive power expected by the time dependent pointing theorem when used to analyze microwave energy, as described above. Previous attempts to construct composite antennas with TE and TM radiators that are electrically perpendicular to one another have relied on a three dimensional array of these devices. Such a design can not be easily commercialized. In addition, the previously proposed composite antenna designs were supplied with separate power sources located at two or more positions in each loop. In embodiments of the various antennas as described above, the magnetic loop and the electric field radiator (s) are still co-planar and supplied with power from a single location, . This results in a two-dimensional array that can reduce the complexity of the physical arrangement and also enhance commercial viability. Alternatively, the electric field radiator (s) may be located on the magnetic loop at a point where a current flowing through the magnetic loop is located at the reflection minimum.

Embodiments of the antennas described herein have greater efficiency than conventional antennas that partially cancel reactive power. In addition, embodiments have large antenna openings for their respective physical sizes. For example, a half-wavelength antenna with an omnidirectional pattern according to an embodiment may have a significantly larger gain than the typical 2.11 dBi gain of simple field dipole antennas.

Yet another embodiment consists of a cross-sectional antenna with a built-in counterpoise for the electric field radiator. 9A illustrates an embodiment of a 2300-2700 MHz cross-section antenna with a built-in counterpoise for a single electric field radiator and the electric field radiator. The antenna 900 comprises a magnetic loop 902 having an electric field radiator 904 directly coupled to the magnetic loop 902 without gain of the electrical traces. The electric field radiator 904 is physically located on the inside of the magnetic loop 902. As in other embodiments, the electric field radiator 904 may be bound to the magnetic loop 902 at a 90/270 connection point or at a point where the current flowing through the magnetic loop 902 is located at the reflection minimum . In an alternative embodiment, the electric field radiator 904 may be bound to a magnetic loop 902 having an electrical trace. Also, while the antenna 900 has been described as an electric field radiator, alternative embodiments may include one or more electric field radiators. Alternate embodiments may also include one or more electric field radiators physically located on the outside of the magnetic loop 902.

An alternative embodiment of the self-contained antenna may also include a first electric field radiator having a first length and a second electric field radiator having a second length different from the first length. As with the antenna embodiments described above, broadband antennas are made possible by using one or more electric field radiators having different lengths.

The antenna 900 includes a transition portion 906 and a counterpoise portion 908 for the electric field radiator 904. The transition portion 906 comprises a portion of the magnetic loop 902 having a width greater than the width of the magnetic loop 902. The transition portion 906 electrically isolates the built-in counterpoise portion 908. The built-in counterpoise portion 908 allows the antenna 900 to be completely independent of the chassis or any ground plane of the product using the antenna 900.

The counterpoise portion 908 is referred to as being built-in since the counterpoise portion is formed from the magnetic loop 902. As described, the built-in counterpoise portion 908 allows the antenna 900 to be completely independent of the ground plane of the product. Embodiments of the cross-sectional antenna shown in Figs. 4A-4C are printed only on a single plane and do not include a ground plane, but require a ground plane to be provided by the device using the antenna. Conversely, a self-contained counterpoise antenna does not require a ground plane to be provided by the device using the antenna.

In the cross-sectional embodiments described above, a device using an antenna may be fabricated by using the chassis or some other metallic component of the device as a ground plane for the cross-sectional antenna, or as a ground plane for the cross- Together with the ground plane of the device acting thereon, provides a ground plane for the antenna. However, any changes to the device's electrical network, either to the device's chassis or to the ground plane of the device, can adversely affect the performance of the antenna. This phenomenon is not specific to the cross-sectional embodiments described in the present invention, but instead applies to antennas that are widely used in research and commercial applications. It would therefore be desirable to have an antenna that does not require a ground plane that would not be affected by any changes made to the device using the antenna.

By not requiring a ground plane, the antenna 900 does not depend on the ground plane outside the antenna. The independence of the self-contained antenna 900 from such an external ground plane means that the performance of the antenna is not affected by the variations formed in the device. This indicates that, in terms of manufacturing and design, a self-contained antenna can be designed for a specific frequency and that the performance level means to integrate and use the antenna independently of the device. For example, the wireless router maker requests a particular antenna based on the set of requirements. These requirements may include, among other requirements, the space available for the antenna, the frequency range for the antenna, and the substrate to be used. At that time, the design and manufacture of the antenna may be performed independently of the design and manufacture of the actual wireless router. Further, any further changes to the wireless router do not affect the performance and efficiency of the antenna since the antenna is self-contained, and it is also possible that changes in the electrical network of the router, the ground plane of the router, Lt; / RTI >

The length of the transition portion 906 may be set based on the operating frequency of the antenna. For a high frequency antenna with a short wavelength, a short transition can be used. Conversely, for a low-frequency antenna with a long wavelength, a long transition 906 can be used. The transition portion 906 may be adjusted independently of the counterpoise portion 908. For example, the transition for the 5.8 GHz antenna may be only half the size of the transition 906 of FIG. 9A, while the counterpoise portion 908 is still the entire left side of the magnetic loop 902 It will be like.

The length of the counterpoise portion 908 may be adjusted according to the need to obtain a predetermined antenna performance. However, it is preferable to have the counterpoise portion 908 as large as possible. For example, in an alternate embodiment, the counterpoise portion 908 is configured to allow the entire length of the left side of the magnetic loop 902, rather than only about 80% of the left side of the magnetic loop 902, . However, the width of the traces of the magnetic loop 902 as described above affects the electrical length of the magnetic loop 902. A magnetic loop having a thin trace all around the magnetic loop is electrically longer than a magnetic loop having portions of the magnetic loop having a wide trace or a wide trace. For example, the magnetic loop 902 is an example of a magnetic loop having a wide trace for the transition portion 906 and the counterpoise portion 908. Therefore, it is preferable to have a counterpoise portion as long as possible, but the length of the counterpoise portion 908 affects the electric length of the magnetic loop 902. Electrically long magnetic loops result in lower frequencies. Conversely, an electrically short magnetic loop results in a higher frequency. For example, using a counterpoise over the entire length of the left side of the magnetic loop will increase the overall width of the magnetic loop, so that the magnetic loop is electrically shortened, Loop. For example, a frequency of 5.8 GHz is formed instead of the intended target frequency of 5.6 GHz.

Embodiments of the antenna 900 may further include a narrow-light-narrow transition portion as described above and a narrow-light-narrow transition portion as described above, in addition to the transition portion and the counterpoise portion, for tuning the electric length of the magnetic loop to a predetermined frequency. / RTI > and / or a light-narrow-light transition. Embodiments of the self-contained antennas may also include magnetic loops having corners cut at an angle as previously described to improve current flow around the corners of the magnetic loops.

As described above, the counterpoise portion 908 for the electric field radiator 904 may be used instead of the ground plane. The electric field radiator 904 is a substantially monopole antenna. The unipolar antenna is formed by replacing one half of the dipole antenna, and the half has a ground plane perpendicular to the other half. In embodiments of the self-contained antennas, the electric field radiator has a large metal piece electrically connected to an electric field radiator usable in place of the ground plane. In the cross-sectional antenna 440 from Figure 4c, the electric field radiator 444 emits an electric field based on the position of the ground plane used for the antenna 440. Such an electric field rotates at right angles to the plane of the electric field radiator, whereas the magnetic field rotates in a generally coplanar manner with the plane. Such a pattern of electric fields has a generally donut shape which is referred to as a nearly complete omnidirectional pattern. As discussed above, embodiments of cross-sectional antennas need not necessarily provide their own ground plane. Thus, if the antenna 440 is used in a device, the device will act as the ground plane of the antenna 440 and the radiation pattern emitted by the electric field radiator 444 will be returned to the device. However, if the cross-shaped self-contained antenna including the counterpoise portion also includes a ground plane, the above-described radiation patterns can be formed into an electric field that rotates with respect to the plane of the electric field radiator, or an electric field that is coplanar with the plane of the electric field radiator The magnetic field rotating at right angles to the plane on the above planes.

The counterpoise portion 908 need not be machined on or above the upper left corner of the magnetic loop 902. [ In alternate embodiments, the counterpoise portion may be located on the upper right corner, with an electric field radiator 904 positioned continuously on the left side of the magnetic loop 902. Regardless of the physical locations of the counterpoise portion 908 and the electric field radiator 904 (or radiators in the case of more than one), the counterpoise portion and the electric field radiator (s) need to be located 180 degrees from the phase . In yet another embodiment, the length of the counterpoise can also be adjusted as needed. The counterpoise 908 may also be located along the right side of the magnetic loop 902 either directly under the electric field radiator 904 or at other locations around the magnetic loop 902.

The antenna 900 also includes a balun 910. A balun is an electrical transformer type that can convert balanced electrical signals to unbalanced (cross-sectional) signals for ground (difference), or vice versa. In particular, baluns represent high impedance as common mode signals and low impedance as other mode signals. The balun 910 acts as a function to cancel the common mode current. In addition, the balloon 910 tunes the antenna 900 to a predetermined input impedance and also tunes the impedance of the entire magnetic loop 902. The balun 910 has a generally triangular shape and is composed of two portions divided by an intermediate gap 912.

The two portions of the balun 910 are magnetically and electrically coupled. The gap 912 of the balun 910 removes the common mode current by magnetically preventing current from flowing in one direction, such as current flowing back through the transmitter to the device using the antenna 900. This is important because the reflection of the current flows through the transmitter and adversely affects the performance of the antenna 900 or the device using the antenna 900 due to the common mode current. In particular, reflection of current through the transmitter causes interference to the electrical network of the device using the antenna. Such negative performance may also prevent the device from meeting Federal Communications Commission (FCC) regulations. The gap 912 of the balun 910 eliminates the common mode current and thus prevents current from returning into the connector of the antenna 900.

The gap 912 can be adjusted based on antenna design and dimensions. In an embodiment, electromagnetic simulation can be used to visualize the current flowing through the antenna 900. Next, the gap 912 may be increased or decreased until the simulation shows that the current is no longer reflected and returns through the transmitter. The elimination of the common mode current causes the current flowing in one direction into the transmitter to cease and flow in the opposite direction with one direction flowing into the antenna 900 and a second direction flowing out of the antenna 900 Can be visualized as the point at which the current starts.

The tapered sides 914 of the balun 910 serve for electrical coupling purposes. The angle of the tapered side surfaces 914 may be adjusted to impedance match the antenna 900. The individual inductors and individual capacitors are connected to the antenna 900 along a supply line (not shown) to match the impedance of the device that supplies the antenna 900 to the impedance of the antenna 900. [ . For example, if an antenna requires an input of 50 OMEGA, but the device's electrical network supplies 150 OMEGA, then 150 OMEGA supplied to the antenna may be transformed into 50 OMEGA required by the antenna, A series of inductors and capacitors are used. In contrast to this common practice in the industry, embodiments of the self-contained antenna 900 may use any number of external components, such as, for example, a series of inductors and capacitors along the line feeding the antenna 900 It does not need to be impedance matched. Instead, the balun 910 is used to match the impedance of the antenna 900 to the connector that supplies the antenna 900 and to match the impedance of the magnetic loop 902.

The height of the balun 910 is a function of the operating frequency of the antenna 900. Thus, the large balun 910 requires a low frequency, while the short balun 910 requires a high frequency. When using a large balun on the antenna, it is important to bring the balun 910 close to the electric field radiator (s). By placing the balun 910 too close to the electric field radiator 904, a capacitive coupling may be formed between the balun 910 and the electric field radiator 904. It is therefore important that the balun 910 is properly spaced from the electric field radiator 904 to prevent capacitive coupling from affecting the performance of the antenna 900. [ If a particular antenna design requires the use of a large balun depending on the operating frequency of the antenna, to properly impedance match the antenna and also to remove the common mode current, the balun may be used as described in antenna 920 of FIG. 9B You can move down. In an alternative embodiment, a self-contained counterpoise antenna 900 may not include the balun 910.

The antenna 900 is an example of a self-contained type counterpoise composite electric field antenna. Embodiments of the antenna 900 may be printed or otherwise laminated on a about 1.6 mm FR-4 substrate. The characteristics and design of the antenna 900 also make it possible to apply to other materials, including materials that are not found suitable for flexible printed circuits, acrylonitrile butadiene styrene (ABS) plastic and microwave frequencies. The operating frequency of the antenna 900 is approximately 2300 to 2700 MHz and is suitable for a variety of embedded applications including mobile phones, access points, PDAs, laptops, PC-cards, sensors, and automotive applications . Embodiments of the antenna 900 have achieved a peak efficiency of about 94% and a peak gain of about + 3dBi. The antenna 900 has a width of about 31 mm and a length of about 31 mm. The antenna 900 has linear polarization and an impedance of about 50?. The antenna 900 also has a voltage standing wave ratio of less than 2: 1 (< 2: 1). The size and efficiency of the antenna 900 are suitable for Wi-Fi applications, and efficiency, size, and gain are important in the Wi-Fi.

Another embodiment of a cross-sectional antenna with a built-in counterpoise is illustrated in Fig. 10a. The antenna 1000 is an example of an antenna having linear polarization. The antenna does not require a ground plane due to built-in counterpoise. The antenna 1000 may be printed or otherwise laminated on a 1.6 mm thick FR-4 substrate. As with the antenna 900, the characteristics and design of the antenna 1000 can also be applied to other materials, including flexible printed circuit, acrylonitrile butadiene styrene (ABS) plastic, and even materials not found suitable for microwave frequencies. Adaptable. The antenna 1000 operates with a peak efficiency of about 92% and a measured peak gain of about + 3dBi in the frequency range of about 882 MHz to 948 MHz. The antenna 1000 has an antenna impedance of about 50 OMEGA, and also has a voltage standing wave ratio of less than 2: 1 (< 2: 1). The antenna 1000 has a width of about 76 mm and a height of about 76 mm.

The antenna 1000 includes a magnetic loop 1002 having a first electric field radiator 1004 directly coupled to the magnetic loop 1002 and a second electric field radiator 1006 coupled directly to the magnetic loop 1002 ). All of the electric field radiators 1004 and 1006 are coupled to the magnetic loop 1002 without the gain of the electrical traces. The electric field radiators 1004 and 1006 are physically located on the inner side of the magnetic loop 1002. By using two electric field radiators instead of one as in the antenna 900 shown in Fig. 9, the gain of the antenna increases. A curved line 1008 separating the two electric field radiators 1004 and 1006 is used to delay the phase between the two electric field radiators 1004 and 1006 to form farfield patterns additive .

The two electric field radiators 1004 and 1006 together with the curved line 1008 form an electric field radiator array 1010 having a phase delay. In particular, the curved line 1008 ensures that the two electric field radiators 1004 and 1006 have phases that are 180 degrees different from each other. The curved line may be used as a space saving technique. For example, if a small antenna is needed, the two electric field radiators are forced to come close together due to the need for a minimum size, where the curved line 1008 indicates that the electric field radiators are consistently continuously 180 degrees apart Phase. ≪ / RTI > The electrical length of the trace of the curved line 1008 can be adjusted as needed based on the required delay. For example, the traces may be short or long while the width remains constant. Alternatively, the length of the trace may remain constant while the width of the trace is formed to be wide or thick. As discussed above, the electrical length of the trace depends on its physical length and physical width. 10B illustrates an alternative embodiment of a self-contained antenna 1020 that does not have the curved line 1008. FIG.

As described in connection with antenna 900, antenna transitions 1012 and counterpoise 1014 may accordingly be adjusted based on a plurality of factors. The transition 1012 is based on operating frequency, but it must also be sufficiently long so that the counterpoise 1014 is electrically insulated. A counterpoise 1014 as large as possible is suitable. Finally, the balun 1016 removes the common mode current and also matches the impedance of the antenna 1000 with the impedance of the transmitter supplying the feed antenna 1000.

In an alternative embodiment of the antenna 1000, the electric field radiator array 1010 may be arranged on the left side of the antenna 1000 instead of on the right side. In such an alternative embodiment, the counterpoise 1040 may be positioned on the upper right side of the magnetic loop 1002. The counterpoise 1014 may also be located directly below the electric field radiators 1004 and 1006 along the right side of the magnetic loop 1002.

Figs. 11A-11C illustrate the 2D radiation pattern for the antenna 900 of Fig. 11A illustrates a 2D radiation pattern on the XZ plane 1100. FIG. Dashed line 1102 represents the actual radiation pattern and dashed line 1104 represents the 3dB beam width and dashed line 1106 represents the maximum intensity of the electric field along one direction, And indicates the place where the strongest electric field is detected. 11B illustrates a 2D radiation pattern for an antenna 900 on an XY plane 1110 and FIG. 11C illustrates a 2D radiation pattern for an antenna 900 on a YZ plane 1120. FIG.

12A to 12C illustrate the 2D radiation pattern for the antenna 1000 of FIG. 10A. 12A illustrates a 2D radiation pattern on the XZ plane 1200. FIG. The broken line 1204 represents the 3dB beam width and the dotted line 1206 represents the maximum intensity of the electric field along one direction, that is, the dotted line 1206 represents the maximum intensity of the electric field along one direction, And indicates the place where the strongest electric field is detected. FIG. 12B illustrates a 2D radiation pattern for antenna 1000 on XY plane 1210, and FIG. 12C illustrates a 2D radiation pattern for antenna 1000 on YZ plane 1220. FIG.

13A illustrates the voltage standing wave ratio (VSWR) for the antenna 900. FIG. The VSWR configuration shows that the antenna 900 is in good impedance match for a frequency range of about 2.34 GHz to about 2.69 GHz. That is, above the frequency range of about 2.34 GHz to 2.69 GHz, most of the energy supplied into the antenna 900 is emitted rather than returned to the transmitter. Specifically, the inside of the two central vertical solid lines represents the frequency range where the VSWR of the antenna 900 is less than 2: 1 (< 2: 1). 13B illustrates the return loss for the antenna 900. Fig. Return loss and VSWR are mathematically related so that a -10.0 return loss on Fig. 13B corresponds to a VSWR of 2.0 on Fig. 13A. The return loss diagram of FIG. 13B shows that the antenna 900 is well impedance matched between points labeled 1 and 2.

14A illustrates the voltage standing wave ratio (VSWR) for the antenna 1000 of FIG. 10A. The VSWR configuration shows that the antenna 1000 is in good impedance match for a frequency range of about 884 MHz to about 947 MHz. That is, over a frequency range of about 884 MHz to 947 MHz, most of the energy supplied into the antenna 1000 is emitted rather than returned to the transmitter. In particular, the inside of the two central vertical solid lines represents the frequency range where the VSWR of the antenna 1000 is less than 2: 1 (< 2: 1). Fig. 14B illustrates return loss for the antenna 1000. Fig. As described above, a -10.0 reflection loss corresponds to a VSWR of 2.0. The return loss diagram of FIG. 14B shows that the antenna 1000 is well impedance matched between points labeled 1 and 2.

Fig. 15 illustrates another embodiment of the self-contained antenna 1500. Fig. The antenna 1500 is an example of a 5.8 GHz antenna. The dimensions of the particular embodiment of the antenna 1500 are about 15 mm in length and about 15 mm in width. The antenna 1500 is comprised of a magnetic loop 1502 having an electric field radiator 1504 directly bonded to a magnetic loop 1502. In contrast to the self-contained antennas 900 and 1000, the antenna 1500 includes a tapered transition 1506 comprised of two sections 1508 and 1510. The first transition section 1508 starts where the width of the magnetic loop changes from a small width to a large width. The first transition section 1508 tapers linearly toward a small width before the width of the magnetic loop increases again at a location where the second transition section 1510 is initiated. The second transition section increases linearly from a small width to a large width. As described above, by adjusting the width of the trace of the magnetic loop, the electric length of the magnetic loop is allowed to be adjusted. Also, the length, width, and number of the transitions used electrically electrically isolate the counterpoise 1512. The transition portion 1506 should be sufficiently long to minimize the magnitude of the current flowing through the counterpoise 1512. Also, by tapering the transition 1506 with respect to the counterpoise 1512, the bandwidth increases in terms of impedance matching. The balun 1514 removes the common mode current and also matches the impedance of the antenna 1500. An alternative embodiment of the antenna 1500 may not include the balun 1514.

An embodiment includes a magnetic loop having a width and located on a plane generating a magnetic field and having a first inductive reactance; An electric field radiator positioned on a plane emitting an electric field and having a first capacitive reactance and directly coupled to the magnetic loop, the electric field being orthogonal to the magnetic field and the physical arrangement between the electric field radiator and the magnetic loop The electric field radiator forming a two-capacitive reactance; A transition formed on the magnetic loop and having a transition width greater than the width of the magnetic loop; And a counterpoise formed on the magnetic loop positioned along a magnetic loop opposite or adjacent to the electric field radiator, the transition portion generally electrically isolating the counterpoise from the magnetic loop, Type antenna.

Unless otherwise expressly stated, each feature described herein (including any accompanying claims, abstract, and drawings) may be replaced with alternative features provided for the same, equivalent, or similar purpose. Thus, unless expressly stated otherwise, each feature described is merely an example of a set of generic equivalents or similar features.

Although the invention has been described and demonstrated herein in terms of several alternative aspects, it should be understood that the techniques described above may have many additional uses and applications. Accordingly, the present invention is not limited to the specific description, examples and various figures contained in the present application, which simply illustrate exemplary embodiments, and is capable of alternatives and applications to the principles of the present invention.

Claims (11)

As a cross-sectional antenna,
A magnetic loop positioned on a plane and configured to generate a magnetic field; And
An electric field radiator coupled to the magnetic loop and configured to emit an electric field orthogonal to the magnetic field, the electric field radiator being located on the plane and within the magnetic loop,
Wherein the electric field radiator is coupled to the magnetic loop at a point where the current flowing through the magnetic loop is a reflective minimum and the point at which the current is minimum is movable based on the geometry of the magnetic loop,
Wherein the electric field radiator further comprises a second electric field radiator configured to emit the electric field at an operating frequency and having an electrical length, the second electric field radiator being located on a plane and within the magnetic loop, the second electric field radiator being coupled to the magnetic loop The second electric field radiator being configured to emit a second electric field orthogonal to the magnetic field and the second electric field radiator having a second electrical length and configured to emit the second electric field at a second operating frequency, Type antenna.
As a cross-sectional antenna,
A magnetic loop positioned on a plane and configured to generate a magnetic field;
An electric field radiator positioned on the plane and within the magnetic loop, the electric field radiator bound to the magnetic loop and configured to emit an electric field orthogonal to the magnetic field; And
And an electrical trace connecting the electric field radiator to the magnetic loop,
Wherein the electric field radiator is coupled to the magnetic loop at a point where the current flowing through the magnetic loop is a reflective minimum and the point at which the current is minimum is movable based on the geometry of the magnetic loop,
Wherein the electric field radiator further comprises a second electric field radiator configured to emit the electric field at an operating frequency and having an electrical length, the second electric field radiator being located on a plane and within the magnetic loop, the second electric field radiator being coupled to the magnetic loop The second electric field radiator being configured to emit a second electric field orthogonal to the magnetic field and the second electric field radiator having a second electrical length and configured to emit the second electric field at a second operating frequency, Type antenna.
As a cross-sectional antenna,
A magnetic loop positioned on a plane and configured to generate a magnetic field;
An electric field radiator positioned on the plane and within the magnetic loop, the electric field radiator bound to the magnetic loop and configured to emit an electric field orthogonal to the magnetic field; And
And an electrical trace connecting the electric field radiator to the magnetic loop,
Wherein the electric field radiator is coupled to the magnetic loop at a point where the current flowing through the magnetic loop is a reflective minimum and the point at which the current is minimum is movable based on the geometry of the magnetic loop,
The electrical trace having a shape selected from a group of flat curves and a shape minimizing the number of bends in the electrical trace,
Wherein the electric field radiator further comprises a second electric field radiator configured to emit the electric field at an operating frequency and having an electrical length, the second electric field radiator being located on a plane and within the magnetic loop, the second electric field radiator being coupled to the magnetic loop The second electric field radiator being configured to emit a second electric field orthogonal to the magnetic field and the second electric field radiator having a second electrical length and configured to emit the second electric field at a second operating frequency, Type antenna.
As a planar antenna,
One or more magnetic loops positioned on a first plane and configured to generate one or more magnetic fields, wherein the one or more magnetic loops have a first inductive reactance in addition to the overall inductive reactance of the plane antenna;
One or more electric field radiators located on the first plane that emit one or more electric fields orthogonal to the one or more magnetic fields, wherein each electric field radiator of the one or more electric field radiators comprises a magnetic Wherein the one or more electric field radiators have a first capacitive reactance in addition to the total capacitive reactance of the planar antenna and the physical arrangement between the one or more electric field radiators and the one or more magnetic loops The at least one electric field radiators causing a second capacitive reactance in addition to the capacitive reactance; And
A wideband element positioned on a second plane and configured to form a ground plane, the broadband element further comprising a second inductive reactance in addition to the overall inductive reactance and a third inductive Wherein the broadband element is configured to match a broad bandwidth based on one or more physical adjustments to the broadband element such that the total inductive reactance matches the overall capacitive reactance. A flat antenna comprising a broadband element. 5. The flat antenna of claim 4, further comprising at least one phase tracker, wherein each phase tracker of the at least one phase tracker is bound to a respective magnetic loop of the at least one magnetic loops. 6. The flat antenna of claim 5, wherein each phase tracker is physically located within each magnetic loop or physically located outside of each of the magnetic loops. 7. The phase tracker of claim 5 or 6, wherein each phase tracker has a triangular shape, each of the magnetic loops further comprising a drive point, and the tip of the triangle- An electrical angular position, an electrical angular position of 270 degrees from the drive point, and a reflective minimal point at which the current flowing through each magnetic loop is located at the reflective minimal. 7. A method according to any one of claims 4 to 6, wherein the broadband element comprises one or more trapezoidal elements, each trapezoidal element of the one or more trapezoidal elements being connected to the second inductive reactance and the second Lt; RTI ID = 0.0 &gt; 3 &lt; / RTI &gt; capacitive reactance over said wide bandwidth. 9. The flat antenna of claim 8, wherein the one or more physical adjustments include varying the slope of the top side of each of the trapezoidal elements. 9. The system of claim 8, wherein the broadband element comprises one or more choke joints and a ground element, the one or more choke connectors are adapted to isolate the one or more trapezoidal elements from the ground element. A flat antenna comprising: 7. The flat antenna according to any one of claims 4 to 6, wherein the at least one magnetic loops have a rectangular shape.

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