CN105789902B - Composite loop antenna - Google Patents

Abstract

Embodiments relate to a planar (double-sided) and printed (single-sided) composite field antenna. The improvements relate particularly, but not exclusively, to a composite loop antenna having a coplanar electric field radiator with an electric field orthogonal to a magnetic field and a magnetic loop, which achieves performance benefits in terms of higher bandwidth (lower Q), greater radiation intensity/power/gain, and higher efficiency. Other embodiments are directed to a self-contained counterpoise composite field antenna that includes a transition formed on a magnetic loop and has a transition width greater than the width of the magnetic loop. The transitions substantially isolate the counterpoise formed on the magnetic loop opposite or adjacent to the electric field radiator.

Description

Composite loop antenna

The present application is a divisional application of patent applications with application number 201180011656.8(PCT/US2011/024634) and application date of 2011, 2 and 11, and named "composite loop antenna".

RELATED APPLICATIONS

This application claims priority to U.S. application nos. 12/878,016, 12/878,018, and 12/878,020, filed on 8/9/2010, which are non-provisional applications claiming priority to U.S. provisional application No.61/303,594, filed on 11/2/2010.

Brief description of the drawings

Embodiments of the present invention relate to planar (double-sided) and printed (single-sided) composite field antennas, and in particular, but not exclusively, to composite loop antennas having coplanar electric field radiators and magnetic loops with electric fields orthogonal to magnetic fields, which achieve performance benefits in terms of higher bandwidth (lower Q), greater radiation intensity/power/gain, and higher efficiency. Other embodiments relate to a self-contained counterpoise composite field antenna that includes transitions formed on a magnetic loop and has a transition width greater than the width of the magnetic loop. The transitions substantially isolate the counterpoise formed on the magnetic loop opposite or adjacent to the electric field radiator.

Background

The ever-decreasing size of modern telecommunication equipment creates a need for improved antenna designs. Antennas known in devices such as handsets/mobile phones offer one of the main limitations in performance and are almost always compromised in one way or another.

In particular, the efficiency of the antenna may have a major impact on the device performance. A more efficient antenna radiates a higher proportion of the energy supplied to the antenna from the transmitter. Likewise, due to the inherent interaction of antennas, more efficient antennas convert more of the received signal into electrical energy for receiver processing.

In order to ensure maximum transfer of energy (in both transmit and receive modes) between the transceiver (the device operating as both a transmitter and a receiver) and the antenna, the impedances of the two should be matched in magnitude. Any mismatch between the two will result in poor performance, in the transmit case, reflecting energy back from the antenna to the transmitter. When operating as a receiver, the poor performance of the antenna results in lower received power than would otherwise be possible.

Known simple loop antennas are typically current fed devices, which mainly generate a magnetic (H) field. Thus, such antennas are not typically suitable as transmitters. This is particularly true for small loop antennas (i.e., those that are smaller than one wavelength or have a diameter that is smaller than one wavelength). In contrast, voltage fed antennas, such as dipole antennas, generate both electrical (E) and H fields and can be used in both transmit and receive modes.

The amount of energy received by or transmitted from the loop antenna is determined in part by the area of the antenna. Typically, each time the area of the ring is halved, the amount of energy that can be received/transmitted is reduced by about 3dB depending on the application parameters such as initial size, frequency, etc. This physical constraint tends to mean that very small loop antennas cannot be used in practice.

The composite antenna is such that: both transverse electromagnetic (TM) and Transverse Electric (TE) modes are excited to obtain higher performance benefits such as higher bandwidth (lower Q), greater radiation intensity/power/gain, and higher efficiency.

In the late 1940 s, Wheeler and Chu first examined the performance of electrically short (ELS) antennas. Through their work, several numerical formulas were created to describe the limitations of antennas as their physical size decreased. One of the limitations of ELS antennas mentioned by Wheeler and Chu is that they have a large radiation quality factor Q, since they store more energy on time than they radiate on average, which is of particular importance. According to Wheeler and Chu, ELS antennas have a high radiation Q, which results in minimal resistive losses in the antenna or matching network and in very low radiation efficiency, typically between 1-50%. As a result, since the 1940 s, it has been accepted by the scientific community that ELS antennas have a narrow bandwidth and poor radiation efficiency. Many of today's efforts in wireless communication systems using ELS antennas have resulted from rigorous experimentation and optimization of modulation schemes and communication network protocols, but the ELS antennas used commercially today still reflect the narrow bandwidth, inefficient nature first proven by Wheeler and Chu.

In the early 1990 s, Dale m.grimes and Craig a.grimes claimed to mathematically discover certain combinations of TM and TE modes operating together in ELS antennas that exceed the low radiation Q limit established by Wheeler and Chu's theory. Grimes and Grimes describe their operation in a journal entitled "bandwidth and Q of antennas radiating TE and TM modes" published in the IEEE proceedings for electromagnetic compatibility, 5 months 1995. These statements initiate much debate and lead to the term "composite field antenna" in which both TM and TE modes are excited, as opposed to a "simple field antenna" which excites either the TM mode or the TE mode alone. The benefits of compound field antennas have been demonstrated mathematically by several respected RF experts, including teams employed by the U.S. department of air war, naval war, weapons, where they concluded evidence of radiation Q below the Wheeler-Chu limit, increased radiation intensity, directivity (gain), radiated power and radiated efficiency (p.l. overfelft, d.r. bowling, d.j.white, "co-located magnetic loop, electric dipole array antennas (preliminary results)," reported in 9 months of 1994).

Complex field antennas have proven to be complex and physically difficult to implement due to the unwanted effects of component coupling and the associated difficulties in designing low loss passive networks to combine electric and magnetic radiators.

There are many examples of two-dimensional non-composite antennas, which are typically constructed from printed metal strips on a circuit board. However, these antennas are voltage feeds. An example of one such antenna is a Planar Inverted F Antenna (PIFA). Most similar antenna designs also consist primarily of quarter-wave (or some multiple of quarter-wave), voltage-fed, dipole antennas.

Planar antennas are also known in the art. For example, U.S. patent 5,061,938 to Zahn et al requires an expensive Teflon (Teflon) substrate or similar material for operation of the antenna. U.S. patent 5,376,942 to Shiga teaches a planar antenna capable of receiving, but not transmitting, microwave signals. The Shiga antenna further requires an expensive semiconductor substrate. U.S. patent 6,677,901 to Nalbandian relates to a planar antenna that requires a substrate to have a permittivity to permeability ratio of 1:1 to 1:3 and to be operable only in the HF and VHF frequency ranges (3 to 30Mhz and 30 to 300 Mhz). Although it is known to print some lower frequency devices on inexpensive glass reinforced epoxy laminates such as FR-4, which are commonly used in ordinary printed circuit boards, the dielectric losses in FR-4 are considered to be too high and the dielectric constant is not sufficiently tightly controlled for such substrates to be used at microwave frequencies. For these reasons, alumina substrates are more commonly used. Furthermore, none of these planar antennas are composite loop antennas.

The basis for the increased performance of a composite field antenna in terms of bandwidth, efficiency, gain and radiation intensity derives from the effect of the energy stored in the near field of the antenna. In RF antenna design, it is desirable to convert as much energy as possible present to the antenna into radiated power. The energy stored in the near field of the antenna has historically been referred to as reactive power and is used to limit the amount of power that can be radiated. When discussing complex power, there is a real part and an imaginary part (often referred to as "reactive"). Real power leaves the source and never returns, while virtual or reactive power tends to oscillate (within a half wavelength) centered at the fixed position of the source and interact with the source, thereby affecting the operation of the antenna. The presence of real power from multiple sources may be directly added, while multiple sources of imaginary power may be added or subtracted (cancelled). The benefit of the composite antenna is that it is driven by both TM and TE sources, which allows engineers to create designs that take advantage of reactive power cancellation previously not available in simple field antennas, thereby improving the real power transmission performance of the antenna.

In order to be able to cancel reactive power in a composite antenna, the electric and magnetic fields should operate orthogonal to each other. Although a large number of configurations of electric field radiators required to emit electric fields and magnetic loops required to generate magnetic fields have been proposed, all of these designs have always stayed on three-dimensional antennas. For example, U.S. patent 7,215,292 to McLean requires a pair of magnetic circuits in parallel planes with an electric dipole in a third parallel plane located between the pair of magnetic circuits. U.S. patent 6,437,750 to Grimes et al requires that two pairs of magnetic loops and electric dipoles be physically arranged orthogonal to each other. U.S. patent application US 2007/0080878 filed by McLean teaches an arrangement in which the magnetic and electric dipoles are also in orthogonal planes.

Drawings

FIG. 1 illustrates a planar implementation of an embodiment;

FIG. 2 shows a circuit layout of an embodiment incorporating four discrete antenna elements;

FIG. 3A shows a detailed view of one of the antenna elements of FIG. 2 including a phase tracker;

FIG. 3B shows a detailed view of one of the antenna elements of FIG. 2 without the phase tracker;

FIG. 4A shows an embodiment of a small single-sided composite antenna;

FIG. 4B shows an embodiment of a small single-sided composite antenna with magnetic loops cut at an angle of approximately 45 degrees;

FIG. 4C illustrates an embodiment of a small single-sided composite antenna with a magnetic loop having two symmetric wide-narrow-wide transitions;

FIG. 5 shows an embodiment of a small two-sided composite antenna;

fig. 6 shows an embodiment of a large composite antenna array consisting of four composite antenna elements;

fig. 7 shows how the size of the phase tracker affects its inductance and capacitance;

figure 8 shows the ground plane of the antenna embodiment of figure 6;

FIG. 9A illustrates an embodiment of a self-contained counterpoise antenna with a balun;

fig. 9B shows an alternative embodiment of the balun-pulled antenna of fig. 9A;

FIG. 10A illustrates an embodiment of a self-contained counterpoise antenna with an array of electric field radiators and curvilinear traces between the electric field radiators;

FIG. 10B illustrates an embodiment of a self-contained counterpoise antenna with an array of electric field radiators but without curvilinear traces;

11A-11C show approximately the 2D radiation patterns for the antenna in FIG. 9;

12A-12C show approximately the 2D radiation patterns for the antenna in FIG. 10A;

FIG. 13A shows approximately a graph of voltage standing wave ratio for the antenna in FIG. 9;

FIG. 13B shows a graph that approximately shows the measured return loss for the antenna in FIG. 9;

FIG. 14A shows approximately a graph of the voltage standing wave ratio for the antenna in FIG. 10;

FIG. 14B shows a graph of measured return loss approximately for the antenna in FIG. 10; and

fig. 15 shows approximately an embodiment of a self-contained counterpoise antenna with conical transitions.

Detailed Description

Embodiments provide an improved planar complex loop (CPL) antenna that is capable of operating in both transmit and receive modes and that is capable of achieving better performance than known loop antennas. The two main components of a CPL antenna are a magnetic loop that generates a magnetic field (H-field) and an electric field radiator that emits an electric field (E-field).

The electric field radiator may be physically located inside or outside the loop. For example, fig. 1 shows an embodiment of a single CPL antenna element with an electric field radiator located inside the loop coupled by the electric traces, while fig. 3A and 3B show two embodiments of a single CPL antenna element with an electric field radiator located outside the loop. As described further below, fig. 3A includes a phase tracker for wideband applications, while fig. 3B does not include a phase tracker and is more suitable for less wideband applications. Fig. 4A, 4B and 4C show other embodiments of small single-sided antennas with electric field radiators located within the magnetic loop. Embodiments of antennas constructed using any of these techniques can be easily assembled into mobile or handheld devices, such as phones, PDAs, laptops, or as separate antennas. Fig. 2 and other figures illustrate embodiments of a CPL antenna array using microstrip construction techniques. Such printing techniques allow for the design and construction of compact and consistent antennas.

The antenna 100 shown in fig. 1 is arranged and printed on a portion of a printed circuit board 101. The antenna comprises a magnetic loop 110, which in this case is substantially rectangular and with a wide open base. The two ends of the wide open base are fed from the coaxial cable 130 at the drive point in a known manner.

Located inside the loop 110 is an electric field radiator or series resonant circuit 120. The series resonant circuit 120 takes the form of a J-shaped trace 122 on the circuit board 101, which is coupled to the loop 100 by means of a meandering trace 124, the meandering trace 124 operating as an inductor, meaning it has an inductive or inductive reactance. The J-shaped trace 122 has substantially capacitive reactance performance dictated by its dimensions and the material used for the antenna. Trace 122, along with meandering trace 124, acts as a series resonant circuit.

Antenna 100 is presented herein for ease of understanding. Actual embodiments may not physically resemble the antennas shown. In this case, shown as feeding from a coaxial cable 130, i.e., one end of the loop 132 is connected to the center conductor of the cable 130 and the other end of the loop 134 is connected to the sheath of the cable 130. The loop antenna 100 differs from known loop antennas in that the series resonant circuit 120 is coupled to a portion of the loop 134 in a manner that surrounds the loop circumference. This coupling position plays an important role in the operation of the antenna as described below.

By carefully positioning the series resonant circuit 120 and the meandering trace 124 relative to the magnetic loop 110, the E and H fields generated/received by the antenna 100 can be made orthogonal to each other, without physically placing the electric field radiator orthogonal to the magnetic loop 110. This orthogonal relationship has the effect of enabling electromagnetic waves emitted by the antenna 100 to propagate efficiently through space. To achieve this effect, the series resonant circuit 120 and the meandering trace 124 are placed at an electrical potential of about 90 degrees or about 270 degrees along the magnetic loop 110. In an alternative embodiment, the meandering trace 124 may be placed at a point along the magnetic loop 110 where the current flowing through the magnetic loop is at a reflective minimum. Thus, the meandering trace 124 may or may not be placed at about a 90 or 270 degree electrical point. The point along the magnetic loop 110 at which the current is at a reflection minimum depends on the geometry of the magnetic loop 110. For example, the point at which the current is at the reflected minimum may initially be identified as the first region of the magnetic loop. After adding or removing metal to the magnetic loop to obtain impedance matching, the point at which the current is at a reflection minimum may be changed from the first region to the second region.

The magnetic circuit 110 may be any of a number of different electrical and physical lengths; however, for more efficient operation of the antenna, the electrical length is a multiple of the wavelength, quarter wavelength and eighth wavelength with respect to the desired frequency band(s). Adding inductance to the magnetic loop may increase the electrical length of the magnetic loop. Adding capacitance to the magnetic circuit has the opposite effect of reducing the electrical length of the magnetic circuit.

The orthogonal relationship between the H-field and the E-field can be achieved by placing the series resonant circuit 120 and the meandering trace 124 at a physical location that is 90 degrees or 270 degrees around the magnetic loop from the drive point, which physical location varies based on the frequency of the signal transmitted/received by the antenna. As noted, this position may be 90 degrees or 270 degrees from the drive point(s) of the magnetic circuit 110, which are determined by the ends 132 and 134, respectively. Thus, if the end 132 is connected to the center conductor of the cable 130, the meandering trace 124 may be positioned at a 90 degree point, as shown in fig. 1, or at a 270 degree point (not shown in fig. 1).

The orthogonal relationship between the H and E fields may also be achieved by placing the series resonant circuit 120 and the meandering trace 124 at a physical location around the magnetic loop where current flowing through the magnetic loop is at a reflective minimum. As noted previously, the location where the current is at the reflection minimum depends on the geometry of the magnetic loop 110.

By arranging the circuit elements in such a way that there is a 90 degree phase relationship between the devices, an orthogonal relationship is created between the E and H fields, which enables the antenna 100 to function more efficiently as both a receive and transmit antenna. The H-field is generated solely (or substantially solely) by the magnetic loop 110 and the E-field is transmitted through the series resonant circuit 120, which transfers the energy transmitted from the antenna across a very large distance in a form suitable for transmission.

The series resonant circuit 120 includes an inductive (L) device and a capacitive (C) device, the values of which are selected to resonate at the operating frequency of the antenna 100 and to cause the inductive reactance to match the capacitive reactance. This is because when the reactance of the capacitive device is equal to the reactance of the inductive device, i.e. when X_L＝X_CWhen in use, togetherVibration occurs most efficiently. The values of L and C may then be selected to give the desired operating range. Other forms of series resonant circuits, for example using a crystal oscillator, may be used to give other operating characteristics. If a crystal oscillator is used, the Q of such a circuit is much larger than that of the simple L-C circuit shown, and this will therefore limit the bandwidth characteristics of the antenna.

As noted above, the series resonant circuit 120 effectively operates as an E-field radiator (which due to the interaction inherent in the antenna means it is also an E-field receiver). As shown, the series resonant circuit 120 is a quarter-wave antenna, but the series resonant circuit may also operate as a multiple of a full-wavelength, a multiple of a quarter-wave, or a multiple of an eighth-wave antenna. If special restrictions prevent the desired wavelength of material from being used as the traces 122, the meandering traces 124 may be utilized as a means of increasing propagation delay in order to achieve an electrically equivalent full, quarter, or eighth wavelength series resonant circuit 120. It is theoretically possible to use only a rod antenna of the desired wavelength instead of a series resonant circuit, but in practice this is generally not the case, as long as it is physically connected to the loop at the point of 90/270 degrees or at the point where the current through the magnetic loop is at the reflection minimum and obeys X_L＝X_CThe requirements are met.

As noted above, the positioning of the series resonant circuit 120 is important: it may be located and coupled to the loop at a point where the phase difference between the E and H fields is 90 degrees or 270 degrees or at a point where the current flowing through the magnetic loop is at a reflection minimum. From here on, the point at which the series resonant circuit 120 is coupled to the magnetic loop 110 will be referred to as the "connection point", the connection point at 90 or 270 degrees electrical point along the magnetic loop will be referred to as the "90/270 connection point", and the connection point at which the current is at a reflective minimum will be referred to as the "reflective minimum connection point".

The amount of variation in the location of the connection point depends to some extent on the intended use of the antenna and the magnetic loop geometry. For example, the optimal connection point may be found by comparing the performance of an antenna using 90/270 connection points with the performance of an antenna using reflective minimum connection points. The connection point that yields the highest efficiency for the intended use of the antenna may then be selected. The 90/270 connection point may not be different from the reflection minimum connection point. For example, an embodiment of the antenna may have a current at a reflection minimum at the 90/270 degree point or near the 90/270 degree point. If an 90/270 degree connection point is used, the amount of change from the exact 90/270 degrees will depend somewhat on the intended use of the antenna, but in general, the closer it is placed to 90/270 degrees, the better the performance of the antenna. Ideally, the magnitudes of the E and H fields should also be identical or substantially similar.

In practice, the point at which the series resonant element 120 is coupled to the loop 110 can be found experimentally by using E and H field probes that define the 90/270 degree position or point at which the current is at the reflection minimum. The point at which the meandering trace 124 should be coupled to the loop 110 may be determined by moving the trace 124 until the desired 90/270 degree difference is observed. Another method for determining the 90/270 connection point and the reflection minimum connection point along the loop 110 is to visualize the surface current in an electromagnetic software simulation program, where the optimal connection point along the loop 110 will be visualized as the area(s) of minimum surface current magnitude(s).

Therefore, it is necessary to ensure the optimum performance of the antenna according to experimental measurements and the degree of successive approximation, even if the principle of the preferential element arrangement is well understood. This is simply due to the nature of the printed circuit, which often requires a degree of "tuning" before the desired performance is achieved.

Known simple loop antennas provide a very wide bandwidth, typically one octave, whereas known antennas such as dipole antennas have a much narrower bandwidth, typically a much smaller fraction of the operating frequency (such as 20% of the operating center frequency).

Printed circuit technology is well known and will not be discussed in detail herein. It is sufficient to arrange and print (typically via etching or laser trimming) copper traces on a suitable substrate with a specific dielectric effect. By careful selection of materials and dimensions, specific values for capacitance and inductance can be achieved without the need for separate discrete components. However, as will be described further below, the design of the present embodiment alleviates the substrate limitations of previous higher frequency planar antennas.

As noted, the present embodiment is arranged and fabricated using known microstrip technology, where the final design is reached as a result of a certain amount of manual calibration, thereby adjusting the physical traces on the substrate. In practice, a calibrated capacitive rod is used, which comprises a metal element with a known capacitive element, for example, 2 picofarads. Capacitive rods may be placed in contact with various portions of the antenna trace, for example, while the performance of the antenna is being measured.

At the hands of a skilled technician or designer, this technique reveals that the traces that make up the antenna should be adjusted in size, equivalent to adjusting capacitance and/or inductance. After a number of iterations, an antenna with the desired performance may be achieved.

The connection point between the series resonant element and the loop is again determined experimentally using E and H field probes. Once the approximate connection location has been determined, bearing in mind the frequencies discussed herein, the slightest disturbance from the test equipment may have a large practical effect, and fine adjustments may be made to the connections and/or the values of L and C by in situ laser trimming of the traces. Once the final design is established, it can be reproduced with good repeatability. Alternatively, the surface current may be visualized using an electromagnetic software simulation program, and the region or regions where the surface current is at a minimum may be selected to determine the connection point between the series resonant element and the loop.

Antennas constructed in accordance with the embodiments discussed herein provide significant efficiency gains over known antennas of similar volume.

In other embodiments, multiple discrete antenna elements may be combined to provide greater performance than can be achieved using a single element.

Fig. 2 shows an antenna 200 arranged and printed on a portion of a circuit board 205 in a known manner. Although the circuit board 205 is shown in plan view, there is a certain amount of thickness to the substrate that makes up the circuit board, and a ground plane (not shown) is printed on the back side of the circuit board 205 in a manner similar to the ground plane area 624 shown in fig. 6 and 8. In fig. 2, the antenna 200 comprises 4 separate, functionally identical antenna elements 210 arranged in two groups, each driven in parallel.

The effect of providing multiple instances of the base antenna element 210 is to improve the overall performance of the antenna 200. In the absence of losses associated with the construction of the antenna, it is theoretically possible to construct an antenna comprising a large number of instances of individual base antenna elements 210, adding 3dB of gain to the antenna per even number of elements. In practice, however, losses-especially dielectric heating effects-mean that it is not possible to add extra components indefinitely. The example of a four-element antenna shown in figure 2 is good to the extent physically possible and adds 6dB (minus any dielectric heating losses) over an antenna consisting of a single element.

The antenna 200 of fig. 2 is suitable for use in a microcellular base station or other part of a fixed wireless infrastructure, while the single element 210 is suitable for use in a mobile device such as a mobile phone or handset, pager, PDA or portable computer. The only really determined problem is size. The components and operation of element 210 are further explained and illustrated in fig. 3A and 3B with reference to antennas 310 and 370, respectively.

Fig. 3A shows a single antenna 310 (an embodiment of one of the elements 210 of fig. 2) that is capable of achieving a larger bandwidth, equivalent to one or one half octave, as described below, by including a phase tracking antenna element 330, the phase tracking antenna element 330 having been specifically adapted to provide a larger operating bandwidth (wider bandwidth) than the narrower bandwidth antenna 100 of fig. 1. In particular, this wider bandwidth is achieved by the combination of the phase tracker 330 with the rectangular electric field radiator 320 and the loop element 350. A rectangular electric field radiator 320 replaces the series resonant circuit 120 shown in figure 1. However, the operating bandwidth of the rectangular electric field radiator 320 is wider than the operating bandwidth of the tuning circuit 120 due to the operation of the phase tracker 330, as explained further below.

An alternative embodiment of an antenna 310 is shown in fig. 3B as antenna 370, having the same rectangular electric field radiator 320, loop element 350 and drive or feed point 340 as antenna 310 of fig. 3A, but lacking phase tracker 330, and thus having a narrower bandwidth of operation than antenna 310. Another approach for combining wide bandwidth operation is described by the CPL antenna element in fig. 4A, which incorporates multiple electric field radiators 404 and 408, as described further below.

In the case of the tuning circuit 120, the connection point between the tuning circuit and the loop is important in determining the overall performance of the antenna 100. In the case of the electric field radiators 320 in the antennas 310 and 370 in fig. 3A and 3B, located outside of the loop 350, while the battery radiators are still generally disposed at the midpoint of 90/270 degrees around the loop 350 at the center frequency or at a point where the current is at a reflective minimum, precise positioning is less important since the connection points are effectively distributed along the length of one side of the electric field radiators. Thus, the edges of the electric field radiator 320 coincide with the ends of the loop 350, along with the size of the loop, to define the operating frequency range of the antennas 310 and 370.

The size of the loop 350 is also important in determining the operating frequency of the antennas 310 and 370. In particular, as previously mentioned, the total length of the loop 350 is a critical dimension. To allow for a wide operating frequency range, a triangular phase tracker element 330 is placed directly opposite the electric field radiator 320 (in one of the two possible positions as shown in fig. 2). Phase tracker 330 effectively acts as an automatic variable length tracking device that lengthens or shortens the electrical length of loop 350 depending on the frequency of the RF signal fed at feed or drive point 340.

The phase tracker 330 behaves as a nearly infinite series of L-C components, only some of which will resonate at a given frequency, thereby automatically changing the effective length of the loop. In this way, a wider bandwidth of operation can be achieved compared to a simple loop with no such phase tracking component.

As shown in fig. 2, phase tracker 330 has two different possible positions. These positions are selected for each antenna element 210 in the group of antenna elements 210 shown in fig. 2 to minimize mutual interference between adjacent antenna elements 210. From an electrical point of view, both configurations are functionally identical.

Larger bandwidths of antennas 310 and 370 (up to 1)¹/₂Octave) is possible because the magnetic loop 350 is a dead short of the signal current. As shown in fig. 3A and 3B, the magnetic circuit is a dead short because it is a half-wave short, but it may also be a dead short between a quarter-wave open and a full-wave short. The phase of the antenna is determined by the dimension 360. Dimension 360 spans the length of the electric field radiator 320 and the length of the left side of the magnetic loop 350. The signals are shorted at the point where the signals are 180 degrees out of phase. A magnetic field with a maximum amplitude is generated by the magnetic loop and there is a magnetic field of smaller amplitude generated by the electric field radiator. Furthermore, the magnetic loop can vary in length from an RF short with very low real impedance to an approximate RF open with very high real impedance. The highest magnitude electric field is emitted by one or more electric field radiator elements. However, the magnetic loop also generates a small electric field opposite to the magnetic field that is lower in magnitude than the electric field emitted by the electric field radiator.

The efficiency of the antenna is achieved by maximizing the current in the magnetic loop to produce the highest possible H-field. This is achieved by designing the antenna such that the current moves into the E-field radiator and is reflected back in the opposite direction, as further described below in fig. 6. The maximized H-field extends from the antenna in all directions, which maximizes the efficiency of the antenna since more current is available for transmission purposes. The maximum H-field energy that can be generated occurs when the magnetic loop is a full RF short or when the magnetic loop has a very low real impedance. However, under normal circumstances, an RF short is undesirable because it will burn out the transmitter driving the antenna. The transmitter produces a set amount of energy at a set impedance. By utilizing an impedance matched to the electric field properties, it is possible to have a near RF short circuit loop without burning the transmitter.

The current flowing through the magnetic loop flows into the electric field radiator. The current is then reflected back into the magnetic loop in the opposite direction by the electric field radiator, causing the electric field to reflect into the magnetic field creating a short circuit of the electric field radiator and creating orthogonal electric and magnetic fields.

Dimension 365 consists of the width of the electric field radiator 320. Dimension 365 does not affect the efficiency of the antenna, but its width determines whether the antenna is narrowband or wideband. Dimension 365 has only a large width to widen the band of antenna 310 shown in fig. 3A.

All trace elements of the magnetic loop shown in fig. 3A, for example, can be made very thick without affecting the performance or efficiency of the antenna. However, making these loop element traces thicker makes it possible to accept larger input powers and additionally modifies the physical dimensions of the antenna to fit the desired space, such as may be required by many different portable devices (such as mobile phones) operating within a specified frequency range.

It will be clear to the skilled person that any form of E-field radiator may be used in the multi-element configuration shown in figures 2, 3A and 3B, of which the rectangular electric field radiator 320 is merely an example. Likewise, single element embodiments may use rectangular electric field radiators, tuned circuits, or any other suitable form of antenna. The multi-element version shown in fig. 2 uses four discrete elements 210, but this can vary up and down depending on the exact system requirements and available space, as will be explained, with some limitations on the upper limit of the elements 210.

Embodiments of the present disclosure allow for the use of single or multi-element antennas, operable over a much increased bandwidth and having superior performance characteristics compared to known antennas of similar size. Furthermore, no complex components are required, resulting in inexpensive devices applicable to a wide range of RF devices. Embodiments of the present disclosure find particular application in mobile telecommunications devices, but may be used in any device requiring an efficient antenna.

Embodiments consist of small single-sided composite antennas ("single-sided antennas" or "printed antennas"). "single-sided" means that the antenna elements are located or printed on a single layer or plane when desired. As used herein, the term "printed antenna" applies to any single-sided antenna disclosed herein, whether the elements of the printed antenna are printed or produced in some other manner, such as etching, depositing, sputtering, or coating a metal layer on a surface or placing a non-metallic material around a metal layer. Multiple layers of single-sided antennas may be combined into a single device to allow wider bandwidth operation in a smaller physical volume, but each device will still be single-sided. The single-sided antennas described below have no ground plane on the back or lower plane and, as such, are short-circuited devices in nature, which represent a new concept in antenna design. A single-sided antenna is balanced but may be driven with a balanced line or an unbalanced line if a significant ground plane is present in the intended application. The physical dimensions of such antennas may vary significantly depending on the performance characteristics of the antenna, but the antenna 400 shown in fig. 4A is approximately 2cm by 3 cm. Smaller or larger embodiments are possible.

The single-sided antenna 400 consists of two electric field radiators physically located within a magnetic loop. In particular, as shown in fig. 4A, the single-sided antenna 400 is comprised of a magnetic loop 402, a first electric field radiator 404 connected to the magnetic loop 402 via a first electrical trace 406, and a second electric field radiator 408 connected to the magnetic loop 402 via a second electrical trace 410. Electrical traces 406 and 410 connect electric field radiators 404 and 408 to magnetic loop 402 at respective 90/270 degree electrical locations relative to the feed or drive point. Optionally, electrical traces 406 and 410 may connect electric field radiators 404 and 408 to the magnetic loop in areas where current flowing through the magnetic loop is at a reflective minimum. As discussed above, the connection or coupling point of traces 406 and 410 changes for different frequencies, explaining why radiator 404 of one frequency is shown connected to loop 402 at a different point than radiator 408 at a different frequency. At lower frequencies, the wave takes longer to reach the 90/270 degree point; the physical location of the 90/270 degree point is thus higher along the magnetic loop than the higher frequency waves. At higher frequencies, less time is spent reaching the 90/270 degree point, resulting in the 90/270 degree point being physically located lower along the magnetic loop than the lower frequency wave. Similarly, the point along the magnetic loop where the current is at a reflection minimum may also depend on the frequency of the electric field radiator. Finally, alternative embodiments of the antenna 400 may consist of one or more electric field radiators coupled directly to the magnetic loop 402 without electrical traces.

The electric field radiators 404 also have different dimensions than the electric field radiators 408 because each electric field radiator emits waves of a different frequency. A smaller electric field radiator 404 will have a smaller wavelength and thus a higher frequency. The larger electric field radiator 408 will have a longer wavelength and a lower frequency.

The physical arrangement of the electric field radiator(s) physically located within the magnetic loop may reduce the size of the overall antenna, while at the same time providing a broadband device, compared to other embodiments in which the electric field radiator(s) and the magnetic loop are physically located outside each other. Alternative embodiments may have a different number of electric field radiators, each arranged at a different location around the loop. For example, the first embodiment may have only one electric field radiator located inside the magnetic loop, while the second embodiment with two electric field radiators may have one electric field radiator located inside the magnetic loop and a second electric field radiator located outside the magnetic loop. Alternatively, more than two electric field radiators may be physically located inside the magnetic circuit. As with the other antennas described above, the single-sided antenna 400 is a transducer due to electric and magnetic fields.

As noted, the use of multiple electric field radiators allows for broadband functionality. Each electric field radiator may be configured to emit waves of a different frequency, resulting in the electric field radiator covering a broadband range. For example, the single-sided antenna 400 may be configured to cover the standard IEEE802.11b/g radio frequency range using two electric field radiators configured at two frequency ranges. The first electric field radiator 404 can be configured to cover, for example, a 2.41GHz frequency, and the second electric field radiator 408 can be configured to cover, for example, a 2.485GHz frequency. This would allow the single-sided antenna 400 to cover a frequency band of 2.41GHz to 2.485GHz, which corresponds to the ieee802.11b/g standard. The use of two or more electric field radiators allows for wideband operation without the use of phase trackers (as shown in figures 2 and 3), as shown with reference to the physically larger antenna embodiments described above. In an alternative embodiment, a wideband antenna may also be implemented by tapering the plurality of electric field radiators using logarithmic indexing, similar to a YAGI antenna.

The length of the electric field radiator generally determines the frequency that the electric field radiator will cover. Frequency is inversely proportional to wavelength. Thus, a small electric field radiator will have a smaller wavelength, resulting in a higher frequency wave. On the other hand, a large electric field radiator will have a longer wavelength, resulting in lower frequency waves. However, these generalizations are also specific implementations.

For optimum efficiency, the electric field radiator should have an electrical length of about a multiple, quarter or eighth of a wavelength at the frequency it generates. As previously mentioned, if the amount of physical space available limits the electrical length of the electric field radiator to less than the desired wavelength, the meandering trace can be used to increase the propagation delay and electrically lengthen the electric field radiator.

In fig. 4A and 4B, electrical traces 406 and 410 are inductive and their respective lengths determine their inductance to their shape or other characteristics. For optimum efficiency, the inductive reactance of the electrical traces should be matched to the capacitive reactance of the corresponding electric field radiator. Electrical traces 406 and 410 are bent in order to reduce the overall size of the antenna. For example, the curve of the electrical trace 406 may be closer to the magnetic loop 402 than to the electric field radiator 404, or the curve of the trace 406 may face down than up, similar to the electrical trace 410. Shaping the electrical trace to extend its length is not because the shape has any particular meaning other than in this context. For example, instead of having straight electrical traces, curves may be added to the electrical traces in order to increase their length and increase their inductive reactance accordingly. However, sharp corners on the electrical traces and the sinusoidal shape of the electrical traces can negatively impact the efficiency of the antenna. In particular, electrical traces with sinusoidal shapes cause the electrical traces to emit small electric fields that are out of phase with the electric field radiator portions, thereby reducing the efficiency of the antenna. Thus, the efficiency of the antenna can be improved by using electrical traces shaped as curves that are flexible and graceful and with as few bends as possible.

The spacing between elements in the single-sided antenna 400 adds capacitance to the overall antenna. For example, the spacing between the top of the electric field radiator 404 and the magnetic loop 402, the spacing between the two electric field radiators 404 and 408, the spacing between the left sides of the electric field radiators 404 and 408 and the magnetic loop 402, the spacing between the right sides of the electric field radiators 404 and 408 and the magnetic loop 402, and the spacing between the bottom of the electric field radiator 408 and the magnetic loop 402 all affect the capacitance of the antenna 400. As previously described, in order for the antenna 400 to resonate with an optimal frequency, the inductive and capacitive reactance of the entire antenna should be matched over the desired frequency band(s). Once the inductive reactance has been determined, the distance between the various elements may be determined based on the capacitive reactance value required to match the inductive reactance value for the antenna.

Given a set of equations to find the spacing between elements and the associated fringe capacitance, a multi-objective optimization method can be used to determine the optimal spacing between elements. Linear programming can be used to optimize the optimal spacing between elements or between any two adjacent antenna elements. Optionally, a non-linear programming such as a genetic algorithm may be used to optimize the interval values.

As noted previously, the size of the single-sided antenna 400 depends on many factors, including the desired operating frequency, narrowband-to-wideband functionality, and tuning of capacitance and inductance.

In the case of the antenna element 400 in fig. 4A, the length of the magnetic loop 402 is one wavelength (360 degrees), which is designed for the optimum frequency, although multiples of other wavelengths may also be used. When designed for optimal frequencies, a portion of the magnetic loop will also act as an electric field radiator, and the electric field radiator will generate a small magnetic field, increasing the directivity and efficiency of the antenna. The length of the magnetic circuit may also be arbitrary, or on the order of multiple wavelengths, quarter wavelengths or eighth wavelengths, with some lengths being increased more frequently than others. One wavelength is open circuit for voltage and short circuit for current. Optionally, the length of the magnetic loop 402 may be physically smaller than the wavelength but additional inductance may be added by increasing the propagation delay to electrically lengthen the loop. The width of the magnetic loop 402 is based primarily on its desired effect on the inductance of the magnetic loop 402 and its capacitance. For example, making the magnetic loop 402 physically shorter will make the wavelength smaller, resulting in a higher frequency. In designing for the optimum frequency of the magnetic loop 402, the inductance and capacitance should satisfy the equation of w ═ 1/sqrt (lc), where w is the wavelength of the loop 402. Thus, the magnetic loop 402 can be tuned by changing the inductance and capacitance that affect the electrical length. Reducing the width of the magnetic loop also increases the inductance. In a thinner magnetic loop, more electrons have to be pushed through a smaller area, increasing the delay.

The top 412 of the magnetic loop 402 is thinner than any other portion of the magnetic loop 402. This allows the size of the magnetic circuit to be adjusted. The top 412 may be reduced since it has minimal impact on the 90/270 degree connection point. Further, trimming the top 412 of the magnetic loop 402 increases the electrical length of the magnetic loop 402 and increases the inductance, which may help match the inductive reactance to the total capacitive reactance of the antenna. Optionally, the height of the top 412 may be increased to increase capacitance (or equivalently, decrease inductance). As mentioned previously, the reflection minimum connection point depends on the geometry of the magnetic loop. Thus, changing the loop geometry by cutting back on top 412 or adding on top 412, or by changing any other aspect of the magnetic loop, would require identifying the point at which the current is at the reflective minimum after modifying the loop geometry.

The magnetic loop 402 need not be square as shown in fig. 4A. In an embodiment, the magnetic loop 402 may be rectangular or malformed and the two electric field radiators 404 and 408 may be placed at the corresponding 90/270 degree connection points or at the reflective minimum connection points. For optimum efficiency, the electrical length of the malformed circuit will be on the order of a multiple of a wavelength, or on the order of a multiple of a quarter or an eighth wavelength over the desired frequency band(s). The electric field radiator can be placed inside or outside the malformed magnetic loop. Again, it is critical to identify connection points along the magnetic loop that maximize the efficiency of the antenna. The connection point may be an 90/270 degree electrical point along the magnetic loop or a point where the current flowing through the magnetic loop is at a reflective minimum.

For example, in a smartphone, a malformed antenna design may be fitted into the available malformed space, such as the back cover of the mobile device. Instead of a square magnetic circuit, it may be rectangular in shape, circular in shape, oval in shape, generally E-shaped, generally S-shaped, or the like. Likewise, small malformed antennas may fit into non-uniform spaces on a portable computer or other portable electronic device.

As discussed above, the location of the electrical trace may be at an electrical point of about 90/270 degrees along the magnetic loop or at a reflective minimum connection point such that the electric field emitted by the electric field radiator is orthogonal to the magnetic field generated by the magnetic loop. The 90/270 connection point and the reflection minimum connection point are important because these points allow reactive power (virtual power) to be sent away from the antenna and not returned. Reactive power is typically generated around the antenna near field and stored. The reactive power oscillates centered around a fixed location near the source and affects the operation of the antenna.

With respect to fig. 4A, dashed line 414 indicates where the most significant region of the fringe capacitance phenomenon occurs. Two pieces of metal within the antenna, such as the magnetic loop and the electric field radiator, separated by a distance, can create a level of fringe capacitance. By using fringe capacitance, the single-sided antenna embodiment allows all of the elements of the antenna to be printed on one side of almost any type of suitable substrate material, including inexpensive dielectric materials. Examples of inexpensive dielectric materials that can be used as the substrate include glass reinforced epoxy laminate FR-4, which has a dielectric constant of about 4.7 ± 0.2. In the single-sided antenna 400, for example, no back or ground plane is required. Of course, wires are connected to each end of the magnetic circuit, with one of the wires being grounded. As noted previously, this full wavelength antenna design implies a composite loop antenna that is optimally and effectively shorted. In fact, a single-sided antenna will function almost optimally in the presence of a counterpoise ground plane, as is common in embedded antenna designs where the counterpoise is provided by the object to which the antenna is mounted.

The 2D design of the single-sided antenna embodiment has several advantages. By using an appropriate substrate or dielectric base, which can be very thin, the traces of the antenna can be exactly sprayed or printed on the surface and still function as a composite loop antenna. Furthermore, the 2D design allows the use of antenna materials that are typically not seen as suitable for microwave devices, such as very inexpensive substrates. Other advantages are that the antenna can be placed on a malformed surface, such as the back of a mobile phone cover, the edge of a laptop, and so forth. Embodiments of a single-sided antenna may be printed on a dielectric surface with an adhesive placed on the back of the antenna. The antenna may then be attached to various computing devices with wires connected to the antenna to provide the required power and ground. For example, as noted above, according to this design, an IEEE802.11b/g wireless antenna may be printed on a surface that is about the size of a postage stamp. The antenna may be adhered to a lid of a laptop computer, a case of a desktop computer, or a back cover of a cell phone or other portable electronic device.

Various dielectric materials may be used with the single-sided antenna embodiments. The advantage of FR-4 as a substrate over other dielectric materials such as Polytetrafluoroethylene (PTFE) is that it has a lower cost. Dielectrics typically used for higher frequency antenna designs have much lower loss characteristics than FR-4, but they can cost much more than FR-4.

The single-sided antenna embodiment can also be used for narrowband applications. Narrowband refers to channels where the bandwidth of the message does not exceed the coherence bandwidth of the channel. In wideband, the message bandwidth significantly exceeds the coherence bandwidth of the channel. Narrowband antenna applications include Wi-Fi and point-to-point remote microwave links. In accordance with the embodiments described above, for example, an array of narrowband antennas may be printed on a sticker, which may then be placed on a laptop for Wi-Fi access at distance and signal strength that is good compared to standard Wi-Fi antennas.

Fig. 4B shows an alternative embodiment of a single-sided antenna 420 with magnetic loops 422 cut away with corners at an angle of about 45 degrees. Cutting off the corners of the magnetic loop 422 at an angle may improve the efficiency of the antenna. A magnetic circuit having corners forming an angle of approximately 90 degrees affects the flow of current through the magnetic circuit. When the current flowing through the magnetic loop encounters a corner of a 90 degree angle, causing the current to bounce, the reflected current flows against the primary current or forms an eddy current pool. This is particularly true in smaller antenna embodiments, since the energy loss at the 90 degree corner negatively impacts the performance of the antenna. Cutting off the corners of the magnetic circuit at an angle of approximately 45 degrees improves current flow around the corners of the magnetic circuit. Thus, the angled corners enable electrons in the current to be less obstructed as they flow through the magnetic circuit. While cutting off the corners at a 45 degree angle is preferred, alternative embodiments are possible that may cut off at angles other than 45 degrees.

Fig. 4C illustrates an alternative embodiment of a single-sided antenna 440 that uses transitions of various widths in the magnetic loop 442 to add inductance or add capacitance to the magnetic loop 442. The corners of the magnetic loop 442 have been cut away at approximately a 45 degree angle in order to improve the flow of current as it flows around the corners of the magnetic loop 442, thereby increasing the efficiency of the antenna. A single electric field radiator 444 is physically located inside the magnetic loop 442. The electric field radiator 444 is connected to the magnetic loop 442 with an electrical trace 446 having a flexible curved shape. As previously discussed, having electrical traces 446 with a flex curve that is not sinusoidal and minimizes the number of bends in the traces may improve the efficiency of the antenna.

The term jump is used to refer to a change in the width of the magnetic circuit. In fig. 4C, the magnetic loop 442 is generally rectangular in shape and includes a first transition on the left side and a second transition on the right side. In the embodiment shown in fig. 4C, the first transition is symmetrical to the second transition. The transitions on the left and right sides of the magnetic loop 442 include a middle narrow section 448 or a middle narrow section that is thinner than the remainder of the magnetic loop 442 and that is located between and adjacent to the first and second wide sections 450, 452, the first and second wide sections 450, 452 having a greater width than the narrow section 448. Specifically, the magnetic circuit transitions from a first wide section 450 to an intermediate narrow section 448, and the intermediate narrow section 448 transitions to a second wide section 452. The wide-narrow-wide transition in the magnetic loop produces a pure inductance, thereby increasing the electrical length of the magnetic loop. Thus, using a wide-narrow-wide transition in the magnetic loop is a method of increasing the electrical length of the magnetic loop 442 by adding inductance to the magnetic loop 442. The length of the intermediate narrow section 448 can also be increased or decreased as needed to add a desired inductance to the magnetic circuit. For example, in fig. 4C, the middle narrow section 448 spans approximately one-quarter of the left and right of the magnetic loop 442. However, the intermediate narrow section 448 may be increased to about half or some other ratio across the left and right sides of the magnetic loop 442, thereby increasing the inductance of the magnetic loop 442.

The transitions are not limited to sections or segments having a width less than the remainder of the magnetic circuit 442. The selective transition may include an intermediate wide section or intermediate wide segment that is wider than the remainder of the magnetic circuit 442 and located between and adjacent to the intermediate wide section of the first and second narrow sections, the first and second narrow sections having a width that is less than the wide section. Specifically, in such an alternative embodiment, the magnetic circuit transitions from a first narrow section to an intermediate wide section, which then transitions to a second narrow section. The narrow-wide-narrow transitions in the magnetic circuit create capacitance, thereby shortening the electrical length of the magnetic circuit. The length of the middle wide section may be increased or decreased to add capacitance to the magnetic loop.

Using transitions in the magnetic circuit, that is, varying the width of the magnetic circuit over one or more sections or segments of the magnetic circuit, serves as a method for tuning impedance matching. The transition in the magnetic loop that changes width may also be tapered to further increase the inductance or capacitance to ensure that the reactive inductance and reactive capacitance of all elements in the antenna are matched. For example, in a wide-narrow-wide transition, the first wide section may taper from its larger width to the smaller width of the intermediate narrow section. Likewise, the intermediate narrow section may taper from its narrow width to the larger width of the first wide section or the second wide section, or both. The sections in the narrow-wide-narrow transition and the sections in the wide-narrow-wide transition may taper independently of each other. For example, in a first narrow-wide-narrow transition, only the middle wide section may taper, while in a second narrow-wide-narrow transition only the first narrow section may taper. The taper may be linear, stepped or curved.

The actual difference in width between the parts of the magnetic loop will depend on the amount of inductance or capacitance required to ensure that the total reactive capacitance of the antenna matches the total reactive inductance of the antenna. The embodiment shown in fig. 4C shows two wide-narrow-wide transitions located opposite each other and symmetrical. However, alternative embodiments may have transitions only on one side of the magnetic circuit 442. Furthermore, if more than one transition is used in the magnetic circuit, the transitions need not be symmetrical. For example, a malformed shape magnetic loop may have two transitions, each transition having a different length and width. Furthermore, different types of transitions may also be used on a single magnetic loop. For example, the magnetic loop may have one or more narrow-wide-narrow transitions and one or more wide-narrow-wide transitions.

Fig. 5 shows an embodiment of a small double-sided or planar antenna 500. The planar antenna 500 uses a second plane on the back side that includes a tunable patch, shown by dashed line 502, that produces a capacitive reactance for a particular frequency to match the inductive reactance of the magnetic loop 504. Tunable patch 502 is a generally square piece of metal having a flexible position relative to the other elements of antenna 500. In an embodiment, the tunable patch 502 should be located at a point away from the 90/270 degree electrical point along the magnetic loop, or a point away from the region where the current is at a reflective minimum, such as the upper left corner of the antenna 500, as shown in fig. 5. The electric field radiator 506 is located inside the magnetic loop 504 to reduce the overall size of the dual-sided antenna 500. For optimum efficiency, the electric field radiator 506 should have an electrical length equal to approximately one-quarter wavelength at its respective operating frequency. If the electric field radiator is made smaller, this will result in smaller wavelengths at higher frequencies. The electric field radiator 506 is bent into a generally J-shape to fit its entire length inside the magnetic loop 504. Alternatively, the electric field radiator 506 may be stretched out so as to be positioned on a straight line, instead of being bent in a J-shape, or bent in an alternative shape. Although such embodiments are contemplated herein, the antenna may be made wider and the overall size of the antenna increased.

The electrical trace 508 connects the electric field radiator 506 to the magnetic loop 504 at the 90/270 connection point or at a minimum reflection connection point. The top 510 of the magnetic circuit 504 is smaller compared to the other sides of the magnetic circuit 504. This is done to increase the inductance and lengthen the electrical length of the magnetic loop 504. Increasing the inductance further enables the inductive reactance to be matched to the total capacitive reactance of the antenna 500, as is the case in the small, single-sided antenna 400, and may be adjusted as discussed above.

The tunable patch 502 may also be located anywhere along the top 510 of the magnetic loop 504. However, a tunable patch 502 having a point away from where the magnetic loop 504 connects to the electric field radiator 506 yields better performance. The size of the tunable patch 502 may also be increased by changing the depth, length, and height of the patch. Increasing the depth of the tunable patch 502 will result in more space being occupied by the antenna design. Optionally, the tunable patch 502 can be made very thin, but its length and height can be adjusted accordingly. Instead of having a tunable patch 502 covering the upper left corner of the antenna 500, the length and height may be increased so as to cover the left half of the antenna 500. Optionally, the length of the tunable patch 502 may be increased, allowing the patch to extend the upper half of the antenna 500. Likewise, the height of the tunable patch 502 may be increased, allowing the patch to extend the left side of the antenna 500. Tunable patches can also be made smaller.

Similar to the single-sided antenna, various dielectric materials may be used with embodiments of the dual-sided antenna 500. Dielectric materials that can be used include FR-4, PTFE, cross-linked polystyrene, and the like.

Fig. 6 shows an embodiment of a large antenna 600, consisting of an array of four antenna elements 602, with bandwidths of almost one and one half octave. Each antenna element 602 is composed of a TE mode (transverse electric) radiator, or a magnetic field (H-field) radiator, or a magnetic loop dipole 604 (roughly indicated by a dashed line and referred to as magnetic loop 604) and a TM mode (transverse magnetic) radiator, or an electric field (E-field) radiator, or an electric field dipole 606 (indicated by a rectangular shaded area and referred to as electric field radiator 606) outside the magnetic loop 604. The magnetic loop 604 must be electrically one wavelength which creates a short circuit. Although the magnetic loop 604 may be physically smaller than one wavelength, adding additional inductance, as discussed below, will electrically lengthen the magnetic loop 604. The physical width of the magnetic loop 604 may also be adjusted in order to obtain the proper inductance/capacitance of the magnetic loop 604 so that it will resonate at the desired frequency. As noted below, the physical parameters of the magnetic loop 604 are not dependent on the quality of the dielectric material used for the antenna element 602.

As previously discussed, the magnetic loop 604 is a dead short in order to maximize the amount of current in the magnetic loop and in order to generate the highest H-field. At the same time, impedance is matched from the emitter to the load in order to prevent the emitter from being burned out due to a short circuit. The current moves from the magnetic loop 604 into the electric field radiator 606 in the direction of arrow 607 and is reflected back in the opposite direction (from the electric field radiator 606 into the magnetic loop 604 in the direction of arrow 609).

In an embodiment, each antenna element 602 is approximately 4.45 centimeters wide by approximately 2.54 centimeters high, as shown in fig. 6. However, as previously described, the dimensions of all components are determined by the operating frequency and other characteristics. For example, the traces of the magnetic loop 604 may be made very thick, which increases the gain of the antenna element 602 and allows the physical dimensions of the antenna element 602, and thus the dimensions of the antenna 600, to be modified to fit in any desired physical space, yet still be at resonance, while maintaining some of the same increased gain and maintaining a similar level of efficiency, none of which may be with prior art voltage fed antennas. The antenna can be tuned to almost any size as long as the modified design maintains (1) a magnetic loop with surface currents that inherit in a closed form, (2) energy reflection from the E-field radiator into the magnetic loop, and (3) the matched impedance of the components. Although the gain will vary based on the particular size and shape selected for the antenna, a similar level of efficiency can be achieved.

Phase tracker 608 (indicated by the triangular shaded area) makes antenna 600 wideband and can be eliminated for narrowband designs. The tip of the phase tracker 608 is ideally located at 90/270 degrees electrical along the magnetic loop 604. However, in alternative embodiments, the tip of the phase tracker may be located at the minimum reflection connection point. The dimensions 610 of the electric field radiator 606 are of no practical importance to the overall operation of the antenna element 602. Dimension 610 only has a width that makes antenna element 602 wideband, and dimension 610 may be reduced if antenna element 602 is intended to be a narrowband device. As shown, antenna element 602 is intended to be wideband in that it includes a phase tracker 608. The dimension 612 is determined by the center frequency of operation and determines the phase of the antenna element 602. Dimension 612 spans the length of the electric field radiator 606 and the length of the left side of the magnetic loop 604. Dimension 612 will typically be a quarter wavelength with slight adjustments to the dielectric material used as the substrate. The electric field radiator 606 has a length that represents approximately one-quarter wavelength at the frequency of interest. The length of the electric field radiator 606 can also be sized to be a multiple of a quarter wavelength at the frequency of interest, but these variations can reduce the effectiveness of the antenna.

The width of the top 614 of the magnetic loop 604 is intended to be smaller than any other portion of the magnetic loop 604, although this difference may not be apparent in the diagram of fig. 6. This size difference is similar to the smaller antenna embodiments previously discussed, where the top portion 614 can be trimmed to increase electrical length and increase inductance. The top 614 of the magnetic loop 604 may be clipped since it has minimal impact on the 90/270 degree electrical position. Increasing the inductance by clipping the top portion 614 makes the magnetic loop 604 electrically appear longer.

The dimensions 616, 617, and 618 of the magnetic loop 604 are all determined by the wavelength dimensions. Dimension 616 consists of the width of the magnetic loop 604. Dimension 617 consists of the length of the left portion of the bottom side of magnetic loop 604. In other words, dimension 617 consists of the length from the bottom of magnetic circuit 604 to the left of magnetic circuit opening 619. Dimension 618 consists of the full length of the magnetic loop 604. Best antenna performance is achieved when dimension 616 is equal in size to dimension 618, resulting in a square loop. However, rectangular or irregularly shaped magnetic loops 604 may also be used.

As previously noted, phase tracker 608 is included for wideband operation of antenna 600, and removal of phase tracker 608 can make antenna 600 less wideband. The antenna 600 can optionally be made narrowband by reducing the physical vertical dimensions of the phase tracker 608 and the dimensions of the electric field radiator 606. The support of the phase tracker 608 and its wideband operation in the antenna has the potential to reduce the total number of antennas used in various devices such as handsets. The size of the phase tracker 608 also affects its inductance and capacitance, as shown in fig. 7. The capacitance and inductance ranges of phase tracker 608 may be tuned by adjusting the physical dimensions of phase tracker 608. The inductance (L) of phase tracker 608 is based on the height of phase tracker 608. The capacitance (C) of phase tracker 608 is based on the width of phase tracker 608.

The antenna element 602 and the plurality of pairs of antenna elements 602 have a set of gaps formed therebetween. The two antenna elements 602 on the left side of the antenna 600 constitute a first pair of antenna elements 602, and the two antenna elements 602 on the right side of the antenna 600 constitute a second pair of antenna elements 602. There is a first gap 620 between each pair of antenna elements 602 and a second gap 622 between each of the pairs of antenna elements 602. The first gap 620 between each pair of elements 602 and the second gap 622 between each set of each pair of antenna elements 602 are designed to align the far-field radiation patterns produced by the antenna elements 602 in the most efficient manner so that the far-field radiation patterns are additive rather than subtractive. Known phased antenna array techniques can be used to determine the optimal spacing between the multi-CPL antenna elements 602 so that the far field radiation patterns of each element can be summed.

In an embodiment, the far field radiation pattern may be simulated on a computer based on the relationship of the different components of the antenna element 602. For example, the size of the antenna elements 602, the spacing between the antenna elements 602, and the spacing between each pair of antenna elements 602, as well as the relationship of the components, may be adjusted until additive orientation and alignment of the far-field radiation patterns is achieved. Optionally, the far field radiation pattern may be measured using an electrical device, thereby adjusting the relationship of the components based thereon.

Referring back now to fig. 6, the antenna element 602 is fed by a microstrip feed line represented by dashed line 624. The feed line within dashed line 624 is matched to the network to drive the impedance and depends on the dielectric material used. The symmetry of the feed line is also important to avoid unnecessary phase delays that would result in the far field radiation patterns produced by the antenna elements subtracting rather than adding.

With respect to fig. 6, the embodiment uses a common combiner/splitter 626 to split the input signal into two portions for feeding both sets of antenna elements and combining the return signals. The second and third combiner/splitters 628 then split the resulting signal into two portions to feed each pair of antenna elements 602 and combine the return signals. Combiners/ splitters 626 and 628 are desirable because they result in nearly perfect impedance matching over a wide frequency range along the feeder line and prevent power from reflecting along the feeder line, which can result in performance loss.

Fig. 8 shows a bottom layer 800 of an antenna 600 that includes elements 802, 812, 814, and 816, each of which includes a trapezoidal element 804, a choke joint area 806, and a riser 808. Elements 802, 812, 814, and 816 act as capacitors, although elements 812 and 814 also set the phase angle of antenna 600 by reflecting a signal or RF energy to the bottom of bridge unit 820. If the resulting pattern produced by the antenna 600 is expected to be spherical, the distance 826 from the bottom of the trapezoidal shaped element 804 to the bottom of the bridge unit 820 cannot be greater than a quarter wavelength. By varying the distance 826 for each element 802, 812, 814, and 816, different shaped radiation patterns can be produced. Finally, the current interruptive elements 822 and 824 represent locations where trace material has been removed from the bottom left and right corners of the bridge unit 820 to prevent reflection by elements 802 and 816, which will in turn change the phase angle set by elements 812 and 814.

The trapezoid elements 804 keep the magnetic loops 604 of each respective antenna element 602 consistent by virtue of the fact that each trapezoid element 804 is logarithmically driven in size. The slope of each ladder element 804, and in particular the slope of the top side of the ladder element 804, serves to add varying inductance and capacitance to help match the inductive reactance to the capacitive reactance in the antenna 600. The electrical length of each respective magnetic loop 604 on the other side of the antenna 600 can be adjusted by adding capacitance through the ladder element 804. The trapezoid elements 804 are aligned with the upper traces 614 of the magnetic loops 604 on the other side of the antenna 600. Choke joint 806 serves to isolate the ladder element 804 from the ground and thereby prevent leakage of the resulting signal. Sides 809 and 810 of the trapezoid element 804 are counterpoise to the electric field radiator 606 on the other side of the antenna 600, which requires the ground to set the polarization. The side 809 consists of the right side of the trapezoid elements 804 and the upper right portion of the riser 808 above the choke joint 806. That is, side 809 consists of the right side of each element 802, 812, 814, and 816 located above choke joint 806. Side 810 is comprised of the left side of the trapezoid elements 804 and the left side of the riser 808. That is, side 810 is comprised of the left side of each element 802, 812, 814, and 816 located above ground plane element 828. The ground nets 809 and 810 increase the transmission/reception efficiency of the antenna 600. The ground plane element 828 is standard for microstrip antenna designs, e.g., a 50 ohm trace on a 4.7 dielectric is about 100 mils wide.

As previously noted, the ladder element 804 can be trimmed to change the capacitance or inductance of the corresponding magnetic loop. The trimming process includes shrinking or expanding the sections of the trapezoid elements 804. For example, it may be determined that additional capacitive reactance is needed in order to match the inductive reactance of the magnetic loop. The ladder element 804 can be enlarged to increase capacitance. An optional fine tuning step is to change the slope of the trapezoid elements 804. For example, the slope may change from a 15 degree angle to a 30 degree angle. Alternatively, if the magnetic loop 604 is modified by increasing the area or by cutting the width of the upper trace 614 of the magnetic loop 604, the metal on the ground plane corresponding to the modified magnetic loop 604 must be adjusted accordingly. For example, the top side of the ladder element 804 or the overall length of the ladder element 804 may be clipped or increased based on whether the upper trace 614 of the magnetic loop 604 is clipped or increased.

As described herein, the simultaneous excitation of TM and TE radiators results in a time-dependent poynting arrangement of the predicted zero reactive power when used to analyze microwave energy. Previous attempts to construct a composite antenna with TE and TM radiators electrically orthogonal to each other relied on a three-dimensional arrangement of these elements. Such a design is not easily commercialized. Furthermore, previously proposed composite antenna designs have been fed by separate power supplies at two or more locations in each loop. In various embodiments of the antenna as disclosed herein, the magnetic loop and the electric field radiator(s) are positioned at 90/270 electrical angles to each other still lying on the same plane and are fed with power from a single location. This results in a two-dimensional arrangement which reduces the physical arrangement complexity and increases commercialization. Alternatively, the electric field radiator(s) may be positioned at a point on the magnetic loop where the current flowing through the magnetic loop is at a reflective minimum.

Embodiments of the antennas disclosed herein have greater efficiency than conventional antennas due in part to the cancellation of reactive power. Furthermore, embodiments have large antenna apertures for their respective physical dimensions. For example, a half-wave antenna with a full pattern according to an embodiment will have a significantly larger gain than the typical 2.11dBi gain of a simple field dipole antenna.

Yet another embodiment consists of a single-sided antenna with an embedded counterpoise for the electric field radiator. Figure 9A shows an embodiment of a single-sided 2300 to 2700MHz antenna with a single electric field radiator and an embedded counterpoise for the electric field radiator. The antenna 900 is comprised of a magnetic loop 902 with an electric field radiator 904 coupled directly to the magnetic loop 902 without utilizing electrical traces. The electric field radiator 904 is physically located inside the magnetic loop 902. As with other embodiments, the electric field radiator 904 may be coupled to the magnetic loop 902 at the 90/270 connection point or at a point where the current flowing through the magnetic loop 902 is at a reflective minimum. In an alternative embodiment, the electric field radiator 904 may be coupled to the magnetic loop 902 with an electrical trace. Further, although antenna 900 is shown with one electric field radiator, alternative embodiments may include one or more electric field radiators. Alternative embodiments may also include one or more electric field radiators physically located outside of the magnetic loop 902.

Alternative embodiments of the self-contained antenna may also include a first electric field radiator having a first length and a second electric field radiator having a second length different from the first length. Similar to the antenna embodiments described in the previous text, a broadband antenna can be realized using one or more electric field radiators having different lengths.

The antenna 900 includes transitions 906 and counterpoise 908 to the electric field radiator 904. Transition 906 is comprised of a portion of magnetic loop 902 having a width greater than the width of magnetic loop 902. The transitions 906 electrically isolate the embedded counterpoise 908. The embedded counterpoise 908 allows the antenna 900 to be completely independent of any ground plane or chassis of the product in which the antenna 900 is used.

The counterpoise 908 is referred to as embedded because the counterpoise is formed by the magnetic circuit 902. As noted, the embedded counterpoise 908 allows the antenna 900 to be completely independent of the product's ground plane. The embodiment of a single-sided antenna shown in fig. 4A-4C, although printed on only a single plane and does not include a ground plane, requires that the ground plane be provided by the device using the antenna. In contrast, a self-contained counterpoise antenna does not require a ground plane to be provided by the device using the antenna.

In the single-sided embodiments described above, the device using the antenna provides a ground plane for the antenna, the ground plane of the device acting as a ground plane for the single-sided antenna, or the chassis or some other metallic component of the device acting as a ground plane for the single-sided antenna. However, any modification to the device circuitry, the device chassis, or the device ground plane can negatively impact the performance of the antenna. This phenomenon is not specific to the single-sided embodiment disclosed herein, but is applied to antennas that are widely used in research and commerce. It is therefore desirable to have an antenna that does not require a ground plane and will not be affected by any changes made to the device using the antenna.

By not requiring a ground plane, antenna 900 does not rely on a ground plane external to the antenna. The independence of the self-contained antenna 900 from the external ground plane means that the performance of the antenna is not affected by changes made to the device. In terms of manufacturing and design, this means that a self-contained antenna can be designed for a given frequency and independent of the performance level of the device intended to incorporate and use the antenna. For example, a wireless router manufacturer may request a specified antenna based on a set of requirements. These requirements may include, among others, the space available for the antenna, the frequency range for the antenna, the substrate used. The design and manufacture of the antenna can be done independently of the design and manufacture of the actual wireless router. Furthermore, any future changes to the wireless router will not affect the performance and efficiency of the antenna because the antenna is self-contained and is not affected by changes to the circuitry of the router, the ground plane of the router, or the chassis of the router.

The length of the transition 906 may be set based on the operating frequency of the antenna. For higher frequency antennas, where the wavelength is shorter, shorter transitions may be used. On the other hand, for lower frequency antennas, where the wavelength is longer, longer transitions 906 may be used. The transitions 906 may be adjusted independently of the counterpoise 908. For example, the transitions for a 5.8GHz antenna may be only half the size of the transitions 906 in FIG. 9A, while the counterpoise 908 may still be as long as the entire left side of the magnetic loop 902.

The counterpoise 908 length may be adjusted as necessary to achieve desired antenna performance. However, it is preferred to have the counterpoise 908 as large as possible. For example, in an alternative embodiment, the counterpoise 908 may span the total length of the left side of the magnetic loop 902, rather than only about 80% of the left side of the magnetic loop 902. However, as previously described, the width of the traces of the magnetic loop 902 affects the electrical length of the magnetic loop 902. A magnetic loop having a thin trace all the way around the magnetic loop is electrically longer than a magnetic loop with a wider trace or having a portion of a magnetic loop with a wider trace. For example, the magnetic loop 902 is an example of a magnetic loop with wider traces for the transition 906 and for the counterpoise 908. Thus, while it is desirable to have as long a counterpoise as possible, the length of the counterpoise 908 affects the electrical length of the magnetic circuit 902. Electrically longer magnetic loops result in lower frequencies. On the other hand, a magnetic loop that is electrically short results in a higher frequency. For example, using a counterpoise that spans the entire length of the left side of the magnetic loop will increase the overall width of the magnetic loop, electrically shortening the magnetic loop and resulting in a magnetic loop with a higher than desired frequency. For example, resulting in a frequency of 5.8GHz instead of the desired target frequency of 5.6 GHz.

In addition to transitions and counterpoise, embodiments of the antenna 900 may also include narrow-wide-narrow transitions and/or wide-narrow-wide transitions, as previously described herein, in order to tune the electrical length of the magnetic loop to a desired frequency. In addition, embodiments of the self-contained antenna may also include magnetic loops that are cut off at an angle as previously described in order to improve the flow of current around the corners of the magnetic loops.

As previously described, the counterpoise 908 to the electric field radiator 904 is used instead of the ground plane. The electric field radiator 904 is effectively a monopole antenna. A monopole antenna is formed by replacing one half of the dipole antenna with the ground plane at right angles to the remaining half. In the self-contained antenna embodiment, the electric field radiator looks for a large piece of metal that it can use to electrically connect to the electric field radiator instead of the ground plane. In the single-sided antenna 400 of fig. 4C, the electric field radiator 444 radiates an electric field based on the position of the ground plane for the antenna 440. The electric field rotates perpendicular to the plane of the electric field radiator, while the magnetic field rotates in a manner that is substantially coplanar with the plane. The pattern of this electric field is generally circular, which is also referred to as a nearly perfect full pattern. As previously discussed, the single-sided antenna embodiment does not necessarily provide its own ground plane. Thus, if the antenna 440 is being used in a device, the device will act as a ground plane for the antenna 440 and the radiation pattern emitted by the electric field radiator 444 can be reflected back into the device. However, if the single-sided, self-contained antenna including the counterpoise also includes a ground plane, the radiation pattern described above will effectively switch, with the electric field rotating about or on one or more planes coplanar with the plane of the electric field radiator, and the magnetic field rotating perpendicular to that plane.

The counterpoise 908 need not be positioned or machined on the upper left corner of the magnetic circuit 902. In an alternative embodiment, the counterpoise may be positioned at the upper right corner, with the electric field radiator 904 in turn positioned at the left side of the magnetic loop 902. Regardless of the physical location of the counterpoise 908 and the electric field radiator 904 (or several radiators if there is more than one), the counterpoise and the electric field radiator(s) need to be 180 degrees out of phase. In a further embodiment, the length of the counterpoise may also be adjusted if necessary. The counterpoise 908 may also be located along the right side of the magnetic loop 902, directly below the electric field radiator 904, or at other locations around the magnetic loop 902.

The antenna 900 also includes a balun 910. A balun is an electrical transformer that can convert an electrical signal balanced (differential) with respect to ground into an unbalanced (single-ended) signal and vice versa. In particular, baluns cause high impedance to common mode signals and low impedance to differential mode signals. The balun 910 acts to cancel the common mode current. In addition, the balun 910 tunes the antenna 900 to a desired input impedance and tunes the impedance of the entire magnetic loop 902. The balun 910 is generally triangular and consists of two parts separated by a mid-gap 912.

The two portions of the balun 910 are magnetically and electrically coupled. The gap 912 in the balun 910 eliminates common mode current by magnetically preventing current flow in one direction, such as back through the transmitter and to the device using the antenna 900. This is important because reflections of current through the transmitter due to common mode currents can negatively impact the performance of the antenna 900 and the performance of the device in which the antenna 900 is used. In particular, the reflection of the current through the transmitter may cause interference in the circuitry of the device in which the antenna is used. This negative performance can also cause devices to fail Federal Communications Commission (FCC) regulations. The gap 912 in the balun 910 cancels the common mode current, preventing the current from being reflected back into the connector of the antenna 900.

The gap 912 may be adjusted based on antenna design and size. In an embodiment, electromagnetic simulation may be used to visualize the current flowing through the antenna 900. The gap 912 may then be increased or decreased until the simulation shows that the current is no longer reflected and flows back through the emitter. The cancellation of the common mode current can be visualized as the point where the current stops flowing in one direction into the transmitter and begins flowing in the opposite direction, one direction into the antenna 900 and the second direction out of the antenna 900.

The purpose of the tapered sides 914 of the balun 910 is to be electrically coupled. The angle of the tapered side 914 may be adjusted to impedance match the antenna 900. Typically, various inductors and various capacitors are placed along the feed line (not shown) to the antenna 900 to match the impedance of the device feeding the antenna 900 to the impedance of the antenna 900. For example, if the antenna expects a 50 ohm input, but the circuitry of the device is feeding the antenna 150 ohms, then a series of inductors and capacitors are used to balance the mismatch problem by converting the 150 ohms fed to the antenna to the 50 ohms expected for the antenna. In contrast to common practice in these industries, embodiments of the self-contained antenna 900 do not require impedance matching via any external components, such as using a series of inductors and capacitors along the line feeding the antenna 900. Instead, the balun 910 is used to match the impedance of the antenna 900 to the connector feeding the antenna 900 and to the impedance of the magnetic loop 902.

The height of the balun 910 is a function of the operating frequency of the antenna 900. Thus, a higher balun 910 is required for lower frequencies, and a shorter balun 910 is required for higher frequencies. The proximity of the balun 910 to the electric field radiator is important when using a high balun in the antenna. Positioning the balun 910 too close to the electric field radiator 904 results in capacitive coupling between the balun 910 and the electric field radiator 904. Therefore, it is important for the balun 910 to be properly spaced from the electric field radiator 904 to prevent capacitive coupling from affecting the antenna 900 performance. If a particular antenna design requires the use of a high balun due to the operating frequency of the antenna to properly impedance match the antenna and cancel the common mode current, the balun may be moved downward, as shown in antenna 920 in fig. 9B. In alternative embodiments, the self-contained counterpoise antenna 900 may not include the balun 910.

Antenna 900 is an example of a self-contained counterpoise composite field antenna. Embodiments of the antenna 900 may be printed or otherwise deposited on an approximately 1.6 millimeter FR-4 substrate. The performance and design of the antenna 900 also makes it adaptable to other materials, including flexible printed circuits, Acrylonitrile Butadiene Styrene (ABS) plastic, and even materials not considered suitable for microwave frequencies. The operating frequency of the antenna 900 is approximately 2300 to 2700MHz, making it suitable for various embedded applications including mobile phones, access points, PDAs, laptops, PC cards, sensors and automotive applications. Embodiments of antenna 900 achieve a peak efficiency of approximately 94% and a peak gain of approximately +3 dBi. The antenna 900 has a width of about 31 millimeters and a length of about 31 millimeters. Antenna 900 has a linear polarization and an impedance of approximately 50 ohms. The antenna 900 also has a voltage standing wave ratio of less than 2:1 (< 2: 1). The size and efficiency of the antenna 900 makes it suitable for Wi-Fi applications where efficiency, size and gain are important.

An alternative embodiment of a single-sided antenna with an embedded counterpoise is shown in fig. 10A. Antenna 1000 is an example of an antenna with linear polarization. Due to the embedded counterpoise, the antenna does not require a ground plane. The antenna 1000 may be printed or otherwise deposited on a 1.6 mm thick FR-4 substrate. Similar to antenna 900, the performance and design of antenna 1000 makes it suitable for other materials, including flexible printed circuits, Acrylonitrile Butadiene Styrene (ABS) plastic, or even materials not considered suitable for microwave frequencies. The antenna 1000 operates over a frequency range of about 882MHz to 948MHz, with a measured peak gain of about +3dBi and a peak efficiency of about 92%. The antenna 1000 has an antenna impedance of about 50 ohms and a voltage standing wave ratio of less than 2:1 (< 2: 1). Antenna 1000 has a width of about 76 millimeters and a height of about 76 millimeters.

The antenna 1000 is comprised of a magnetic loop 1002 with a first electric field radiator 1004 coupled directly to the magnetic loop 1002 and a second electric field radiator 1006 coupled directly to the magnetic loop 1002. Both electric field radiators 1004 and 1006 are coupled to the magnetic loop 1002 without the use of electrical traces. The electric field radiators 1004 and 1006 are physically located inside the magnetic loop 1002. The use of two electric field radiators, instead of one electric field radiator as in the antenna 900 in fig. 9, increases the gain of the antenna. The curve 1008 separating the two electric field radiators 1004 and 1006 acts to delay the phase between the two electric field radiators 1004 and 1006 so that the far field patterns of the two may add.

The two electric field radiators 1004 and 1006, together with the curve 1008, form an electric field radiator array 1010 with phase delay. Specifically, the curve 1008 ensures that the two electric field radiators 1004 and 1006 are 180 degrees out of phase with each other. Curves can be used as a space saving technique. For example, if a small antenna is desired, then the curve 1008 can be used to ensure that the electric field radiators are still 180 degrees out of phase with each other, as the need to minimize size forces the two electric field radiators closer together. The electrical length of the trace of curve 1008 can be adjusted, when necessary, based on the desired delay. For example, the traces can be made longer or shorter while keeping the width constant. Alternatively, the length of the traces may be kept constant while making the width of the traces wider or thicker. As described above, the electrical length of a trace depends on its physical length and its physical width. FIG. 10B illustrates an alternative embodiment of a self-contained antenna 1020 without the curve 1008.

As discussed with respect to the antenna 900, the antenna transitions 1012 and counterpoise 1014 may be adjusted accordingly based on a number of factors. The transition 1012 depends on the operating frequency, but it must also be long enough to ensure that the counterpoise 1014 is electrically isolated. It is preferable that the earth mat 1014 be as large as possible. Finally, the balun 1016 cancels the common mode current and matches the impedance of the antenna 1000 to the impedance of the transmitter feeding the antenna 1000.

In alternative embodiments of the antenna 1000, the electric field radiator array 1010 may be disposed on the left side of the antenna 1000 instead of the right side. In such an alternative embodiment, the counterpoise 1014 would be positioned on the upper right side of the magnetic circuit 1002. The counterpoise 1014 can also be located along the right side of the magnetic loop 1002, directly below the electric field radiators 1004 and 1006.

Fig. 11A-11C illustrate 2D radiation patterns of the antenna 900 of fig. 9. Fig. 11A shows a 2D radiation pattern in the XZ plane 1100. The solid line 1102 represents the actual radiation pattern, the dashed line 1104 represents the 3dB beamwidth, and the dotted line 1106 represents the maximum strength of the field along one direction, i.e. the line 1106 represents where the strongest field is detected in the illustrated 2D radiation pattern. Fig. 11B shows the 2D radiation pattern of the antenna 900 on the XY plane 1110, and fig. 11C shows the 2D radiation pattern of the antenna 900 on the YZ plane 1120.

Fig. 12A-12C show the 2D radiation pattern of the antenna 1000 in fig. 10A. Fig. 12A shows a 2D radiation pattern in the XZ plane 1200. The solid line 1202 represents the actual radiation pattern, the dashed line 1204 represents the 3dB beamwidth, and the dotted line 1206 represents the maximum strength of the field along one direction, i.e. the line 1206 represents where the strongest field is detected in the illustrated 2D radiation pattern. Fig. 12B shows the 2D radiation pattern of the antenna 1000 on the XY plane 1210, and fig. 12C shows the 2D radiation pattern of the antenna 1000 on the YZ plane 1220.

Fig. 13A shows the Voltage Standing Wave Ratio (VSWR) of the antenna 900. The VSWR plot shows that the antenna 900 is a good impedance match for the frequency range of about 2.34GHz to about 2.69 GHz. That is, throughout the frequency range of about 2.34GHz to 2.69GHz, most of the energy fed into the antenna 900 will be radiated out, rather than reflected back into the transmitter. Specifically, the interior of the two central vertical solid lines represents a frequency range where the VSWR of the antenna 900 is less than 2:1 (< 2: 1). Fig. 13B shows the return loss of the antenna 900. The return loss and VSWR are mathematically related such that the-10.0 return loss on fig. 13B corresponds to 2.0 in VSWR on fig. 13A. The return loss plot in fig. 13B shows that the antenna 900 is a good impedance match between the points labeled 1 and 2.

Fig. 14A illustrates a Voltage Standing Wave Ratio (VSWR) of the antenna 1000 in fig. 10A. The VSWR plot shows that the antenna 1000 is a good impedance match for the frequency range of about 884MHz to about 947 MHz. That is, over a frequency range of approximately 884MHz to 947MHz, most of the energy fed into the antenna 1000 will be radiated out, rather than being reflected back into the transmitter. Specifically, the frequency range within the two central vertical solid lines representing the VSWR of the antenna 1000 less than 2:1 (< 2: 1). Fig. 14B shows the return loss of the antenna 1000. As previously mentioned, the-10.0 return loss corresponds to 2.0 in VSWR. The return loss plot in fig. 14B shows that the antenna 1000 is a good impedance match between the points labeled 1 and 2.

FIG. 15 illustrates another embodiment of a self-contained antenna 1500. Antenna 1500 is an example of a 5.8GHz antenna. A particular embodiment of antenna 1500 has dimensions of a length of about 15 millimeters and a width of about 15 millimeters. The antenna 1500 is comprised of a magnetic loop 1502 and an electric field radiator 1504 coupled directly to the magnetic loop 1502. In contrast to the self-contained antennas 900 and 1000, the antenna 1500 includes a cone transition 1506 made up of two segments 1508 and 1510. The first transition section 1508 begins with the width of the magnetic loop changing from a small width to a large width. The first transition section 1508 tapers linearly toward a smaller width before the location where the width of the magnetic circuit where the second transition section 1510 begins increases again. The second transition section increases linearly from the small width to the larger width. As previously discussed, adjusting the width of the trace of the magnetic loop allows the electrical length of the magnetic loop to be adjusted. In addition, the length, width, and number of transitions used electrically isolate the counterpoise 1512. The transition 1506 must be long enough to minimize the magnitude of the current flowing through the counterpoise 1512. Furthermore, tapering the transition 1506 to the counterpoise 1512 may increase bandwidth in terms of impedance matching. The balun 1514 cancels the common mode current and matches the impedance of the antenna 1500. Alternative embodiments of the antenna 1500 may not include the balun 1514.

An embodiment consists of a single-sided antenna comprising: a magnetic loop having a width, the magnetic loop being located on a plane where the magnetic field is generated and having a first inductive reactance; an electric field radiator located on a plane of the emitted electric field and having a first capacitive reactance, the electric field radiator being directly coupled to the magnetic loop, wherein the electric field is orthogonal to the magnetic field, and wherein a physical arrangement between the electric field radiator and the magnetic loop results in a second capacitive reactance; a transition formed on the magnetic circuit and having a width greater than a width of the magnetic circuit; and a counterpoise formed on the magnetic loop and located opposite or adjacent to the electric field radiator along the magnetic loop, wherein the transitions substantially electrically isolate the counterpoise from the magnetic loop.

Each feature disclosed in this specification (including any accompanying claims, abstract and drawings) may be replaced by alternative features serving the same, equivalent or similar purpose, unless expressly stated otherwise. Thus, unless expressly stated otherwise, each feature disclosed is one example only of a generic series of equivalent or similar features.

While the invention has been shown and described herein with respect to several alternatives, it will be appreciated that the techniques described herein may have many additional uses and applications. Accordingly, the present invention should not be limited to the particular descriptions, embodiments and various figures contained in this specification, which illustrate only examples of embodiments, alternatives and applications of the principles of the invention.

Claims (26)

1. A method for tuning a complex loop (CPL) antenna, the method comprising:

connecting at least one electric field radiator to a magnetic loop at a connection point along the magnetic loop to form a CPL antenna, wherein the magnetic loop and the at least one electric field radiator lie on a plane, and wherein the connection point comprises one of an electrical angular position of about 90 degrees and an electrical angular position of about 270 degrees relative to a drive point of the magnetic loop;

matching the magnetic loop and the at least one electric field radiator to a source impedance connected to a drive point of the magnetic loop; and

after impedance matching, the connection point is adjusted to a second connection point at which the current flowing through the magnetic loop is at a reflective minimum.

2. The method of claim 1, wherein the magnetic circuit is configured to generate a magnetic field, and wherein the at least one electric field radiator is configured to emit an electric field orthogonal to the magnetic field.

3. The method of claim 1, wherein impedance matching the magnetic loop and the at least one electric field radiator to the source further comprises adding or removing metal to the magnetic loop.

4. The method of claim 1, wherein the second connection point at which the current flowing through the magnetic circuit is at a reflection minimum depends on a geometry of the magnetic circuit.

5. The method of claim 1, wherein the second connection point substantially corresponds to the other of an electrical angular position of about 90 degrees and an electrical angular position of about 270 degrees along the magnetic circuit.

6. The method of claim 1, wherein connecting the electric field radiator to the magnetic loop at the connection point comprises placing an electrical trace between the electric field radiator and the magnetic loop.

7. The method of claim 6, wherein the electrical trace has a shape that is substantially smooth curve or a shape that minimizes the number of bends in the electrical trace.

8. The method of any one of claims 1 to 5, wherein the at least one electric field radiator is directly coupled to the magnetic loop.

9. The method of any one of claims 1 to 7, wherein the magnetic circuit has an electrical length approximately equal to a multiple of a wavelength, approximately equal to a multiple of a quarter wavelength, or approximately equal to a multiple of an eighth wavelength.

10. The method of any one of claims 1 to 7, wherein the at least one electric field radiator has an electrical length approximately equal to a multiple of a wavelength, approximately equal to a multiple of a quarter wavelength, or approximately equal to a multiple of an eighth wavelength.

11. The method of any one of claims 1 to 7, wherein current flowing through the magnetic loop flows into the at least one electric field radiator and reflects the current into the magnetic loop in an opposite direction, causing the electric field to reflect into a magnetic field and generate an electric field orthogonal to the magnetic field.

12. The method of any one of claims 1 to 7, wherein the magnetic circuit has a substantially rectangular shape.

13. The method of claim 12, wherein four corners of the generally rectangular shape of the magnetic circuit are cut at an angle.

14. The method of any one of claims 1 to 7, wherein the magnetic loop is formed by a plurality of consecutively connected segments, wherein at least one segment of the plurality of consecutively connected segments is formed by a first segment having a first width, an intermediate segment having an intermediate width, and a second segment having a second width, wherein a first end of the first segment is connected to and adjacent to a first end of the intermediate segment and wherein a second end of the intermediate segment is connected to and adjacent to a first end of the second segment, and wherein the first width and the second width are different from the intermediate width.

15. The method of claim 2, wherein the at least one electric field radiator has an electrical length and is configured to emit the electric field at an operating frequency, and wherein the at least one electric field radiator comprises a second electric field radiator located on the plane and within the magnetic loop, the second electric field radiator coupled to the magnetic loop, the second electric field radiator configured to emit a second electric field orthogonal to the magnetic field, the second electric field radiator having a second electrical length and configured to emit the second electric field at a second operating frequency.

16. The method of any of claims 1 to 7, further comprising:

forming a transition on the magnetic loop, wherein the transition has a width greater than a width of the magnetic loop; and

forming a counterpoise on the magnetic loop positioned along the magnetic loop opposite or adjacent to the at least one electric field radiator, wherein the transitions are configured to substantially electrically isolate the counterpoise from the magnetic loop.

17. The method of claim 16, wherein the counterpoise has a counterpoise width that is greater than a width of the magnetic circuit.

18. The method of any of claims 1 to 7, further comprising: adding a balun to the magnetic loop, the balun configured to cancel a common mode current and tune the CPL antenna to a desired input impedance.

19. A method for tuning a planar composite loop (CPL) antenna, the method comprising:

connecting at least one electric field radiator to at least one magnetic loop at a connection point along each of the at least one magnetic loop to form a planar CPL antenna, wherein each of the at least one magnetic loop and the at least one electric field radiator lie on a first plane, wherein the planar CPL antenna comprises a broadband element lying on a second plane and configured to create a ground plane, and wherein the connection point comprises an electrical angular position of about 90 degrees or an electrical angular position of about 270 degrees relative to a driving point of each of the at least one magnetic loop;

matching the at least one magnetic loop and the at least one electric field radiator to a source impedance connected to a drive point of the at least one magnetic loop; and

after impedance matching, the connection point is adjusted to a second connection point at which current flowing through the at least one magnetic loop is at a reflective minimum.

20. The method of claim 19, further comprising: one or more phase trackers are coupled to each of the at least one magnetic loop.

21. The method of claim 20, wherein each phase tracker is physically located inside or physically located outside each magnetic loop.

22. The method of claim 20 or 21, wherein each phase tracker is substantially triangular in shape, and wherein the tip of each substantially triangular shaped phase tracker is aligned with: an electrical angle position of about 90 degrees relative to the drive point, an electrical angle position of about 270 degrees relative to the drive point, or a reflective minimum point at which current flowing through each magnetic loop is at a reflective minimum.

23. The method of any one of claims 19 to 21, wherein the broadband element comprises one or more trapezoidal elements.

24. The method of claim 23, further comprising: the slope of the top edge of each trapezoid element is varied.

25. The method of claim 23, wherein the broadband element comprises one or more choke joints and a ground element, wherein the one or more choke joints are configured to isolate the one or more trapezoid elements from the ground element.

26. The method of any one of claims 19 to 21, wherein the one or more magnetic circuits have a substantially rectangular shape.

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