KR102057880B1 - Compound loop antenna - Google Patents

Abstract

Embodiments of the present invention relate to planar (duplex) and printed (sectional) composite field antennas. The present embodiments relate to improvements and, in particular, are not exclusive, but achieve co-field radiators and performance gains at high bandwidth (low Q), large radiation intensity / power / gain, and high efficiency. A composite loop antenna having magnetic loops having an electric field orthogonal to magnetic fields. Embodiments also relate to a self-contained counterpoise compound field antenna formed on the magnetic loop and comprising a transition having a transition width that is greater than the width of the magnetic loop. The transition portion generally insulates the counterpoise formed on the magnetic loop opposite or adjacent to the electric field radiator.

Description

Compound Loop Antenna {COMPOUND LOOP ANTENNA}

Before and after notes on related applications

This application takes priority from U.S. Provisional Application No. 61 / 303,594, filed Feb. 11, 2010, and U.S. App. Claims are made from No. 12 / 878,020.

Embodiments of the present invention relate to planar (duplex) and printed (sectional) composite field antennas, in particular, but not exclusively, field radiators on the same plane and performance gains at high bandwidth (low Q), A composite loop antenna having magnetic loops having a large radiation intensity / power / gain and an electric field orthogonal to magnetic fields achieving high efficiency. Embodiments also relate to a self-contained counterpoise compound field antenna formed on the magnetic loop and including a transition having a transition width greater than the width of the magnetic loop. The transition portion generally insulates the counterpoise formed on the magnetic loop opposite or adjacent to the electric field radiator.

The ever-decreasing size of modern telecommunication devices raises the need for improved antenna designs. Known antennas in devices such as mobile / cellular telephones represent one of the serious limitations in terms of performance and are included in any way with almost no exception.

In particular, the efficiency of the antenna can seriously affect the performance of the device. The more efficient the antenna, the higher the rate of energy supplied from the transmitter. Similarly, due to the inherent compatibility of the antenna, the more efficient the antenna, the more received signals are converted into electrical energy for processing by the receiver.

In order to ensure maximum energy transfer (both in the receive and transmit modes) between the transceiver (the device acting as both the transmitter and the receiver) and the antenna, the impedances of both must be matched on a scale with each other. If any mismatching occurs between them, in the case of transmission, the performance of the lane is reflected where energy is reflected from the antenna into the transmitter. When operating as a receiver, the lane's performance of the antenna will result in a lower reception than otherwise possible.

Known simple loop antennas are generally current supply devices, which primarily generate a magnetic (H) field. As such they are generally not suitable as transmitters. This is especially true for small loop antennas (i.e. smaller ones with a diameter smaller than one wavelength). In contrast, voltage supply antennas, such as dipoles, generate both an electric (E) field and a magnetic (H) field and can be used in both receive and transmit modes.

The amount of energy received by or transmitted from the loop antenna is partly determined by its area. In general, when the area of the loop is halved each time, the amount of energy that can be received / transmitted is reduced by about 3 dB based on application parameters, such as initial magnitude, frequency, and the like. This physical constraint is assumed to mean the fact that very small loop antennas cannot actually be used.

Composite antennas have high performance gains such as high bandwidth (low Q), large radiation intensity / power / gain, and high efficiency in both transverse-electromagnetic (TM) and transverse electric (TE) modes Are the antennas that are excited to achieve this.

In the late 1940s, Wheeler and Chu tested for the first time the characteristics of electrical near field (ELS) antennas. Through his work, some numerical formulas have been formed that account for the limitations of antennas as their physical size is reduced. One particularly important one of the limitations of the ELS antennas mentioned by Wheeler and Weight is that they have large radiation quality factors (Q) and store more energy in the time average than they emit. According to Wheeler and Chu, ELS antennas have a high radiation quality factor (Q), which results in minimal resistance loss to the antenna or matching network, and generally results in very low radiation efficiency of between 1-50%. . As a result, since the 1940's, the fact that ELS antennas have a narrow bandwidth and poor radiation efficiency has been accepted by Science World. Most of the achievements in wireless communication systems using modern ELS antennas have resulted from thorough experimentation and optimization of modulation schemes and on air protocols. It still reflects the wheeler and further established narrow bandwidth, low efficiency characteristics.

In the early 1990s, Grimsdale M. And Dale M. Grimes and Craig A. Grimes mathematically describe the specific combination of TM and TE modes operating together in ELS antennas that surpass the low radiation Q limits formed by Wheeler and Weight's theory. Claimed to be found. They described their work in a publication entitled, "Q and Bandwidth of Antennas Emitting TE and TM Modes," published in the IEEE Journal of Electromagnetic Compatibility in May 1995. This argument sparked a lot of debate and led to the concept of a "complex field antenna" in which both TE and TM modes are excited, very different from the "simple field antenna" in which TM or TE modes are solely excited. The gain of the composite field antenna is lower radiation Q, increased radiation intensity, directivity (gain) than the wheeler and further limiting. U.S. Pat. Proven mathematically by renowned RF experts, including groups employed by the Naval Air Warfare Center Weapons Division [P. L. Overfelft, D. R. Bowling, D. J. White, “Magnetic Loop, Electronic Dipole Array Antenna (Preliminary Results), Co-located,” Interim rept. September 1994].

Composite field antennas have proved to be complex and difficult to implement physically due to undesirable device coupling effects and difficulties associated with designing low loss passive circuits for combining electrical and magnetic radiators.

There are many examples of two-dimensional non-composite antennas consisting of generally printed metal strips on a circuit board. However, such antennas are supplied with voltage. One example of such an antenna is a planar inverted-F antenna (PIFA). Most similar antenna designs also consist primarily of 1/4 wavelength (or a multiple of 1/4 wavelength), supply voltage, and dipole antennas.

Planar antennas are also known in the art. For example, US Pat. No. 5,061,938 to Zahn et al. Requires an expensive Teflon substrate or similar material to operate the antenna. US Pat. No. 5,376,942 to Shiga discloses a planar antenna capable of receiving but not transmitting microwave signals. The cigar antennas also require expensive semiconductor substrates. U.S. Patent No. 6,677,901 to Nalbandian relates to a planar antenna requiring a substrate having a permittivity to permeability ratio of 1: 1 to 1: 3, which is limited to HF and VHF frequency ranges ( 3 to 30 MHz and 30 to 300 MHz). Techniques for printing rather low frequency devices on expensive glass reinforced epoxy laminated sheets such as FR-4, which are generally used for conventional printed circuit boards, are known, but dielectric losses in FR-4 are too high. It is also contemplated to have a dielectric constant that cannot sufficiently control such a substrate to be used at microwave frequencies. For these reasons, more alumina substrates are generally used. Also, none of these planar antennas is a composite loop antenna.

In terms of bandwidth, efficiency, gain and radiation intensity, the basis for the increased performance of composite field antennas is formed from the energy effects stored in the near field of the antenna. In RF antenna designs, it is desirable to transfer as much energy present in the antenna as possible to radiated power. The energy stored in the antenna near field has historically been referred to as reaction force and also acts to limit the amount of power that can be radiated. When discussing complex power, there are real and virtual (sometimes referred to as "reactive") parts. The actual power never leaves the source and never returns, while the virtual or reactive power oscillates with respect to the fixture (within half wavelength) of the source and also tends to interact with the source, thus affecting the operation of the antenna. . The presence of real power is added directly from multiple sources, but multiple sources of virtual power can be appended or subtracted (erased). The gain of the composite antenna is driven by both TM and TE sources, allowing engineers to obtain designs that use reactive power cancellation that was previously unobtainable in simple magnetic antennas, thus improving the antenna's actual power transmission characteristics. Is in point.

In order to be able to cancel the reactive power in the composite antenna, the electric and magnetic fields must operate at right angles to each other. Many arrangements of the electric field radiator (s) needed to emit the electric field, and the magnetic loops necessary to generate the magnetic field have been proposed, but all such designs have remained unchanged based on a three-dimensional antenna. For example, US Pat. No. 7,215,292 to McLean requires a pair of magnetic loops in parallel planes with electrical dipoles on a third parallel plane located between the pair of magnetic loops. . U. S. Patent No. 6,437, 750 to Grimes et al. Requires two pairs of magnetic loops and electrical dipoles to be physically arranged at right angles to one another. US application US2007 / 0080878, filed by McLean, discloses an arrangement in which the magnetic dipole and the electric dipole are located in orthogonal planes.

1 shows a planar realization of an embodiment.
2 shows a circuit layout of an embodiment incorporating four separate antenna elements.
FIG. 3A is a detailed view of one of the antenna elements of FIG. 2 including a phase tracker. FIG.
FIG. 3B is a detailed view of one of the antenna elements of FIG. 2 without a phase tracker. FIG.
4A illustrates an embodiment of a small cross-sectional composite antenna.
4B illustrates an embodiment of a small cross-sectional composite antenna having a magnetic loop in which corners are cut at about 45 degrees.
4C illustrates an embodiment of a small cross-sectional composite antenna with a magnetic loop having two symmetrical wide-narrow-wide transitions.
5 is a view for explaining an embodiment of a small double-sided composite antenna;
FIG. 6 illustrates an embodiment of a large composite antenna array consisting of four composite antenna elements. FIG.
FIG. 7 shows how the dimensions of the phase tracker affect inductance and capacitance.
8 illustrates a ground plane of the antenna embodiment of FIG.
FIG. 9A illustrates an embodiment of a self-supporting counterpoise antenna with a balun. FIG.
9b is a view for explaining another embodiment of the antenna according to FIG. 9a with the balun removed;
FIG. 10A illustrates an embodiment of a self-supporting counterpoise antenna having an array of electric field radiators and a curved trace between electric field radiators. FIG.
10B illustrates an embodiment of a self-contained counterpoise antenna with an array of electric field radiators but without the curved traces.
11A-11C are schematic diagrams illustrating 2D radiation patterns for the antenna according to FIG. 9.
12A-12C are schematic diagrams illustrating 2D radiation patterns for an antenna according to FIG. 10A.
FIG. 13A is a schematic diagram illustrating a diagram of voltage standing wave ratio for an antenna according to FIG. 9. FIG.
FIG. 13B is a schematic diagram illustrating a plot of measured return loss for the antenna according to FIG. 9. FIG.
14A is a schematic diagram illustrating a diagram of a voltage standing wave ratio for an antenna according to FIG. 10.
14b is a schematic diagram illustrating a plot of measured return loss for an antenna according to FIG. 10.
Fig. 15 is a schematic diagram illustrating an embodiment of a self-supporting counterpoise antenna having a tapered transition portion.

Embodiments provide an improved planar composite loop (CPL) antenna that can operate in both transmit and receive modes and can also give much better performance than known loop antennas. The two main elements of a CPL antenna are a magnetic loop that generates a magnetic field (H field) and an electric field radiator that emits an electric field (E field).

The electric field radiator may be physically located inside the loop or outside the loop. For example, FIG. 1 shows an embodiment of a single CPL antenna element with an electric field radiator located on the inside of the loop coupled by electrical traces, while FIGS. 3A and 3B are located on the outside of the loop. Two embodiments of a single CPL antenna element with electric field radiator are shown. 3A includes a phase tracker for broadband applications, as further described below, while FIG. 3B does not include a phase tracker and is also more suitable for less broadband applications. 4A, 4B and 4C illustrate other embodiments of a small cross-sectional antenna in which the electric field radiator (s) are located within the magnetic loop. Embodiments of antennas formed using any of these techniques can be easily assembled within mobile portable devices such as, for example, telephones, PDAs, laptops, and can also be assembled as separate antennas. 2 and other figures show an embodiment of a CPL antenna array using a microstrip forming technique. Such printing techniques make the antenna to be designed or manufactured compact and consistent.

The antenna 100 shown in FIG. 1 is arranged and printed on a section of the printed circuit board 101. The antenna includes a magnetic loop 110, in which case the loop is necessarily rectangular and generally has an open base. The two ends of the open base portion are fed from the coaxial cable 130 to the drive point in a generally known manner.

An electric field radiator or series of resonant circuits 120 is located inside the loop 110. The series of resonant circuits 120 are J-shaped traces on the circuit board 101 that are coupled to the loop 100 by a dead trace 124 acting as an inductor, meaning having an inductance or an inductive reactance. (122) takes the form. The J-shaped trace 122 necessarily has capacitive reactance characteristics dictated by its dimensions and the materials used for the antenna. Trace 122 serves as a series of resonant circuits with the sand trace 124.

Here, the antenna 100 is shown to be easily understood. The actual embodiment may not be physically similar to the antenna shown. In this case, the feed from coaxial cable 130 is shown, ie one end of the loop 132 is connected to the central conductor of the cable 130, while the other end of the loop 134 is connected to the cable. Is connected to the outer shield of 130. The loop antenna 100 differs from known loop antennas in that the series of resonant circuits 120 are coupled to the loop 134 midway around the loop. This coupling position serves as an important part of the operation of the antenna, as described below.

By carefully placing the series of resonant circuits 120 and the sand traces 124 with respect to the magnetic loop 110, the electric and magnetic fields generated / received by the antenna 100 may cause an electric field radiator to It may be formed to be perpendicular to each other, without having to be arranged to be orthogonal to the loop 110 physically. This orthogonal relationship has the effect of enabling the electromagnetic waves radiated by the antenna 100 to effectively propagate through space. To achieve this effect, the series of resonant circuits 120 and the sand traces 124 are positioned at about 90 degrees or about 270 degrees electrical position along the magnetic loop 110. In an alternate embodiment, the sand trace 124 may be located at a point where the current flowing through the magnetic loop, along the magnetic loop 110, is a reflective minimum. Thus, the sand trace 124 may or may not be located at about 90 degrees or 270 degrees electrical points. The point of the magnetic loop 110 at which the current is a reflective minimum depends on the geometry of the magnetic loop 110. For example, the point at which the current is a reflective minimum can be recognized as the first region of the magnetic loop. After adding or removing metal to the magnetic loop to achieve impedance matching, the point at which the current is a reflective minimum may change from the first region to the second region.

The magnetic loop 110 may have many different electrical and physical lengths; However, with respect to certain frequency band (s), for more efficient operation of the antenna, electrical lengths are provided that are multiples of wavelength, quarter wavelength and eighth wavelength. By adding an inductance to the magnetic loop, the electrical length of the magnetic loop is increased. By adding capacitance to the magnetic loop, the electrical length of the magnetic loop is reduced.

The vertical relationship between the magnetic field and the electric field is such that the series of resonant circuits at a physical position that is 90 degrees or 270 degrees around the magnetic loop from the drive point, the physical position varies based on the frequency of the signals transmitted / received by the antenna. By positioning 120 and the sand trace 124. As mentioned, this position may be 90 degrees or 270 degrees from the drive point (s) of the magnetic loop 110, as determined by the ends 132, 134, respectively. Thus, if the end 132 is connected to the central conductor of the cable 130, the sand trace 124 may be located at a 90 degree point as shown in FIG. 1 or at a 270 degree point. (Not shown in FIG. 1).

The vertical relationship between the magnetic field and the electric field is also achieved by placing the series of resonant circuits 120 and the sand traces 124 at a physical location around the magnetic loop where the current flowing through the magnetic loop is located at a reflective minimum. Can be. As mentioned above, the position at which the current reaches the reflection minimum depends on the geometry of the magnetic loop 110.

By arranging the circuit elements in this manner so that the phase relationship between the elements is 90 degrees, there is a vertical relationship between the electric and magnetic fields, which makes the antenna 100 more effectively function as both a receiving and transmitting antenna. Is formed. The magnetic field is generated alone (or necessarily alone) by the magnetic loop 110, while the electric field is provided in a series of resonances in which the energy delivered from the antenna is provided in a form suitable for transmission over a greater distance. Emitted by the circuit 120.

The series of resonant circuits 120 is a value selected to resonate at the operating frequency of the inductive (L) and capacitive (C) device (s), antenna 100 and the inductive reactance to match the capacitive reactance. Include them. This is because when the reactance of the capacitive element is equal to the reactance of the inductive element, i.e. when X _L = X _C , resonance produces the best efficiency. Thus, the values of L and C can be selected to provide a predetermined operating range. For example, other types of resonant circuits using crystal oscillators can be used to provide different operating characteristics. If a crystal oscillator is used, the Q-value of such a circuit is much larger than that of the simple LC circuit shown, thus limiting the bandwidth characteristics of the antenna.

As described above, the series of resonant circuits 120 effectively operate as electric field radiators (which also mean as electric field receivers thanks to the reciprocity inherent in the antenna). As shown, the series of resonant circuits 120 is a quarter wave antenna, but the series of resonant circuits is also a multiple of a full wavelength antenna, a multiple of a quarter wavelength antenna, or a multiple of a 1/8 wavelength antenna. Can work. If certain limitations interfere with certain wavelengths of the material to be used as traces 122, propagation delays are achieved to achieve a series of resonant circuits 120 that correspond to electrically full, quarter or eighth wavelengths. It is possible to use the sand traces 124 as meaning increasing < RTI ID = 0.0 > It may theoretically be possible, but in practice, instead of the series of resonant circuits, simply the current flowing through the magnetic loop is physically connected to the loop at the point where the reflection minimum is located or at a point at 90/270 degrees and also X _L It is not common to use a rod antenna of a predetermined wavelength, in accordance with the requirement of = X _C.

As mentioned above, the position of the series of resonant circuits 120 is important: it is at the position where the phase difference between the electric and magnetic fields is 90 degrees or 270 degrees or at the point where the current flowing through the magnetic loop is at the reflection minimum. It can be located and combined in a loop. Therefrom, the point at which the series of resonant circuits 120 is coupled to the magnetic loop 110 is referred to as a "connection point" and the connection point at 90 degrees or 270 degree electrical points along the magnetic loop is "90 /. 270 connection point ", and also the connection point at which the current is located at the reflective minimum is referred to as the" reflection minimum connection point ".

The amount of change in position of the connection point depends to some extent on the intended use of the antenna and the magnetic loop geometry. For example, the optimal connection point can be found by comparing the performance of the antenna using the 90/270 connection point to the performance of the antenna using the reflection minimum connection point. The connection point can then be chosen which results in the highest efficiency for the intended use of the antenna. The 90/270 connection point may not be different from the reflective minimum connection point. For example, an embodiment of an antenna may have a current at a reflection minimum at or near the 90/270 degree point. If a 90/270 degree connection point is used, the amount of change from the exact 90/270 degree is based to some extent on the intended use of the antenna, but is generally located close to 90/270 degree and the performance of the antenna Improves. The magnitude of the electric and magnetic fields should also ideally be generally the same or similar.

In practice, the point at which the series of resonating elements 120 couples to the loop 110 can be found empirically through the use of electric and magnetic field probes that define the point where the current is at the reflection minimum or the 90/270 degree position. have. The point at which the dead trace 124 can be coupled to the loop 110 can be determined by moving the trace 124 until a certain 90/270 degree difference is observed. As another method for determining the reflection minimum connection point and the 90/270 connection point along the loop 110, the best connection point along the loop 110 is the area (s) of the minimum surface current magnitude (s). There is a method of visualizing the surface current of an electromagnetic software simulation program to be visualized.

Thus, empirical measurements, the degree of trial and error, require a guarantee of the sinister performance of the antenna, although the principles under the arrangement of the elements are well understood. This is simply due to the nature of the printed circuits, often requiring a degree of 'tuning' before the desired performance is achieved.

Known simple loop antennas offer a very wide bandwidth, usually one octave, whereas known antennas, such as dipoles, have very narrow bandwidths--typically a much smaller fraction of the operating frequency (such as 20% of the operating center frequency). Has

Printed circuit technology is well known and is not described in detail herein. It is sufficient only that the copper traces are arranged and printed (usually through etching or laser trimming) on a suitable substrate with special dielectric effects. With careful selection of materials and dimensions, specific values of capacitance and inductance can be achieved without requiring separate separate elements. However, as will be described further below, the design of this embodiment relaxes the substrate limitations of previous high frequency planar antennas.

As described above, the present embodiments are arranged and manufactured using known microstrip techniques in which the final design is reached as a result of a certain amount of manual calibration and the physical traces on the substrate are adjusted accordingly. In practice, calibrated capacitance sticks are used and include metal elements with known capacitance elements of 2 picoFarad, for example. For example, a capacitance stick can be placed in contact with various locations of the antenna trace, where the performance of the antenna is measured.

Under the management of skilled technicians or designers, such techniques indicate that the traces that make up the antenna can be scaled, such as by adjusting the capacitance and / or inductance. After a number of iterations, an antenna with some performance may be achieved.

The connection point between the series of resonating elements and the loop is again determined empirically using electric and magnetic field probes. Once the proper connection location has been determined, fine-tuning the L and C values and / or connections by laser trimming the in-situ traces, keeping in mind that minimal intervention from the test equipment at the frequencies discussed can have a large practical effect. Can be done. Once the final design is formed, it can be recycled with the product repeatedly. Alternatively, the connection point between the series of resonating elements and the loop can be determined using an electromagnetic software simulation program for visualizing the surface current, and also selecting the area or regions where the surface current is located at the minimum.

Antennas formed in accordance with the above-described embodiments generally provide for improved efficiency of antennas of similar volume or larger known in the art.

In a further embodiment, a plurality of separate antenna elements can be combined to provide greater performance than would be achieved by the use of a single element.

2 shows antenna 200 arranged and printed on a section of circuit board 205 in a known manner. Although the circuit board 205 has been described in a plane, there is a certain amount of thickness on the substrate constituting the circuit board, and the ground plane area 624 shown in FIGS. 6 and 8 has a ground plane (not shown). Printed on the back side of the circuit board 205 in a similar manner. In FIG. 2, the antenna 200 comprises four separate antenna elements 210 that are functionally identical, arranged as two sets, each driven in parallel.

The effect of providing many examples of the basic antenna element 210 is to improve the overall performance of the antenna 200. If there is no loss associated with the configuration of the antenna, it may be possible in theory to construct an antenna comprising very many individual instances of the basic antenna elements 210, the number of elements each adding 3 dB of gain to the antenna. Redundant In practice, however, the loss-in particular the dielectric heating effect-means that it is not possible to add extra elements indefinitely. The example shown as a four-element antenna in FIG. 2 suffices to be physically possible, adding 6 dB of gain (with little dielectric heating loss) over an antenna composed of a single element.

The antenna 200 of FIG. 2 is suitable for use in an article of a micro-cellular base station or other fixed wireless infrastructure, while the single element 210 is a cellular or mobile handset, a pager, a PDA or a laptop. Suitable for use in mobile devices such as computers. The only real decision issue is size. The components and operation of the elements 210 are further described and illustrated in FIGS. 3A and 3B for antennas 310 and 370, respectively.

FIG. 3A is one or as described below, through the inclusion of the phase tracking antenna element 300, specifically adapted to provide a larger operating bandwidth (optical bandwidth) than the narrow bandwidth antenna 100 of FIG. 1. A single antenna 310 (an embodiment of one of the elements 210 of FIG. 2) that can achieve a large bandwidth up to 1/2 octave is described. In particular, this optical bandwidth is achieved by combining the phase tracker 330 with the rectangular electric field radiator 320 and the loop element 350. The rectangular electric field radiator 320 replaces the series of resonant circuits 120 shown in FIG. However, the operating bandwidth of the rectangular electric field radiator 320 is also wider than that of the tuning circuit 120 due to the operation of the phase tracker 330 as described below.

For an alternative embodiment of the antenna 310, the same rectangular electric field radiator 320, the loop element 350, and the drive or feed point 340 are provided, but without the phase tracker 330. 3B is described as an antenna 370, such as antenna 310 of FIG. 3A, having a narrower operating bandwidth than antenna 310. Another method for incorporating wide band operation is illustrated by the CPL antenna element of FIG. 4A incorporating multiple electric field radiators 404 and 408 as described further below.

In the case of the tuning circuit 120, the connection point between the tuning circuit and the loop was important in determining the overall performance of the antenna 100. In the case of electric field radiator 320 in the antennas 310, 370 from FIGS. 3A and 3B, located on the outside of the loop 350, although the connection point is generally still at the center frequency, the loop ( 350) Even if placed at a 90/270 degree midway around or at a point where the current is located at the reflective minimum, the exact location is as described above, since the connection point is effectively distributed along the length of one side of the electric field radiator. Less important. As such, the end points where the edges of the electric field radiator 320 meet the loop 350 determine the operating frequency range of the antennas 310, 370 along with the dimensions of the loop.

The dimensions of the loop 350 are also important in determining the operating frequency of the antennas 310, 370. In particular, the overall length of the loop 350 is a key dimension as described above. To allow a wide operating frequency range, a triangular phase tracker element 330 is provided directly opposite the electric field radiator 320 (located in one of two possible locations as shown in FIG. 2). The phase tracker 330 effectively acts as an automatic variable length tracking device that lengthens or shortens the electrical length of the loop 350 based on the frequency of the RF signal fed into it at the feed or drive point 340.

The phase tracker 330 is like a near-infinite series of L-C elements, only a portion of which resonates at a given frequency, thus automatically changing the effective length of the loop. In this way, a wider operating bandwidth can be achieved than with a simple loop without any such phase tracking elements.

The phase trackers 330 shown in FIG. 2 have two different possible positions. These locations are selected for each antenna element 210 in the group of antenna elements 210 shown in FIG. 2 to minimize mutual intervention between adjacent antenna elements 210. In electrical terms, the two configurations are functionally identical.

Since the magnetic loop 350 is a completely short signal current, a large bandwidth (octave less than 1½) of the antennas 310 and 370 is possible. As shown in FIGS. 3A and 3B, the magnetic loop is completely short as it is half wave short, but it can also be completely short at quarter wave open and full short wave. The phase of the antenna is determined by the dimension 360. The dimension 360 spans between the length of the electric field radiator 320 and the length of the left side of the magnetic loop 350. The signal is shortened at the point where the signal is located 180 degrees from the phase. The magnetic field having the largest magnitude is generated by the magnetic loop, and the small magnitude of the magnetic field is generated by the electric field radiator. In addition, the magnetic loop can vary from RF short whose length is very low to near RF open with very high real impedance. The largest magnitude electric field is emitted by one or more electric field radiator elements. However, the magnetic loop also produces a small electric field which is smaller in magnitude than the electric field emitted by the electric field radiator and which is opposite to the magnetic field.

The efficiency of the antenna is achieved by minimizing the current in the magnetic loop to generate the highest possible magnetic field. This is accomplished by designing the antenna so that current flows into the electric field radiator and also reflects in the opposite direction, as further described in FIG. 6 below. Since more current can be used for transmission purposes, the maximum magnetic field protrudes from the antenna in all directions to maximize the efficiency of the antenna. The maximum magnetic field energy that can be generated occurs when the magnetic loop is a full RF short or when the magnetic loop has a very low real impedance. However, under normal circumstances, RF short is undesirable because it causes the transmitter driving the antenna to fail. The transmitter produces an energy set amount at the set impedance. By using the impedance matching characteristic of the electric field, it is possible to have a near RF short loop without causing the transmitter to fail.

Current flowing through the magnetic loop flows into the electric field radiator. The current is then reflected by the electric field radiator along the opposite direction into the magnetic loop, whereby the electric field reflects into the magnetic field to form a short of the electric field radiator and also forms an orthogonal electric and magnetic field.

The dimension 365 consists of the width of the electric field radiator 320. The dimension 365 does not affect the efficiency of the antenna, but its width determines whether the antenna is narrowband or wideband. The dimension 365 only has a large width to widen the band of the antenna 310 described in FIG. 3A.

For example, all of the trace elements of the magnetic loop described in FIG. 3A can be made very thin without affecting the performance or efficiency of the antenna. However, by making these loop trace elements thicker, some of them may be able to accommodate larger input power, or may be required by many other portable devices, such as mobile phones operating within certain frequency ranges. It is possible to change the physical size of the antenna to match the space of the antenna.

Those skilled in the art will clearly appreciate that any form of electric field radiator may be used in the multi-element configuration shown in FIGS. 2, 3A and 3B, with a rectangular electric field radiator 320 configured merely by way of example. Likewise, single element embodiments may use rectangular electric field radiators, tuning circuits or any other suitable type of antenna. The multi-element version shown in FIG. 2 uses four separate antennas 210, but this is an available space and accurate system with some limitations on the upper range of the elements 210 as described below. It can be changed up and down based on the requirements.

Embodiments of the present description have superior performance characteristics, compared to known antennas of similar size, are operable over much increased bandwidth, and allow the use of single or multi-element antennas. It also creates a low cost device that does not require any complex components and is applicable to a wide range of RF devices. Embodiments of the present description find particular use in mobile communication devices, but may be used in any device where an efficient antenna is desired.

One embodiment consists of a small, cross-sectional composite antenna ("cross-section antenna" or "print antenna"). By "cross-section" it is meant that the antenna elements are positioned or printed on a single layer or plane if desired. As used, the phrase “print antenna” refers to whether the elements of the printed antenna are printed or etched, stacked, sputtered, or some other method of providing a metal layer on a surface or placing a non-metallic material around the metal layer. Regardless of whether it is formed as or not, it applies to any cross-sectional antenna described herein. Multiple layers of the cross-sectional antennas may be combined in a single device to provide wider bandwidth operation at smaller physical volumes, but each of the devices is still cross-sectional. The above-described cross-sectional antenna has no ground plane on the rear side or the bottom plane and is necessarily a shortened device, which itself represents a new concept in antenna designs. The cross-sectional antenna is balanced but can be driven with balanced or unbalanced lines if an important ground plane is present in the device of the intended application. The physical size of such an antenna can vary greatly based on the performance characteristics of the antenna, but the antenna 400 described in FIG. 4A is about 2 cm x 3 cm. Similar or larger implementations are possible.

The cross-sectional antenna 400 is composed of two electric field radiators which are physically located inside the magnetic loop. In particular, as shown in FIG. 4A, the cross-sectional antenna 400 consists of a magnetic loop 402, where the first electric field radiator 404 has a magnetic loop 402 having a first electric trace 406. ), And the second electric field radiator 408 is connected to the magnetic loop 402 having the second electric trace 410. The electrical traces 406, 410 connect the electric field radiators 404, 408 to the magnetic loop 402 at corresponding 90/270 degree electrical locations with respect to a feed or drive point. Alternatively, the electrical traces 406 and 410 may connect the electric field radiators 404 and 408 to the magnetic loop in the region where the current flowing through the magnetic loop is located in the reflection minimum. As discussed above, for other frequencies, the connection or connection point of the traces 406, 410 changes, which is why at one frequency the radiator 404 is different from the radiator 408 where the radiator 404 is located at another frequency. It will be described in the connection to the loop 402. At low frequencies, it takes a long time for the wave to reach the 90/270 degree point; In conclusion, the physical position of the 90/270 degree point will be higher along the magnetic loop compared to the high frequency. At high frequencies, reaching the 90/270 degree point takes less time, resulting in the physical location of the 90/279 degree point being lowered along the magnetic loop compared to the low frequency. Likewise, the points along the magnetic loop where the current is located in the reflection minimum may also depend on the frequency of the electric field radiator. Finally, alternative embodiments of the antenna 400 may consist of one or more electric field radiators that are directly coupled to the magnetic loop 402 without electrical traces.

The electric field radiator 404 also has a different size than the electric field radiator 408 because each electric field radiator emits waves at different frequencies. Small electric field radiator 404 may have a short wavelength and thus a high frequency. Large field radiator 408 may have long wavelengths and low frequencies.

The physical arrangement of the electric field radiator (s) physically located inside the magnetic loop will reduce the size of the entire antenna as compared to the physical location of the electric field radiator (s) and other embodiments where the magnetic loops are external to each other. Can and at the same time provide a broadband device. Alternative embodiments may have different numbers of electric field radiators, each arranged at different locations around the loop. For example, the first embodiment may have only one electric field radiator located inside of the magnetic loop, while the second embodiment with two electric field radiators draws one electric field radiator on the inside of the magnetic loop. It may have a second electric field radiator on the outside of the magnetic loop. Alternatively, more than two electric field radiators may be physically located inside the magnetic loop. Like the other antennas described above, the cross-sectional antenna 400 acts as a transducer thanks to the electric and magnetic fields.

As mentioned above, the use of multiple electric field radiators is allowed for broadband functionality. Each electric field radiator can be configured to emit waves at different frequencies, with the result that the electric field radiators cover a wide range. For example, the cross-sectional antenna 400 may be configured to cover the standard IEEE 802.11b / g radio frequency range with the use of two electric field radiators configured in two frequency ranges. The first electric field radiator 404 may be configured to cover the 2.41 GHz frequency, for example, while the second electric field radiator 408 may be configured to cover the 2.485 GHz frequency, for example. This allows the cross-sectional antenna 400 to cover a frequency band of 2.41 GHz to 2.485 GHz corresponding to the IEEE 802.11b / g standard. The use of two or more electric field radiators results in wideband operation without the use of a phase tracker (as shown in FIGS. 2 and 3), as described in connection with the physically large antenna embodiments described above. In an alternative embodiment, wideband antennas can also be achieved by taping multiple electric field radiators using a log scale, similar to YAGI antennas.

In general, the length of the electric field radiators determines the frequencies they cover. Frequency is inversely proportional to wavelength. Thus, small electric field radiators have a small wavelength and result in high frequencies. On the other hand, large electric field radiators have long wavelengths and result in low frequencies. However, such generalizations are also practical characterizations.

For optimum efficiency, the electric field radiator may have an electrical length of multiples of about 1 wavelength, 1/4 wavelength or 1/8 wavelength of the frequency generated. As mentioned above, if the electrical length of the electric field radiator is limited such that the amount of available physical space is less than a predetermined wavelength, a dead trace is used to electrically lengthen the electric field radiator and also add propagation delay. Can be.

4A and 4B, the electrical traces 406 and 410 are inductors and their individual length versus their shape or other features determine their inductance. For optimal efficiency, the inductive reactance of the electrical trace can be matched with the capacitive reactance of the corresponding electric field radiator. The electrical traces 406 and 410 are bent to reduce the overall size of the antenna. For example, the curve of the electrical trace 406 is formed closely with the magnetic loop 402 instead of being formed closely with the electric field radiator 404, or the curve of the trace 406 is with the electrical trace. Similar to 410, instead of rising it can be lowered. The electrical traces are shaped to extend their length, but that is not because the shape has any particular significance in that context. For example, instead of having a straight electrical trace, a curve can be added to the electrical trace to increase its length and correspondingly increase its inductive reactance. However, sharp corners on the electrical trace and sinusoidal shape of the electrical trace can adversely affect the efficiency of the antenna. In particular, an electrical trace in the form of a sinusoidal wave causes the electrical trace to emit a small electric field that partially outphases the electric field radiator and reduces the efficiency of the antenna. Thus, the efficiency of the antenna can be improved by using electrical traces molded into soft and elegant curves or with as few bends as possible.

The spacing between elements in the cross-sectional antenna 400 adds capacitance to the entire antenna. For example, the spacing between the top of the electric field radiator 404 and the magnetic loop 402, the spacing between the two electric field radiators 404 and 408, the left side of the electric field radiators 404 and 408, and The spacing between the magnetic loops 402, the right side of the electric field radiators 404, 408 and the magnetic loop 402, and the bottom of the electric field radiator 408 and the magnetic loop 402. The spacing between all affects the capacitance of the antenna 400. As described above, in order to resonate the antenna 400 with optimum efficiency, the inductive reactance and the capacitive reactance of the entire antenna are matched in a given frequency band (s). Once the inductive reactance is determined, the distance between the various elements can be determined based on the capacitive reactance value needed to match the inductive reactance value for the antenna.

Formula sets are given to find the space between the devices and the associated edge capacitance, and the optimal space between the devices can be determined using multi-objective optimization. The optimal space between elements or between any two adjacent antenna elements can be optimized using linear programming. Alternatively, nonlinear programming such as a genetic algorithm can be used to optimize the spatial values.

As mentioned above, the size of the cross-sectional antenna 400 is based on a number of factors including the desired operating frequency, narrow band to wide band function, and tuning of capacitance and inductance.

In the case of the antenna element 400 of FIG. 4A, the length of the magnetic loop 402 is one wavelength (360 degrees) designed for optimal efficiency, although multiples of other wavelengths may also be used. When designed for optimum efficiency, part of the magnetic loop also acts as an electric field radiator, and the electric field radiator also generates a small magnetic field in addition to the directivity and efficiency of the antenna. The length of the magnetic loop can be arbitrary, or can be a multiple of approximately 1 wavelength, 1/4 wavelength or 1/8 wavelength, which is a particular length that can increase efficiency over other cases. One wavelength is an open circuit for voltage and a short circuit for current. Alternatively, the length of the magnetic loop 402 may be physically less than one wavelength, but extra inductance may be added to electrically lengthen the loop by increasing the propagation delay. The width of the magnetic loop 402 is primarily based on the capacitance as well as the desired effect on the inductance of the magnetic loop 402. For example, by physically shortening the magnetic loop 402, the wavelength is made small to cause a high frequency. In the design for optimal efficiency of the magnetic loop 402, inductance and capacitance must satisfy the equation w = l / sqrt (LC), where w is the wavelength of the loop 402. Thus, the magnetic loop 402 can be tuned by varying inductance and capacitance affecting the electrical length. By reducing the width of the magnetic loop, it also adds inductance. In thinner magnetic loops, more electrons must be forced through a small area, additional delay.

The upper portion 412 of the magnetic loop 402 is thinner than any other portion of the magnetic loop 402. This allows the size of the magnetic loop to be adjusted. The upper portion 412 can be reduced since it has a minimal impact on the 90/270 degree connection point. In addition, the upper portion 412 of the magnetic loop 402 is trimmed to increase the electrical length of the magnetic loop 402 and also increase the inductance such that the inductive reactance matches the total capacitive reactance of the antenna. Can help. Alternatively, the height of the upper portion 412 can be increased to increase capacitance (or equally reduce inductance). As mentioned above, the reflection minimum connection point depends on the geometry of the magnetic loop. Thus, by trimming the upper portion 412 or increasing the upper portion 412 or changing any other aspect of the magnetic loop, the loop is changed geometrically to recognize after the loop geometry has been modified. This would require a point where the current would be located at the reflection minimum.

The magnetic loop 402 should not be square as described in FIG. 4A. In an embodiment, the magnetic loop 402 may be rectangular or unusual in shape, and two electric field radiators 404 and 408 may be located at corresponding 90/270 degree connection points or reflective minimum connection points. . For optimum efficiency, the electrical length of the odd shaped loop can be a multiple of approximately one wavelength or a multiple of approximately 1/4 or 1/8 wavelength in a given frequency band (s). The electric field radiators may be located on the inside or outside of the singularly shaped magnetic loop. The key is also to identify the connection point along the magnetic loop that maximizes the efficiency of the antenna. The connection point may be a 90/270 degree electrical point along the magnetic loop or a point where the current flowing through the magnetic loop is located at the reflection minimum.

For example, in a smartphone, the singular shaped antenna design can be matched in the available singular shaped space, such as the back cover of the mobile device. Instead of the magnetic loop having a square shape, it may have a rectangular shape, a circular shape, an elliptical shape, a generally E-shaped shape, a generally S-shaped shape, or the like. Likewise, small singular shaped antennas can be matched in a non-uniform space on a laptop computer or other portable electronic device.

As described above, the location of the electrical trace is about 90/270 degree electrical point or reflective minimum connection point along the magnetic loop such that the electric field radiated by the electric field radiator is perpendicular to the magnetic field generated by the magnetic loop. It can be located at The 90/270 degree connection point and the reflective minimum connection point are important because these points prevent reactive power (virtual power) from being transmitted away from the antenna and not returned. Reactive power is generally generated and stored around the near electric field of the antenna. Reactive power oscillates around a fixed location near the source and affects the operation of the antenna.

Referring to FIG. 4A, the broken line 414 shows that the most important areas of the edge capacitance phenomenon occur. A piece of metal in the antenna, such as magnetic loops and electric field radiators, at a certain distance away may form the level of edge capacitance. Through the use of edge capacitance, embodiments of cross-sectional antennas allow all the elements of the antenna to be printed on one side of almost any type of suitable substrate materials, including inexpensive dielectric materials. Examples of inexpensive dielectric materials that can be used as the substrate include glass reinforced epoxy laminates (FR-4) having a dielectric constant of about 4.7 ± 0.2. For example, in the cross-sectional antenna 400, there is no need for a back or ground plane. Rather, lead is connected to each end of the magnetic loop and one of the leads is grounded. As mentioned above, this full wavelength antenna design means a shorted composite loop antenna with optimal efficiency. In practice, the cross-sectional antenna will perform most optimally in the presence of a counterpoise ground plane, as is commonly seen in buried antenna designs in which counterpoise is provided by the object on which the antenna is mounted.

The 2D design of the embodiment of the cross section antenna has several advantages. With the use of a suitable substrate or dielectric base that can be very thin, the traces of the antenna are actually sprayed or printed onto the surface and still function as a composite loop antenna. The 2D design also allows the use of antenna materials that are not generally seen as suitable for microwave devices, such as very inexpensive substrates. Another advantage is that the antenna can be located on singularly shaped surfaces, such as the back of the cell phone case cover, the edges of the ramtop, and the like. Embodiments of the cross-sectional antenna may be printed on the dielectric surface with an adhesive located on the back side of the antenna. The antenna can then be attached on various computing devices with leads connected to the antenna to provide the necessary power and ground. For example, as described above, with such a design, an IEEE 802.11b / g wireless antenna may be printed on the surface approximately in stamp size. The antenna may be attached to the cover of a laptop, the case of a desktop computer, or the back cover of a cell phone or other portable electronic device.

Various dielectric materials can be used as an embodiment of the cross-sectional antenna. The advantage of FR-4 as a substrate on other dielectric materials, such as polytetrafluoroethylene (PTFE), is that it has a low cost. In general, dielectrics used for high frequency antenna designs have much lower loss characteristics than FR-4, but they are generally more expensive than FR-4.

Embodiments of the cross-sectional antenna may also be used for narrowband applications. Narrowband is associated with a channel where the bandwidth of the message does not exceed the channel correlation bandwidth. For wideband, the message bandwidth greatly exceeds the channel correlation bandwidth. Narrowband antenna applications include Wi-Fi and point-to-point long distance microwave links. According to the embodiments described above, for example, an array of narrowband antennas can be printed on a sticker, and then for Wi-Fi access beyond good signal strength and a large distance compared to standard Wi-Fi antennas. May be located on a laptop.

4B illustrates another embodiment of a cross-sectional antenna 420 with a magnetic loop 422 having corners cut at an angle of about 45 degrees. By cutting the corners of the magnetic loop 422 at a certain angle, the efficiency of the antenna is improved. By having a magnetic loop with corners that are formed at an angle of about 90 degrees, it affects the flow of current flowing through the magnetic loop. When the current flowing through the magnetic loop strikes a 90 degree angle corner, it causes the current to recoil, and the reflected current flows against the main current flow or forms an eddy pool. Energy loss as a result of the 90 degree corners can adversely affect the performance of the antenna, in particular of the small antenna embodiment. Cutting the coaters of the magnetic loop at an angle of about 45 degrees improves the current flow around the corners of the magnetic loop. Thus, the angled corners make the electrons in the current less imminent as they flow through the magnetic loop. It is suitable to cut the corners at a 45 degree angle, but alternative embodiments of cutting at angles other than 45 degrees are also possible.

4C illustrates an alternative embodiment of a cross-sectional antenna 440 that uses transitions of various widths in the magnetic loop 442 to add inductance or capacitance to the magnetic loop 442. The corners of the magnetic loop 442 were cut at an angle of about 45 degrees to improve the flow of current as the current flows around the corners of the magnetic loop 442, thereby increasing the efficiency of the antenna. A single electric field radiator 444 is physically located inside the magnetic loop 442. The electric field radiator 444 is connected to the magnetic loop 442 with an electric trace 446 having a soft curve shape. As discussed above, the efficiency of the antenna is improved by having an electrical trace 446 that is not sinusoidal and also has soft curves that minimize the number of bends in the trace.

The term transition is used to refer to a change in the width of the magnetic loop. In FIG. 4C, the magnetic loop 442 takes a generally rectangular shape and includes a first transition portion on the left side and a second transition portion on the right side. In the embodiment shown in FIG. 4C, the first transition portion is symmetrical with the second transition portion. The transition portion on both the left and right sides of the magnetic loop 442 is thinner than the remainder of the magnetic loop 442 and is between the first wide section 450 and the second wide section 452. A middle narrow section 448 or a middle narrow segment, positioned adjacent to all of them, wherein the first wide section 450 and the second wide section 452 are less than the narrow section 448. Have a large width. In particular, the magnetic loop transitions from the first wide section 450 to the middle narrow section 448, and the middle narrow section 448 transitions to the second wide section 452. The wide-narrow-wide transition in the magnetic loop produces a pure inductance, thus increasing the electrical length of the magnetic loop. Thus, the use of the wide-narrow-wide transitions in the magnetic loop is a method of increasing the electrical length of the magnetic loop 442 by adding inductance to the magnetic loop 442. The length of the intermediate narrow section 448 can also be increased or decreased as needed to add some inductance to the magnetic loop. For example, in FIG. 4C, the intermediate narrow section 448 spans about one quarter of the right side and left side of the magnetic loop 442. However, the intermediate narrow section 448 can be increased to span about one half of the right and left side portions of the magnetic loop 442, thus increasing the inductance of the magnetic loop 442. do.

The transitions are not limited to sections or segments having a width smaller than the remainder of the magnetic loop 442. An alternative transition portion may comprise a medium wide section or a medium wide segment, wider than the remainder of the magnetic loop 442 and located between, and adjacent to, the first narrow section and the second narrow section, wherein The first narrow section and the second narrow section have a smaller width than the wide section. In particular, in such an alternative embodiment, the magnetic loop transitions from the first narrow section to the middle narrow section, which later transitions to the second narrow section. Narrow-width-narrow transitions in the magnetic loop produce capacitance, so the electrical length of the magnetic loop is shortened. The length of the intermediate broad can be increased or decreased by adding capacitance to the magnetic loop.

Acts as a method for tuning impedance matching by using a transition in the magnetic loop, ie by changing the width of the magnetic loop to more than one or more sections or segments of the magnetic loop. Transitions that change the widths in the magnetic loop may also be tapered to further add inductance or capacitance to ensure that the reactive inductance and reactive capacitance of all the elements of the antenna are matched. For example, in the wide-narrow-wide transition, the first wide section may taper from its large width to the small width of the intermediate narrow section. Similarly, the intermediate narrow section may taper from its narrow width to a greater width of or both of the first wide section or the second wide section. Sections in the narrow-width-narrow transition and the wide-narrow-wide transition may be tapered independently of each other. For example, in the first narrow-width-narrow transition, only the intermediate wide section may be tapered, whereas in the second narrow-width-narrow transition, only the first narrow section may be tapered. have. The tapering may be linear, stepped or curved.

The actual difference in width between the portions of the magnetic loop will depend on the amount of inductance or capacitance required to reliably match the antenna's total reactive capacitance with the antenna's total reactive inductance. The embodiment described in FIG. 4C shows two wide-narrow-wide transitions symmetrically positioned opposite one another. However, alternative embodiments may have a transition on only one side of the magnetic loop 442. Also, if more than one transition is used in the magnetic loop, these transitions do not need to be symmetrical. For example, the singularly shaped magnetic loop may have two transitions, which transitions have different lengths and widths. In addition, other types of transitions may also be used on a single magnetic loop. For example, the magnetic loop may have one or more narrow-width-narrow transitions or both one or more wide-width-wide transitions.

5 illustrates an embodiment of a small double-sided antenna or planar antenna 500. The planar antenna 500 is on the back side comprising a tunable patch represented by dashed line 502 that generates a capacitive reactance to match the inductive reactance of the magnetic loop 504 for a particular frequency. Use a second plane. The variable patch 502 is a generally square metal piece having a fixed position relative to other elements of the antenna 500. In one embodiment, the variable patch 502, as shown in Figure 5, at a point away from the 90/270 degree electrical point along the magnetic loop or reflects a current such as the upper left corner of the antenna 500 It may be located at a point away from the area located at the minimum. The electric field radiator 506 is located inside the magnetic loop 504 to reduce the overall size of the double-sided antenna 500. For optimum efficiency, the electric field radiator 506 may have an electrical length that corresponds to nearly a quarter wavelength at its corresponding operating frequency. If the electric field radiator is small, it causes smaller wavelengths at higher frequencies. The electric field radiator 506 is bent in a generally J shape to match the entire length inside the magnetic loop 504. Alternatively, the electric field radiator 506 may be stretched to be placed on a straight line or bent in another form, rather than bent into a J shape. Such an embodiment has been considered herein, but it makes the antenna wider and also increases the overall size of the antenna.

The electrical trace 508 connects the electric field radiator 506 to the magnetic loop 504 at the 90/270 degree connection point or at the minimum reflective connection point. The upper portion 510 of the magnetic loop 504 is smaller compared to the other sides of the magnetic loop 504. This serves for the purpose of lengthening the electron length of the magnetic loop 504 and increasing the inductance. The increase in inductance also allows matching the inductive reactance with the total capacitive reactance of the antenna 500, as in the case of the small cross-sectional antenna 400, and can also be adjusted as described above.

The variable patch 502 may also be located anywhere along the upper portion 510 of the magnetic loop 504. However, when the variable patch 502 is located away from the point where the magnetic loop 504 is connected to the electric field radiator 506, the best performance is achieved. The size of the variable patch 502 can also be increased by changing its depth, length, and height. Increasing the depth of the variable patch 502 results in an antenna design that can take up more space. Alternatively, the variable patch 502 can be made very thin, but its length and height can be adjusted correspondingly. Instead of having the variable patch 502 covering the upper left corner of the antenna 500, the length and height may be increased to cover the left half of the antenna 500. Alternatively, the length of the variable patch 502 can be increased, allowing the upper half of the antenna 500 to expand. Similarly, the height of the variable patch 502 can be increased, allowing the left side of the antenna 500 to expand. The variable patch can also be made small.

Similar to the cross-sectional antenna, various dielectric materials may be used in embodiments of the double-sided antenna 500. Dielectric materials that can be used include FR-4, PTFE, crosslinked polystyrene, and the like.

FIG. 6 describes an embodiment of a large antenna 600, consisting of an array of four antenna elements 602 and having a bandwidth of one or half octaves. Each antenna element 602 is a TE mode (electric wave) radiator, or a magnetic field (H field) radiator, or a magnetic loop dipole 604 (schematically indicated by a dashed line) outside the magnetic loop 604. 604] and TM mode (transverse magnetic) radiator, or electric field (E-field) radiator, or electric field dipole 606 (represented as a rectangular shaped area and also referred to as electric field radiator 606). The magnetic loop 604 should be electrically one wavelength causing a short circuit. On the other hand, while the magnetic loop 604 may be physically smaller than one wavelength, the magnetic loop 604 is electrically stretched by adding extra inductance as discussed below. The physical width of the magnetic loop 604 can also be adjusted to obtain the appropriate impedance / capacitance of the magnetic loop 604 so that it can also resonate at a predetermined frequency. As described below, the physical parameters of the magnetic loop 604 do not depend on the amount of dielectric material used for the antenna elements 602.

As described above, the magnetic loop 604 is fully shorted to maximize the amount of current in the magnetic loop and also generate the highest magnetic field. At the same time, impedance is matched to the load from the transmitter to prevent the transmitter from becoming unusable as a result of a short. Current moves from the magnetic loop 604 into the electric field radiator 606 in the direction of arrow 607 and in the opposite direction (from the electric field radiator 606 into the magnetic loop 604 in the direction of arrow 609). Is returned.

In an embodiment, each of the antenna elements 602 is about 4.45 cm wide by about 2.54 cm high, as described in FIG. 6. However, as mentioned above, the size of all components is determined by the operating frequency or other characteristics. For example, the traces of the magnetic loop 604 are made very thick, thereby increasing the gain of the antenna element 602 and also increasing the physical size of the antenna element 602 and thus the size of the antenna 600. It can be changed to match any given physical space, remains in resonance, sustains some of the same increased gain, maintains similar levels of efficiency, and the voltages available in the prior art to the antenna It is not supplied at all. As long as the modified design has (1) a magnetic loop with inherited hermetically sealed surface current, (2) energy reflection from the electric field radiator into the magnetic loop, and (3) the matched impedance of the components, the antenna is nearly It can be adjusted to any size. Although gain may vary based on the particular size and shape selected for the antenna, a similar level of efficiency can be achieved.

Phase tracker 608 (denoted by the triangular shaped region) makes antenna 600 wideband and can also be removed for narrowband design. The tip of the phase tracker 608 is ideally positioned at a 90/270 degree electrical position along the magnetic loop 604. However, in an alternative embodiment, the tip of the phase tracker may be located at the minimum reflective connection point. The dimension 610 of the electric field radiator 606 does not really matter in the overall operation of the antenna element 602. Dimension 610 only has a width to make antenna element 602 wideband, and dimension 610 may be reduced if antenna element 602 tends to be a narrowband device. As described, antenna element 602 tends to be broadband because it includes phase tracker 608. Dimension 612 is determined by the center operating frequency and also determines the phase of antenna element 602. The dimension 612 is located over the length of the electric field radiator 606 and the length of the left side of the magnetic loop 604. Dimension 612 may generally be a quarter wavelength and is slightly adjusted for the dielectric material used as the substrate. The electric field radiator 606 has a length representing about a quarter wavelength at the frequency of interest. The length of the electric field radiator 606 may also be sized to be a multiple of a quarter wavelength at the frequency of interest, but such a change may reduce the effectiveness of the antenna.

The width of the upper portion 614 of the magnetic loop 604 is intended to be smaller than any other portion of the magnetic loop 604, but such differences may not appear in the figure of FIG. 6. This difference in size is similar to the small antenna embodiments already described, and the upper portion 614 can be removed to increase the electrical length and add inductance. The upper portion 614 of the magnetic loop 604 can be removed as it has a minimal impact on the 90/270 degree electrical position. When inductance is added by removing the upper portion 614, the magnetic loop 604 exhibits an electrically long life.

The dimensions 616, 617, 618 of the magnetic loop 604 are all determined by the wavelength dimension. Dimension 616 consists of the width of the magnetic loop 604. Dimension 617 consists of the length of the left portion of the bottom side of the magnetic loop 604. That is, the dimension 617 consists of the length of the bottom portion of the magnetic loop 604 relative to the left side of the magnetic loop opening 619. Dimension 618 consists of the entire length of the magnetic loop 604. Best antenna performance is achieved when the dimension 616 is the same size as the dimension 618, resulting in a rectangular loop. However, magnetic loops 604 having a rectangular or irregular shape may also be used.

As mentioned above, the phase tracker 608 is included for wideband operation of the antenna 600, and the removal of the phase tracker 608 causes the antenna 600 to be less broadband. The antenna 600 may be formed narrowband by selectively reducing the physical vertical dimension of the phase tracker 608 and the dimension of the electric field radiator 606. The support of broadband operation in the phase tracker 608 and antenna has the potential to reduce the total number of antennas used in various devices such as cell phones. The dimensions of the phase tracker 608 also affect its inductance and capacitance as described in FIG. The range of capacitance and inductance of the phase tracker 608 may be tuned by adjusting the physical dimensions of the phase tracker 608. The inductance L of the phase tracker 608 is based on the height of the phase tracker 608. The capacitance C of the phase tracker 608 is based on the width of the phase tracker 608.

The antenna elements 602 and pairs of antenna elements 602 have a set of gaps formed between them. The two antenna elements 602 located on the left side of the antenna 600 constitute the first pair of antenna elements 602 while the two antenna elements located on the right side of the antenna 600 Elements 602 make up a second pair of antenna elements 602. A first gap 620 exists between each pair of antenna elements 602, and a second gap 622 exists between each set of antenna elements 602. The first gap 620 between each pair of elements 602 and the second gap 622 between each set of pairs of antenna elements 602 allow far-field radiation patterns to be reduced. Rather, it is designed to align the far-field radiation patterns generated by the antenna elements 602 in the most effective manner. Well known phase antenna array techniques can be used to determine the optimal spacing between multiple CPL antenna elements 602, resulting in the far-field radiation pattern of each of the elements.

In an embodiment, the far-field radiation patterns may be modeled on a computer based on the relationship of other components of the antenna elements 602. For example, the size of the antenna elements 602, the space between the antenna elements 602 or between the pairs of antenna elements 602, and the relationship of the components may further affect the further orientation of the far-field radiation patterns. And can be adjusted until energetics are achieved. Alternatively, the far field radiation patterns can be measured using electrical equipment having a relationship of components controlled on such a base.

Returning to FIG. 6, the antenna elements 602 are supplied by a microstrip supply line, shown by dashed lines 624. The supply line in dashed line 624 is matched with the network to drive the impedance and is based on the dielectric material used. The symmetry of the supply lines is also important to avoid unnecessary phase delays that result in far-field radiation patterns generated by antenna elements that are removed instead of additives.

6, the embodiment uses a common synthesizer / separator 626 to separate the incoming signal into two to supply two sets of antenna elements and combine the return signals. A second and third combiner / separator 628 then separates the return signals into two to supply each pair of antenna elements 602 and to combine the return signals. The synthesizers / separators 626, 628 are preferred because they form a near perfect impedance match along the supply lines over a wide frequency range and also prevent power from being returned along the supply line resulting in performance loss. Do.

FIG. 8 illustrates a bottom layer 800 of an antenna 600 comprising elements 802, 812, 814, 816, each of which is a trapezoidal element 804, choke joint. ) Region 806 and laser 808. Elements 802, 812, 814, 816 may also set the phase angle of the antenna 600 by reflecting the signal or RF energy to the bottom of the bridge element 820, even though the elements 812, 814 reflect the signal or RF energy to the bottom of the bridge element 820. , Acts as capacitors. The distance 826 from the bottom of the trapezoidal elements 804 to the bottom of the bridge element 820 is 1/4 if a spherical shape for the derived pattern generated by the antenna 600 is desired. It cannot be larger than the wavelength. By varying the distance 826 for each of the elements 802, 812, 814, 816, other types of radiation patterns can be formed. Finally, cutting elements 822, 824 are provided where trace materials are removed from the bottom left corner and bottom right corner of the bridge element 820 to prevent reflection of the elements 802, 816. As a result, the phase angle set by the elements 812 and 814 is changed.

The trapezoidal elements 804 hold the magnetic loop 604 of each corresponding antenna element 602 due to the fact that each trapezoidal element 804 is dimensionally log driven. Add inductance and capacitance in which the slope of each trapezoidal element 804, in particular the slope of the upper side of the trapezoidal element 804, is changed to match the inductive reactance to the capacitive reactance of the antenna 600. It is used to By adding capacitance through the trapezoidal elements 804, the electrical length of each corresponding magnetic loop 604 on the other side of the antenna 600 can be adjusted. The trapezoidal elements 804 are aligned with the upper trace 614 of the magnetic loop 604 on the other side of the antenna 600. The choke connectors 806 act to insulate the trapezoidal elements 804 from ground, thus preventing leakage of the derived signal. Sides 809 and 810 of the trapezoidal elements 804 are counterpoised to electric field radiators 606 on the other side of the antenna 600, which need to be grounded to establish polarization. The side 809 consists of a right side of the trapezoidal elements 804 and an upper right portion of the laser 808 overlying the choke connector 806. That is, the side 809 consists of the right side of each element 802, 812, 814, 816 overlying the choke connector 806. Side 810 is comprised of the left side of the trapezoidal elements 804 and the left side of the laser 808. That is, the side 810 consists of the left side of each element 802, 812, 814, 816 overlying the ground plane element 828. The counterpoises 809 and 810 increase the transmit / receive efficiency of the antenna 600. The ground plane element 828 becomes the standard for microstrip antenna designs, for example, with a 50 ohm trace on a 4.7 dielectric being about 100 mils wide.

As described above, the trapezoidal elements 804 may be fine tuned to change the inductance of the corresponding magnetic loop or to change the capacitance. The fine tuning process includes a process of shrinking or enlarging sections of the trapezoidal elements 804. For example, it may determine if additional capacitive reactance is needed to match the inductive reactance of the magnetic loop. Thus, the trapezoidal elements 804 can be enlarged to increase the capacitance. An alternative fine tuning step is to change the slope of the trapezoidal elements 804. For example, the inclination may vary from a 15 degree angle to a 30 degree angle. Alternatively, if the magnetic loop 604 is changed by increasing its area or by reducing the width of the upper trace 614 of the magnetic loop 604, corresponding to the modified magnetic loop 604 The metal on the ground plane must also be adjusted accordingly. For example, the upper side of the trapezoidal element 804 or the overall length of the trapezoidal element 804 is reduced or increased based on whether the upper trace 614 of the magnetic loop 604 is decreased or increased. Can be.

Simultaneous excitation of the TM and TE radiators, as described above, results in zero reactive power expected by the time dependent pointing theorem when used to analyze microwave energy. Previous attempts to build composite antennas with TE and TM radiators that are electrically perpendicular to each other have relied on the three-dimensional arrangement of these devices. Such a design cannot be easily commercialized. In addition, previously proposed composite antenna designs have been supplied with separate power sources located at two or more locations in each loop. In embodiments for the various antennas as described above, the magnetic loop and the electric field radiator (s) are still coplanar and supplied with power from a single location, such that they are 90/270 electrical angles to each other. Located in This results in a two-dimensional arrangement that can reduce the complexity of the physical arrangement and also enhance commercial viability. Alternatively, the electric field radiator (s) may be located on the magnetic loop at the point where the current flowing through the magnetic loop is at the reflective minimum.

Embodiments of the antennas described herein have greater efficiency than conventional antennas that partially cancel reactive power. Embodiments also have large antenna openings for their respective physical sizes. For example, a half-wave antenna having an omnidirectional pattern according to an embodiment may have a gain significantly greater than the general 2.11 dBi gain of simple magnetic dipole antennas.

Another embodiment consists of a cross-sectional antenna with a built-in counterpoise for the electric field radiator. 9A illustrates an embodiment for a 2300 to 2700 MHz cross-sectional antenna having a single electric field radiator and a built-in counterpoise for the electric field radiator. The antenna 900 is composed of a magnetic loop 902 with an electric field radiator 904 coupled directly to the magnetic loop 902 without the gain of an electrical trace. The electric field radiator 904 is physically located on the inside of the magnetic loop 902. As in other embodiments, the electric field radiator 904 may be coupled to the magnetic loop 902 at a 90/270 connection point or at a point where the current flowing through the magnetic loop 902 is at a reflective minimum. . In an alternative embodiment, the electric field radiator 904 may be coupled to a magnetic loop 902 having an electrical trace. In addition, although the antenna 900 has been described as one electric field radiator, alternative embodiments may include one or more electric field radiators. Alternative embodiments may also include one or more electric field radiators that are physically located on the outside of the magnetic loop 902.

Alternative embodiments of a self-supporting antenna may also include a first electric field radiator having a first length, and a second electric field radiator having a second length different from the first length. As with the antenna embodiments described above, broadband antennas are enabled by using one or more electric field radiators of different lengths.

The antenna 900 includes a transition portion 906 and a counterpoise portion 908 for the electric field radiator 904. The transition portion 906 is composed of a portion of the magnetic loop 902 having a width larger than the width of the magnetic loop 902. The transition part 906 electrically insulates the built-in counterpoise part 908. The built-in counterpoise portion 908 allows the antenna 900 to be completely independent of the chassis or any ground plane of the product using the antenna 900.

The counterpoise portion 908 is referred to as being built-in because the counterpoise portion is formed from the magnetic loop 902. As described, the built-in counterpoise portion 908 allows the antenna 900 to be completely independent from the ground plane of the product. The embodiments of the cross-sectional antenna shown in FIGS. 4A-4C print only on a single plane and do not include a ground plane, but require a ground plane to be provided by a device using the antenna. In contrast, self-contained counterpoise antennas do not require a ground plane to be provided by a device using the antenna.

In the cross-sectional embodiments described above, a device using an antenna may be used as the ground plane for the cross-sectional antenna by using the chassis of the device or some other metal component, or as a ground plane for the cross-sectional antenna. Together with the ground plane of the device that acts, it provides a ground plane for the antenna. However, any change to the electrical circuitry of the device, either to the chassis of the device or to the ground plane of the device, can adversely affect the performance of the antenna. This phenomenon is not specific to the cross-sectional embodiments described herein, but instead applies to antennas widely used in research and commercial fields. Thus, it is desirable to have an antenna that does not require a ground plane that will not be affected by any changes made to the device using the antenna.

By not requiring a ground plane, the antenna 900 does not depend on the ground plane outside the antenna. The independence of the self-sustaining antenna 900 from this external ground plane means that the performance of the antenna is not affected by changes made in the device. This indicates from the standpoint of manufacture and design that a self-sustaining antenna can be designed for a particular frequency and that the performance level means incorporating and using the antenna independently of the device. For example, a wireless router maker requests a specific antenna based on set of requirements. Such requirements may include, among other requirements, the space available for the antenna, the frequency range for the antenna, and the substrate to be used. At that time, the design and manufacture of the antenna may be performed independently of the design and manufacture of the actual wireless router. Also, any further changes to the wireless router do not affect the performance and efficiency of the antenna as the antenna is self-sufficient, and also changes to the electrical network of the router, the ground plane of the router, or the chassis of the router. Are not affected by them.

The length of the transition unit 906 may be set based on the operating frequency of the antenna. For high frequency antennas with short wavelengths, short transitions can be used. In contrast, for a low frequency antenna having a long wavelength, the long transition portion 906 may be used. The transition unit 906 may be adjusted independently of the counterpoise unit 908. For example, the transition portion for a 5.8 GHz antenna may be only half the size of the transition portion 906 of FIG. 9A, while the counterpoise portion 908 is still at the full left side of the magnetic loop 902. Will be together.

The length of the counterpoise portion 908 may be adjusted according to the need for obtaining a predetermined antenna performance. However, it is desirable to have the counterpoise portion 908 as large as possible. For example, in an alternative embodiment, the counterpoise portion 908 may be less than about 80% of the left side of the magnetic loop 902, rather than the entire length of the left side of the magnetic loop 902. Can wear. However, the width of the trace of the magnetic loop 902 as described above affects the electrical length of the magnetic loop 902. A magnetic loop having a thin trace all around the magnetic loop is electrically longer than a magnetic loop having portions of the magnetic loop having a wide or wide trace. For example, the magnetic loop 902 is an example of a magnetic loop having a wide trace for the transition portion 906 and the counterpoise portion 908. Thus, it is suitable to have a counterpoise portion as long as possible, but the length of the counterpoise portion 908 affects the electrical length of the magnetic loop 902. Electrically long magnetic loops result in lower frequencies. In contrast, electrically short magnetic loops result in higher frequencies. For example, using a counterpoise over the entire length of the left side of the magnetic loop will increase the overall width of the magnetic loop, so that the magnetic loop will be electrically short, causing the magnetic to have a higher frequency than expected. Results in a loop. For example, a frequency of 5.8 GHz is formed instead of a predetermined target frequency of 5.6 GHz.

Embodiments of the antenna 900 also include a narrow-light- narrow transition portion as described above, separately from the transition portion and counterpoise portion, for tuning the electrical length of the magnetic loop to a predetermined frequency. And / or light-narrow-light transitions. In addition, embodiments of self-supporting antennas may also include magnetic loops with corners cut at an angle as described above to improve current flow around corners of the magnetic loops.

As described above, the counterpoise portion 908 for the electric field radiator 904 may be used instead of the ground plane. The electric field radiator 904 is substantially a monopole antenna. The monopole antenna is formed by replacing one half of the dipole antenna with one half having a ground plane perpendicular to the other half. In embodiments of the self-sustaining antennas, the electric field radiator has a large piece of metal electrically connected to an electric field radiator that can be used instead of the ground plane. In the cross-sectional antenna 440 from FIG. 4C, the electric field radiator 444 emits an electric field based on the position of the ground plane used for the antenna 440. This electric field rotates at right angles to the plane of the electric field radiator, while the magnetic field rotates in a substantially coplanar manner with the plane. This pattern of electric field has a generally donut shape, referred to as an almost complete omnidirectional pattern. As discussed above, embodiments of cross-sectional antennas do not necessarily provide their own ground plane. Thus, if the antenna 440 is used in a device, the device will act as the ground plane of the antenna 440 and the radiation pattern radiated by the electric field radiator 444 will be returned into the device. However, if the cross-sectional self-supporting antenna comprising the counterpoise portion also comprises a ground plane, the above-mentioned radiation patterns are either in the electric field rotating with respect to the plane of the electric field radiator or in the same plane as the plane of the electric field radiator. The above-described planes can be effectively converted into a magnetic field that rotates at right angles to the plane.

The counterpoise portion 908 need not be located on or machined above the upper left corner of the magnetic loop 902. In alternative embodiments, the counterpoise portion may be located on an upper right corner, along with an electric field radiator 904 located continuously on the left side of the magnetic loop 902. Regardless of the physical positions of the counterpoise portion 908 and the electric field radiator 904 (or radiators if one or more), the counterpoise portion and the electric field radiator (s) need to be located 180 degrees from the phase. . In another embodiment, the length of the counterpoise can also be adjusted as needed. The counterpoise 908 may also be located along the right side of the magnetic loop 902, just below the electric field radiator 904 or at other locations around the magnetic loop 902.

The antenna 900 also includes a balloon 910. A balun is a type of electrical transformer that can convert balanced electrical signals to ground (difference) into unbalanced (cross-sectional) signals, or vice versa. In particular, the balun exhibits high impedance as common mode signals and low impedance as other mode signals. The balloon 910 serves to cancel a common mode current. The balloon 910 also tunes the antenna 900 to a predetermined input impedance and also tunes the impedance of the entire magnetic loop 902. The balloon 910 has a generally triangular shape and consists of two parts divided by an intermediate gap 912.

Two portions of the balloon 910 are magnetically and electrically coupled. The gap 912 of the balloon 910 removes common mode current by magnetically preventing current from flowing in one direction, such as backflowing through the transmitter to a device using the antenna 900. This is important because the reflection of the current flows through the transmitter and, due to the common mode current, adversely affects the performance of the antenna 900 or of the device using the antenna 900. In particular, reflection of current through the transmitter causes radio interference in the electrical network of the device using the antenna. Such negative performance can also prevent the device from meeting Federal Communications Commission (FCC) regulations. The gap 912 of the balloon 910 eliminates common mode current, thus preventing current from returning into the connector of the antenna 900.

The gap 912 can be adjusted based on the antenna design and dimensions. In an embodiment, electromagnetic simulation can be used to visualize the current flowing through the antenna 900. The gap 912 may then be increased or decreased until the simulation shows that the current is no longer reflected and also returns through the transmitter. The removal of the common mode current flows in the opposite direction, in which the current flowing in one direction into the transmitter stops and also has one direction flowing into the antenna 900 and a second direction flowing out of the antenna 900. It can be visualized as the starting point of the current.

The tapered sides 914 of the balloon 910 serve for electrical coupling. The angle of the tapered sides 914 may be adjusted to impedance match the antenna 900. Typically an individual inductor and an individual capacitor are arranged along the supply line (not shown) to match the impedance of the device supplying the antenna 900 to the impedance of the antenna 900. Is located in. For example, if an antenna requires an input of 50 ohms but the electrical network of the device supplies 150 ohms to the antenna, such a mismatching problem may be achieved by modifying the 150 ohms supplied to the antenna to the 50 ohms required by the antenna. A series of inductors and capacitors are used to balance the voltage. Contrary to this common practice in the industry, embodiments of self-supporting antenna 900 may employ some external components, for example by using a series of inductors and capacitors along the line supplying the antenna 900. There is no need for impedance matching. Instead, the balun 910 is used to match the impedance of the antenna 900 to the connector supplying the antenna 900 and also to match the impedance of the magnetic loop 902.

The height of the balloon 910 is a function of the operating frequency of the antenna 900. Thus, the larger balun 910 requires a lower frequency, while the short balun 910 requires a higher frequency. When using a large balun for an antenna, it is important to bring the balun 910 close to the electric field radiator (s). Positioning the balloon 910 too close to the electric field radiator 904 can form a capacitive coupling between the balloon 910 and the electric field radiator 904. Thus, it is important that the balun 910 is properly spaced from the electric field radiator 904 to prevent capacitive coupling from affecting the antenna 900 performance. If a particular antenna design requires the use of a large balun according to the operating frequency of the antenna, the balun is described as described in antenna 920 of FIG. 9B to properly impedance match the antenna and also remove common mode current. You can move it down. In alternative embodiments, the self-supporting counterpoise antenna 900 may not include the balun 910.

The antenna 900 is an example of a self-supporting counterpoise composite field antenna. Embodiments of the antenna 900 may be printed or otherwise stacked on about 1.6 mm FR-4 substrates. The characteristics and design of the antenna 900 also make it applicable to other materials, including flexible printed circuits, acrylonitrile butadiene styrene (ABS) plastics, and even materials that cannot be considered suitable for microwave frequencies. The operating frequency of the antenna 900 is about 2300-2700 MHz, suitable for a variety of embedded applications including mobile phones, access points, PDAs, laptops, PC-cards, sensors, and automotive applications. Done. Embodiments of the antenna 900 have achieved a peak efficiency of about 94% and a peak gain of about +3 dBi. The antenna 900 has a width of about 31 mm and a length of about 31 mm. The antenna 900 has linear polarization and an impedance of about 50 Ω. The antenna 900 also has a voltage standing wave ratio of less than two to one (<2: 1). The size and efficiency of the antenna 900 is suitable for Wi-Fi applications, where efficiency, size and gain are important.

Another embodiment of a cross-sectional antenna with built-in counterpoise is described in FIG. 10A. Antenna 1000 is one example of an antenna having linear polarization. The antenna does not require a ground plane because of the built-in counterpoise. The antenna 1000 may be printed or otherwise stacked on a 1.6 mm thick FR-4 substrate. As with the antenna 900, the characteristics and design of the antenna 1000 may be applied to other materials, including flexible printed circuits, acrylonitrile butadiene styrene (ABS) plastic, and even materials that cannot be considered suitable for microwave frequencies. Make it adaptable. The antenna 1000 operates with a peak efficiency of about 92% and a measured peak gain of about +3 dBi in the frequency range of about 882 MHz to 948 MHz. The antenna 1000 has an antenna impedance of about 50 Ω and has a voltage standing wave ratio of less than 2 to 1 (<2: 1). The antenna 1000 has a width of about 76 mm and a height of about 76 mm.

The antenna 1000 has a first electric field radiator 1004 coupled directly to the magnetic loop 1002 and a second electric field radiator 1006 coupled directly to the magnetic loop 1002. It is composed of Both field radiators 1004 and 1006 are coupled to the magnetic loop 1002 without the benefit of an electrical trace. The electric field radiators 1004 and 1006 are physically located on the inside of the magnetic loop 1002. By using two electric field radiators instead of one, as in the antenna 900 shown in FIG. 9, the gain of the antenna is increased. Curved line 1008 separating the two electric field radiators 1004 and 1006 delays the phase between the two electric field radiators 1004 and 1006 to form a farfield patterns additive. To provide the ability to

The two electric field radiators 1004 and 1006 form an electric field radiator array 1010 having a phase delay with the curve line 1008. In particular, the curved line 1008 ensures that the two electric field radiators 1004 and 1006 are 180 degrees out of phase with each other. The curved line can be used as a space saving technique. For example, if a small antenna is needed, the two electric field radiators are forced to close together due to the need for a minimum size, with the curved line 1008 being consistently different from each other by the field radiators. It can be used to take a phase. The electrical length of the trace of the curve line 1008 can be adjusted as needed based on the required delay. For example, the trace can be formed short or long while the width remains constant. Alternatively, the length of the trace can be kept constant while the trace is wide or thick. As mentioned above, the electrical length of the trace depends on its physical length and physical width. 10B illustrates an alternative embodiment of a self-supporting antenna 1020 without the curved line 1008.

As described with respect to the antenna 900, the antenna transition portion 1012 and the counterpoise 1014 may be adjusted accordingly based on a plurality of factors. The transition portion 1012 is based on the operating frequency, but it must also be long enough that the counterpoise 1014 is electrically isolated. As large a counterpoise 1014 as possible is suitable. Finally, the balun 1016 removes common mode current and also matches the impedance of the antenna 1000 with the impedance of the transmitter supplying the supply antenna 1000.

In alternative embodiments of the antenna 1000, the electric field radiator array 1010 may be arranged on the left side of the antenna 1000 instead of the right side. In such alternative embodiments, the counterpoise 1040 may be located on the upper right side of the magnetic loop 1002. The counterpoise 1014 may also be located directly below the electric field radiators 1004 and 1006 along the right side of the magnetic loop 1002.

11A-11C illustrate a 2D radiation pattern for the antenna 900 of FIG. 9. 11A illustrates the 2D radiation pattern on the XZ plane 1100. The solid line 1102 represents the actual radiation pattern, the dashed line 1104 represents the 3 dB beamwidth, and the dashed line 1106 represents the maximum intensity of the electric field along one direction, i.e. the dashed line 1106 represents the 2D radiation pattern described. Indicates the location where the strongest electric field is detected. FIG. 11B illustrates the 2D radiation pattern for the antenna 900 on the XY plane 1110, and FIG. 11C illustrates the 2D radiation pattern for the antenna 900 on the YZ plane 1120.

12A-12C illustrate a 2D radiation pattern for the antenna 1000 of FIG. 10A. 12A illustrates a 2D radiation pattern on the XZ plane 1200. The solid line 1202 represents the actual radiation pattern, the dashed line 1204 represents the 3 dB beamwidth, and the dashed line 1206 represents the maximum intensity of the electric field along one direction, i.e. the dashed line 1206 represents the 2D radiation pattern described. Indicates the location where the strongest electric field is detected. 12B illustrates the 2D radiation pattern for the antenna 1000 on the XY plane 1210, and FIG. 12C illustrates the 2D radiation pattern for the antenna 1000 on the YZ plane 1220.

13A illustrates the voltage standing wave ratio VSWR for the antenna 900. The VSWR configuration shows that the antenna 900 has good impedance matching for the frequency range of about 2.34 GHz to about 2.69 GHz. That is, above the frequency range of about 2.34 GHz to 2.69 GHz, most of the energy supplied into the antenna 900 is radiated rather than returned into the transmitter. In particular, the two center vertical solid lines inside represent a frequency range in which the VSWR of the antenna 900 is less than 2: 1 (<2: 1). 13B illustrates the return loss for antenna 900. Return loss and VSWR are mathematically related such that -10.0 return loss on FIG. 13B corresponds to VSWR of 2.0 on FIG. 13A. The return loss diagram of FIG. 13B shows that the antenna 900 is well impedance matched between points classified as 1 and 2. FIG.

FIG. 14A illustrates the voltage standing wave ratio VSWR for the antenna 1000 of FIG. 10A. The VSWR configuration shows that the antenna 1000 is good impedance matched for a frequency range of about 884 MHz to about 947 MHz. That is, above the frequency range of about 884 MHz to 947 MHz, most of the energy supplied into the antenna 1000 is radiated rather than returned into the transmitter. In particular, the two center vertical solid lines inside represent a frequency range in which the VSWR of antenna 1000 is less than 2: 1 (<2: 1). 14B illustrates the return loss for the antenna 1000. As described above, the -10.0 return loss corresponds to a VSWR of 2.0. The return loss diagram of FIG. 14B shows that the antenna 1000 is well impedance matched between points classified as 1 and 2. FIG.

15 illustrates another embodiment of a self-supporting antenna 1500. The antenna 1500 is an example of a 5.8 GHz antenna. The dimensions of certain embodiments of the antenna 1500 are about 15 mm long and about 15 mm wide. The antenna 1500 is composed of a magnetic loop 1502, with an electric field radiator 1504 coupled directly to the magnetic loop 1502. In contrast to self-contained antennas 900 and 1000, the antenna 1500 includes a tapered transition 1506 consisting of two sections 1508 and 1510. The first transition section 1508 starts where the width of the magnetic loop varies from small to large. The first transition section 1508 tapers linearly toward a smaller width before the width of the magnetic loop increases again at the location where the second transition section 1510 begins. The second transition section increases linearly from small to large. As described above, by adjusting the width of the trace of the magnetic loop, the electrical length of the magnetic loop is allowed to be adjusted. In addition, the length, width, and number of the transition portions used electrically insulate the counterpoise 1512. The transition unit 1506 should be long enough to minimize the magnitude of the current flowing through the counterpoise 1512. Also, by tapering the transition portion 1506 relative to the counterpoise 1512, the bandwidth is increased in terms of impedance matching. The balloon 1514 removes common mode current and also matches the impedance of the antenna 1500. An alternative embodiment of the antenna 1500 may not include the balun 1514.

Embodiments include a magnetic loop having a width, located on a plane generating a magnetic field, and having a first inductive reactance; An electric field radiator located on a plane that emits an electric field and having a first capacitive reactance and directly coupled to a magnetic loop, the electric field is orthogonal to the magnetic field and the physical arrangement between the electric field radiator and the magnetic loop is first Said electric field radiator forming a capacitive reactance; A transition portion formed on the magnetic loop and having a transition width greater than the width of the magnetic loop; And a counterpoise formed on the magnetic loop located along the magnetic loop opposite or adjacent to the electric field radiator, wherein the transition portion comprises the counterpoise to electrically insulate the counterpoise from the magnetic loop. Type antenna.

Unless expressly stated otherwise, each feature described herein (including the appended claims, abstract and drawings) may be replaced by alternative features provided for the same, equivalent, or similar purpose. Thus, unless expressly stated otherwise, each feature described is one example of a series of generic equivalent or similar features.

While the invention has been described and illustrated herein in some alternative respects, it should be understood that the techniques described above may have many additional uses and applications. Accordingly, the present invention is not limited to the specific descriptions, embodiments, and various drawings contained in the present application that merely describe exemplary embodiments, but alternatives and applications to the principles of the present invention are possible.

Claims (38)

delete delete delete delete delete delete delete delete delete delete delete delete As a cross section antenna,
A magnetic loop located on a plane and configured to generate a magnetic field; And
An electric field radiator located on the plane and within the magnetic loop, the electric field radiator is configured to emit an electric field coupled to the magnetic loop and orthogonal to the magnetic field, the amplitude of the surface current on the magnetic loop being minimal At which point the electric field radiator is coupled to the magnetic loop, and the point is based on a change in geometry of the magnetic loop based on impedance matching between the feed for the cross-sectional antenna and the cross-sectional antenna. A moveable, including the electric field radiator,
The electric field radiator has an electrical length and is configured to emit the electric field at the frequency of operation and further comprises a second electric field radiator located on the plane and in the magnetic field, the second electric field radiator being connected to the magnetic loop. Coupled, the second electric field radiator is configured to emit a second electric field orthogonal to the magnetic field, and the second electric field radiator is configured to emit the second electric field at a second frequency of operation with a second electrical length , Cross-sectional antenna. delete delete As a flat antenna,
One or more magnetic loops located on the first plane and configured to generate one or more magnetic fields;
One or more electric field radiators located on the first plane that emit one or more electric fields orthogonal to the one or more magnetic fields, wherein each electric field radiator of the one or more electric field radiators is a magnetic one of the one or more magnetic loops; One or more electric field radiators coupled to a loop; And
A wideband element located on a second plane and configured to form a ground plane, the wideband element having a total inductive reactance coupled with a wide bandwidth based on one or more physical adjustments to the wideband element. And the wideband element configured to be capable of matching a total capacitive reactance. The method of claim 16,
Further comprising one or more phase trackers, wherein each phase tracker of the one or more phase trackers is coupled to a respective magnetic loop of the one or more magnetic loops. The method of claim 17,
Wherein each phase tracker is physically located inside each magnetic loop or physically located outside each magnetic loop. 19. The device of claim 17 or 18, wherein each of the phase trackers has a triangular shape, each of the magnetic loops further comprises a driving point, and the tip of the triangular shaped phase tracker is 90 degrees from the driving point. And an electrical angle position, an electrical angle position of 270 degrees from the drive point, and a position selected from the group consisting of an amplitude minimum point at which the amplitude of surface current on each magnetic loop is minimum. The method according to any one of claims 16 to 18,
And the wideband element comprises one or more trapezoidal elements. The method of claim 20,
The one or more physical adjustments include varying the slope of the top side of each trapezoidal element. The method of claim 20,
The broadband element comprises one or more choke joints and a ground element, wherein the one or more choke connectors are configured to isolate the one or more trapezoidal elements from the ground element. 19. The planar antenna of any one of claims 16-18, wherein the one or more magnetic loops have a rectangular shape. delete delete delete delete delete delete delete delete delete delete delete delete delete delete delete

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