JP5916019B2 - Compound loop antenna - Google Patents

Description

[Cross-reference of related applications]
This application enjoys priority from U.S. Application Nos. 12 / 878,016, 12 / 878,018 and 12 / 878,020 filed September 8, 2010, and these applications are This is a non-provisional application that enjoys priority from US Provisional Application No. 61 / 303,594, filed on Feb. 11, 2010.
Embodiments of the present invention relate to planar (both sides) and printed (single sided) composite area antennas, and more particularly, but not exclusively, to a composite loop antenna, and the composite loop The antenna has a co-planar electric field radiator and magnetic loop with an electric field perpendicular to the magnetic field, performance benefits for higher bandwidth (low Q), greater line strength / capacity / gain As well as greater efficiency.
A further embodiment relates to an independent counterpoise composite area antenna including transitions formed on the magnetic loop and having a transition width greater than the width of the magnetic loop.
The transition substantially separates the counterpoise formed on the magnetic loop opposite or adjacent to the electric field radiator.

The ever-decreasing size of modern telecommunication devices creates the need for improved antenna design. Known antennas in devices such as mobile / cell phones provide one of the main limitations in performance and almost always compromise in either way.

The efficiency of the antenna can have a major influence on the performance of the device in particular. A more efficient antenna will radiate the energy supplied from the transmitter at a higher rate. Similarly, due to the inherent interaction of the antennas, a more efficient antenna will convert more of the received signal into electrical energy for processing by the receiver.

To ensure maximum transfer of energy (to both transfer and receive modes) between the transceiver (device acting as both transmitter and receiver) and antenna, both impedances match each other in magnitude Should do. Any incorrect combination between the two will result in sub-optimal performance, and when transmitting, energy is reflected back from the antenna to the transmitter. When operating as a receiver, antenna sub-optimal performance results in lower received power than otherwise possible.

Known simple loop antennas are typically currently supplied devices. It primarily creates a magnetic (H) region. As such, they are typically not as suitable as transmitters. This is especially true for small loop antennas (ie, having a smaller or less than one wavelength diameter). In contrast, a voltage supply antenna such as a dipole generates both an electrical (E) field and an H field and can be used in both transmit and receive modes.

The amount of energy received by or transmitted from the loop antenna is measured, in part, by that portion. Typically, each time the portion of the loop is halved, the amount of energy that can be received / transmitted is reduced by approximately 3 dB, depending on the application parameters such as initial size, frequency, etc. This physical constraint tends to mean that very small loop antennas cannot be used in implementation.

Composite antennas achieve transverse performance (TM mode and transverse electrical (TE) mode higher performance benefits such as higher bandwidth (low Q), greater line strength / capacity / gain and greater efficiency) To be activated.

At the end of the 1940s, Wheeler and Chu first studied the characteristics of an electrically short (ELS) antenna. Through their work, several equations were created to describe the limitations of the antenna that would decrease in physical size. One of the limitations of ELS antennas mentioned by Wheeler and Chu (which is of particular importance) stores a large radiation quality factor in that it stores a larger time average energy than radiates. Having Q. According to Wheeler and Chu, ELS antennas have a high radiation Q, resulting in the least resistive damage of the antenna or matching network, typically very low radiation efficiency between 1-50% Leads to. As a result, since the 1940s, it has generally been accepted by the scientific community that ELS antennas have a narrow bandwidth and poor radiation efficiency. Much of the current achievements in wireless communication systems utilizing ELS antennas have arisen from precise experimentation and optimization of modulation schemes and over air protocols. However, commercially used ELS antennas still reflect narrow bandwidth and low efficiency, resulting in wheelers and chews.

In the early 1990s, Dale M. Grimes and Craig A.M. Grimes claimed to have certain combinations of mathematically found TM and TE modes that work together in ELS antennas that exceed the low emission Q limits established by Wheeler and Chu theory. Grimes and Grimes describe their work in a journal entitled “Bandwidth and Q of Antenna Radiating TE and TM Models” published in the IEEE transaction of the Electromagnetic Compatibility Group in May 1995. These requests sparked much debate, and if either TM mode or TE mode is activated alone, both TM mode and TE mode are activated as opposed to "simple field antennas". The term “composite field antenna” led. The benefits of multiple area antennas have been mathematically proven by several reputable RF experts, including a group employed by the Weapon Division of the US Naval Wear Fair Centre. They concluded that evidence of a radiation Q below the Wheeler-Chu limit, ie increased radiation intensity, directivity (gain), radiated power, and radiated efficiency (PL Overelf, DR Bowling, DJ White, “Collocated Magnetic Loop, Electric Dipole Array Antenna (Preliminary Results)”, September 1994).

Complex field antennas are complex and difficult to physically implement due to the undesired effects of element connections and the associated difficulties in designing low-loss passive networks to combine electrical and magnetic radiators I understood.

There are many examples of two-dimensional non-composite antennas, but they generally consist of a printed strip of metal on a circuit board. However, these antennas are voltage supply type. An example of such an antenna is a planar inverted F antenna (PIFA). The majority of similar antenna designs also consist mainly of quarter wavelength (or multiple of quarter wavelength), voltage supply, and dipole antenna.

Planar antennas are also known in the art. For example, Patent Document 1 issued to Tsahn et al. Requires an expensive Teflon substrate or similar material for the antenna to operate. Patent Document 2 issued to Shiga teaches a planar antenna that can receive a microwave signal, but does not transmit it. Shiga antennas require more expensive semiconductor substrates. U.S. Pat. No. 6,057,038 issued to Nalbandian relates to a planar antenna, requires a substrate having a dielectric constant for a permeability of 1: 3 to 1: 1, and has a frequency range of 3 HF and VHF (3-30 MHz and 30 Only 300 MHz). Although it is known to print some low frequency devices on an inexpensive glass reinforced epoxy laminate sheet such as FR-4, it is commonly used for normal printed circuit boards, and FR-4 The dielectric loss at is considered to be too high, and the dielectric constant is not sufficiently controlled for such a substrate to be used at microwave frequencies. For these reasons, alumina substrates are more commonly used. In addition, none of these planar antennas are composite loop antennas.

The basis for the increased performance of composite field antennas stems from the effect of energy stored in the near area of the antenna, due to bandwidth, efficiency, gain and radiation strength. In RF antenna design, it is desirable that the energy imparted to the antenna be transferred as much as possible during the radiated power. The energy stored in the near area of the antenna has historically been called reactive power and serves to limit the amount of ability that can be radiated. When discussing complex abilities, there are real and imaginary parts (often referred to as “invalid”). Real power leaves the source and does not return. However, imaginary or reactive power tends to swing with respect to the fixed position (within half wavelength) of the source and interacts with the source, thereby affecting antenna operation. The presence of real power from multiple sources is directly additive. However, multiple sources of imaginary capabilities can be additive or negative (cancelled). The benefits of composite antennas allow engineers to create designs that utilize reactive power cancellation that was not previously available in simple field antennas, or be driven by both TM and TE sources, It is to improve the actual power transmission quality of the antenna.

In order to be able to cancel reactive power in the composite antenna, the electric and magnetic fields should operate at right angles to each other. Many configurations of electric field radiators necessary to generate a magnetic field, as well as many configurations of magnetic loops necessary to generate a magnetic field, have been proposed, but all such designs must be made with a three-dimensional antenna. It has been resolved. For example, U.S. Patent No. 6,057,028 issued to MacLaine requires a set of magnetic loops in a plane parallel to an electric dipole on a third parallel plane disposed between the sets of magnetic loops. U.S. Pat. No. 6,057,031 issued to Grimes et al. Requires two sets of magnetic loops and electric dipoles to be physically aligned at right angles to each other. U.S. Pat. No. 6,057,096 filed by McLane teaches the arrangement when the magnetic and electric dipoles are also in orthogonal planes.

US Pat. No. 5,061,938 US Pat. No. 5,376,942 US Pat. No. 6,677,901 US Pat. No. 7,215,292 US Pat. No. 6,437,750 US Patent Application Publication No. 2007/0080878

FIG. 1 illustrates the achievement of the plane of the embodiment. FIG. 2 shows a circuit arrangement for an embodiment incorporating four discrete antenna elements. FIG. 3A shows a detailed view of one of the antenna elements that does not include the phase tracker of FIG. FIG. 3B shows a detailed view of one of the antenna elements that does not include the phase tracker of FIG. FIG. 4A shows an embodiment of a small single-sided composite antenna. FIG. 4B shows an embodiment of a small single-sided composite antenna with a magnetic loop, with the corners of the magnetic loop being cut at an angle of approximately 45 degrees. FIG. 4C shows an embodiment of a small single-sided composite antenna with a magnetic loop having two symmetric wide-narrow-wide transitions. FIG. 5 shows an embodiment of a small double-sided composite antenna. FIG. 6 shows an embodiment of a large composite antenna array composed of four composite antenna elements. FIG. 7 shows how the dimensions of the phase tracker affect its inductance and capacitance; FIG. 8 shows the ground plane of the antenna embodiment. FIG. 9A shows a counterpoise antenna antenna embodiment with an independent balun. FIG. 9B shows an alternative embodiment of the antenna of FIG. 9A with a pull-down balun. FIG. 10A shows an embodiment of an independent counterpoise antenna with a curved trace between the array of electric field radiators and the electric field radiator. FIG. 10B shows an embodiment of an independent counterpoise antenna with an array of electric field radiators but no bent traces. FIG. 11A approximately shows a 2D radiation pattern for the antenna from FIG. FIG. 11B approximately shows the 2D radiation pattern for the antenna from FIG. FIG. 11C approximately shows the 2D radiation pattern for the antenna from FIG. FIG. 12A approximately shows a 2D radiation pattern for the antenna from FIG. 10A. FIG. 12B approximately shows the 2D radiation pattern for the antenna from FIG. 10A. FIG. 12C approximately shows a 2D radiation pattern for the antenna from FIG. 10A. FIG. 13A approximately shows a plot of voltage standing wave ratio for the antenna from FIG. 9; FIG. 13B approximately shows a plot of the return loss measured for the antenna from FIG. FIG. 14A approximately shows a plot of voltage standing wave ratio for the antenna from FIG. 10; FIG. 14B approximately shows a plot of the return loss measured for the antenna from FIG. FIG. 15 schematically illustrates an embodiment of an independent counterpoise antenna with a tapered transition.

Embodiments are improved planar composite loop (CPL) antennas that can operate in both transmit and receive modes and allow greater performance than known loop antennas. A composite loop (CPL) antenna. The two main components of the CPL antenna are a magnetic loop that generates a magnetic field (H region) and an electric field radiator that emits an electric field (E region).

The electric field radiator may be physically located inside the loop or outside the loop. For example, FIG. 1 shows an embodiment of a single CPL antenna element with an electric field radiator placed inside a loop connected by electrical traces, while FIGS. 3A and 3B are placed outside the loop. Figure 2 shows two embodiments of a single CPL antenna element with an electric field radiator. FIG. 3A includes a phase tracker for broadband applications, as described further below, while FIG. 3B does not include a phase tracker and is more suitable for less broadband applications. 4A, 4B and 4C show another embodiment of a small single-sided antenna with an electric field radiator disposed in the magnetic loop. Built-in antenna embodiments using any of these technologies can be easily assembled into mobile or handheld devices (eg, phones, PDAs, laptops) or as separate antennas. FIG. 2 and other figures show an embodiment of a CPL antenna array using microstrip structure technology. Such printing technology allows a compact and consistent antenna to be designed and constructed.

The antenna (100) shown in FIG. 1 is arranged and printed on a section of the printed circuit board (101). The antenna includes a magnetic loop (110), which is in this case essentially rectangular with an open base. The two ends of the generally open base are fed from the coaxial cable (130) at the drive point in a known manner.

An electric field radiator or a series resonance circuit (120) is disposed inside the loop (110). The series resonant circuit (120) takes the form of a J-shaped trace (122) on the circuit board (101) and is connected to the loop (100) by a bent trace (124) operating as an inductor. That means having an inductance or inductive reactance. The J-shaped trace (122) has an essentially capacitive reactance property determined by the dimensions and materials used for the antenna. Trace (122) along with bent trace (124) functions as a series resonant circuit.

The antenna (100) is shown here for ease of understanding. The actual embodiment may not physically resemble the antenna shown. In this case, it is shown that it is fed from the coaxial cable (130), that is, one end of the loop (132) is connected to the central conductor of the cable (130) and the other end of the loop (134) is connected to the cable (130). Connected to the outer sheath. The loop antenna (100) differs from the known loop antenna in that a series resonant circuit (120) is connected to a loop (134) portion of a passage around the periphery of the loop. As will be described below, this connection part plays an important part in the operation of the antenna.

Perpendicular to the magnetic loop (110) by carefully positioning the series resonant circuit 120 and the bending trace (124) relative to the magnetic loop (110) generated and received by the antenna (100), E and H regions. The electric field radiators can be made perpendicular to each other without the need to physically arrange them. This right angle relationship has the effect of allowing electromagnetic waves emitted by the antenna (100) to propagate effectively over time. As a result, achieving the series resonant circuit (120) and the bent trace (124) is placed at an electrical position of approximately 90 degrees or approximately 270 degrees along the magnetic loop (110). In an alternative embodiment, the bent trace (124) may be placed at a point along the magnetic loop (110) where the current flowing through the magnetic loop is the minimum that reflects. Thus, the bent trace (124) may or may not be placed at an electrical point of approximately 90 or 270 degrees. The point along the magnetic loop (110) at which the current is reflected will depend on the geometrical point of the magnetic loop (110). For example, the point at which the current is reflected can be initially recognized as being the first region of the magnetic loop. To achieve impedance matching, after adding or removing metal from the magnetic loop, the point at which the current is reflected may change from the first region to the second region.

The magnetic loop (110) can be any one of many different electrical and physical lengths;
However, for a desired frequency band, an electrical length that is a wavelength, a quarter wavelength, and a multiple of 1/8 wavelength provides more efficient operation of the antenna. Adding inductance to the magnetic loop increases the electrical length of the magnetic loop. Adding capacitance to the magnetic loop has the opposite effect and reduces the electrical length of the magnetic loop.

The right-angle relationship between the H and E regions can be achieved by placing the series resonant circuit (120) and the bent trace (124) at a physical location that is either 90 or 270 degrees around the magnetic loop from the drive point. The physical position depends on the frequency of the signal transmitted / received by the antenna. As noted, this position can be either 90 or 270 degrees from the drive point of the magnetic loop (110) and is determined by terminals (132) and (134), respectively. Thus, if the end (132) is connected to the central conductor of the cable (130), the bent trace 124 can be placed at a 90 degree point or at a 270 degree point, as shown in FIG. Can be done.

The right-angle relationship between the H and E fields also causes the series resonant circuit (120) and the bent trace (124) to be in a physical position around one magnetic loop where the current flowing through the magnetic loop is minimized. This can be achieved by installing. As noted earlier, the position at which the current is minimized depends on the geometry of the magnetic loop (110).

By inserting circuit elements so that there is a 90 degree phase relationship between the components in this way, a right angle relationship is created between the E field and the H field, which allows the antenna (100) to receive and transmit. Can function more effectively as both antennas. The H field is generated by the magnetic loop (110) alone (or essentially alone), while the E field is emitted by the series resonant circuit (120) and is shaped to be suitable for transmission over far distances. Gives the energy transmitted from the antenna.

The series resonant circuit (120) comprises an inductive (L) element and a capacitive (C) element, the values of which are selected to resonate at the frequency of operation of the antenna (100), resulting in inductive Reactance is consistent with capacitive reactance. This is because resonance occurs most efficiently when the reactance of the capacitive element is equal to the inductive reactance. That is, when XL = Xc, the values of L and C can be selected to provide the desired operating range in this way. Other forms of series resonant circuits that use crystal oscillators can be used, for example, to provide other operating characteristics. If a crystal oscillator is used, the Q value of such a circuit will be much larger than the simple LC circuit shown and will therefore limit the bandwidth point of the antenna.

As noted earlier, the series resonant circuit (120) is effectively operating as an E-field radiator (which means that it is an E-field receiver due to the inherent nature of the antenna). As shown, the series resonant circuit (120) is a 1/4 wavelength antenna, but the series resonant circuit also operates as an antenna with multiples of all wavelengths, multiples of 1/4 wavelength, or multiples of 1/8 wavelength. Can do. If special restrictions impede the desired wavelength of the material used as trace (122), an electrically equivalent full wavelength, 1/4 wavelength, 1/8 wavelength series resonant circuit (120) is achieved. Therefore, a bent trace (124) can be used as a means to increase the propagation delay. It would be possible in theory, but not in general, so simply using a rod antenna of the desired wavelength instead of a series resonant circuit would result in the rod antenna having a 90 degree / 270 degree point or magnetic loop. It is theoretically possible if it is physically connected to the loop at the point where the current flowing through it is the minimum that is reflected, and meets the requirement of XL = Xc, but is generally not possible in practice.

As noted before, the positioning of the series resonant circuit (120) is important. That is, the series resonant circuit (120) is such that the phase difference between the E field and the H field is either 90 degrees or 270 degrees, or the current that flows through the magnetic loop is the minimum that is reflected. Can be positioned and can be connected to a loop. From this specification, the point where the series resonant circuit (120) is connected to the magnetic loop (110) is referred to as the “connection point” and is the connection point of electrical points of 90 degrees or 270 degrees along the magnetic loop. Is referred to as the “90 ° / 270 ° connection point” and the minimum connection point at which the flow reflects is referred to as the “minimum connection point”.

The amount of change in the location of the connection point depends to some extent on the intended use of the antenna and magnetic loop geometry. For example, the optimal connection can be found by comparing the performance of an antenna using 90/270 connection points with the performance of an antenna using the smallest reflection point. The connection point that produces the highest efficiency for the intended use of the antenna can then be selected. The 90/270 connection point may not differ from the smallest connection point that reflects. For example, an antenna embodiment can have a current at a minimum of 90 degrees / 270 degrees of reflection, or can approach a 90 degrees / 270 degrees point. When using a 90/270 degree connection point, the exact change from 90/270 degree depends to some extent on the intended use of the antenna, but when positioned closer to 90/270 degree, The performance of the antenna is improved accordingly. The size of the E field and the H field should also ideally be the same or substantially similar.

In practice, the point where the series resonant element (120) is connected to the loop (110) is a 90 degree / 270 degree position, or an E field probe and an H field flow that define the point at which the current is minimized. It can be found empirically through the use of parts. The point at which the bent trace (124) is to be connected to the loop (110) can be measured by moving the trace (124) until the desired 90/270 degree difference is observed. Another way to measure the 90/270 junction and the minimum junction to reflect along the loop (110) is to visualize the surface current with an electromagnetic software simulation program, and the loop (110 ) Along with the best connection point is visualized as the region with the smallest surface current magnitude.

Therefore, even if the principles underlying element placement are well understood, a level of empirical decision and a level of trial and error are required to ensure optimal antenna performance. This is simply due to the nature of the printed circuit, which often requires a degree of “tuning” before the desired performance is achieved.

Known simple loop antennas present a very wide bandwidth (typically 1 octave), while known antennas such as dipoles have a much narrower bandwidth-typically operating It has a much smaller portion of the frequency (such as 20% of the center frequency of operation).

Printed circuit technology is well known and will not be described in detail herein. Suffice it to say that copper traces are arranged and printed (usually via etching or laser trimming) on a suitable substrate with a particular dielectric effect. With careful selection of materials and dimensions, specific values of capacitance and inductance can be achieved without the need for separate individual elements. However, as will be described further below, the design of the current embodiment relaxes the limitations of the previous higher frequency planar antenna substrate.

As noted, the current embodiment is arranged and manufactured using known microstrip technology where the final design is the result of a specific amount of manual calibration, thereby adjusting the physical traces on the substrate. Is done. In practice, a calibrated capacitance stick is used, which includes a metal element having a known capacitance element (eg, 2 picofarads). Capacitance sticks can be placed against various portions of the antenna trace, for example, while antenna performance is being measured.

Under the control of a person skilled in the art, it is clarified that the traces making up the antenna should be adjusted with dimensions equivalent to adjusting the capacitance and / or inductance. After multiple iterations, an antenna with the desired performance can be achieved.

The connection point between the series resonant element and the loop is empirically determined again using E-field and H-field probes. Once the approximate connection location is determined, even the slightest interference with the test equipment can have a large practical effect, and by connecting and / or by laser trimming the traces in-situ. Fine adjustment of the values of L and C can be made. Once the final design is established, it can be reproduced with good repeatability. Alternatively, the location of the connection between the series resonant element and the loop can be determined using an electromagnetic software simulation program that visualizes the surface current and selecting the region where the surface current is minimized.

An antenna built in accordance with the embodiments described herein presents a substantial efficiency gain over known antennas of similar capacity.

In further embodiments, multiple discrete antenna elements can be combined to provide greater performance that can be achieved through the use of a single element.

FIG. 2 shows the antenna (200) while being aligned and printed on a section of the circuit board (205) in a known manner. Although the circuit board (205) is shown in plan view, there is a specific thickness amount for the board that makes up the circuit board, and the ground plane (not shown) is the contact plane shown in FIGS. Printed on the back of the circuit board (205) in a manner similar to the ground area (624). In FIG. 2, the antenna (200) includes four separate, functionally identical antenna elements (210) arranged in two sets, each set driven in parallel.

The effect of providing multiple stages of the basic antenna element (210) is to improve the overall performance of the antenna (200). Without the losses associated with the structure of the antenna, in theory it would be possible to build an antenna that includes many individual stages of the basic antenna element (210), and each would have a gain of 3 dB on the antenna. This is twice the number of elements to be added. In practice, however, the loss (especially the dielectric heating effect) means that extra elements cannot be added indefinitely. The embodiment shown in FIG. 2 of a four-element antenna is appropriate within the scope of what is physically possible and has a gain of 6 dB (less any arbitrary dielectric heating loss) over a single element antenna. Add.

While the antenna (200) of FIG. 2 is suitable for use in a fixed wireless infrastructure micro-cellular base station or other item, the single element (210) can be a cellular or mobile handset, pager, Suitable for use with mobile devices such as PDAs or laptop computers. The only real critical issue is the size.
The components and operation of element (210) are further described and shown in FIGS. 3A and 3B for antennas (310) and (370), respectively.

FIG. 3A shows a single antenna (310) (one embodiment of element 210 of FIG. 2) and includes 1 by including a phase tracking antenna element (330), as described below. Greater bandwidth can be achieved, up to .5 octaves, and is particularly suitable to provide a larger operational bandwidth (wider bandwidth) than the narrower bandwidth antenna (100) of FIG. . This wider bandwidth is achieved in particular by the rectangular electric field radiator (320) and the combination of the loop element (350) and the phase tracker (330). A rectangular electric field radiator (320) replaces the series resonant circuit (120) shown in FIG. However, as further described below, the operating bandwidth of the rectangular electric field radiator (320) is wider than the operating bandwidth of the tuning circuit (120) due to the operation of the phase tracker (330).

An alternative embodiment of antenna (310) is shown in FIG. 3B as antenna (370), which is the same rectangular electric field radiator (320), loop element (3) as antenna (310) of FIG. 3A. 350), and a drive or feed point (340), but lacks a phase tracker (330) and therefore has a narrower bandwidth of operation than the antenna (310). Another way of incorporating a wide bandwidth implementation is depicted by the CPL antenna element in FIG. 4A, and incorporates multiple electric field radiators (404) and (408) as described further below.

In the case of the tuning circuit (120), the connection point between the tuning circuit and the loop was important in determining the overall performance of the antenna (100). From FIGS. 3A and 3B, the electric field radiator (320) at antennas (310) and (370) is placed outside of loop (350), but the exact position is less important. Because it is generally arranged in the middle of 90 ° / 270 ° around the loop (350) at the center frequency, or the point where the current is minimized, the connection point is the length of one side of the electric field radiator. It is because it is distributed effectively along. Thus, the termination point at which the edge of the electric field radiator (320) encounters its loop (350), along with the dimensions of the loop, determines the operating frequency range of the antennas (310) and (370).

The dimensions of the loop (350) are also important in determining the operating frequency of the antennas (310) and (370). As previously mentioned, the overall length of the loop (350) is a particularly important dimension. To allow for a wider operating frequency range, the triangular phase tracker element (330) is connected to the electric field radiator (320) (within one of two possible positions as shown in FIG. 2). Provided directly opposite. The phase tracker (330) effectively acts as an automatic variable length tracking device, extending the electrical length of the loop (350) depending on the frequency of the RF signal supplied or drive point (340) supplied. ,shorten.

The phase tracker 330 is equivalent to a nearly infinite series of L-C elements, only some of which will resonate at a particular frequency. This automatically alternates the effective length of the loop. In this way, a wider bandwidth of operation can be achieved than with a simple loop that does not have such a phase tracking element.

The phase tracker (330) shown in FIG. 2 has two different possible positions. These positions are chosen for each antenna element (210) within the group of antenna elements (210) shown in FIG. 2 to minimize mutual interference between adjacent antenna elements (210). From an electrical perspective, the two settings are functionally identical.

Larger bandwidths (up to 1 * octave) of antennas (310) and (370) are possible. This is because the magnetic loop (350) is a complete short of the signal current. As shown in FIGS. 3A and 3B, the magnetic loop is a complete short circuit. Because it is a half-wave short. However, 1/4 wave open, full wave short, and perfect short. The antenna phase is determined by the dimension (360). The dimension (360) spans the length of the electric field radiator (320) and the length of the left side of the magnetic loop (350). The signal is shorted at the point where the signal is 180 degrees out of phase. The magnetic field with the largest amount is generated by the magnetic loop. There is also a smaller amount of magnetic field generated by the electric field radiator. Also, the magnetic loop can vary in length from an RF short with a very low real impedance to a near RF open with a very high real impedance. The highest amount of electric field is emitted by one or more electric field radiator elements. However, the magnetic loop also produces a small electric field and the opposite magnetic field in quantity than the electric field emitted by the electric field radiator.

Antenna efficiency is achieved by maximizing the current in the magnetic loop so as to produce as high an H field as possible. This is accomplished by designing the antenna so that the current moves to the E-field radiator and reflects back in the opposite direction, as described further below in FIG. The maximized H field protrudes from the antenna in all directions, maximizing the efficiency of the antenna. This is because more current can be utilized for transmission purposes. If the magnetic loop is a perfect RF short, or if the magnetic loop has a very low real impedance, the maximum H-region energy that can be generated will occur. Under normal circumstances, however, one RF short is undesirable because the RF short consumes the transmitter driving the antenna. The transmitter erases the set amount of energy with set impedance. By using the impedance matching property of the electric field, it is possible to have a close RF short loop without burning out the transmitter.

The current flowing through the magnetic loop flows into the electric field radiator. The current is then reflected back into the magnetic loop by the electric field radiator back in the opposite direction and reflected back into the magnetic field, causing a short circuit of the electric field radiator, creating a right-angle electric and magnetic field.

The dimension (365) consists of the width of the electric field radiator (320). The dimension (365) does not affect the efficiency of the antenna. However, its width determines whether the antenna is narrowband or wideband. The dimension (365) has a larger width only to increase the bandwidth of the antenna (310) shown in FIG. 3A.

All trace elements of the magnetic loop shown in FIG. 3A can be very thick, for example, without affecting the performance or efficiency of the antenna. Making these loop element traces thicker, however, is desirable as it can be required by many different portable devices such as mobile phones that receive larger input power or otherwise operate within a specific frequency range. It is possible to modify the physical dimensions of the antenna to fit the space.

It will be apparent to those skilled in the art that any shape of E-field radiator may be used in the multiple element configurations shown in FIGS. 2, 3A and 3B, with a rectangular electric field radiator (320) being merely an example. I will. Similarly, single element embodiments can use rectangular electric field radiators, tuning circuits or other suitable shapes of antennas. The multiple element version shown in FIG. 2 uses four individual elements (210). However, this may vary up or down depending on the exact system requirements and available space, as will be described for some limitations to the upper range of the element (210).

Embodiments of the present disclosure can operate over greatly increased bandwidth, have superior performance characteristics compared to known antennas with similar dimensions, and single or multiple element antennas Allows the use of either. Furthermore, no complex elements are required, resulting in an inexpensive device that can be applied to a wide range of RF devices. While embodiments of the present disclosure find particular use in mobile telecommunications devices, they can be used in any device where an efficient antenna is desired.

One embodiment consists of a small, single-sided composite antenna (“single-sided antenna” or “printed antenna”). “Single-sided antenna” means that the antenna elements are placed or printed on a single layer or surface, if desired. As used herein, the term “print antenna” refers to whether the elements of the print antenna are printed or made by some other method such as etching, deposit, sputtering, Or to any single-sided antenna described herein, whether created with some other method of applying a metal layer to the surface or placing a non-metallic material around the metal layer. Apply. Multiple layers of single side antennas can be combined into a single device to allow for wider bandwidth in a smaller physical volume. However, each of the means will still be single-sided. The single-sided antenna described below does not have a ground plane on the back side or on the bottom side, and is itself a short-circuited device, representing a new concept in antenna design. Single-sided antennas are balanced. However, if an important ground plane is present in the intended application device, it can be driven by either balanced or unbalanced lines. The physical dimensions of the antenna may vary considerably depending on the performance characteristics of the antenna. However, the antenna (400) shown in FIG. 4A is approximately 2 cm × 3 cm. Smaller or larger implementations are possible.

The single-sided antenna (400) consists of two electric field radiators physically located inside the magnetic loop. In particular, as shown in FIG. 4A, the single-sided antenna (400) includes a magnetic loop (402), and a first electric field radiator (404) is coupled to the magnetic loop (402) with a first electrical trace (406). Connected, the single-sided antenna (400) further comprises a second electric field radiator (408), the second electric field radiator (408) being coupled to the magnetic loop (402) along with a second electrical trace (420). Connected. Electrical traces (406) and (410) connect the electric field radiators (404) and (408) to the magnetic loop (402) at the corresponding 90/270 degree electrical position with respect to the supply or drive point. . Alternatively, the electrical traces (406) and (410) may connect the electric field radiators (404) and (408) to the magnetic loop at the point where the current flowing through the magnetic loop is minimized. As noted above, the different frequencies of traces (406) and (410), the connection or connection point of the traces change, and the radiator (404) is at a particular frequency, the radiator (408) (which is at a different frequency). The reason why it is shown to be connected to the loop (402) in different points will be explained. At lower frequencies, it takes longer for the wave to arrive at the 90/270 degree point. Thus, the physical position of the 90/270 degree point along the magnetic loop will be higher compared to higher frequency waves. At higher frequencies, it takes less time to arrive at the 90/270 degree point, resulting in a lower 90/270 degree physical location along the magnetic loop compared to the lower frequency wave. . Similarly, the point along the magnetic loop where the current is minimized can also depend on the frequency of the electric field radiator. Finally, an alternative embodiment of the antenna (400) may consist of one or more electric field radiators connected directly to the magnetic loop (402) without electrical traces.

Since the electric field radiators emit waves at different frequencies, the electric field radiator (404) also has different dimensions than the electric field radiator (408). A smaller electric field radiator (404) will have a smaller wavelength and thus a higher frequency. A larger electric field radiator (408) will have longer wavelengths and lower frequencies.

The physical configuration of the electric field radiator reduces the overall size of the antenna compared to other embodiments when the physical position of the electric field radiator disposed inside the magnetic loop and the magnetic loops are outside each other While providing a broadband device. Alternative embodiments may have a different number of electric field radiators, each arranged at a different location around the loop. For example, the first embodiment may have only one electric field radiator, while the second embodiment with two electric field radiators has one electric field radiator inside the magnetic loop and the outside of the magnetic loop. Second electric field radiator. Alternatively, more than two electric field radiators can be physically placed inside the magnetic loop. Like the other antennas described above, the single-sided antenna (400) is a transducer with electric and magnetic fields.

As noted, the use of multiple electric field radiators allows for broadband functionality. The electric field radiators can each be configured to emit waves at different frequencies, resulting in an electric field radiator that covers a wide range. For example, the single-sided antenna (400) can be configured to cover the radio frequency range defined in IEEE 802.11b / g, a standard for the use of two electric field radiators configured in two frequency ranges. . The first electric field radiator (404) may be configured, for example, to cover a frequency of 2.41 GHz, while the second electric field radiator (408), for example, covers a frequency of 2.485 GHz. Can be configured. This would allow the single-sided antenna (400) to cover the 2.41 GHz to 2.485 GHz frequency band. It corresponds to the standard defined by IEEE 802.11b / g. As shown with respect to the physically larger antenna embodiments described above, the use of two or more electric field radiators produces broadband operation without the use of a phase tracker (as shown in FIGS. 2 and 3). In an alternative embodiment, a wideband antenna can also be achieved by using a log scale similar to a YAGI antenna to taper a number of electric field radiators.

The length of the electric field radiator generally determines the frequency that they cover. The frequency is inversely proportional to the wavelength. Thus, a small electric field radiator will have a smaller wavelength and will result in a higher frequency wave. On the other hand, large electric field radiators will have longer wavelengths, resulting in a few waves resulting in lower frequencies. However, these summaries are also specific implementations.

For optimum efficiency, the electric field radiator should have an electrical length that is approximately one wavelength, one quarter wavelength, or a multiple of one eighth wavelength at the frequency it generates. As previously mentioned, if the available physical space limits the electrical length of the electric field radiator to less than the desired wavelength, the bent trace adds propagation delay and electrically lengthens the electric field radiator. May be used to

In FIGS. 4A and 4B, electrical traces (406) and (410) are inductors, and their lengths versus their form or other characteristics determine their inductance.
For optimal efficiency, the inductive reactance of the electrical trace should match the capacitive reactance of the corresponding electric field radiator. Electrical traces (406) and (410) are bent to reduce the external dimensions of the antenna. For example, the curve of the electrical trace (406) can approach the magnetic loop (402) instead of being close to the electric field radiator (404), or the curve of the trace (406) faces up. Instead of getting, it can face down, but resembles an electrical trace (410). Electrical traces are shaped to extend their length, and the form does not have any particular effectiveness factor other than in its context. For example, instead of having a straight electrical trace, a curve can be added to the electrical trace to increase its length and correspondingly increase its inductive reactance. However, the sharp corners on the electrical trace and the sinusoidal shape of the electrical trace may negatively affect the efficiency of the antenna. Electrical traces with sinusoidal morphology result in electrical traces that emit a small electric field, in particular partially out-synchronizing the electric field radiator, thereby reducing the efficiency of the antenna. Thus, antenna efficiency may be improved by the use of flexible and elegant curves and electrical traces shaped with as few bends as possible.

The spacing between the elements of the single-sided antenna (400) adds capacitance across the antenna. For example, the spacing between the top of the electric field radiator (404) and the magnetic loop (402), the spacing between the two electric field radiators (404) and (408), the left side of the electric field radiators (404) and (408) and the magnetic loop The spacing between (402), the spacing between the right side of the electric field radiators (404) and (408) and the magnetic loop (402), and the spacing under the electric field radiator (408) and the magnetic loop (402), all Affects the capacitance of the antenna (400). In order for the antenna (400) to resonate with optimal efficiency, the inductive reactance and capacitive reactance of the entire antenna should match in the desired frequency band, as described above. Once the inductive reactance is determined, the distance between the elements can be determined based on the capacitive reactance value required to match the inductive reactance value of the antenna.

Given a set of formulas to find the spacing between an element and the associated edge capacitance, the optimal spacing between elements can be determined using multipurpose optimization. The optimal spacing between elements or between any two adjacent antenna elements can be optimized using linear programming. Alternatively, non-linear programming such as a genetic algorithm can be used to optimize the interval value.

As previously noted, the size of the single-sided antenna (400) depends on many factors including the desired frequency, narrowband to wideband functionality, and capacitance and inductance tuning.

For the antenna element (400) in FIG. 4A, the length of the magnetic loop (402) is one wavelength (360 degrees) and multiples of the other wavelengths can be used, for optimal efficiency. Designed. When designed for optimal efficiency, part of the magnetic loop will also act as an electric field radiator. An electric field radiator will also increase the directivity and efficiency of the antenna and generate a small magnetic field. The length of the magnetic loop can also be arbitrary, or can be approximately one wavelength, quarter wavelength, multiple of 1/8 wavelength, so that certain lengths increase efficiency over others. One wavelength is an open for voltage and a short for current. Alternatively, the length of the magnetic loop (402) can be physically less than a wavelength. However, extra inductance can be added to lengthen the electrical loop by increasing the propagation delay. The width of the magnetic loop (402) is based primarily on the desired effect it has on the capacitance as well as the inductance of the magnetic loop (402). For example, physically shortening the magnetic loop (402) will result in higher frequencies and will result in smaller wavelengths. In the optimal efficiency design of the magnetic loop (402), the inductance and capacitance should satisfy the factor w = 1 / sqrt (LC), where w is the wavelength of the loop (402). Thus, the magnetic loop (402) can be adjusted by changing its inductance and capacitance which affect the electrical length. Reducing the width of the magnetic loop also adds inductance. In thinner magnetic loops, more electrons must pass through a small area with a delay.

The top (412) of the magnetic loop (402) is thinner than any other part of the magnetic loop (402). This allows for the size of the magnetic loop to be adjusted. The top (412) can be small because it has a minimal effect at the 90/270 degree connection. In addition, scraping the top (412) of the magnetic loop (402) increases the electrical length of the magnetic loop (402) and increased inductance so that the inductance matches the sum of the reactance and the capacitive reactance of the antenna. Can help. Alternatively, the height of the top (412) can be increased to increase capacitance (or equivalently reduced inductance). As described above, the minimum connection point that reflects depends on the geometry of the magnetic loop. Changing the loop geometry by scraping the top (412) or increasing the top (412) or changing any other aspect of the magnetic loop will modify the loop geometry. It will require a point where the current to be identified afterwards is the minimum that will reflect.

As shown in FIG. 4A, the magnetic loop (402) need not be square. In an embodiment, the magnetic loop (402) is formed in a rectangle or formed on one side (odd) and the two electric field radiators (404) and (408) have corresponding 90 degree / 270 degree connection points or reflections. Can be placed at the minimum connection point. For optimum efficiency, the electrical length of the loop formed on one side will be approximately a multiple of one wavelength, or approximately a quarter wavelength or a multiple of approximately 1/8 wavelength in the desired frequency band. The electric field radiator can be arranged inside or outside the magnetic loop formed on one side. Again, the key is to identify the connection points along the loop that maximize the efficiency of the antenna. The connection point may be a 90/270 degree electrical point along the magnetic loop, or a point where the current flowing through the magnetic loop is minimized.

For example, in a smartphone, an antenna design formed on one side can be put into an available space formed on one side, such as the back cover of a mobile device. Instead of a magnetic loop formed in a square, it is formed in a rectangle, a circle, an ellipse, a substantially E shape, or a substantially S shape. Or may be formed. Similarly, an antenna formed on one small side can be placed in a non-uniform space on a laptop computer or other portable electronic device.

As described above, the position of the electrical trace is approximately 90 degrees / 270 degrees electrical along the magnetic loop so that the electric field emitted by the electric field radiator is perpendicular to the magnetic field generated by the magnetic loop. It can be a point, or a point that is the smallest connection point that reflects. The 90 ° / 270 connection point and the minimum reflection point are important. This is because these points allow reactive power (imaginary capability) to be transmitted away from the antenna and not returned. Reactive power is typically generated and stored around an antenna near the field. The reactive power swings with respect to a certain position near the source and affects the operation of the antenna.

With respect to FIG. 4A, the dashed line (414) shows where the most significant portion of the phenomenon of edge capacitance occurs. The two metals in the antenna, such as the magnetic loop and the electric field radiator, can generate edge capacitance levels separately at a fixed distance. Through the use of edge capacitance, single-sided antenna embodiments allow for all elements of the antenna to be printed on one side of almost any form of suitable substrate material, including cheap dielectric materials. . An example of a cheap dielectric material that can be used with the substrate containing glass reinforced epoxy laminate FR-4. It has a dielectric constant of about 4.7 ± 0.2. For a single-sided antenna (400), for example, there is no need for a backside or ground plane. Rather, the lead connects to one of the leads that are grounded at each end of the magnetic loop. As noted earlier, this full wavelength antenna design implies the most efficient, shorted, composite loop antenna. In practice, single-sided antennas are typical in embedded antenna designs where the counterpoise is provided by the antenna being mounted, but perform optimally in the presence of a counterpoise ground plane.

The 2D design of the single-sided antenna embodiment has several advantages. With the use of a suitable substrate or dielectric base (which can be very thin), the antenna trace can literally be sprayed or printed on the surface as a composite loop antenna, as a composite loop antenna Can function. In addition, 2D designs typically allow the use of antenna materials without being considered suitable for microwave devices such as very cheap substrates. A further advantage is that the antenna can be placed on a surface formed on one side, such as the back of the edge of a laptop, such as a cell phone housing cover. Single-sided antenna embodiments can be printed on a dielectric surface with an adhesive placed behind the antenna. The antenna can then be attached on various computing devices with leads connected to the antenna to provide the required power and ground. For example, as noted earlier, with this design, an IEEE 802.11b / g radio antenna can be printed on the surface in the size of a postage stamp. It may be possible to attach the antenna to a laptop cover, a desktop computer case or the back cover of a cell phone or other portable electronic device.

Various dielectric materials can be used with the single-sided antenna embodiment. The advantage of FR-4 as a substrate on other dielectric materials such as polytetrafluoroethylene (PTFE) is that it has a low cost. The dielectric typically used for higher frequency antenna designs has many lower loss properties than FR-4. However, they can be substantially more expensive than FR-4.

Single-sided antenna embodiments can also be used for narrowband applications. Narrowband refers to a path where the bandwidth of the message does not exceed the connection bandwidth of the path. At wide bandwidth, the message bandwidth significantly exceeds the path connection bandwidth. Narrowband antenna applications include Wi-Fi and point-to-point long range microwave repeaters. According to the above-described embodiment, for example, an array of narrowband antennas can be printed on a stick, which is then compared to a standard Wi-Fi antenna at a long distance and good signal strength Wi-. It can be placed on a laptop for Fi access.

FIG. 4B shows an alternative embodiment of a single-sided antenna (420) with a magnetic loop (422) whose corners are cut at a 45 degree angle. Cutting the corners of the magnetic loop (422) at a particular angle improves the efficiency of the antenna. Having a magnetic loop with corners forming an angle of approximately 90 degrees affects the flow of current flowing through the magnetic loop. When the current flowing through the magnetic loop hits a corner at a 90 degree angle, it is a reflected current that either flows against the main current flow or forms an eddy pool, causing the current to bounce. create. The energy loss as a result of the 90 degree corner may negatively affect antenna performance most significantly in smaller antenna embodiments. Cutting the corner of the magnetic loop at an angle of approximately 45 degrees improves current flow near the corner of the magnetic loop. Thus, as they flow through the magnetic loop, electrons in the current cannot be disturbed by angled corners. Although it is preferred to cut the corners at a 45 degree angle, alternative embodiments are also possible, with alternative embodiments being cut at an angle different from 45 degrees.

FIG. 4C shows an alternative embodiment of a single-sided antenna (440) that uses various width transitions in the magnetic loop (442) to add inductance or add capacitance to the magnetic loop (442). The corner of the magnetic loop (442) cuts at an angle of approximately 45 degrees to improve the current flow as it flows near the corner of the magnetic loop (442), thereby increasing the efficiency of the antenna. It was done. A single electric field radiator (444) is physically located inside the magnetic loop (442). The electric field radiator (444) is connected to the magnetic loop (442) with an electrical trace (446) having a flexible curved configuration. As mentioned above, having electrical traces (446) that are not formed in a sinusoid and have a flexible curve that minimizes the number of bends in the trace improves the efficiency of the antenna.

The term change is used to refer to the change in the width of the magnetic loop. In FIG. 4C, the magnetic loop (442) is formed in a substantially rectangular shape, which includes a first transition on the left and a second transition on the right. In the embodiment shown in FIG. 4C, the first transition is symmetric with the second transition. Both the left and right transitions of the magnetic loop (442) include an intermediate narrow section (448), or an intermediate narrow portion, which is thinner than the rest of the magnetic loop (442) and the first wide section (450 ) And the second wide section (452) and adjacent to the first wide section (450) and the second wide section (452), the first wide section (450) and the second wide section (452) The wide section (452) has a larger width than the narrow section (448). In particular, the transition from the first wide section (450) to the middle narrow section (448), the middle narrow section (448) transitions to the second wide section (452). The wide-narrow-wide transition of the magnetic loop creates a pure inductance, thereby increasing the electrical length of the magnetic loop.
Thus, the use of a wide-narrow-wide transition within the magnetic loop is a way to increase the electrical length of the magnetic loop (442) by adding inductance to the magnetic loop (442). The length of the central narrow section (448) may also be increased or decreased when necessary to add the desired inductance to the magnetic loop. For example, in FIG. 4C, the middle narrow section (448) extends about ¼ of the left and right sides of the magnetic loop (442). However, the middle narrow section (448) may be increased to cover about half or some other ratio on the left and right sides of the magnetic loop (442), so that the magnetic loop (442) Increase the inductance.

The transition is not limited to sections or portions having a width less than the width of the remaining portion of the magnetic loop (442). Alternative transitions may include an intermediate wide section, or an intermediate wide section, which is wider than the rest of the magnetic loop (442) and is disposed between the first narrow section and the second narrow section. And having a width smaller than the width of the first narrow section, the second narrow section, and the wide section. In particular, in such an alternative embodiment, the magnetic loop transitions from a first narrow section to an intermediate wide section, and then the intermediate wide section transitions to a second narrow section. The narrow-wide-narrow transition in the magnetic loop creates a capacitance, thereby shortening the electrical length of the magnetic loop. The length of the middle wide section can be increased or decreased, adding capacitance to the magnetic loop.

Using transitions in the magnetic loop, i.e., changing the width of the magnetic loop for one or more sections or portions of the magnetic loop, serves as a way to tune impedance matching. Transitions that change the width in the magnetic loop can also be tapered, and to ensure that the responsive inductance and responsive capacitance of all elements of the antenna match, the inductance or capacitance can be reduced. Add. For example, in a wide-narrow-wide transition, the first wide section can taper from its larger width to the smaller width of the intermediate narrow section. Similarly, the intermediate narrow section can taper from its narrow width to a larger width of either the first wide section or the second wide section, or both. The narrow-wide-narrow transition section and the wide-narrow-wide transition section may taper independently of each other. For example, in the first narrow-wide-narrow transition, only the middle wide section can be tapered, while in the second narrow-wide-narrow transition, the first narrow section. Only can be tapered.
The taper can be linear, stepped or curved.

The actual difference in width between portions of the magnetic loop depends on the amount of inductance or capacitance, which ensures that the total responsive capacitance of the antenna matches the total responsive inductance of the antenna. Is needed to do. The embodiment shown in FIG. 4C shows two wide-narrow-wide transitions. The transitions are arranged opposite to each other and are symmetric. However, alternative embodiments can have transitions on only one side of the magnetic loop (442). In addition, if more than one transition is used in the magnetic loop, these transitions need not be symmetric. For example, a magnetic loop formed on one side can have two transitions, which have different lengths and widths. In addition, different types of transitions can be used for a single magnetic loop. For example, a magnetic loop can have both one or more narrow-wide-narrow transitions and one or more wide-narrow-wide transitions.

FIG. 5 shows an embodiment of a small, double-sided or planar planar antenna (500).
The planar antenna (500) uses a rear second side including a tunable patch, indicated by a dashed line (502), and the inductive reactance of the magnetic loop (504) for a particular frequency. Produces a capacitive reactance consistent with The tunable patch (502) is a substantially square piece of metal that has a flexible position relative to the other elements of the antenna (500). In an embodiment, the tunable patch (502) should be located at a point away from the area where current is reflected, such as the upper left corner of the antenna (500), as shown in FIG. is there. The electric field radiator (506) is located inside the magnetic loop (504) to reduce the external dimensions of the double-sided antenna (500). For optimal efficiency, the electric field radiator (506) should have an electrical length approximately equal to a quarter wavelength at its corresponding operating frequency. If the electric field radiator is smaller, it will result in smaller wavelengths at higher frequencies. The electric field radiator (506) is bent substantially in a J shape to fit the entire length inside the magnetic loop (504). Alternatively, the electric field radiator (506) can be stretched and is therefore in a straight line rather than bent in a J-shape. While such embodiments are contemplated herein, such embodiments make the antenna wider and increase the outer dimensions of the antenna.

An electrical trace (508) connects the magnetic loop (504) and the electric field radiator (506) at the 90/270 junction or the least reflective junction. Compared to the opposite side of the magnetic loop (504), the top (510) of the magnetic loop (504) is smaller. This serves the purpose of increasing the inductance and extending the electrical length of the magnetic loop (504). As was the case with a small single-sided antenna (400), the inductive reactance can be matched to the total capacitive reactance of the antenna (500) by further increasing the inductance and can be adjusted as described above. .

The tunable patch (502) can also be located anywhere along the top (510) of the magnetic loop (504). However, having the magnetic loop (504) away from connecting the tunable patch (502) to the electric field radiator 506 yields better performance. The dimensions of the tunable patch (502) may also be increased by changing its depth, length and height. Increasing the depth of the tunable patch (502) will result in an antenna design that occupies more space.
Alternatively, the tunable patch (502) can be very thin. However, its length and height can be adjusted accordingly. Instead of having a tunable patch (502) that covers the upper left corner of the antenna (500), the length and height may be increased to cover the left half of the antenna (500). Alternatively, the length of the tunable patch (502) may be increased allowing it to expand the upper half of the antenna (500). Similarly, the height of the tunable patch (502) may be increased allowing it to extend the left side of the antenna (500). Tunable patches can also be smaller.

Because it resembles a single-sided antenna, various dielectric materials can be used with the double-sided antenna (500) embodiments. Dielectric materials that can be used include FR-4, PTFE, cross-linked polystyrene, and the like.

FIG. 6 shows an embodiment of a large antenna (600), which consists of an array of four antenna elements (602) with a bandwidth of 1 + 1/2 (1.5) octaves. The antenna elements (602) are each from a TE mode (transverse electric) radiator, a magnetic (H field) radiator, or a magnetic loop dipole (604) (shown roughly by a dashed line and called a magnetic loop (604)). Constructed TM mode (transverse magnetic) radiator, electric (E-field) radiator, or electric field dipole (606) outside the magnetic loop (604) (shown by hatched and shaded rectangular areas, electric field radiator (Referred to as (606)). The magnetic loop (604) must be electrically one wavelength, which creates a short circuit. While the magnetic loop (604) may be physically less than one wavelength, adding extra inductance will electrically extend the magnetic loop (604), as will be described below. Also, the physical width of the magnetic loop (604) can be adjusted to obtain the proper magnetic loop (604) inductance / capacitance, so that it resonates at the desired frequency. As noted below, the physical parameters of the magnetic loop (604) are independent of the quality of the dielectric material used for the antenna element (602).

As previously described, the magnetic loop (604) is a perfect short, thus maximizing the amount of current in the magnetic loop and producing the highest H field. At the same time, the impedance is adapted from the transmitter to the lead, preventing the transmitter from burning out as a result of a short circuit.
The current moves from the magnetic loop (604) to the electric field radiator (606) in the direction of the arrow (607) and in the opposite direction (from the electric field radiator (606) to the magnetic loop (604) in the direction of the arrow (609)). Reflect and return.

In an embodiment, as shown in FIG. 6, each antenna element (602) is about 4.45 centimeters wide and about 2.54 centimeters high. However, as mentioned above, the dimensions of all elements are determined by the frequency of operation and other characteristics. For example, the trace of the magnetic loop (604) can be made very thick, increasing the gain of the antenna element (602) to allow the physical size of the antenna element (602), and then the antenna Allows for 600 magnitudes and is modified to fit any desired physical space in which additional resonances exist, while maintaining some of the same increased gain and at similar levels of efficiency However, neither is possible with prior art voltage supply antennas. The modified design maintains (1) a magnetic loop that inherits a closed-form surface current, (2) reflection of energy from the E-field radiator to the magnetic loop, and (3) a matched impedance among the elements As long as the antenna can be adjusted to almost any size. Although the gain varies depending on the particular dimensions and configuration chosen for the antenna, a similar level of efficiency can be achieved.

The phase tracker (608) (indicated by the hatched area formed in the triangle shape) makes the antenna (600) broadband and can be removed for narrowband systems. The top of the phase tracker (608) is ideally placed at a 90/270 degree electrical location along the magnetic loop (604). However, in an alternative embodiment, the top of the phase tracker can be placed at the least reflective connection location. The dimension (610) of the electric field radiator (606) is actually not critical to the overall operation of the antenna element (602). Only dimension (610) has a width to make antenna element (602) broadband. Also, if the antenna element (602) is intended to be a narrow band device, the dimension (610) can be reduced. As shown, antenna element (602) is intended to be broadband. Because it contains a phase tracker (608). The dimension (612) is determined by the center frequency of operation and determines the phase of the antenna element (602). The dimension (612) spans the length of the electric field radiator (606) and the length of the left side of the magnetic loop (604). The dimension (612) will be slightly adjusted for the dielectric material used as the substrate and will typically be a quarter wavelength. The electric field radiator (606) has a length representing approximately one quarter of the wavelength at the frequency of interest. The length of the electric field radiator (606) can also be sized to be a multiple of a quarter wavelength at the frequency of interest. However, these changes can reduce the effectiveness of the antenna.

Although this difference may not be apparent in the drawing of FIG. 6, the width of the top (614) of the magnetic loop (604) is intended to be smaller than any other part of the magnetic loop (604). This dimensional difference is similar to the smaller antenna embodiment described above when the electrical length is increased and the top (614) can be trimmed to add inductance. The top (614) of the magnetic loop (604) can be shaved because it has minimal effect on the electrical position of 90/270 degrees. Adding inductance by scraping the top (614) causes the magnetic loop (604) to appear electrically longer.

The dimensions (616), (617) and (618) of the magnetic loop (604) are all determined by the wavelength magnitude. The dimension (616) consists of the width of the magnetic loop (604). The dimension (617) consists of the length of the lower left part of the magnetic loop (604). That is, the dimension (617) consists of the length of the lower portion of the magnetic loop (604) on the left side of the magnetic loop opening (619). The dimension (618) consists of the entire length of the magnetic loop (604). A dimension (616) equal to dimension (618) results in a square loop and the best antenna performance is achieved. However, rectangular or irregularly formed magnetic loops (604) can also be used.

As noted before, phase tracker (608) is included for wideband operation of antenna (600). Also, the deletion of the phase tracker (608) makes the antenna (600) less broadband. The antenna (600) can alternatively be narrowband by reducing the physical vertical dimensions of the phase tracker (608) and the dimensions of the electric field radiator (606). The phase tracker (608) and the support of the phase tracker (608) for broadband operation at the antenna have the potential to reduce the total number of antennas used in various devices such as cell phones. The size of the phase tracker (608) also affects its inductance and capacitance, as shown in FIG. The capacitance and inductance ranges of the phase tracker (608) can be adjusted by adjusting the physical dimensions of the phase tracker (608). The inductance (L) of the phase tracker (608) is based on the height of the phase tracker (608). The capacitance (C) of the phase tracker (608) is based on the width of the phase tracker (608).

The antenna element (602) and the pair of antenna elements (602) form a set of gaps between them. The two antenna elements (602) arranged on the left side of the antenna (600) constitute a first set of antenna elements (602). However, the two antenna elements (602) placed on the right side of the antenna (600) constitute a second set of antenna elements (602). There is a first gap (620) between each set of antenna elements (602) and a second gap (622) between each set of pairs of antenna elements (602). A first gap (620) between each pair of elements (602) and a second gap (622) between each pair of antenna elements (602) are generated by the antenna elements (602) in the most efficient manner. The far-field radiation pattern is designed to be aligned so that the far-field radiation pattern is additive rather than negative. Well-known phased antenna array techniques can be used to determine the optimal distance between multiple CPL antenna elements (602) so that the far field radiation pattern of each element is additive. It is.

In embodiments, the far field radiation pattern can be modeled on a computer based on the relationship of the different configurations of the antenna elements (602). For example, until additional orientation and alignment of the far field radiation pattern is achieved, the dimensions of the antenna elements (602), the spacing between the antenna elements (602) and between the pair of antenna elements (602), and Element relationships can be adjusted. Alternatively, the far field radiation pattern can be measured using electrical equipment, with the association of elements adjusted on that basis.

Referring now back to FIG. 6, the antenna element (602) is provided by a microstrip feedline represented by the dashed line (624). The feed line in dashed line (624) coincides with the network and depends on the dielectric material used to drive the impedance. Also, the symmetry of the feed line is important in avoiding unnecessary phase lag resulting in the far field radiation pattern generated by the antenna elements becoming negative instead of being additive.

Referring to FIG. 6, the embodiment uses a common combiner / splitter (626) to split the incoming signal into two, feed and return signals to two sets of antenna elements (602). Make a pair. The second and third combiner / splitter (628) then splits the resulting signal in two to feed the antenna element (602) and combine the returning signals. Combiners / splitters (as they result in an almost perfect impedance match along the feedline over a wide frequency range, preventing power from being reflected back along the feedline and resulting in performance loss 626) and (628) are desirable.

FIG. 8 shows the bottom layer (800) of the antenna (600), which includes elements (802), (812), (814) and (816), each of which is trapezoidal. It includes an element (804), a choke junction region (806), and a riser (808). Elements (812) and (814) also reflect the signal (or RF energy) to the bottom of the bridge element (820) to set the phase angle of the antenna (600), but elements (802), (812) ), (814) and (816) act as capacitors. If a spherical shape is desired for the resulting pattern generated by the antenna (600), the distance (826) from the bottom of the trapezoidal element (804) to the bottom of the bridge element (820) is a quarter wavelength. It cannot be over. By changing the distance (826), the radiation patterns formed in different shapes can be obtained by changing the distance (826) for each of the elements (802), (812), (814) and (816). Can be generated. Finally, cutout elements (822) and (824) represent the trace material being removed from the lower left and lower right corners of bridge element (820) to prevent reflection of elements (802) and (816). However, this in turn changes the phase angle set by elements (812) and (814).

A trapezoidal element (804) holds the magnetic loop (604) of each corresponding antenna element (602) in tuning due to the fact that each trapezoidal element (804) is log driven in size.
The slope of each trapezoidal element (804) (especially the slope on the top side of the trapezoidal element (804)) has a variable inductance and capacitance that helps the antenna (600) match the inductive reactance to the capacitive reactance. Used to add. By adding capacitance through the trapezoidal element (804), the electrical length of each corresponding magnetic loop (604) on the opposite side of the antenna (600) can be adjusted. The trapezoidal element (804) adjusts the position with the top and trace (614) of the magnetic loop (604) opposite the antenna (600). The choke joint (806) serves to separate the trapezoidal element (804) from ground and consequently prevent leakage of the composite signal. Sides (809) and (810) of the trapezoidal element (804) are counterpoises for the electric field radiator (606) on the opposite side of the antenna (600). It requires a basis for setting polarity formation. The side (809) consists of the right side of the trapezoidal element (804) and the upper right part of the razor (808) located on the choke joint (806).
That is, the side (809) consists of the right side of each element (802), (812), (814) and (816) located on the choke junction (806). The side (810) consists of the left side of the trapezoidal element (804) and the left side of the razor (808). That is, the side (810) consists of the left side of each element (802), (812), (814) and (816) located on the ground plane element (828). Counterpoises (809) and (810) increase the transmit or receive efficiency of antenna (600). The ground plane element (828) is standard for microstrip antenna designs, for example, a 50 ohm trace on a 4.7 dielectric is about 100 mils wide.

As noted previously, the trapezoidal element (804) can be fine tuned to change the capacitance or changing inductance of the corresponding magnetic loop. The fine tuning procedure involves shrinking or enlarging the section of the trapezoidal element (804). For example, it is determined that additional capacitive reactance is required to match the inductive reactance of the magnetic loop. Accordingly, the trapezoidal element (804) may be expanded to increase capacitance. An alternative fine tuning step is to change the slope of the trapezoidal element (804). For example, the tilt may be changed from an angle of 15 degrees to an angle of 30 degrees. Alternatively, if the magnetic loop (604) is modified either by increasing its portion or by cutting the width of the top trace (614) of the magnetic loop (604), the modified magnetic The metal on the ground plane corresponding to the loop (604) must be adjusted. For example, the top side of the trapezoidal element (804), or the total length of the trapezoidal element (804), can be trimmed based on whether the top trace (614) of the magnetic loop (604) has been trimmed or increased. May be increased.

The simultaneous excitation of the TM and TE radiators, as described herein, has zero reactive power as expected by the time-dependent pointing theorem when used to analyze microwave energy. As a result. Prior attempts to build a composite antenna with TE and TM radiators that were electrically perpendicular to each other depended on the three-dimensional configuration of these elements. Such a design cannot be easily commercialized. In addition, the previously proposed composite antenna design was powered by separate power sources at more than one location in each loop. In various embodiments of the antenna as disclosed herein, the magnetic loop and the electric field radiator are positioned at an electrical angle of 90/270 of each other on the same plane and powered from a single location. . This reduces the complexity of the physical arrangement, improves commercialization and results in a two-dimensional arrangement. Alternatively, the electric field radiator can be placed in the magnetic loop at the point where the flow flowing through the magnetic loop is minimized.

The antenna embodiments described herein have greater efficiency than conventional antennas due in part to reactive power cancellation. In addition, the embodiments have large antenna apertures for each physical size. For example, a half-wave antenna with an omni-directional pattern according to an embodiment will have a gain that is significantly greater than the normal 2.11 dBi gain of a simple field dipole antenna.

However, another embodiment consists of a single-sided antenna with a built-in counterpoise for the electric field radiator. FIG. 9A illustrates a single-sided 2300-2700 MHz antenna embodiment with a single electric field radiator and a built-in counterpoise for the electric field radiator. The antenna (900) consists of a magnetic loop (902) with an electric field radiator (904) connected directly to the magnetic loop (902) without the benefit of electrical traces. The electric field radiator (904) is physically placed inside the magnetic loop (902). As in other embodiments, the electric field radiator (904) can be connected to the magnetic loop (902) at a 90/270 connection point, or at a point where the flow flowing through the magnetic loop 902 is minimal. In an alternative embodiment, the electric field radiator (904) may be coupled to the magnetic loop (902) by an electrical trace. In addition, while antenna (900) is shown with one electric field radiator, alternative embodiments may include one or more electric field radiators. Alternative embodiments may also include one or more electric field radiators physically located outside the magnetic loop (902).

Alternative embodiments of independent antennas may also include a first electric field radiator having a first length and a second electric field radiator having a second length different from the first length. Broadband antennas are possible using one or more electric field radiators with different lengths, similar to the antenna embodiments previously described herein.

Antenna (900) includes a transition (906) and an electric field radiator (counterpoise (908) for 904. Transition (906) has a width greater than the width of magnetic loop (902), a magnetic loop ( 902), the transition 906 separates the electrically embedded counterpoise (908), which is the antenna (900) or product using the antenna (900). Allows the chassis to be completely independent of any ground plane.

Since the counterpoise is formed from a magnetic loop (902), the counterpoise (908) is said to be built-in. As noted, the built-in counterpoise (908) allows the antenna (900) to be completely independent of the product ground plane. The single-sided antenna embodiment shown in FIGS. 4A-4C is simply printed on a single stage and does not include a ground plane, but requires a ground plane to be provided by the device using the antenna. And In contrast, an independent counterpoise antenna does not require a ground plane to be provided by the device using the antenna.

In the single-sided embodiment described above, the device using the antenna is for the antenna, at the ground plane of the device that serves as the ground plane for the single-sided antenna, or the chassis of the device or some other metal component as the ground plane. A ground plane is provided by use with a single-sided antenna. However, any modifications to the device circuitry, device chassis, or device ground plane may negatively affect antenna performance. This phenomenon is not unique to the single-sided embodiments disclosed herein, but instead applies to antennas that are widely used in research and commerce. It is therefore desirable to have an antenna that does not require a ground plane and that is not affected by any changes made to the device using the antenna.

By not requiring a ground plane, the antenna (900) does not rely on a ground plane outside the antenna. This independence of the antenna (900) independent of the external ground plane means that the performance of the antenna is not affected by changes made to the device. In manufacturing and design terms, this suggests that an independent antenna can be designed for a specific frequency, and that the level of performance means that using the antenna is independent of the device. . For example, a wireless router manufacturer can require a specific antenna based on a set of requirements. These requirements can include, among other requirements, the space available for the antenna, the frequency range for the antenna, and the substrate to be used. The antenna design and product can then be made independent of the actual wireless router design and product. In addition, any future changes to the wireless router will not affect the performance and efficiency of the antenna because the antenna is independent and unaffected by changes to the router circuitry, router ground plane or router chassis.

The length of the transition (906) can be set based on the frequency of operation of the antenna. For higher frequency antennas, shorter transitions can be used where the wavelength is shorter. On the other hand, for lower frequency antennas, longer transitions (906) can be used where the wavelength is longer. Transition (906) may be adjusted independently of counterpoise (908). For example, the transition for a 5.8 GHz antenna is only half the size of the transition (906) in FIG. 9A, while the counterpoise (908) is as much as the entire left side of the magnetic loop (902). Still long.

The counterpoise (908) length can be adjusted as necessary to obtain the desired antenna performance. However, it is preferable to have as large a counterpoise (908) as possible. For example, in an alternative embodiment, the counterpoise (908) may span not only about 80% of the left side of the magnetic loop (902) but also the entire length of the left side of the magnetic loop (902). However, as noted above, the trace width of the magnetic loop (902) affects the electrical length of the magnetic loop (902). A magnetic loop with a thin trace has a portion of the magnetic loop that is either halfway electrically longer than a magnetic loop with a wider trace around the magnetic loop or with a wider trace. For example, magnetic loop (902) is an example of a magnetic loop with a transition (906) and a wider trace of counterpoise (908). Thus, while it is desirable to have a counterpoise that is as long as possible, the length of the counterpoise (908) affects the electrical length of the magnetic loop (902). Electrically longer magnetic loops are lower in frequency. On the other hand, electrically shorter magnetic loops are higher in frequency. For example, the use of a counterpoise that spans the entire length of the left side of the magnetic loop increases the overall width of the magnetic loop, thereby electrically shortening the magnetic loop, resulting in a magnetic loop having a higher frequency as desired. For example, a frequency of 5.8 GHz results instead of a target frequency of 5.6 GHz.

An embodiment of the antenna (900) is a narrow-wide-narrow in addition to transitions and counterpoise, as previously described herein, to match the electrical length of the magnetic loop to the desired frequency. Transitions and / or wide-narrow-wide transitions can be included. In addition, independent antenna embodiments may also include magnetic loops with isolated corners at angles as described above to improve flow flow near the corners of the magnetic loop.

As previously mentioned, a counterpoise (908) for the electric field radiator (904) is used in place of the ground plane. The electric field radiator (904) is effectively a monopole antenna.
A monopole antenna is formed by replacing half of a dipole antenna with a ground plane that is perpendicular to the other half of the dipole antenna. In an independent antenna embodiment, the electric field radiator looks like a large piece of metal that is electrically connected to the electric field radiator, but it can be used instead of a ground plane. From FIG. 4C, in the single-sided antenna (440), the electric field radiator (444) radiates an electric field based on the position of the ground plane used for the antenna (440). This electric field rotates perpendicular to the plane of the electric field radiator, but the magnetic field rotates substantially flush with that plane. This electric field pattern is self-contained in a donut shape, which is also called an almost perfect omnidirectional pattern. As mentioned above, single-sided antenna embodiments do not necessarily provide their own ground plane. Thus, if an antenna (440) is used in the device, the device will serve as a ground plane for the antenna (440). Also, the radiation pattern emitted by the electric field radiator (444) can be reflected back into the device. However, if the single-sided independent antenna including the counterpoise includes a ground plane, the radiation pattern described above is effectively switched, and the electric field is one or more planes that are flush with the plane of the electric field radiator or the plane of the electric field radiator. And the magnetic field rotates perpendicular to the plane.

There is no need to place or machine counterpoise (908) in the upper left corner of the magnetic loop (902). In an alternative embodiment, the counterpoise may be placed in the upper right corner with the electric field radiator (904) placed on the left side of the magnetic loop (902). Regardless of the physical position of the counterpoise (908) and electric field radiator (904) (or multiple radiators if more than one), the counterpoise and electric field radiators must be 180 degrees out of phase. In still other embodiments, the length of the counterpoise can be adjusted as needed. A counterpoise (908) is also placed along the right side of the magnetic loop (902), just below the electric field radiator (904), or at other locations around the magnetic loop (902).

The antenna (900) further includes a balun (910). A balun is a type of electrical transformer that can convert an electrical signal that is balanced with respect to ground (difference) to an unbalanced (single-ended) signal and vice versa. Specifically, the balun provides a high impedance to the common mode signal and a low impedance to the differential signal. The balun (910) serves the function of canceling the common mode current. In addition, the balun (910) tunes the antenna (900) to the desired input impedance and tunes the impedance of the entire magnetic loop (902). The balun (910) is formed in a substantially triangular shape and consists of two parts divided by an intermediate gap (912).

The two baluns (910) are magnetically and electrically coupled. The gap (912) in the balun (910) returns through the transmitter and removes the common mode current by magnetically preventing it from flowing in one direction, such as to a device using the antenna (900). This is important because the mode current causes the current flow reflection by the transmitter to negatively affect the performance of the antenna (900) and the device using the antenna (900) due to the common mode current. . Current reflection by the transmitter causes interference to the circuit of the device, particularly using antennas. Such negative performance or devices may be prevented from complying with Federal Communication Commission (FCC) rules. The gap (912) in the balun (910) cancels the common mode current, thereby preventing the current from being reflected behind the connector of the antenna (900).

The gap (912) can be adjusted based on the antenna design and dimensions. In embodiments, electromagnetic simulation can be used to visualize the current flowing through the antenna (900). Thereafter, the gap (912) may be increased or decreased until the simulation indicates that current is no longer reflected and is flowing back through the transmitter. Canceling the common-mode current means that the current flowing into the transmitter in one direction stops, starts flowing in the opposite direction, flows into the antenna (900) in one direction, and flows out of the antenna (900) in the second direction. Can be made into a picture.

The tapered side (914) of the balun (910) serves the purpose of being electrically connected. The angle of the tapered side (914) can be adjusted to match the impedance with the antenna (900). Typically, individual inductors and individual capacitors are placed along a feed line (not shown) to the antenna (900) to power the antenna up to the impedance of the antenna (900) to match the impedance of the device Let For example, if the antenna expects an input of 50 ohms, but the device's circuitry is providing the antenna with 150 ohms, a series of inductors and capacitors are supplied to the antenna up to 50 ohms expected by the antenna. Used to balance mismatch problems by transforming ohms. In contrast to general industry practice, the independent antenna (900) embodiment is optional, such as by using a series of inductors and capacitors along the wiring that feeds the antenna (900). There is no need for impedance matched by. Instead, the balun (910) is used to match the connector feeding the antenna to the impedance of the antenna (900) and to the impedance of the magnetic loop (902).

The height of the balun (910) is a function of the operating frequency of the antenna (900). Thus, a higher balun (910) is required for lower frequencies, while a shorter balun (910) is required for higher frequencies. When using a high balun in the antenna, the proximity of the balun (910) to the electric field radiator is important. Positioning the balun (910) too close to the electric field radiator 904 can create a capacitive coupling between the balun (910) and the electric field radiator (904). Thus, it is important for the balun (910) spaced from the electric field radiator (904) to prevent capacitive coupling from adversely affecting the performance of the antenna (900). If the particular antenna design requires the use of a higher balun at the operating frequency of the antenna to properly match the impedance to the antenna and cancel the common mode current, the antenna (920) in FIG. As shown, the balun can be moved down. In an alternative embodiment, the independent counterpoise antenna (900) may not include a balun (910).

Antenna (900) is an example of an independent counterpoise multi-field antenna. Embodiments of the antenna (900) can be printed or otherwise deposited on an approximately 1.6 millimeter FR-4 substrate. The characteristics and design of the antenna (900) will make the antenna (900) a flexible printed circuit, acrylonitrile butadiene styrene (ABS) plastic, and other materials including materials that do not appear to be suitable for microwave frequencies. It can be adapted. The operating frequency of the antenna 900 is approximately 2300-2700 MHz and is suitable for various embedded applications including mobile phones, access points, PDAs, laptops, PC cards, sensors and automotive applications. The antenna (900) embodiment achieved a peak efficiency of about 94% and a peak gain of about +3 dBi. The antenna (900) has a width of about 31 millimeters and a length of about 31 millimeters. The antenna (900) has linear polarization and an impedance of about 50 ohms. The antenna (900) also has a voltage standing wave ratio of less than 2: 1 (<2: 1). The size and efficiency of the antenna (900) make it suitable for Wi-Fi applications where efficiency, size and gain are important.

An alternative embodiment of a single-sided antenna with a built-in counterpoise is shown in FIG. 10A. The antenna (1000) is an example of an antenna with linear polarization. The antenna does not require a ground plane due to the built-in counterpoise. The antenna (1000) can be printed or otherwise deposited on a 1.6 millimeter thick FR-4 substrate. Similar to antenna (900), the properties and design of antenna (1000) do not appear to be suitable for flexible printed circuits, acrylonitrile butadiene styrene (ABS) plastic, and microwave frequencies. Applicable to other materials including materials. The antenna (1000) operates in a frequency range of about 882 MHz to 948 MHz with a measured peak gain of about +3 dBi and a maximum efficiency of about 92%. The antenna (1000) has an antenna impedance of about 50 ohms and a voltage standing wave ratio of less than 2: 1 (<2: 1). The antenna (1000) has a width of about 76 millimeters and a height of about 76 millimeters.

The antenna (1000) consists of a magnetic loop (1002), the first electric field radiator (1004) is directly coupled to the magnetic loop (1002), and the second electric field radiator (1006) is directly coupled to the magnetic loop (1002). ing. Both electric field radiator (1004) and electric field radiator (1006) are coupled to magnetic loop (1002) without the benefit of electrical tracing. The electric field radiators (1004) and (1006) are physically disposed inside the magnetic loop (1002). The use of two electric field radiators increases the gain of the antenna instead of that in the antenna (900) of FIG. The curve (1008) separating the two electric field radiators (1004) and (1006) shows the two electric field radiators (1004) and (1006) in order to add their far field patterns to each other. It serves to delay the phase between the radiators (1004) and (1006).

Two electric field radiators (1004) and (1006) along with curve (1008) create an electric field radiator array (1010) with a phase lag. Specifically, curve (1008) ensures that the two electric field radiators (1004) and (1006) are 180 degrees out of phase with each other. The curve can be used as a space saving technique. For example, if a small antenna is required and two electric field radiators are forced to be closer together due to the need to minimize dimensions, the curve (1008) shows that the electric field radiators are still 180 degrees out of phase with each other. Can be used to guarantee.
The electrical length of the trace of curve (1008) can be adjusted as needed based on the required delay. For example, the trace can be longer or shorter while keeping the width constant. Alternatively, the length of the trace can be kept constant as the width of the trace is made wider or thicker. As mentioned above, the electrical length of the trace depends on its physical length and its physical width. FIG. 10B shows an alternative embodiment of an independent antenna (1020) without a curve (1008).

As described with respect to antenna (900), antenna transition (1012) and counterpoise (1014) can be adjusted based on a number of factors. The transition (1012) depends on the frequency of operation. However, the transition must be long enough to ensure that the counterpoise (1014) is electrically isolated. The largest possible counterpoise (1014) is preferred. Finally, the balun (1016) cancels the common mode current and matches the impedance of the antenna (1000) to the impedance of the transmitter supplied to the antenna 1000.

In an alternative embodiment of the antenna (1000), the electric field radiator array (1010) can be arranged on the left side of the antenna (1000) instead of being arranged on the right side of the antenna (1000). In such an alternative embodiment, the counterpoise (1014) would be placed on the upper right side of the magnetic loop (1002). A counterpoise (1014) can also be positioned along the right side of the magnetic loop (1002), just below the electric field radiators (1004) and (1006).

11A-11C show a 2D radiation pattern for the antenna (900) from FIG. FIG. 11A shows a 2D radiation pattern on the XZ plane (1100). The solid line (1102) represents the actual antenna radiation pattern and the dashed line (1104) represents a 3 dB beamwidth. The dotted line (1106) represents the maximum force of the field along one direction. That is, the dotted line (1106) represents that the strongest field has been detected in the 2D radiation pattern shown.
FIG. 11B shows a 2D radiation pattern for the antenna (900) on the XY plane (1110). FIG. 11C also shows a 2D radiation pattern for the antenna (900) on the YZ plane (1120).

12A-12C show 2D radiation patterns for the antenna (1000) from FIG. 10A. FIG. 12A shows a 2D radiation pattern on the XZ plane (1200). The solid line (1202) represents the actual radiation pattern and the dashed line (1204) represents the 3 dB beam width. The dotted line (1206) represents the maximum force of the field along one direction. That is, the dotted line (1206) represents that the strongest field is detected in the 2D radiation pattern shown. FIG. 12B shows a 2D radiation pattern for the antenna (1000) on the XY plane (1210). FIG. 12C also shows a 2D radiation pattern for the antenna (1000) on the YZ plane (1220).

FIG. 13A shows the voltage standing wave ratio (VSWR) for the antenna (900). The VSWR plot shows that the antenna (900) is a good impedance match for a frequency range of approximately 2.34 GHz to approximately 2.69 GHz. That is, most of the energy supplied to the antenna (900) will be radiated, rather than being reflected back to the transmitter, in a frequency range of approximately greater than 2.34 GHz to 2.69 GHz. Inside the two central vertical solid lines, in particular, represents a frequency range where the VSWR of the antenna (900) is less than 2: 1 (<2: 1). FIG. 13B shows the return loss for the antenna (900). The return loss and the VSWR are mathematically related so that the return loss of −10.0 on FIG. 13B corresponds to 2.0 in the VSWR on FIG. 13A. The return loss diagram from FIG. 13B shows that between the points represented by 1 and 2, the antenna (900) is a good impedance match.

FIG. 14A shows the voltage standing wave ratio (VSWR) for the antenna (1000) from FIG. 10A. The VSWR plot shows that the antenna 1000 has a good impedance match for the frequency range from approximately 884 MHz to 947 MHz. That is, the frequency range from about 884 MHz to 947 MHz, which is the majority of the energy supplied to the antenna (1000), is not reflected back to the transmitter, but most of the energy supplied to the antenna (1000) is radiated. Will be done. The inside of the two central vertical solid lines represents the frequency range, especially when the VSWR of the antenna (1000) is less than 2: 1 (<2: 1). FIG. 14B shows the return loss for the antenna (1000). As described above, the return loss of −10.0 corresponds to 2.0 in VSWR. The return loss diagram from FIG. 14B shows that between the points indicated by 1 and 2, the antenna (1000) is a good impedance match.

FIG. 15 shows yet another embodiment of an independent antenna (1500). The antenna (1500) is an example of a 5.8 GHz antenna. Certain embodiments of the antenna (1500) have dimensions of about 15 millimeters in length and about 15 millimeters in width. The antenna (1500) consists of a magnetic loop (1502), and the electric field radiator (1504) is directly coupled to the magnetic loop (1502). In contrast to independent antennas (900) and (1000), antenna (1500) includes a tapered transition (1506) made up of two sections (1508) and (1510). The first transition section (1508) begins where the width of the magnetic loop changes from a small width to a large width. The first transition section (1508) tapers linearly toward a smaller width before the width of the magnetic loop increases again, before the second transition section begins. The second transition section increases linearly from a small width to a larger width. As previously mentioned, adjustment of the width of the magnetic loop traces allows it to be adjusted for electrical length. In addition, the length, width, and number of transitions used electrically isolate the counterpoise (1512). The transition (1506) must be long enough so that the current flowing through the counterpoise (1512) is minimized in quantity. In addition, tapering the transition (1506) relative to the counterpoise (1512) increases bandwidth due to impedance matching. Balun (1514) cancels the common mode current. The balun (1514) matches the impedance of the antenna (1500). Alternative embodiments of antenna (1500) may not include balun (1514).

Embodiments of a single-sided antenna are arranged on a magnetic loop having a width, a magnetic loop that generates a magnetic field and is disposed on a surface having a first inductive reactance, an electric field is generated on a surface having a first capacitive reactance. An electric field radiator, wherein the electric field radiator is directly coupled to the magnetic loop, wherein the electric field is perpendicular to the magnetic field, and the physical arrangement between the electric field radiator and the magnetic loop results in a second capacitive reactance. The embodiment of the single-sided antenna further comprises a transition formed on the magnetic loop and having a transition width greater than the width of the magnetic loop and a magnetic loop facing or adjacent to the electric field field radiator A counterpoise formed on the magnetic loop, wherein the transition is from the magnetic loop to the counterpoise Substantially electrically isolated.

Features disclosed in this specification (including any appended claims, abstracts and drawings) are intended to be alternative, serving the same, equivalent or similar purposes, unless expressly stated otherwise. Can be replaced with features. Thus, unless expressly stated otherwise, each feature disclosed is one example only of a generic series of equivalent or similar features.

While the invention is described herein by several alternative terms, it should be understood that the techniques described herein may have many additional uses and applications. It is. Accordingly, the present invention principles should not be limited to the mere specific descriptions, embodiments, and various figures contained herein, which are merely illustrative of exemplary embodiments, alternatives, choices and applications of the principles of the present invention. .

Claims (1)

A single-sided antenna,
The single-sided antenna comprises a magnetic loop positioned in a plane and configured to generate a magnetic field;
The single-sided antenna further comprises at least one electric field radiator positioned in the plane and placed inside the magnetic loop, the at least one electric field radiator being coupled to the magnetic loop and against the magnetic field A single-sided antenna, wherein the electric field radiator is coupled to the magnetic loop at a point where the current flowing through the magnetic loop is a minimum to be reflected.