KR100659423B1 - Motor driving device - Google Patents

Abstract

The present invention is to detect the motor load state in the simplified sensorless sinusoidal drive, converts the AC power 1 to the DC power by the rectifier circuit 2, the motor 4 by the inverter circuit 3 Drive the output current of the inverter circuit 3 by detecting the output current of the inverter circuit 3 by controlling the reactive current to a predetermined value at the set rotation speed, and controlling the load state from the inverter circuit output voltage or output power. Determine.

Description

Motor Drive Device {MOTOR DRIVING DEVICE}             

1 is a block diagram of a motor drive device according to a first embodiment of the present invention;

2 is an inverter circuit diagram of the same motor drive device;

3 is a current detection timing chart of the same motor driving apparatus;

4 is a block diagram of a control means of the same motor drive device;

5 is a control vector diagram of the same motor drive device;

6 is a control vector diagram at the time of disassembly of the same motor drive device;

7 is a waveform of each part and a timing chart of the control means of the same motor drive device;

8 is a flowchart of a motor control program of the same motor driving apparatus;

9 is a flowchart of a carrier signal interrupt subroutine of a motor control program of the same motor drive apparatus;

10 is a flowchart of a rotation speed control subroutine of a motor control program of the same motor drive device;

11 is a timing chart of start control of the same motor drive device;

12 is a sectional view of the motor driving apparatus of the dish washing machine in the second embodiment of the present invention;

13 is a block diagram of a control means of the same motor drive device;

14 is a control timing chart for detecting drainage air inflow of the same motor drive device;

15 is a control timing chart of the cleaning air inflow detection of the same motor drive device;

16 is a flowchart of a rotation speed control subroutine of the same motor drive device;

Fig. 17 is a block diagram of control means of the motor drive device in the third embodiment of the present invention.

Explanation of symbols for the main parts of the drawings

1: AC power supply 2: rectification circuit

3: inverter circuit 4: motor

5: current detection means 6: control means

The present invention relates to a motor drive device for sensorless sinusoidal drive.

Conventionally, this kind of motor drive apparatus has reduced the vibration and noise of a motor by omitting a rotor position sensor and driving a sensorless sinusoidal wave, and improved reliability (for example, refer patent document 1).

[Patent Document 1] Japanese Unexamined Patent Publication No. 2000-350489

However, in the above-described conventional configuration, in order to estimate the rotor position, the processor must calculate the motor constants, the circuit parameters, or the motor load in advance, detect the error between the predetermined calculated value and the measured current value, and calculate the error to minimize the error. Therefore, the computation was very complicated, and a processor having a high speed and high performance computational function was required. Moreover, when motor load fluctuations are large, there existed a problem that it was easy to remove.

The present invention solves the above-mentioned problems, and operates stably against load fluctuations. For example, even if output fluctuations or outages occur due to load fluctuations, the load condition or outage is detected to correct control parameters or starting conditions. It is to restart, to provide a sensorless sinusoidal drive motor drive device which simplifies the calculation of the processor and operates stably against load fluctuations.

In order to solve the above-mentioned problems, the motor drive device converts AC power into DC power by a rectifier circuit, drives a motor by an inverter circuit, and detects and sets the output current of the inverter circuit by current detection means. The inverter circuit is PWM controlled so that the rotation speed is controlled, the output voltage overcurrent phase or the reactive current of the inverter circuit is controlled to be a predetermined value, and the load state is determined by the output voltage or output power of the inverter circuit.

The first invention includes an AC power source, a rectifier circuit for converting AC power of the AC power source to DC power, an inverter circuit for converting DC power of the rectifier circuit to AC power, and a motor driven by the inverter circuit. And a current detecting means for detecting an output current of the inverter circuit, and control means for controlling the motor so as to set a rotation speed by PWM controlling the inverter circuit by an output signal of the current detecting means. Is to control the output voltage and output current phase of the inverter circuit or the reactive current to a predetermined value, and determine the load state by the output voltage or output power of the inverter circuit. It is possible to realize the sensorless sinusoidal drive that operates.

According to a second aspect of the present invention, the control means of the first aspect of the present invention is configured to perform V / f control so that the phase of the output voltage and the output current of the inverter circuit, or the reactive current becomes a predetermined value, thereby determining the load state from the V / f control value. This makes it easy to detect the rotation state such as when the motor is stopped by rotation and stop or detect the magnitude of the load.

In the third invention, the control means in the first invention detects the motor load state by the output voltage or the output power of the inverter circuit, and changes the phase or reactive current of the output voltage and output current of the inverter circuit. Since the output current phase or the reactive current can be changed according to the load condition, even when the load fluctuates large or the load fluctuates during operation, the rotation can be stably rotated. Therefore, even when the load fluctuation is large, it is possible to operate and to realize a highly reliable motor drive device.

According to a fourth aspect of the present invention, the control means of the first aspect of the present invention detects the motor load state by the output voltage or the output power of the inverter circuit, and changes the motor driving condition. In the case of fluctuation, it is possible to stably rotate by changing the driving conditions such as the rotation speed and the motor current, or to reduce the noise by changing the rotation speed.

In the fifth aspect of the present invention, the control means of the first aspect of the invention allows the motor load torque to be determined by the output power and the drive frequency of the inverter circuit, and the rotation speed control and the output control corresponding to the motor load torque are possible.

According to a sixth aspect of the present invention, the control means of the first aspect of the invention determines the load torque based on the output power and the drive frequency of the inverter circuit, and controls the motor by the variation of the load torque. Since the optimum motor current, current phase, or rotational speed can be set by calculating the value, optimum control such as stopping the motor in case of overload and lowering the motor current in case of light load can be controlled. .

In the seventh invention, the control means in the first invention determines the load torque based on the output power and the drive frequency of the inverter circuit, and controls the phase of the output voltage and output current of the inverter circuit or the reactive current according to the load torque. The motor efficiency can be maximized by setting the current according to the load torque.

In the eighth invention, the control means in the first invention detects an abnormal rotation of the motor by the output voltage or the output power of the inverter circuit, changes the phase of the output voltage and the output current or the reactive current of the inverter circuit, The motor is restarted. When the motor stops rotating, the current can be set properly by increasing the output current and restarting, so that the motor can be stably rotated.

In the ninth invention, the control means in the first invention detects an abnormal rotation of the motor by the output voltage or the output power of the inverter circuit, changes the starting condition of the inverter circuit, and restarts the motor. It is possible to set the optimum starting condition according to the above, and to rotate stably.

(Example 1)

1 is a block diagram of a motor drive device according to the first embodiment. In Fig. 1, AC power is added to the rectifier circuit 2 from the AC power supply 1 and converted into DC power, and the inverter circuit 3 converts DC power into 3-phase AC power to drive the motor 4. do. The rectifier circuit 2 connects the capacitors 21a and 21b in series to the direct current output terminal of the full-wave rectifier circuit 20, connects the connection points of the capacitors 21a and 21b to one terminal of the AC power input, and provides a direct current. The voltage circuit is configured to increase the voltage applied to the inverter circuit 3.

The current output means 5 of the inverter circuit 3 is connected to the current detection means 5 to detect the current flowing in each of the three arm lower arms of the inverter circuit 3, thereby outputting the output current of the inverter circuit 3, That is, each phase current of the motor 4 is detected.

The control means 6 calculates the output current of the inverter circuit 3 from the output signal of the current detection means 5, and applies the predetermined frequency and the predetermined voltage according to the set rotation speed to rotate the motor 4 by rotation. The motor 4 can be rotationally driven at a set synchronous speed by controlling the motor load to be in phase with the output current or the reactive current according to the motor load.

Fig. 2 is a detailed circuit diagram of the inverter circuit 3 of the motor drive device, and is composed of a three-phase full bridge inverter circuit composed of six transistors and diodes. Here, the U-phase arm 30A which is one of the three-phase arms will be described. The parallel connection of the upper arm transistor 31a1 and the anti-parallel diode 32a1 made of an insulated gate bipolar transistor (hereinafter referred to as IGBT) and The parallel connection of the lower arm transistor 31a2 and the anti-parallel diode 32a2 made of IGBTs is connected in series, and the collector terminal of the upper arm transistor 31a1 is connected to the positive potential terminal Lp of the DC power supply. The emitter terminal of the transistor 31a1 is connected to the output terminal U, and the emitter terminal of the lower arm transistor 31a2 is the Ln terminal side of the DC power supply via the shunt resistor 50a constituting the current detecting means 5. Connect to

The upper arm transistor 31a1 is driven by the upper arm gate driving circuit 33a1 according to the upper arm driving signal Up, and the lower arm transistor 31a2 is driven to the lower arm gate driving circuit 33a2 according to the lower arm driving signal Un. By on and off switching control. The upper arm gate driving circuit 33a1 incorporates an RS flip-flop circuit set and reset by the differential signal to turn on the upper arm transistor 31a1 when the upper arm driving signal Up rises, and the upper arm driving signal Up The upper arm transistor 31a1 is turned off at the falling of. No RS flip-flop is required for the lower arm gate drive circuit.

The gate applied voltage of the IGBT is required to be 10 to 15 V. When the lower arm transistor 31a2 is turned on, the negative arm transistor 31a2 is turned on via the boost strap resistor 34a and the boost strap diode 35a from the + terminal B1 of the 15 V DC power supply. Since the strap capacitor 36a is charged, the upper arm transistor 31a can be switched on and off by the accumulated energy of the boost strap capacitor 36a. The boost strap capacitor 36a is similarly charged even when the antiparallel diode 32a2 of the lower arm is turned on.

The V-phase arm 30B and the W-phase arm 30C are the same connection, and the emitter terminals of the lower arm transistors of each arm are connected to the shunt resistors 50b and 50c constituting the current detecting means 5, and the shunt The other terminal of the resistors 50b and 50c is connected to the DC power supply negative potential terminal Ln. If the lower arm transistor is formed by IGBT or power MOSFET, switching control can be performed by controlling the gate voltage. Therefore, the voltage of the shunt resistor connected to the emitter terminal in the case of IGBT and the source terminal in the case of power MOSFET When the resistance value is selected to be 1 V or less, the on / off switching control can be controlled by voltage control with little effect on the switching operation, and the inverter circuit output current, by detecting the voltages veu, vev, vew of the shunt resistor, That is, the motor current can be detected.

Figure 3 is an inverter circuit output current detection timing chart, PWM control by triangular wave modulation, in order to reduce the influence of switching noise, high-speed A / D conversion to avoid the switching timing of the upper and lower arms IGBT to a motor control processor such as a microcomputer By detecting current. In FIG. 3, ck denotes a peak value of the triangular wave modulated signal Vt, that is, a synchronization signal generated at time t3, and vu is a U phase voltage control signal, and the triangular wave modulated signal Vt and the U phase voltage control signal vu are compared with each other. The drive signal Up of the arm transistor 31a and the drive signal Un of the U-phase lower arm transistor 31a2 are generated. The period t1 to t2 and the period t5 to t6 are referred to as dead time Δt in the non-conduction period of the upper and lower arm transistors, and the A / D conversion timing is the time t3 when the upper arm transistor is off and the lower arm transistor is on. It is good to carry out within the range of the time t4 to which dead time (DELTA) t time is left from t3.

Fig. 4 is a block diagram of a control means according to the present invention to realize sensorless sinusoidal drive by a high speed processor such as a microcomputer or a digital signal processor.

A basic control method will be described using a control vector diagram according to the present invention shown in FIG. Fig. 5 is a vector diagram of the dq coordinate system of the surface permanent magnet motor (abbreviated as SPM motor) in which the permanent magnet is provided on the rotor surface. The motor induced voltage Vr is coaxial with the q axis, and the induced voltage Vr is rotated with the induced voltage constant kr. It is proportional to the number N, i.e., the motor drive frequency f. In other words, the ratio (Vr / f) of the motor induced voltage Vr to the frequency f is controlled substantially constant.

Controlling the motor current I coaxially with the q-axis is equivalent to vector control, but assumes that the angle γ is advancing since the q-axis cannot be detected because there is no rotor position sensor. Since the voltage equation of the motor is represented by equation (1), when the driving frequency f is fixed, the motor applied voltage vector Vi is fixed by fixing the current vector I in the d-q coordinate system. In contrast, when the motor applied voltage vector Vi is fixed, the current vector I is fixed. The same applies to the case where coordinate transformation is performed on the a-r axis having the motor applied voltage Vi (the main axis) as the main axis. When the current vector I is fixed, the motor induced voltage vector Vr is fixed. In other words, if the motor constant is known in advance, the phase of the induced voltage Vr and the current I can be constantly controlled by fixing the current vector I, so that the q-axis current Iq (i.e., torque current) can be controlled almost constant. Almost the same control as vector control becomes possible.

By selecting the reactive current Isinφ to an appropriate value and decreasing the advance angle γ, the motor current I becomes almost equal to the torque current (q-axis current) Iq, enabling high efficiency operation and reducing the motor loss, thereby increasing the temperature of the motor. Can be reduced and the motor can be downsized.

In addition, in normal operation, as shown in FIG. 5, by setting the motor current I to γ, the phase γ with the q-axis is delayed even if the phase φ is changed due to a sudden load change, so that the torque decreases rapidly and there is no case of stepping out. . In particular, since the rotational speed suddenly decreases and the phase γ is delayed with respect to the q-axis, and the phase φ becomes 90 degrees or more, the possibility of demodulation becomes high, so that the case of delayed phase is reduced by advancing the control. Stable performance is improved.

In addition, since the field control (d-axis current is negative) is weakened by the advance control, the torque current can be reduced because the sum of the voltage vector Vo of the sum of the motor induced voltage Vr and the coil winding voltage jωLI can be reduced. Increasing Iq enables high speed rotation.

As described above, if the motor constant (winding resistance R, winding inductance L, motor induced voltage constant kr) and the torque current Iq corresponding to the motor load are known, the motor current I with respect to the motor applied voltage Vi to control the motor current vector. Since the absolute value of and the phase phi may be controlled, the r-axis current Ir (= Isinφ) or the a-axis current Ia (= Icosφ) after the bus axis coordinate transformation is controlled from the dq coordinate in the vector diagram of FIG. 5.

In FIG. 4, the drive condition setting means 60 obtains the drive rotation speed, the torque current, the true angle γ according to the motor driving condition, sets the drive frequency f, the reactive current Isinφ, and the like, and the rotation speed setting means 61. The set signal is sent to the reactive current setting means 62. The carrier signal generating means 63 generates a triangular wave signal Vt and a synchronization signal ck for PWM modulation, and the carrier frequency (switching frequency) is set to an ultrasonic frequency of 15 Hz or more in order to reduce motor noise. The synchronization signal ck is sent to each operation block, and each operation block operates in synchronization with the synchronization signal ck.

In order to set the motor drive frequency f, the rotation speed setting means 61 obtains the phase angle Δθ of the carrier signal period, adds it to the electric angle calculation means 64, and sends a set frequency signal to the V / f setting means 65. . The electric angle calculating means 64 obtains the phase θ in synchronization with the synchronous signal ck, and adds the phase signal θ to the storage means 66 or the coordinate conversion means for storing the normalized sine wave table.

The V / f setting means 65 sets a value corresponding to the rotational speed or the load torque by setting the applied voltage constant kvn corresponding to the drive frequency f and the load torque. As described later, when the washing operation in the forward rotation and the drainage operation in the reverse rotation are performed by the one motor two pumps or the one motor one pump system, the torque current required for the motor changes, respectively, so that the applied voltage constant kvn is determined. It is necessary to change the setting value by rotating and reverse rotation.

The storage means 66 stores in the storage area a standardized sinusoidal table necessary for calculating a trigonometric function corresponding to the phase angle, for example, sinusoidal data from -1 to +1 from phase 0 degrees to 360 degrees. Have

As shown in the timing chart of FIG. 3, the high-speed A / D conversion means 67 matches the output signals veu, vev, vew of the current detection means 5 to the inverter output current by the peak value of the triangular wave modulation signal Vt. A / D conversion is performed for several microseconds or less with the digital signals Iu, Iv, and Iw, and an instantaneous value of each phase current is added to the three-phase, two-phase, bus-axis conversion means 68.

As shown in Fig. 5, the three-phase / 2-phase bus-axis converting means 68 converts the instantaneous value of the inverter circuit output current by three-phase / 2-phase to convert the inverter circuit output voltage axis, that is, the motor bus axis (ar axis). By converting the coordinates to, the absolute transformation is performed using Equation 2 to obtain the a-axis component Ia and the r-axis component Ir. Ir corresponds to Isin φ and becomes an reactive current component when viewed from the inverter output (bus voltage). By the coordinate conversion, not only the reactive current component Ir is instantaneously obtained from the instantaneous output current value, but also the instantaneous output current vector Im can be obtained instantaneously from the squared average shown in equation (3). In addition, since the current phase φ seen from the inverter output (bus voltage) is instantaneously obtained from equation (4), the responsiveness is remarkably improved compared to the phase detection by providing the current zero cross detection means.

The reactive current comparing means 69 compares the output signal Ir of the three-phase / 2-phase busbar axis converting means 68 with the setting signal Irs of the reactive current setting means 62, outputs an error signal ΔIr, and outputs an error signal. The amplification calculating means 70 amplifies or integrates the applied voltage constant change signal kv to the control voltage comparison setting means 71.

The control voltage comparison setting means 71 generates an inverter output voltage control signal Va by comparing the output signal kvn of the V / f setting means 65 with the output signal kv of the error signal amplification calculating means 70, thereby generating a reactive current. By controlling the inverter output voltage so that the component Ir becomes a predetermined value, the inverter output voltage control signal Va is applied to the two-phase / 3-phase bus-axis inverse conversion means 72.

The two-phase / 3-phase bus-axis inverse transformation means 72 generates a three-phase sine wave voltage signal using the inverse transformation formula shown in equation (5). Since the inverter output voltage is in phase with the a-axis, only Va needs to be calculated, and the three-phase voltages vu, vv, and vw are output to the PWM control means 73.

The load state discriminating means 74 determines the motor load state by comparing the output signals of the V / f setting means 65 and the error signal amplifying calculating means 70. In normal operation, there is almost no difference between the output signal kvn of the V / f setting means 65 and the output signal kv of the error signal amplifying calculation means 70, and the motor is rotationally controlled so as to have a predetermined reactive current Isinφ. However, when the motor is out of phase and the motor stops rotating, as shown in the vector diagram of FIG. 6, the motor induced voltage Vr becomes 0. Therefore, the motor applied voltage becomes small so as to reach a predetermined reactive current Isinφ, and the V / f setting means. The difference between the output signal kvn of 65 and the output signal kv of the error signal amplifying calculation means 70 becomes much larger than the value of normal operation. That is, since the output signal kvn of the V / f setting means 65 is constant, the output signal kv of the error signal amplification calculating means 70 becomes smaller than the value of normal operation, so that the stop of the motor or abnormal rotation can be detected. Can be. The abnormal stop restart means 75 stops the operation of the inverter circuit 3 by determining an abnormality from the output signal of the load state determining means 74, and then restarts the motor. The drive condition change means 76 It restarts by changing the starting time set by the drive condition setting means 60, starting conditions, such as starting current, or reactive current Isinphi set value.

The motor may be out of phase when the torque current Iq is insufficient, the phase is delayed with respect to the q-axis, or the phase is excessively advanced, and eventually, when the reactive current set value Irs in the normal state is small, or starting up. Since the reactive current set value Irs at the time is small, when the step out determination is made, the reactive current set value Irs may be changed and restarted. At the time of start-up, if the start-up time is accelerated, the acceleration increases and a larger torque current is required. Therefore, one option is to change the start-up time and lengthen the start-up time.

7 shows timing charts of waveforms of respective parts by PWM control. Eu is a motor induced voltage waveform seen from the neutral point, and Iu is slightly progressing from the motor induced voltage Eu, which is a U-phase current waveform. vu, vv, vw are the PWM control input signals of the U phase, V phase, and W phase, that is, the output signals of the two-phase, three-phase, bus-axis inverse conversion means 72, and the PWM control by comparing with the triangular wave modulated signal Vt. Generate the output signal Up. The signal vu and the U phase output voltage phase are the same, and the phase of the U phase current Iu is delayed in phase φ from the signal vu.

8 is a flowchart showing the operation of the motor driving apparatus according to the present invention. In step 100, the motor drive program starts, and in step 101, various settings such as drive rotation speed, V / f setting, reactive current, and the like are made. Next, the process proceeds to step 102 to determine whether it is a start operation, and if it is a start operation, it proceeds to step 103 to execute a start control subroutine.

The start control subroutine 103 linearly raises the drive frequency f to the start time t1a from the rotational speed 0 to the set rotational speed fs as shown in the start control timing chart of FIG. Change the reactive current set value Irs according to f. In the case of a fluid load such as a pump or a fan, the torque is changed by the third power of the rotational speed. Therefore, the torque current Iq corresponding to the rotational speed is precisely determined by an experiment or the like. Stable starting is possible. However, at start-up, the torque current needs to be increased for acceleration, and the reactive current set value Irs must be set slightly larger than the value corresponding to the torque in order to prevent outage.

In the normal starting operation, the V / f set value and the reactive current set value Irs can be started even if they are started at the set values in the steady state.

In the case of starting out detection or abnormal rotation detection and restarting, the starting time is changed to t1b, and the starting torque is increased by making the reactive current set value Irsb larger than the initial set reactive current set value Irsa.

Next, the flow advances to step 104 to determine whether there is a carrier signal interrupt, and if there is a carrier signal interrupt, the carrier signal interrupt subroutine of step 105 and the rotation speed control subroutine of step 106 are executed.

9 is a flowchart of a carrier signal interrupt subroutine. The program starts from step 200, and in step 201 it is determined whether the count number k of carrier synchronization signal ck is the number of carriers kc in one cycle of the motor drive frequency f, and if it is the same, the process proceeds to step 202 to erase the carrier count number k. . The number of carriers kc in one cycle of the motor drive frequency f is obtained beforehand when the drive frequency is set.

For example, the driving frequency f at the rotational speed 4040 rpm of the 8-pole motor is 269.3 Hz, and the period T is 3.712 msec. When the carrier period Tc is 64 µsec (carrier frequency 15.6 Hz), the pulse number kc is 58. The phase Δθ of one carrier period Tc becomes Δθ = 2π / kc when one period of the driving frequency f is 2π.

In step 203, the number of counts of the carrier synchronization signals is incremented, and then, in step 204, the electric angle θ is calculated from the phase number? Of the number of carriers k and one carrier period Tc. Next, the flow advances to step 205 to detect a signal from the current detecting means 5 to detect inverter output currents Iu, Iv, and Iw. Next, the flow advances to step 206 to perform three-phase, two-phase and bus-axis coordinate transformation according to equation (2) to obtain the reactive current Ir and the effective current Ia, and proceeds to step 207 to store Ir and Ia.

Next, the process proceeds to step 208 to obtain the vector absolute value Im of the motor current by the following equation (3), and then proceeds to step 209 to determine whether or not the calculated value Im is greater than or equal to the overcurrent set value Imax.

If the calculated value Im is equal to or larger than the overcurrent set value Imax, the process proceeds to step 210, where the driving of the power semiconductor of the inverter circuit 3 is stopped to stop the motor driving, and the process proceeds to step 211 to emit an overcurrent abnormal flag.

If the calculated value Im is less than the overcurrent set value Imax, the process proceeds to step 212, in which the inverter output control signal Va from the rotation speed control subroutine is called, and then the process proceeds to step 213, according to equation (5), according to equation (5), two-phase / three-phase • The bus axis coordinate transformation is performed to obtain control signals vu, vv, and vw of each inverter. The control proceeds to step 214 to perform PWM control, and the control proceeds to step 215 and returns.

10 is a flow chart of the speed control subroutine. Since the rotation speed control subroutine does not necessarily need to be executed for each carrier signal, for example, it may be executed for every two carrier signals. When the carrier frequency becomes the ultrasonic frequency, the program processing time in the carrier period becomes a problem. Therefore, the process must be performed for each carrier such as phase calculation, current detection operation, or PWM control, and must be executed for each carrier shown in FIG. 10. It is possible to execute a sequential program such as a dish washing machine other than the motor control by dividing the unnecessary processing and dividing the processing not necessarily executed for each carrier into a plurality of processes.

The rotation speed control subroutine starts from step 300, calls up the drive frequency setpoint fs by step 301, and then proceeds to step 302 to call the reactive current setpoint Irs corresponding to the frequency setpoint fs, and step 303 Next, the reactive current Ir obtained from the three-phase / 2-phase / coaxial coordinate transformation is called, and the flow proceeds to step 304 to call the applied voltage constant set value V / f. Next, in step 305, Irs and Ir are compared to calculate the applied voltage constant kv from the error signal ΔIr. Then, in step 306, the difference Δkv between the applied voltage constant set value V / f and the applied voltage constant kv is calculated. Calculate Next, the flow proceeds to step 307 to calculate the bus-axis applied voltage signal Va from Δkv, and stores the Va. After the determination, the flagging flag is turned on, and the flow proceeds to step 310 to return the subroutine.

Returning to the motor drive program shown in FIG. 8 again, it is determined in step 107 whether there is a step out flag, and if there is a step out flag, the motor drive is stopped in step 108, and in step 109 the reactive current Isinφ is changed to step 110. Proceeds to execute the restart subroutine, and proceeds to step 111 to return the motor drive subroutine.

(Example 2)

Hereinafter, Embodiment 2 of the present invention will be described with reference to FIGS. 12, 13, 14, and 15.

12 is a cross-sectional view of a motor driving apparatus of the dish washing machine in the second embodiment of the present invention. Pump motor is 1 motor 1 pump type.

Tap water is supplied to the washing tank 7 from the water supply valve 8, and the washing water 9 is stored in the washing tank 7. A flat DC brushless motor 4a is disposed below the cleaning tank 7 so that the axial direction is vertical, a pump case 10 is disposed below the motor 4a, and the impeller 11 is rotated. Thereby applying pressure from the axial direction to the centrifugal direction.

When rotating in the forward rotation direction, the washing water is sprayed on the dish (not shown) from the powder 12b having the spray nozzle 12a to clean it. When the motor is rotated forward, the internal pressure of the pump case 10 is increased, and the drain valve 13 provided on the side of the pump case 10 is closed, so that the direction of water flow is toward the fine particle 12b. When the impeller 11 is rotated in reverse, pressure is applied from the side of the impeller 11 in the vertical direction, and the drain valve 13 is opened, so that the water flow in the vertical direction flows in the direction of the drain pipe 14, so it is cleaned with one motor and one pump. And drainage becomes possible.

Even in the case of the one-motor two-pump system in which an impeller and a pump case are provided for washing and draining, respectively, it is possible to drain in the forward rotation and reverse rotation, but the height of the pump is increased and the lower volume of the washing tank 7 is achieved. There is a problem that cannot be made small.

Since the efficiency of the drain pump is very poor in the one-motor one-pump system, the noise when air enters after draining the washing water in the drainage operation is large, and the drain valve 13 In the case of not being completely closed, there are two major problems that the washing water disappears by slightly draining the washing water so that the heating heater (not shown) of the washing water is operated even without water.

Since the water of the pump disappears and the load becomes drastically light, both of them can be solved by changing the motor rotation speed by detecting the motor input or the load torque, or by stopping and feeding the motor.

Fig. 13 is a block diagram of control means of the motor drive device according to the second embodiment. The basic idea of the present invention is to calculate the load state or the load torque from the inverter output, that is, the motor input, and the motor input pin is expressed by the product of the inverter output voltage Va and the motor current I and cosφ, and the motor efficiency η When multiplying, it becomes the motor output Po, motor efficiency (eta) is determined roughly by rotation speed, motor torque T is represented by the product of torque constant kt and torque current Iq, and Formula (6) is established.

That is, the torque current Iq can be calculated from equation (6) as long as the rotational speed ωr and the motor input (= Va · Icosφ) are known. Since Ia = Icos phi and obtained from Equation 2, the motor torque can always be calculated by calculation even if the phase shift from the q-axis is not known. Since the torque current Iq can be calculated, the phase shift from the q-axis can be estimated in reverse.

FIG. 13 is a partial change of the block diagram of FIG. 4, and only the changed part will be described. FIG. Other configurations are the same as those in the first embodiment, and the operation and action thereof are also the same, so detailed description thereof is omitted.

The inverter output, i.e., the motor input, is calculated by adding the applied voltage signal Va and the a-axis signal Ia of the three-phase, two-phase, bus-shaft axis converting means 68 to the output power calculating means 77 of the inverter, and thereby determining the load state. A motor input signal and a drive frequency signal f are added to 74a to calculate the load torque to determine the motor load state. If the drive frequency is constant, the load variation can be determined from the motor input. In order to reduce noise by reducing the load torque by reducing the rotational speed, a control signal is sent to the driving condition changing means 78, and the rotational speed signal from the driving condition changing means 78 to the driving condition setting means 60a. To control the set speed. The drive condition changing means 78 not only changes the drive rotational speed, but also changes the reactive current Isinφ or various conditions at startup.

Fig. 14 shows changes in the motor input W and the motor rotation speed N as the time elapses at the start of drainage operation of the dishwasher. The motor input when the washing water is accumulated in the washing tank is almost constant, The motor input or torque drops sharply. Therefore, it is possible to detect the air inflow from the motor input or the torque change, and when the predetermined value falls from the constant drain output W1 (W3), it is determined as the air inflow, and the rotation speed is decreased from N1 to N2 at time t3. As a result, the noise during inflow of drainage air can be reduced.

Fig. 15 shows the change in the motor input W with the passage of time when the drain valve of the dishwasher is not completely closed. When the washing pump motor rotates, washing water leaks from the drain valve, causing air inflow, and the motor load becomes light. Therefore, when the inverter output, that is, the motor input is smaller than the predetermined value Wd, the washing water is detected to stop the driving of the motor and supply water. Do

16 is an embodiment of a rotation speed control subroutine that detects a change in motor input. Steps 300 to 306a are basically the same as those in FIG. 10 in the first embodiment, and description thereof is omitted. In step 311, the effective current Ia obtained from the carrier signal interrupt subroutine is called, and then the flow proceeds to step 312 to obtain the motor input pin, and then the flow proceeds to step 313 to determine the motor input drop, and the motor input is lowered by the predetermined value. If so, return the subroutine with the fallback flag turned on.

When the motor input drop flag is turned on, it is determined that air is introduced, and when the washing operation is performed, the motor driving is stopped to supply the replenishment water. If air inflow is detected even after several times of operation, it is possible to prevent the heater from operating when there is no water by determining abnormality and stopping the operation, or by operating the motor by rotating the motor to remove garbage from the drain valve. have.

In the case of drainage operation, the noise during drainage can be reduced by determining the air inflow from the predetermined input drop and controlling the rotation speed.

Since the present invention can instantly detect motor load changes such as motor input and torque fluctuations, it is also suitable for detecting an imbalance in clothing by detecting torque fluctuations during dehydration operation of a washing machine or a laundry dryer.

(Example 3)

Fig. 17 is a block diagram of the control means of the motor drive device according to the third embodiment, which detects the motor torque and controls the reactive current Isinφ so as to achieve maximum efficiency.

FIG. 17 is a partial modification of the block diagram of FIG. 13 according to the second embodiment, and only the changed portion is described below. The output power calculating means 77 of the inverter calculates the inverter output power, that is, the motor input from the inverter output voltage Va and the active current Ia, and adds the motor input signal and the driving frequency signal to the torque current calculating means 79 to supply the motor. The torque current Iq is obtained from equation (6).

The effective current calculating means 80 obtains the motor current vector absolute value Im by equation (2), compares the torque current Iq and the motor current vector absolute value Im by the motor current comparison means 81, and compares the difference between Im and Iq. The reactive current changing means 82 changes the reactive current set value Isinφ in accordance with the signal of. If Im increases with respect to Iq, the setting value of Isinφ decreases, and if Im decreases with respect to Iq, the setting value of Isinφ is increased, so that Im and Iq are controlled to be almost the same. Since Isin phi is controlled so that I and Iq shown in FIG. 5 become substantially the same, it means that I becomes coaxial with a q-axis, and maximum efficiency operation like vector control is attained.

In order to control the motor current vector absolute value Im and the torque current Iq to be almost the same, it becomes difficult to control during high torque operation at the start, so that the control loop is executed after stopping at the start and the rotation speed becomes substantially constant. Is preferred. Moreover, it is more suitable for the control of the compressor of a air conditioner, the control of a fan motor, etc., or the drum rotation control of a drum type washing machine than the dish washing pump motor with a large load torque fluctuation.

As described above, the present invention is a DC brushless motor by detecting a three-phase inverter output current, coordinate conversion of the inverter circuit output voltage busbar axis after the three-phase / two-phase conversion to control the motor reactive current or current phase The sensorless sine wave driving of the permanent magnet synchronous motor is enabled, and the load state is detected from the inverter output power or the output voltage.

According to the present invention, the inverter output power, i.e., the motor input or the load state of the motor can be detected in an instant and control equivalent to the vector control can be performed, so that the optimum efficiency according to the maximum efficiency operation or the load can be achieved.

In addition, the instantaneous load fluctuation can be detected, so that the inflow of the pump or the unbalance of the load such as the dehydration of the washing machine and the washing tank or the rotating drum can be detected from the fluctuation of the motor torque.

In addition, since the removal of the motor is easy to be detected, the rotation of the motor can be stabilized by notifying abnormality or restarting by changing the driving conditions such as the reactive current.

In addition, in the conventional sensorless sinusoidal drive, computation for position estimation becomes complicated, and the burden on the processor is large, and various tests for obtaining motor parameters required for position estimation computation require time, but according to the present invention, Since the estimation is unnecessary, the computational steps of the processor can be reduced, the number of bits of the computation data can be reduced, the motor parameters are hardly required, and the maximum efficiency operation can be performed automatically. Control equivalent to the above can be achieved, and a motor driving device capable of driving a sensorless sinusoidal wave can be realized.

In particular, the motor control and the sequence control of the laundry dryer or dishwasher require complicated programs, and the carrier frequency needs to be reduced to an ultrasonic frequency to reduce noise. Although the burden on program capacity and arithmetic performance was so great that an expensive processor was required, according to the present invention, an inexpensive processor can achieve the same performance as the sensorless vector control. It can be realized.

Moreover, although this invention mainly demonstrated the SPM motor, it is clear that it is applicable to the IPM motor which embedded the permanent magnet in the iron core rotor.

In addition, even if the inverter output voltage and the phase of the output current or the effective current Icosφ are constantly controlled, the equivalent effect can be obtained.

As described above, the motor drive device according to the present invention converts AC power into DC power by a rectifier circuit, drives the motor by an inverter circuit, detects the output current of the inverter circuit by current detection means, and rotates the setting. PWM control of the inverter circuit to the number, control of the inverter circuit output voltage overcurrent phase or reactive current to a predetermined value, and to determine the load state from the inverter circuit output voltage or output power, It is easy to detect torque fluctuations, and is applicable not only to the pump motor of the dishwasher shown in the embodiment, but also to the compressor motor, fan motor of an air conditioner, or the dehydration of a washing machine or a laundry dryer, and the rotation control of a washing tank or a rotating drum. can do.

The motor driving apparatus of the present invention is to determine the motor load state by the output voltage or the output power of the inverter circuit at the set rotation speed, and can detect the step out and the load torque without the position sensor, Even if it is possible to stably restart and set the current according to the load, it is possible to realize the sensorless sinusoidal drive which operates stably even if the load fluctuation is large.

Claims (9)

With AC power, A rectifier circuit for converting AC power of the AC power source into DC power; An inverter circuit for converting DC power of the rectifier circuit into AC power; A motor driven by the inverter circuit, Current detecting means for detecting an output current of the inverter circuit; A control means for controlling the motor to set a set speed by PWM controlling the inverter circuit by the output signal of the current detecting means, The control means controls the output voltage of the inverter circuit and the phase of the output current or the reactive current to be a predetermined value, and determines the load state by the output voltage or output power of the inverter circuit. Motor-drive unit. The method of claim 1, The control means is a motor drive device which controls V / f so that the output voltage of an inverter circuit, the phase of an output current, or reactive current may become predetermined value, and determines a load state from a V / f control value. The method of claim 1, The control means detects the motor load state by the output voltage or the output power of the inverter circuit, and changes the phase of the output voltage and the output current or the reactive current of the inverter circuit. The method of claim 1, The control means detects the motor load state by the output voltage or the output power of the inverter circuit, and changes the motor driving conditions. The method of claim 1, The control means is a motor drive device to determine the motor load torque based on the output power and the drive frequency of the inverter circuit. The method of claim 1, The control means determines the load torque based on the output power and the drive frequency of the inverter circuit, and controls the motor by the variation of the load torque. The method of claim 1, The control means determines the load torque based on the output power and the driving frequency of the inverter circuit, and controls the phase of the output voltage and output current of the inverter circuit or the reactive current according to the load torque. The method of claim 1, The control means detects an abnormal rotation of the motor by the output voltage or the output power of the inverter circuit, and restarts the motor by changing the output voltage and output current of the inverter circuit or the reactive current. The method of claim 1, The control means detects an abnormality in rotation of the motor by the output voltage or the output power of the inverter circuit, and changes the starting condition of the inverter circuit to restart the motor.

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