JP2005253163A - Motor driving unit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To obtain the same effect as vector control in simplified sensorless sine wave drive. <P>SOLUTION: This motor driving unit converts AC power 1 into DC power by a rectifying circuit 2, and drives a motor 4 by an inverter circuit 3, and detects the output current of the inverter circuit 3 by a current detection means 5, and controls a reactive current to a specified value at a set number of revolutions, and controls the phase of a motor current by the reactance voltage or the reactance power of the motor. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a motor drive device that performs sensorless sine wave drive.

Conventionally, this type of motor driving device has reduced the vibration and noise of the motor and improved the reliability by driving the sensorless sine wave without the rotor position sensor (see, for example, Patent Document 1).
JP 2000-350489 A

  However, in the conventional configuration, in order to estimate the rotor position, a motor constant, a circuit parameter, or a motor load is grasped in advance, and an error between a predetermined calculated value and a measured current value is detected to minimize the error. Since the processor needs to perform calculations, the calculation is very complicated, and a processor having a high-speed and high-performance calculation function is required. Furthermore, there is a problem that the motor can easily be stepped out when the motor load fluctuation is large.

  The present invention solves the above-described conventional problems, and is a sensorless sine wave that operates stably with respect to load fluctuations, simplifies the calculation of the processor, and controls the maximum motor current or the optimum motor current phase. A motor driving device for driving is provided.

  In order to solve the above-mentioned conventional problems, the motor drive device converts AC power into DC power by a rectifier circuit, drives the motor by an inverter circuit, detects the output current of the inverter circuit by current detection means, and sets rotation. The inverter circuit is PWM controlled so that the output voltage and current phase of the inverter circuit and the reactive current are controlled so that the reactive current becomes a predetermined value, and the motor current phase is controlled by the reactance voltage or reactance power of the motor. It is what you do.

  The motor driving device of the present invention controls the motor current phase by the reactance voltage or reactance power of the motor, and can control the motor induced voltage and current phase optimally. Omitted, the processor processing can be reduced, and maximum efficiency operation equivalent to sensorless vector control or field-weakening control is possible.

  A first invention is an AC power source, a rectifier circuit that converts AC power of the AC power source into DC power, an inverter circuit that converts DC power of the rectifier circuit into AC power, and a motor driven by the inverter circuit And current detecting means for detecting the output current of the inverter circuit, and control means for controlling the motor so that the inverter circuit is PWM-controlled by the output signal of the current detecting means to achieve a set rotational speed, The control means controls the phase of the output voltage and output current of the inverter circuit or the reactive current to be a predetermined value, and controls the motor current phase by the reactance voltage or reactance power of the motor. Thus, the arithmetic processing of the processor is reduced, and the maximum efficiency operation or field weakening control becomes possible.

  According to a second aspect of the invention, the control means in the first aspect of the invention compares the reactive voltage of the inverter circuit with the reactance voltage of the motor to control the motor current phase, and the motor reactance with respect to the reactive voltage of the inverter circuit. The motor current phase can be controlled to a desired value by controlling the voltage ratio.

  In the third aspect of the invention, the control means in the second aspect of the invention is such that the reactive voltage of the inverter circuit and the reactance voltage of the motor are substantially equal, so that the motor current and the induced voltage phase can be made substantially the same, and maximum efficiency operation is possible. It becomes.

  According to a fourth invention, the control means in the first invention controls the motor current phase by comparing the reactive power of the inverter circuit and the reactance power of the motor, and the reactance of the motor with respect to the reactive power of the inverter circuit. The motor current phase can be controlled to a desired value by controlling the power ratio.

  In the fifth aspect of the invention, the control means in the fourth aspect of the invention is such that the reactive power of the inverter circuit and the reactance power of the motor are substantially equal, and the motor current and the induced voltage phase can be made substantially the same, and maximum efficiency operation is possible. It becomes.

  In the sixth aspect of the invention, the control means in the first aspect of the invention controls the reactance voltage or the reactance power of the motor to be a set value, and the current phase with respect to the motor induced voltage can be controlled to a desired value. Maximum efficiency operation or field weakening control is possible.

  In the seventh aspect of the invention, the control means in the first aspect of the invention controls the motor current phase after starting the motor. At the time of starting, the phase of the output voltage and the output current of the inverter circuit or the reactive current corresponds to the starting. By controlling to the predetermined value, the starting stability can be improved, the motor current phase can be controlled to a desired value after the starting, and the maximum efficiency operation and the field weakening control can be performed.

(Embodiment 1)
FIG. 1 is a block diagram of the motor drive device according to the first embodiment. In FIG. 1, AC power is applied to a rectifier circuit 2 from an AC power source 1 to convert it into DC power, and the inverter circuit 3 converts DC power into three-phase AC power to drive a motor 4. In the rectifier circuit 2, capacitors 21a and 21b are connected in series to the DC output terminal of the full-wave rectifier circuit 20, and a connection point of the capacitors 21a and 21b is connected to one terminal of the AC power supply input to constitute a DC voltage doubler circuit. The voltage applied to the inverter circuit 3 is increased.

  The current detection means 5 is connected to the negative voltage side of the inverter circuit 3, and the current flowing through the lower arms of the three phases of the inverter circuit 3 is detected, whereby the output current of the inverter circuit 3, that is, the phase current of the motor 4 is detected. To detect.

  The control means 6 calculates the output current of the inverter circuit 3 from the output signal of the current detection means 5, applies a predetermined frequency and a predetermined voltage according to the set rotational speed, and drives the motor 4 to rotate. Accordingly, the motor 4 can be driven to rotate at a set synchronous speed by controlling the phase so as to be a reactive current or phase with respect to the output voltage.

  FIG. 2 is a detailed circuit diagram of the inverter circuit 3 of the motor drive device, which is constituted by a three-phase full-bridge inverter circuit comprising six transistors and diodes. Here, one U-phase arm 30A of the three-phase arm will be described. A parallel connection body of an upper arm transistor 31a1 made of an insulated gate bipolar transistor (hereinafter abbreviated as IGBT) and an antiparallel diode 32a1, and a lower arm made of IGBT. A parallel connection body of the transistor 31a2 and the antiparallel diode 32a2 is connected in series, the collector terminal of the upper arm transistor 31a1 is connected to the positive potential terminal Lp of the DC power supply, and the emitter terminal of the upper arm transistor 31a1 is connected to the output terminal U. The emitter terminal of the lower arm transistor 31a2 is connected to the Ln terminal side of the DC power supply via the shunt resistor 50a constituting the current detecting means 5.

  The upper arm transistor 31a1 is driven by the upper arm gate drive circuit 33a1 in accordance with the upper arm drive signal Up, and the lower arm transistor 31a2 is subjected to on / off switching control by the lower arm gate drive circuit 33a2 in accordance with the lower arm drive signal Un. The upper arm gate drive circuit 33a1 incorporates an RS flip-flop circuit that is set and reset by a differential signal, turns on the upper arm transistor 31a1 at the rise of the upper arm drive signal Up, and rises at the fall of the upper arm drive signal Up. The arm transistor 31a1 is turned off. The RS flip-flop is not necessary for the lower arm gate driving circuit.

  The gate application voltage of the IGBT needs 10 to 15V. When the lower arm transistor 31a2 is turned on, the bootstrap capacitor 36a is charged from the + terminal B1 of the 15V DC power supply via the bootstrap resistor 34a and the bootstrap diode 35a. Therefore, the upper arm transistor 31a can be switched on and off by the energy stored in the bootstrap capacitor 36a. Similarly, the bootstrap capacitor 36a is charged when the lower arm antiparallel diode 32a2 is turned on.

  The V-phase arm 30B and the W-phase arm 30C are similarly connected, and the emitter terminals of the lower arm transistors of each arm are connected to the shunt resistors 50b and 50c constituting the current detecting means 5, and the other of the shunt resistors 50b and 50c is connected. The terminal is connected to the DC power source negative potential terminal Ln. If the lower arm transistor is constituted by an IGBT or a power MOSFET, switching can be controlled by controlling the gate voltage. Therefore, the voltage of the shunt resistor connected to the emitter terminal in the case of IGBT and the source terminal in the case of power MOSFET is 1V or less. If the resistance value is selected, the on / off switching control can be performed by voltage control with little influence on the switching operation, and the inverter circuit output current, that is, the motor current can be determined by detecting the shunt resistance voltages veu, vev, and vew. There are features that can be detected.

  FIG. 3 is an inverter circuit output current detection timing chart. PWM control is performed by triangular wave modulation, and the switching timing of the upper and lower arm IGBTs is removed to reduce the influence of switching noise. Current detection is performed by a motor control processor. In FIG. 3, ck is a peak value of the triangular wave modulation signal Vt, that is, a synchronization signal generated at time t3, vu is a U-phase voltage control signal, and the triangular wave modulation signal Vt and the U-phase voltage control signal vu are compared. A drive signal Up for the U-phase upper arm transistor 31a and a drive signal Un for the U-phase lower arm transistor 31a2 are generated. The period between t1 and t2 and the period between t5 and t6 is called the dead time Δt in the non-conducting period of the upper and lower arm transistors, and the A / D conversion timing is the time t3 when the upper arm transistor is off and the lower arm transistor is on, It may be performed within a range of time t4 that is shifted from t3 by dead time Δt.

  FIG. 4 is a block diagram of the control means of the motor drive device according to the present invention, which realizes sensorless sine wave drive by a high speed processor such as a microcomputer or a digital signal processor.

  A basic control method will be described with reference to a control vector diagram according to the present invention shown in FIG. FIG. 5 is a vector diagram of the dq coordinate system of a surface permanent magnet motor (abbreviated as SPM motor) having a permanent magnet on the rotor surface. The motor induced voltage Vr is coaxial with the q axis, and the induced voltage Vr is induced. It is proportional to the voltage constant kr and the rotation speed N, that is, the motor drive frequency f. In other words, the ratio (Vr / f) between the motor induced voltage Vr and the frequency f is controlled to be substantially constant.

  If the motor current I is controlled coaxially with the q-axis, it is equivalent to vector control. However, since there is no rotor position sensor and the q-axis cannot be detected, it is assumed that the angle γ is advanced. Since the voltage equation of the motor is expressed by Equation 1, when the drive frequency f is fixed, when the current vector I is fixed in the dq coordinate system, the motor applied voltage vector Va is fixed. Conversely, when the motor applied voltage vector Va is fixed, the current vector I is fixed. The same applies to the case where the coordinate conversion is performed on the a-r axis with the motor applied voltage Va (bus axis) as the main axis. When the current vector I is fixed, the motor induced voltage vector Vr is fixed. In other words, if the motor constant is known in advance, the phase of the induced voltage Vr and the current I can be controlled to be constant by fixing the current vector I, so that the q-axis current Iq (that is, the torque current) can be controlled to be substantially constant. Almost the same control as vector control is possible.

  As shown in FIG. 6, by controlling the reactive current Isinφ to an appropriate value so that the reactive voltage component Va · sinφ of the applied voltage is substantially the same as the reactance voltage ωLI, the induced voltage phase and the motor current phase are substantially the same. Then, the motor current I becomes the same as the torque current (q-axis current) Iq, enabling high-efficiency operation and reducing motor loss, thereby reducing the temperature rise of the motor and miniaturizing the motor.

  Also, as shown in FIG. 5, the motor current I is set to advance by controlling the reactive current Isinφ (= Ir) so that the reactance voltage ωLI is larger than the reactive voltage component Va · sinφ of the applied voltage. Therefore, even if the phase φ changes due to a sudden load fluctuation, the phase γ with the q-axis is delayed and the torque is suddenly reduced to prevent step-out. In particular, if the rotational speed is suddenly decreased and the phase γ is delayed with respect to the q-axis and the phase φ is 90 degrees or more, the possibility of step-out increases. The number of cases is reduced, and the stability performance of the rotation control is improved.

  Further, since the field-weakening control (d-axis current is negative) is achieved by the advance angle control, the torque current Iq can be increased to enable high-speed rotation.

  As described above, if the motor constants (winding resistance R, winding inductance L) are known, the reactive voltage component Va · sinφ and reactance voltage ωLI of the applied voltage are calculated in order to control the motor current vector. Thus, the absolute value and phase φ of the motor current I with respect to the motor applied voltage Va may be controlled.

  In FIG. 4, the drive condition setting means 60 obtains the drive rotation speed, torque current, and advance angle γ according to the motor drive conditions, and sets the drive frequency f, the reactive current Isinφ, etc. Then, a setting signal is sent to the reactive current setting means 62. The carrier signal generator 63 generates a triangular wave signal Vt and a synchronization signal ck for PWM modulation, and the carrier frequency (switching frequency) is set to an ultrasonic frequency of several kHz to 15 kHz or more in order to reduce motor noise. . The synchronization signal ck is sent to each computation block, and each computation block operates in synchronization with the synchronization signal ck.

  The rotation speed setting means 61 obtains the phase angle Δθ of the carrier signal period in order to set the motor drive frequency f, sends the set frequency signal to the V / f setting means 65 in addition to the electrical angle calculation means 64. The electrical angle calculation means 64 obtains the phase θ in synchronization with the synchronization signal ck, and adds the phase signal θ to the storage means 66, the coordinate conversion means, etc. that store the standardized sine wave table.

  The V / f setting means 65 sets an applied voltage constant kvn corresponding to the drive frequency f and the load torque, and a value corresponding to the rotational speed or the load torque is set.

  The storage means 66 stores a standardized sine wave table necessary for performing a trigonometric function corresponding to the phase angle in the storage area. For example, the phase is 0 to 360 degrees, and −1 to +1. I have sine wave data.

  As shown in the timing chart of FIG. 3, the high-speed A / D conversion means 67 converts the output signals veu, vev and vew of the current detection means 5 into the digital signal Iu corresponding to the inverter output current at the peak value of the triangular wave modulation signal Vt. , Iv and Iw are A / D converted within a few microseconds or less, and instantaneous values of the respective phase currents are added to the three-phase / 2-phase / bus-axis converting means 68.

  As shown in FIG. 5, the three-phase / two-phase / busbar axis conversion means 68 converts the instantaneous value of the inverter circuit output current into three-phase / two-phase to convert the inverter circuit output voltage axis, that is, the motor busbar axis (ar The coordinate is converted to (axis), and absolute conversion is performed using Equation 2 to obtain the a-axis component Ia and the r-axis component Ir. Ir corresponds to Isinφ and becomes a reactive current component when viewed from the inverter output (bus voltage). By performing coordinate conversion, not only the reactive current component Ir can be obtained instantaneously from the output current instantaneous value, but also the output current vector absolute value I can be obtained instantaneously by the root mean square shown in Equation 3. Further, since the current phase φ or the reactive current Ir seen from the inverter output (bus voltage) can be detected at high speed in synchronization with the carrier cycle, the response is significantly improved compared to the case where the phase is detected by providing the current zero cross detection means. .

  The reactive current comparing means 69 compares the output signal Ir of the three-phase / two-phase / bus axis converting means 68 with the setting signal Irs of the reactive current setting means 62 and outputs an error signal ΔIr, which is amplified by the error signal amplification calculating means 70. Alternatively, the applied voltage constant change signal kv is integrated and output to the control voltage comparison setting means 71.

  The control voltage comparison setting unit 71 compares the output signal kvn of the V / f setting unit 65 with the output signal kv of the error signal amplification calculation unit 70 to generate the inverter output voltage control signal Va. The reactive voltage component Ir is The inverter output voltage is controlled so as to be a predetermined value, and the inverter output voltage control signal Va is applied to the 2-phase / 3-phase / bus shaft reverse conversion means 72.

  The two-phase / three-phase / bus axis reverse conversion means 72 generates a three-phase sinusoidal voltage signal using the reverse conversion formula shown in Formula 4. Since the inverter output voltage is in phase with the a-axis, only Va needs to be calculated, and the three-phase voltages vu, vv, vw are output to the PWM control means 73.

  The start control means 74 changes the drive frequency, phase, or reactive current Isinφ (= Ir) setting conditions only at start-up, and adds a set signal to the drive condition setting means 60 to increase the drive frequency with time at start-up. And set the current required for acceleration. At startup, the driving frequency is forcibly increased with time according to the load, and the reactive current Ir is set to a constant value or a value corresponding to the acceleration, thereby enabling stable startup. Among them, it has the best startup stability.

  The current calculation means 75 obtains the output current vector absolute value I from Ia and Ir by the root mean square shown in Equation 3.

  The reactive voltage calculation means 76 calculates the reactive voltage component Va · sinφ of the inverter output voltage Va. Since sinφ can be calculated by Equation 5, the inverter reactive voltage can be calculated from Va · Ir / I.

  The reactance voltage calculation means 77 calculates the voltage drop ωLI from the drive frequency f, the motor inductance L, and the current I.

  The voltage comparison means 78 compares the reactive voltage Va · sinφ of the inverter output voltage Va with the reactance voltage ωLI, adds the output signal of the voltage comparison means 78 to the power factor changing means 79, and according to the signal of the difference in reactive voltage. A signal is applied to the drive condition setting means 60 to control the reactive current set value Irs. As shown in Expression 6, when the reactive voltage Va · sinφ of the inverter output voltage Va is controlled to be equal to the reactance voltage ωLI, the motor current is reduced and the maximum efficiency operation is possible.

  Further, when the reactance voltage ωLI is increased with respect to the reactive voltage Va · sinφ, the advance angle control is performed. Therefore, the advance angle control can be performed by setting the ratio of the reactance voltage ωLI to the reactive voltage Va · sinφ to 1 or more.

  FIG. 7 shows a timing chart of each part waveform by PWM control. Eu is a motor-induced voltage waveform as viewed from the neutral point, and Iu has a U-phase current waveform and is slightly advanced from the motor-induced voltage Eu. vu, vv, and vw are PWM control input signals of U-phase, V-phase, and W-phase, that is, output signals of the 2-phase / 3-phase / bus-axis reverse conversion means 72 and compared with the triangular wave modulation signal Vt. An output signal Up is generated. The signal vu and the U-phase output voltage phase are the same, and the phase of the U-phase current Iu is delayed from the signal vu by the phase φ.

  FIG. 8 is a flowchart showing the operation of the motor driving apparatus according to the present invention. In step 100, the motor drive program starts. In step 101, various settings such as drive speed, V / f setting, and reactive current are performed. Next, the routine proceeds to step 102, where it is determined whether or not it is a startup operation. If it is a startup operation, the routine proceeds to step 103 and the startup control subroutine is executed.

  The start-up control subroutine 103 linearly increases the drive frequency f from zero rotation speed to the set rotation speed (drive frequency fs), and changes the reactive current set value Irs according to the drive frequency f. In the case of a fluid load such as a pump or a fan, the torque varies depending on the cube of the rotational speed. Strictly speaking, it is assumed that the torque phase Iq corresponding to the rotational speed is obtained by experiments and the rotor phase lags behind the rotating magnetic field. Thus, stable startup is possible by calculating Isinφ and controlling the startup. At startup, the torque current needs to be increased for acceleration, and the reactive current set value Irs needs to be set larger than the value corresponding to the torque in order to prevent step-out.

  The drive system according to the present invention has good start-up stability, and can often be started without greatly changing the V / f set value and the reactive current set value Irs.

  Next, the routine proceeds to step 104, where it is determined whether there is a carrier signal interrupt. If there is a carrier signal interrupt, the carrier signal interrupt subroutine at step 105 and the rotation speed control subroutine at step 106 are executed.

  FIG. 9 is a flowchart of the carrier signal interrupt subroutine. The program starts from step 200. In step 201, it is determined whether or not the count number k of the carrier synchronization signal ck is the number of carriers kc within one cycle of the motor drive frequency f. Clear. The carrier number kc within one cycle of the motor drive frequency f is obtained in advance when the drive frequency is set.

  For example, when the drive frequency f of an 8-pole motor at a rotational speed of 4040 rpm is 269.3 Hz, the period T is 3.712 msec, and the carrier period Tc is 64 μsec (carrier frequency 15.6 kHz), the pulse number kc is 58. The phase Δθ of one carrier period Tc is Δθ = 2π / kc, where one period of the driving frequency f is 2π.

  In step 203, the count number of the carrier synchronization signal is incremented, and then the process proceeds to step 204, where the electrical angle θ is calculated from the carrier number k and the phase Δθ of one carrier cycle Tc. Next, the routine proceeds to step 205 where signals from the current detection means 5 are detected to detect inverter output currents Iu, Iv, Iw. Next, the process proceeds to step 206, where the three-phase / two-phase / bus axis coordinate conversion is performed according to the equation 2, the reactive current Ir and the effective current Ia are obtained, and the process proceeds to step 207 to store Ir and Ia.

  Next, the routine proceeds to step 208, where the absolute vector value I of the motor current is obtained from Equation 3, and then the routine proceeds to step 209, where sin φ is obtained from the calculated values I and Ir.

  Next, the process proceeds to step 213, and 2-phase / 3-phase / bus axis coordinate conversion is performed in accordance with Equation 5 to obtain inverter phase control signals vu, vv, vw. The process proceeds to step 214 to perform PWM control, and the process proceeds to step 215. Return at.

  FIG. 10 is a flowchart of the rotation speed control subroutine. Since the rotation speed control subroutine does not necessarily have to be executed for each carrier signal, it may be executed for every two carrier signals, for example. When the carrier frequency becomes an ultrasonic frequency, the program processing time within the carrier period becomes a problem. Therefore, processing that is always executed for each carrier, such as phase calculation, current detection calculation, or PWM control, coordinate conversion, and that shown in FIG. A sequence program other than motor control can be executed by dividing a process that is not necessarily executed for each carrier and dividing a process that is not necessarily executed for each carrier into a plurality of processes.

  The rotation speed control subroutine is started from step 300, the drive frequency set value fs is called in step 301, and then the process proceeds to step 302 to call the reactive current set value Irs corresponding to the frequency set value fs. The reactive current Ir obtained from the three-phase / 2-phase / bus axis coordinate conversion is called, and the process proceeds to step 304 to call the applied voltage constant set value V / f. Next, the routine proceeds to step 305, where Irs and Ir are compared, and the applied voltage constant kv is calculated from the error signal ΔIr. Next, the routine proceeds to step 306 where the difference Δkv between the applied voltage constant set value V / f and the applied voltage constant kv is calculated. Calculate. Next, the routine proceeds to step 307, where the bus axis applied voltage signal Va is calculated from Δkv, and Va is stored, and the routine proceeds to step 308 where the subroutine is returned.

  Returning to the motor drive program shown in FIG. 8 again, the presence or absence of the start flag is determined in step 107. If there is no start flag, the process proceeds to step 108 to calculate the invalid voltage Va · sinφ of the inverter output voltage Va. Proceeding to step 109, the reactance voltage ωLI of the motor is calculated, and then proceeding to step 110, the difference or ratio between the reactive voltage Va · sinφ of the inverter output voltage Va and the reactance voltage ωLI of the motor is calculated. Then, the power factor changing subroutine for controlling the reactive current Ir according to the reactive voltage difference or the reactive voltage ratio is executed, and the routine proceeds to step 112 where the motor driving program is returned.

  FIG. 11 is a control characteristic diagram for controlling the reactive current Ir according to the difference between the reactive voltage Va · sinφ and the reactance voltage ωLI of the motor.

  If the reactance voltage ωLI increases with respect to the inverter reactive voltage, the advance angle is excessive. Therefore, in order to reduce the reactive current Ir, the reactive current operation amount δIr is made negative, the reactive current Ir is decreased, and the reactive current ωLI is decreased. The angle value can be controlled to a predetermined value.

  By controlling the reactive current according to the reactive voltage difference, the reactive voltage of the inverter output voltage Va and the reactance voltage or ratio of the motor can be controlled to be constant. Alternatively, field weakening control can be performed by advancing the current phase with respect to the motor induced voltage.

  If there is a start flag in step 107, steps 108 to 111 are not executed. At the time of start-up, the start is performed with a predetermined reactive current, and the phase γ of the induced voltage and current is optimized after the start of the predetermined rotation speed or the predetermined time Since it is controlled to a value, the control at the time of start-up becomes simple and the start-up is stable, and the phase is optimally controlled after the start-up, so that the maximum number of revolutions can be increased by maximum efficiency operation or advance angle control.

  As described above, according to the present invention, the reactive voltage of the inverter bus voltage and the reactance voltage of the motor are compared to control the current phase with respect to the motor induced voltage, and the inverter reactive voltage and the reactance voltage are controlled to be equal. Then, in the case of the SPM motor, the maximum efficiency operation is performed. When the reactance voltage is controlled so as to increase with respect to the inverter reactive voltage, the field weakening control can be performed, and the motor can be driven to rotate at high speed.

  In the case of an IPM motor, the maximum efficiency operation can be achieved by appropriately selecting the advance value, and the maximum torque can be controlled at a high speed by the advance control.

  In addition, during startup, it is possible to stabilize the startup by controlling the inverter bus voltage and current phase, or reactive current, and after the startup, the reactance voltage is calculated and feedback control is performed to control the current phase with respect to the motor induced voltage. So it can be operated at the optimal phase

  Although the embodiment for controlling the reactive current Isinφ has been described above, it is obvious that the phase φ can be controlled by using Ir / I instead of Isinφ as shown in Equation 5.

(Embodiment 2)
Hereinafter, a second embodiment of the present invention will be described with reference to FIG.

  FIG. 12 is a block diagram of the control means of the motor drive apparatus according to the second embodiment of the present invention. FIG. 12 is a partial modification of FIG. 4, and only different blocks will be described.

  Although the vector diagram of FIG. 5 or FIG. 6 is established for the SPM motor, the vector diagram for the embedded magnet motor (IPM motor) in which the permanent magnet is embedded in the rotor iron core is a little complicated. You cannot drive. Therefore, a second embodiment in which reactive power comparison is performed will be described as a method for performing similar control in an IPM motor.

  The reactive power calculation means 76a calculates reactive power from the product of the inverter output voltage Va and the reactive current Ir. The reactance power calculation means 77a calculates the reactance power from the square of the drive frequency f, the inductance L, and the current I, and adds the output signals of the reactive power calculation means 76a and the reactance power calculation means 77a to the power comparison means 78a, Lead angle control is performed by adding the output signal of the comparison means 78a to the power factor changing means 79a to control the reactive current Ir.

  In the case of the SPM motor, if the inverter reactive power and the reactance power are controlled to be equal as shown in Expression 7, the motor induced voltage and the current phase can be made substantially the same phase, and the maximum efficiency operation is possible.

  Further, the advance angle control can be performed by controlling the reactance power to be larger than the inverter reactive power. In the case of an IPM motor, the inductance changes depending on the rotation angle, so the reactive power is calculated based on the average inductance or the maximum inductance.

  As described above, according to the present invention, the inverter reactive power and the motor reactance power are compared to control the current phase with respect to the motor induced voltage. When the inverter reactive power and the reactance power are controlled to be equal, the SPM motor In such a case, the maximum efficiency operation is achieved, and if the reactance power is controlled to be larger than the inverter reactive power, field-weakening control can be performed and the motor can be driven to rotate at high speed.

(Embodiment 3)
Hereinafter, a third embodiment of the present invention will be described with reference to FIG.

  FIG. 13 is a block diagram of the control means of the motor drive device according to the third embodiment of the present invention. FIG. 13 is a partial modification of FIG. 4, and only different blocks will be described.

  FIG. 13 compares the reactive voltage of the motor inductance with a set value and controls the motor reactance voltage to be approximately equal to the set value. Since it is the same even if the reactance power of the motor is controlled to be substantially equal to the set value, only an embodiment for controlling the reactance voltage will be described.

  The reactive voltage setting unit 76b sets a predetermined reactance voltage so as to obtain an optimum advance value in accordance with the driving frequency f. The reactance voltage calculation means 77 is the same as that shown in FIG. 4. The reactance voltage ωLI is calculated from the drive frequency f, motor inductance L, and current I, the voltage difference is obtained by the reactive voltage comparison means 78, and the power factor is changed according to the voltage difference. The reactive current Ir is controlled by means 79.

  FIG. 14 is a control characteristic diagram, in which the reactive current manipulated variable δIr is controlled by comparing the reactance voltage ωLI and the set voltage Vz. When the reactance voltage ωLI increases with respect to the set voltage Vz, the amount of advance is large, so the reactive current manipulated variable δIr is controlled to be negative so as to reduce the reactive current Ir. Further, when the reactance voltage ωLI decreases with respect to the set voltage Vz, the advance amount is small, so the reactive current manipulated variable δIr is positively controlled so as to increase the reactive current Ir.

  As described above, the present invention detects the three-phase inverter output current, converts the phase to the inverter circuit output voltage bus axis after the three-phase / two-phase conversion, and controls the motor reactive current or the current phase. Sensorless sinusoidal drive of DC brushless motor (permanent magnet type synchronous motor) is possible, and motor induced voltage and current phase are controlled by calculating motor reactance voltage or reactance power to be a predetermined value. To do.

  That is, the present invention compares the reactive voltage or power supplied to the motor from the inverter with the reactance voltage or reactive power of the motor, and controls the reactive voltage or reactive power ratio, thereby speeding up the motor induced voltage and current phase. If the current phase is set optimally, maximum efficiency operation is possible, and further, high-speed operation is possible by field weakening control.

  Furthermore, only the reactive current is detected at the start of the motor, the reactive current is controlled to a predetermined value, and the motor induced voltage and the current phase are controlled to a predetermined value after the start, so that the feedback control loop is reduced. And stable startup.

  Further, in the conventional sensorless sine wave drive, the calculation for position estimation is complicated and the burden on the processor is large, and furthermore, various tests for obtaining the motor parameters necessary for the position estimation calculation require time. According to the present invention, since position estimation is not required, the number of computation steps of the processor can be reduced, the number of computation data bits can be reduced, almost no motor parameters are required, and maximum efficiency operation can be performed automatically. Therefore, it is possible to reduce the load on the processor and perform the maximum efficiency operation equivalent to the vector control, and it is possible to realize a motor drive device by sensorless control that is inexpensive and highly reliable.

  In particular, motor control and sequence control for washing and drying machines and dishwashers require complex programs. Furthermore, the carrier frequency must be an ultrasonic frequency to reduce noise. For example, the load on the program capacity and calculation performance for the control processor becomes very large, and an expensive processor is required. However, according to the present invention, an inexpensive processor can obtain the same performance as sensorless vector control. Therefore, a motor driving device that can use an inexpensive and highly reliable motor without a position sensor can be realized.

  In the present invention, the reactive voltage or reactive power is mainly calculated and controlled. However, in principle, the reactive voltage or active power can be calculated and compared. However, although the inverter effective voltage and active power can be calculated easily, the reactive power calculation method has few errors because the calculation error of the motor effective voltage (induced voltage) or active power is large.

  As described above, the motor driving device according to the present invention converts AC power into DC power by the rectifier circuit, drives the motor by the inverter circuit, detects the output current of the inverter circuit by the current detection means, and sets the set rotational speed. The inverter circuit is PWM controlled so that the output voltage and current phase of the inverter circuit or the reactive current becomes a predetermined value, and the motor current phase is controlled by the reactance voltage or reactance power of the motor. Therefore, maximum efficiency operation of the motor or field-weakening control is facilitated, and the dishwasher pump motor, air conditioner compressor motor, fan motor, dewatering / washing tub and rotation of the washer / washer / dryer It can also be applied to drum rotation control.

The block diagram of the motor drive device in the 1st Embodiment of this invention Inverter circuit diagram of the motor drive device Current detection timing chart of the motor drive device Block diagram of control means of the motor drive device Control vector diagram of the motor drive device Vector diagram for current phase control of the motor drive Waveform and timing chart of each part of control means of the motor drive device Motor control program flowchart of the motor drive device Flowchart of carrier signal interrupt subroutine of motor control program of the motor drive device Flowchart of rotation speed control subroutine of motor control program of the same motor drive device Control characteristics diagram by comparing reactive voltage and reactance voltage of the motor drive unit The block diagram of the control means of the motor drive device in the 2nd Embodiment of this invention The block diagram of the control means of the motor drive unit in the 3rd Embodiment of this invention Control characteristics diagram by comparison of set value and reactance voltage of the motor drive unit

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 AC power supply 2 Rectifier circuit 3 Inverter circuit 4 Motor 5 Current detection means 6 Control means

Claims (7)

An AC power source, a rectifier circuit that converts AC power of the AC power source into DC power, an inverter circuit that converts DC power of the rectifier circuit into AC power, a motor driven by the inverter circuit, and the inverter circuit Current detection means for detecting an output current; and control means for controlling the motor so that the inverter circuit is PWM-controlled by an output signal of the current detection means to achieve a set rotational speed, the control means being the inverter A motor driving device that controls a phase of an output voltage and an output current of a circuit or a reactive current to be a predetermined value, and controls a motor current phase by a reactance voltage or reactance power of the motor. 2. The motor drive apparatus according to claim 1, wherein the control means compares the reactive voltage of the inverter circuit with the reactance voltage of the motor to control the motor current phase. 3. The motor driving apparatus according to claim 2, wherein the control means makes the reactive voltage of the inverter circuit substantially equal to the reactance voltage of the motor. 2. The motor driving apparatus according to claim 1, wherein the control means compares the reactive power of the inverter circuit with the reactance power of the motor to control the motor current phase. 5. The motor driving apparatus according to claim 4, wherein the control means makes the reactive power of the inverter circuit substantially equal to the reactance power of the motor. 2. The motor driving apparatus according to claim 1, wherein the control means controls the reactance voltage or the reactance power of the motor to be a set value. 2. The motor driving apparatus according to claim 1, wherein the control means controls the motor current phase after the motor is started.

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