JPWO2011135695A1 - Power converter - Google Patents

Abstract

The power conversion device includes a first period in which a switching element for the upper arm and a switching element for the lower arm are turned on in different phases and current is supplied from the DC power source to the motor, and a switching element for the upper arm in all phases. Alternatively, a PHM control mode in which either one of the switching elements for the lower arm is turned on to maintain the torque with the energy stored in the motor alternately according to the electrical angle, and a sine wave command The sine wave PWM control mode in which the switching element is turned on and current is supplied from the DC power supply to the motor according to the pulse width determined based on the comparison result between the signal and the carrier wave is switched based on a predetermined condition.

Description

  The present invention relates to a power conversion device that converts DC power into AC power or AC power into DC power.

  A power conversion device that receives direct current power and converts the direct current power into alternating current power for supplying to the rotating electrical machine includes a plurality of switching elements, and the switching element repeats a switching operation to supply the direct current supplied. Convert power to AC power. Many of the power converters are also used to convert AC power induced in the rotating electrical machine into DC power by the switching operation of the switching element. The above-described switching element is generally controlled based on a pulse width modulation method (hereinafter referred to as a PWM method) using a carrier wave that changes at a constant frequency. By increasing the frequency of the carrier wave, the control accuracy is improved and the torque generated by the rotating electrical machine tends to be smooth.

  However, the switching element not only increases power loss and heat generation when switching from the interrupted state to the conductive state, or when switching from the conductive state to the interrupted state. A leakage current flows into the motor stray capacitance due to the switching operation at the time of switching to the state or at the time of switching from the energized state to the cut-off state, and conduction noise is generated.

  An example of a power converter is disclosed in Japanese Patent Laid-Open No. 63-234878 (see Patent Document 1).

JP-A 63-234878

  It is necessary to reduce the power loss of the switching element described above and reduce conduction noise due to leakage current to the motor stray capacitance. For this purpose, it is desirable to reduce the number of switching times of the switching element. However, as described above, in the generally used PWM method, when the frequency of the carrier wave is lowered in order to reduce the number of times of switching per unit time of the switching element, the distortion of the current output from the power converter increases, This leads to an increase in torque pulsation.

  An object of the present invention is to reduce the number of switching of the switching element and reduce switching loss and conduction noise while suppressing an increase in torque pulsation as much as possible in the power converter. The embodiment described below reflects research results preferable as a product, and solves various specific problems preferable as a product. Specific problems to be solved by specific configurations and operations in the following embodiments will be described in the following embodiments.

  The present invention comprises at least one of the features described below.

A power converter according to a first aspect of the present invention includes an inverter circuit that receives DC power and generates AC power for driving a rotating electrical machine (or motor), and a smoothing unit for supplying DC power to the inverter circuit. A capacitor, a control circuit for controlling the inverter circuit, and a driver circuit for driving the inverter circuit based on the output of the control circuit are provided. In this power conversion device, the inverter circuit includes a plurality of switching elements for configuring the U-phase, V-phase, and W-phase upper arms, and U-phase, V-phase, and W-phase lower arms. DC power from a smoothing capacitor is supplied to a series circuit that has a plurality of switching elements and is configured by connecting a stator winding of a rotating electrical machine (or motor) between an upper arm and a lower arm Then, each of the U-phase, V-phase, and W-phase switching elements of the upper arm and the lower arm is successively turned on and off to generate AC power for driving the rotating electrical machine (or motor). The control circuit supplies a current to the stator winding of the rotating electrical machine (or motor) with the upper arm connected in parallel in a plurality of phases, and the current from the stator winding flows through one lower arm. In the operating range, the conduction time of the upper arm connected in parallel is made longer than the conduction time of the lower arm, and the conduction time of the inverter circuit is controlled by the conduction time of the lower arm, while the current from one upper arm to the stator winding In the second operation region in which the current from the stator winding flows through the lower arm connected in parallel, the conduction time of the lower arm connected in parallel is made longer than the conduction time of the upper arm, and the inverter circuit The conduction time is controlled by the conduction time of the upper arm. Further, based on the control signal from the control circuit, the driver circuit controls the switching elements that constitute the upper arm and the switching elements that constitute the lower arm.
According to the second aspect of the present invention, in the power conversion device according to the first aspect, the lower arm repeats conduction and shut-off operations a plurality of times within the conduction time of the upper arm in the first operation region, and the lower operation in the second operation region. It is preferable that the upper arm repeats the conduction and interruption operations a plurality of times within the arm conduction time.
According to the third aspect of the present invention, in the power conversion device of the first or second aspect, the control circuit sets the same number of conduction times and interruption times of the switching elements in the control state in which the peak value of the AC output is increased. In a state where the switching element becomes narrower by increasing the conduction width and further increasing the peak value of the AC output and the switching element cannot be shut off, the number of conductions and the interruption are maintained by continuing the conduction state of the switching element. It is preferable to reduce the number of times.
According to the fourth aspect of the present invention, in the power conversion device of the first or second aspect, the control circuit switches at a phase angle determined based on preset data within one cycle of the AC output. The element can be conducted, and the conduction width can be controlled according to the peak value.
According to the fifth aspect of the present invention, in the power conversion device according to any one of the first to fourth aspects, the control circuit is predetermined corresponding to a predetermined phase within one cycle of the AC output. The switching element may be switched a number of times.
According to the sixth aspect of the present invention, in the power conversion device according to any one of the first to fifth aspects, the control circuit is configured such that any two phases of the U phase, the V phase, and the W phase are in the conductive state. At one time, the three-phase short-circuit period is controlled by conduction and interruption of the upper arm of the other one phase, and when the lower arm of any two phases of U phase, V phase, and W phase is in the conduction state, the other one phase The three-phase short-circuit period can be controlled by turning on and off the lower arm.

  According to the present invention, in the power conversion device, it is possible to reduce the number of switching while suppressing an increase in torque pulsation to some extent, further reduce switching loss, and further reduce conduction noise due to leakage current to the motor stray capacitance.

  In the following embodiments, as described later, various problems desirable as a product are solved.

It is a figure which shows the control block of a hybrid vehicle. It is a figure which shows the structure of an electric circuit. It is a figure which shows switching of a control mode. It is a figure explaining PWM control and rectangular wave control. It is a figure which shows the example of the harmonic component produced in rectangular wave control. It is a figure which shows the motor control system by the control circuit which concerns on one Embodiment. It is a figure which shows the structure of a pulse generator. It is a flowchart which shows the procedure of the pulse generation by a table search. It is a flowchart which shows the procedure of the pulse generation by real-time calculation. It is a flowchart which shows the procedure of a pulse pattern calculation. It is a figure which shows the generation method of the pulse by a phase counter. It is a figure which shows an example of the line voltage waveform in PHM control mode. (Example of line voltage with 3rd, 5th and 7th harmonic elimination) It is explanatory drawing in case the pulse width of line voltage is unequal with other pulse trains. It is a figure which shows an example of the line voltage waveform in PHM control mode. (Example of line voltage with 3rd, 5th and 7th harmonic elimination) It is a figure which shows an example of the phase voltage waveform in the PHM control mode of FIG. It is a figure which shows the conversion table | surface of a line voltage and a phase terminal voltage. It is a figure which shows the example which converted the line voltage pulse in the rectangular wave control mode into the phase voltage pulse. It is a figure which shows the example which converted the line voltage pulse in the PHM control mode of FIG. 12 into the phase voltage pulse. It is the figure which showed the magnitude | size of the amplitude of the fundamental wave in the line voltage pulse when changing a modulation degree in FIG. 14 and FIG. 20, and the harmonic component of the deletion object. It is a figure which shows an example of the line voltage waveform in PHM control mode. (Example of line voltage with 3rd and 5th harmonic elimination) It is a figure which shows an example of the phase voltage waveform in the PHM control mode of FIG. It is a figure for demonstrating the production | generation method of a PWM pulse signal. It is a figure which shows an example of the voltage waveform between lines in PWM control mode. It is a figure which shows an example of the phase voltage waveform in PWM control mode. It is a figure which compares the line voltage pulse waveform by a PHM pulse signal with the line voltage pulse waveform by a PWM pulse signal. It is a figure for demonstrating the difference in the pulse shape in PWM control and PHM control. It is a figure which shows the relationship between a motor rotational speed and the line voltage pulse waveform by a PHM pulse signal. It is a figure which shows the relationship between the number of line voltage pulses produced | generated in PHM control and PWM control, a phase voltage pulse, and a motor rotational speed. It is a figure which shows the flowchart of the motor control performed by the control circuit which concerns on 1st Embodiment. It is a figure which approximates the trapezoid wave to the U-phase voltage pulse of the PWM control of FIG. It is a figure which approximates the trapezoid wave to the U-phase voltage pulse of PHM control of FIG. It is a figure which shows the pulse width change with respect to the motor rotation speed of PWM control and PHM control. It is a figure which shows the phase voltage spectrum of PWM control and PHM control. It is a figure which shows the fluctuation | variation of the waveform of a phase voltage pulse and the line voltage pulse in PWM control, and a neutral point voltage. It is a figure which shows the pulse reference angle in PHM control of a neutral point voltage fluctuation suppression pattern. It is a figure which shows the fluctuation | variation of the waveform of a phase voltage pulse and a line voltage pulse, and a neutral point voltage in PHM control of a neutral point voltage fluctuation suppression pattern. It is a figure which shows the pulse production | generation method in PHM control of a neutral point voltage fluctuation suppression pattern. It is a figure which shows a mode that the function f (a) is changed according to the modulation degree a, and the position of a pulse reference angle is released | separated or brought close.

  In addition to the contents described in the column of the problem to be solved by the invention and the column of the effect of the invention, the following embodiments can solve a desirable problem in commercialization and have a desirable effect in commercialization. Play. Some of them will be described next, and in the description of the embodiments, specific solutions to problems and specific effects will be described.

[Reduction of switching frequency of switching elements]
In the power conversion device described in the following embodiments, a PWM control mode that is a pulse width modulation method using a carrier wave that changes at a constant frequency, and an angle or phase of an AC output waveform converted from DC power. In order to control the switching operation of the switching element, a driving signal is supplied from the driving circuit to the switching element, and the switching element performs a switching operation of conduction or cutoff in association with the phase of the AC output to be converted. The rotating electrical machine is driven by appropriately switching between a control mode in which the switching frequency of the switching element is smaller than that in PWM control. With such a configuration and operation, the number of switching operations of the switching element per unit time or the number of switching operations per cycle of AC output can be reduced as compared with a general PWM system. By reducing the number of times of switching, the number of occurrences of leakage current (hereinafter referred to as common mode current) due to fluctuations in the neutral point voltage of the rotating electrical machine can be reduced, and the occurrence of conduction noise (hereinafter referred to as common mode noise) can also be suppressed.

  The switching element is preferably an element having a high operating speed and capable of controlling both conduction and cutoff operation based on a control signal. Examples of such an element include an insulated gate bipolar transistor (hereinafter referred to as IGBT) and a field effect transistor. (MOS transistor), and these elements are desirable in terms of responsiveness and controllability.

  The AC power output from the power conversion device is supplied to an inductance circuit composed of a rotating electrical machine or the like, and an AC current flows based on the action of the inductance. In the following embodiments, a rotating electrical machine that acts as a motor or a generator is described as an example of an inductance circuit. Use of the present invention to generate AC power for driving the rotating electrical machine is optimal from the viewpoint of effect, but it can also be used as a power conversion device that supplies AC power to an inductance circuit other than the rotating electrical machine.

  In the following embodiments, switching of the switching element is performed based on the phase of the AC waveform to be output in the first operation range in which the rotating speed of the rotating electrical machine is high or the AC output frequency to be output by the control circuit is high. On the other hand, in the second operating region where the rotational speed of the rotating electrical machine is slower than the first operating range or the AC voltage frequency that the control circuit tries to output is slower than the first operating range, the switching element The switching element is controlled by a PWM method for controlling the operation. The second operating region may include a stopped state of the rotor of the rotating electrical machine. In the following embodiments, a motor generator used as a rotating electrical machine and a motor generator used as a generator will be described as an example.

[Basic control]
In the embodiment described below, as basic control, in the low-speed operation state of the rotating electrical machine that supplies AC power or in the state where the frequency of the AC output to be supplied is low, the AC power is generated and rotated by PWM control. In a state where the rotation speed of the electric machine is increased or in a state where the frequency of the AC frequency to be supplied is high, the control shifts to AC power generation control by PHM control described below. Thereby, the influence of distortion can be suppressed as much as possible, and the switching frequency of the switching element can be reduced.

  Further, from the viewpoint different from the basic control, as described in the following embodiment, in the high speed operation state or the high output operation of the rotating electrical machine, the switching to the rectangular wave control in which the number of times of switching in the PHM control is minimized. In the PHM control described below, the switching timing is controlled in accordance with the phase of the AC waveform to be output, and the AC output, for example, the half cycle of the AC voltage (from zero to π, or from π of the electrical angle) as the degree of modulation increases. The number of times of switching in 2π) gradually decreases, and finally, the process shifts to rectangular wave control in which conduction only once in a half cycle. Similarly, in the PHM control, when the number of harmonic orders to be deleted to be deleted from the motor line voltage is reduced, switching in an AC output, for example, a half cycle of the AC voltage (electrical angle zero to π or π to 2π). The number of times gradually decreases, and finally, the process shifts to rectangular wave control in which conduction is performed only once in a half cycle. For example, first, third, fifth, seventh, eleventh and thirteenth harmonics are to be deleted, then third, fifth, seventh and eleventh harmonics are to be deleted, and then Next, the 5th and 7th harmonics are to be deleted, the 3rd and 5th harmonics are to be deleted, and finally no harmonics to be deleted (rectangular wave control). It is possible to gradually reduce the number of harmonics to be deleted in the control. Thus, in the following embodiments, there is a merit that it is possible to smoothly shift to the rectangular wave control in which the number of times of switching of the switching elements is minimized, and thus controllability is excellent.

  A power converter according to an embodiment of the present invention will be described below in detail with reference to the drawings. A power conversion device according to an embodiment of the present invention is a power conversion device that generates AC power for driving a rotating electrical machine of a hybrid vehicle (hereinafter referred to as HEV) or a pure electric vehicle (hereinafter referred to as EV). This is an applied example. HEV power converters and EV power converters are common in basic configuration and control, and as a representative example, a control configuration when the power converter according to the embodiment of the present invention is applied to a hybrid vehicle. The circuit configuration of the power converter will be described with reference to FIGS. FIG. 1 is a diagram showing a control block of a hybrid vehicle.

  In the power conversion device according to the embodiment of the present invention, an in-vehicle power conversion device of an in-vehicle electric system mounted on an automobile will be described. In particular, a vehicle drive power conversion device that is used in a vehicle drive electrical system and has a very severe mounting environment and operational environment will be described as an example. The vehicle drive power conversion device is provided in a vehicle drive electrical system as a control device that drives a vehicle drive rotating electrical machine. This power conversion device for driving a vehicle converts DC power supplied from an on-vehicle battery or an on-vehicle power generation device constituting an in-vehicle power source into predetermined AC power, and supplies the obtained AC power to the rotating electrical machine. Drives the rotating electrical machine. Further, since the rotating electrical machine has a function as a generator in addition to the function of an electric motor, the power converter not only converts DC power into AC power according to the operation mode, but also generates the rotating electrical machine. The operation to convert the alternating current power to direct current power is also performed. The converted DC power is supplied to the on-vehicle battery.

  The configuration of the present embodiment is optimal as a power conversion device for driving a vehicle such as an automobile or a truck. However, power converters other than these, such as power converters such as trains, ships, and airplanes, as well as rotating electrical machines that drive factory equipment, such as industrial power conversion for generating AC power supplied to fans and pumps, etc. The present invention is also applicable to a power converter used in a control device for a rotating electrical machine that drives a device, a household photovoltaic power generation system, or a household electrical appliance.

  In FIG. 1, HEV 110 is one electric vehicle and includes two vehicle driving systems. One of them is an engine system that uses an engine 120 that is an internal combustion engine as a power source. The engine system is mainly used as a drive source for HEV. The other is an in-vehicle electric system using motor generators 192 and 194 as a power source. The in-vehicle electric system is mainly used as an HEV drive source and an HEV power generation source. The motor generators 192 and 194 are examples of a rotating electric machine such as a synchronous machine or an induction machine, and operate as both a motor and a generator depending on the operation method.

  A front wheel axle 114 is rotatably supported at the front portion of the vehicle body. A pair of front wheels 112 are provided at both ends of the front wheel axle 114. A rear wheel axle (not shown) is rotatably supported on the rear portion of the vehicle body. A pair of rear wheels are provided at both ends of the rear wheel axle. The HEV of the present embodiment employs a so-called front wheel drive system in which the main wheel driven by power is the front wheel 112 and the driven wheel to be driven is the rear wheel. A four-wheel drive system may be adopted.

  A front wheel side differential gear (hereinafter referred to as “front wheel side DEF”) 116 is provided at the center of the front wheel axle 114. The front wheel axle 114 is mechanically connected to the output side of the front wheel side DEF 116. The output shaft of the transmission 118 is mechanically connected to the input side of the front wheel side DEF 116. The front wheel side DEF 116 is a differential power distribution mechanism that distributes the rotational driving force that is shifted and transmitted by the transmission 118 to the left and right front wheel axles 114. The output side of the motor generator 192 is mechanically connected to the input side of the transmission 118. The output side of the engine 120 and the output side of the motor generator 194 are mechanically connected to the input side of the motor generator 192 via the power distribution mechanism 122. Motor generators 192 and 194 and power distribution mechanism 122 are housed inside the casing of transmission 118.

  Motor generators 192 and 194 are synchronous machines having rotors with permanent magnets. The AC power supplied to the armature windings of the stator is controlled by the power converters 140 and 142, whereby the driving of the motor generators 192 and 194 is controlled. A battery 136 is electrically connected to the power converters 140 and 142. Power can be exchanged between the battery 136 and the power converters 140 and 142.

  The in-vehicle electric machine system of the present embodiment includes two of a first motor generator unit composed of a motor generator 192 and a power converter 140 and a second motor generator unit composed of a motor generator 194 and a power converter 142, They are used properly according to the driving conditions. That is, when the vehicle is driven by the power from the engine 120, when assisting the driving torque of the vehicle, the second motor generator unit is operated by the power of the engine 120 as a power generation unit to generate power, and the power generation The first motor generator unit is operated as an electric unit by the electric power obtained by the above. Further, in the same case, when assisting the vehicle speed of the vehicle, the first motor generator unit is operated by the power of the engine 120 as a power generation unit to generate power, and the second motor generator unit is generated by the electric power obtained by the power generation. Is operated as an electric unit.

  In the present embodiment, the vehicle can be driven only by the power of the motor generator 192 by operating the first motor generator unit as an electric unit by the electric power of the battery 136. Furthermore, in the present embodiment, the battery 136 can be charged by generating power by operating the first motor generator unit or the second motor generator unit as the power generation unit by the power of the engine 120 or the power from the wheels.

  The battery 136 is also used as a power source for driving an auxiliary motor 195. The auxiliary motor is, for example, a motor for driving a compressor of an air conditioner or a motor for driving a control hydraulic pump. DC power is supplied from the battery 136 to the power converter 43, converted into AC power by the power converter 43, and supplied to the motor 195. The power conversion device 43 has the same function as the power conversion devices 140 and 142 and controls the phase, frequency, and power of alternating current supplied to the motor 195. For example, the motor 195 generates torque by supplying an AC output of a leading phase with respect to the rotation of the rotor of the motor 195. On the other hand, by generating an AC output with a delayed phase, the motor 195 acts as a generator and operates in a regenerative braking state. Such a control function of the power conversion device 43 is the same as the control function of the power conversion devices 140 and 142. Since the capacity of the motor 195 is smaller than the capacity of the motor generators 192 and 194, the maximum converted power of the power converter 43 is smaller than that of the power converters 140 and 142. However, the circuit configuration and operation of the power conversion device 43 are basically similar to the circuit configuration and operation of the power conversion devices 140 and 142.

  The power converters 140 and 142, the power converter 43, and the capacitor module 500 are in an electrical close relationship. Furthermore, there is a common point that measures against heat generation are necessary. It is also desired to make the volume of the device as small as possible. From these points, the power conversion device described in detail below includes the power conversion devices 140 and 142, the power conversion device 43, and the capacitor module 500 in the casing of the power conversion device. With this configuration, a small and highly reliable device can be realized.

  Further, by incorporating the power conversion devices 140 and 142, the power conversion device 43, and the capacitor module 500 in one housing, it is effective in simplifying wiring and taking measures against noise. In addition, the inductance of the connection circuit between the capacitor module 500, the power converters 140 and 142, and the power converter 43 can be reduced, the spike voltage can be reduced, heat generation can be reduced, and heat dissipation efficiency can be improved.

  Next, the electric circuit configuration of the power converters 140 and 142 or the power converter 43 will be described with reference to FIG. FIG. 2A shows the entire configuration of the electric circuit in the in-vehicle electric system of this embodiment, and FIG. 2B shows the details of the motor generator part of the electric circuit. In the embodiment illustrated in FIGS. 1 to 2, the case where the power conversion devices 140 and 142 or the power conversion device 43 are individually configured will be described as an example. The power conversion devices 140 and 142 or the power conversion device 43 have the same functions and the same functions with the same configuration. Here, the power converter 140 will be described as a representative example.

  The power conversion device 200 according to the present embodiment includes a power conversion device 140 and a capacitor module 500. The power conversion device 140 includes a power switching circuit 144 and a control unit 170. Further, the power switching circuit 144 has a switching element that operates as an upper arm and a switching element that operates as a lower arm. In this embodiment, an IGBT (insulated gate bipolar transistor) is used as a switching element. The IGBT 328 operating as the upper arm is connected in parallel with the diode 156, and the IGBT 330 operating as the lower arm is connected in parallel with the diode 166. There are a plurality of upper and lower arm series circuits 150 (in the example of FIG. 2, three upper and lower arm series circuits 150, 150 and 150), and an AC terminal is connected from the middle point (connection point 169) of each upper and lower arm series circuit 150. 159 is connected to an AC power line (AC bus bar) 186 to the motor generator 192 through 159. In addition, the control unit 170 includes a driver circuit 174 that drives and controls the power switching circuit 144 and a control circuit 172 that supplies a control signal to the driver circuit 174 via the signal line 176.

  The IGBTs 328 and 330 of the upper arm and the lower arm are switching elements, operate in response to the drive signal output from the control unit 170, and convert DC power supplied from the battery 136 into three-phase AC power. The converted electric power is supplied to the armature winding of the motor generator 192. As described above, power conversion device 140 also performs an operation of converting three-phase AC power generated by motor generator 192 into DC power.

  The power conversion device 200 according to the present embodiment includes power conversion devices 140 and 142, a power conversion device 43, and a capacitor module 500 as shown in FIG. Since the power converters 140 and 142 and the power converter 43 have the same circuit configuration as described above, the power converter 140 is described here as a representative, and the power converter 142 and the power converter 43 are as described above. Omitted.

  The power switching circuit 144 is an inverter circuit configured by a three-phase bridge circuit. A direct current positive electrode terminal 314 and a direct current negative electrode terminal 316 are electrically connected to the positive electrode side and the negative electrode side of the battery 136. Between the DC positive terminal 314 and the DC negative terminal 316, upper and lower arm series circuits 150, 150, 150 corresponding to the respective phases are electrically connected in parallel. Here, the series circuit 150 of the upper and lower arms is referred to as an arm. Each arm includes a switching element 328 and a diode 156 on the upper arm side, and a switching element 330 and a diode 166 on the lower arm side.

  In the present embodiment, the use of IGBTs 328 and 330 as switching elements is exemplified. The IGBTs 328 and 330 include collector electrodes 153 and 163, emitter electrodes (signal emitter electrode terminals 155 and 165), and gate electrodes (gate electrode terminals 154 and 164). Diodes 156 and 166 are electrically connected in parallel between the collector electrodes 153 and 163 of the IGBTs 328 and 330 and the emitter electrode as shown in the figure. The diodes 156 and 166 include two electrodes, a cathode electrode and an anode electrode. The cathode electrode is electrically connected to the collector electrode of the IGBTs 328 and 330 and the anode electrode is electrically connected to the emitter electrodes of the IGBTs 328 and 330 so that the direction from the emitter electrode to the collector electrode of the IGBTs 328 and 330 is the forward direction. A MOSFET (metal oxide semiconductor field effect transistor) may be used as the switching element. In this case, the diode 156 and the diode 166 are unnecessary.

  The series circuit 150 of the upper and lower arms corresponds to each phase of AC power supplied to the three-phase motor generator 192. Each series circuit 150, 150, 150 connects the emitter electrode of the IGBT 328 and the collector electrode 163 of the IGBT 330. The connection point 169 is used to output U-phase, V-phase, and W-phase AC power, respectively. The connection point 169 of each phase is connected to the U-phase, V-phase, and W-phase armature windings (stator winding in the synchronous motor) of the motor generator 192 via the AC terminal 159 and the connector 188, respectively. As a result, U-phase, V-phase, and W-phase currents flow through the armature winding. The series circuits of the upper and lower arms are electrically connected in parallel. The collector electrode 153 of the upper arm IGBT 328 is connected to the positive electrode capacitor electrode of the capacitor module 500 via the positive terminal (P terminal) 157, and the emitter electrode of the lower arm IGBT 330 is connected to the capacitor via the negative electrode terminal (N terminal) 158. The module 500 is electrically connected to a negative electrode capacitor electrode via a DC bus bar or the like.

  Capacitor module 500 is for configuring a smoothing circuit that suppresses fluctuations in DC voltage caused by the switching operation of IGBTs 328 and 330. The positive electrode side of the battery 136 is electrically connected to the positive electrode side capacitor electrode of the capacitor module 500, and the negative electrode side of the battery 136 is electrically connected to the negative electrode side capacitor electrode of the capacitor module 500 via the DC connector 138. Thus, the capacitor module 500 is connected between the collector electrode 153 of the upper arm IGBT 328 and the positive electrode side of the battery 136, and between the emitter electrode of the lower arm IGBT 330 and the negative electrode side of the battery 136. Are electrically connected in parallel to the series circuit 150.

  The control unit 170 functions to control the operation of turning on and off the IGBTs 328 and 330, and the control unit 170 controls the switching timing of the IGBTs 328 and 330 based on input information from other control devices and sensors. And a drive circuit 174 for generating a drive signal for switching the IGBTs 328 and 330 based on the timing signal output from the control circuit 172.

  The control circuit 172 includes a microcomputer for calculating the switching timing of the IGBTs 328 and 330. In this microcomputer, as input information, a target torque value required for the motor generator 192, a current value supplied to the armature winding of the motor generator 192 from the series circuit 150 of the upper and lower arms, and the motor generator 192 The magnetic pole position of the rotor is input. The target torque value is based on a command signal output from a host controller (not shown). The current value is detected based on the detection signal output from the current sensor 180. The magnetic pole position is detected based on the detection signal output from the rotating magnetic pole sensor 193 provided in the motor generator 192. In the present embodiment, the case where the current values of three phases are detected will be described as an example, but the current values for two phases may be detected.

  The microcomputer in the control circuit 172 calculates the d and q axis current command values of the motor generator 192 based on the input target torque value, and the calculated d and q axis current command values are detected. The d and q axis voltage command values are calculated based on the difference between the d and q axis current values, and a pulsed drive signal is generated from the d and q axis voltage command values. The control circuit 172 has a function of generating drive signals of two types as will be described later. These two types of drive signals are selected based on the state of the motor generator 192, which is an inductance load, or based on the frequency of the AC output to be converted.

  One of the two types of methods is a modulation method (which will be described later as a PHM method) that controls the switching operation of the IGBTs 328 and 330 that are switching elements based on the phase of the AC waveform to be output. The other one of the above two types is a modulation method generally called PWM (Pulse Width Modulation).

  When driving the lower arm, the driver circuit 174 amplifies a pulse-like modulated wave signal and outputs it as a drive signal to the gate electrode of the corresponding IGBT 330 of the lower arm. When driving the upper arm, the reference potential level of the pulsed modulated wave signal is shifted to the reference potential level of the upper arm, and then the pulsed modulated wave signal is amplified and used as a drive signal. , Output to the gate electrode of the IGBT 328 of the corresponding upper arm. As a result, each IGBT 328, 330 performs a switching operation based on the input drive signal. In this way, by the switching operation of the IGBTs 328 and 330 performed in accordance with the drive signal (drive signal) from the control unit 170, the power converter 140 converts the voltage supplied from the battery 136, which is a DC power supply, to 2π / in electrical angle. The output voltage is converted into U-phase, V-phase, and W-phase output voltages shifted every 3 rads, and supplied to a motor generator 192 that is a three-phase AC motor. The electrical angle corresponds to the rotational state of the motor generator 192, specifically the position of the rotor, and periodically changes between 0 and 2π. By using this electrical angle as a parameter, the switching states of the IGBTs 328 and 330, that is, the output voltages of the U phase, the V phase, and the W phase can be determined according to the rotation state of the motor generator 192.

  In addition, the control unit 170 performs abnormality detection (overcurrent, overvoltage, overtemperature, etc.) and protects the series circuit 150 of the upper and lower arms. For this reason, sensing information is input to the control unit 170. For example, information on the current flowing through the emitter electrodes of the IGBTs 328 and 330 is input to the corresponding drive units (ICs) from the signal emitter electrode terminals 155 and 165 of each arm. Thereby, each drive part (IC) detects overcurrent, and when overcurrent is detected, the switching operation of corresponding IGBT328,330 is stopped, and corresponding IGBT328,330 is protected from overcurrent. Information on the temperature of the series circuit 150 of the upper and lower arms is input to the microcomputer from a temperature sensor (not shown) provided in the series circuit 150 of the upper and lower arms. In addition, voltage information on the DC positive side of the series circuit 150 of the upper and lower arms is input to the microcomputer. The microcomputer performs over-temperature detection and over-voltage detection based on such information, and when an over-temperature or over-voltage is detected, stops the switching operation of all IGBTs 328 and 330, and pulls up and down the series circuit 150 of the upper and lower arms. Thus, the semiconductor module including the circuit 150 is protected from overtemperature or overvoltage.

  In FIG. 2, a series circuit 150 of upper and lower arms is a series circuit of an IGBT 328 and an upper arm diode 156 of the upper arm, and an IGBT 330 and a lower arm diode 166 of the lower arm. IGBTs 328 and 330 are switching semiconductor elements. The conduction and cutoff operations of the IGBTs 328 and 330 of the upper and lower arms of the power switching circuit 144 are switched in a fixed order. The current of the stator winding of the motor generator 192 at the time of switching flows through a circuit formed by the diodes 156 and 166.

  As shown in the figure, the upper and lower arm series circuit 150 includes a positive terminal (P terminal, positive terminal) 157, a negative terminal (N terminal 158, negative terminal), an AC terminal 159 from the connection point 169 of the upper and lower arms, A signal terminal (signal emitter electrode terminal) 155, an upper arm gate electrode terminal 154, a lower arm signal terminal (signal emitter electrode terminal) 165, and a lower arm gate terminal electrode 164 are provided. The power conversion device 200 has a DC connector 138 on the input side and an AC connector 188 on the output side, and is connected to the battery 136 and the motor generator 192 through the connectors 138 and 188, respectively. Further, as a circuit that generates an output of each phase of the three-phase alternating current that is output to the motor generator, a power conversion device having a circuit configuration in which a series circuit of two upper and lower arms is connected in parallel to each phase may be used.

In FIG. 2, if the phase voltages of the respective phases generated between the U-phase, V-phase and W-phase coils of the motor generator and the neutral point 192n are Vu, Vv and Vw, the neutral point voltage Vn is It can be expressed as the following formula (1).
Vn = (Vu + Vv + Vw) / 3 (1)

  The value of the neutral point voltage Vn is based on the above equation (1) as the phase voltages Vu, Vv, and Vw appear in the U, V, and W phase coils when the IGBTs 328 and 330 start the switching operation. Fluctuate. The switching operation of the IGBTs 328 and 330 is started when the drive circuit 174 generates a drive signal for causing the IGBTs 328 and 330 to perform a switching operation based on the timing signal output from the control circuit 172.

  The control circuit 172 of the control unit 170 has a function of generating two types of drive signals. One of the two types of methods is a modulation method (which will be described later as a PHM method) that controls the switching operation of the IGBTs 328 and 330 that are switching elements based on the phase of the AC waveform to be output. The other one of the above two types is a modulation method generally called PWM (Pulse Width Modulation). The fluctuation of the neutral point voltage Vn differs between the PHM method and the PWM method.

  The switching of the control mode performed in the power converter 140 will be described using FIG. The power conversion device 140 switches between a PWM control method and a PHM control method, which will be described later, according to the rotational speed of the motor, that is, the motor generator 192 or the frequency of the AC output to be output. FIG. 3 shows how the control mode is switched in the power converter 140. The horizontal axis of FIG. 3 represents the rotation speed (r / min) of the motor generator or the frequency (Hz) of the AC output to be output, and the vertical axis represents the torque (Nm) of the motor generator.

The frequency of the AC output to be output and the motor generator rotation speed can be expressed as in the following equation (2).
(AC output frequency to be output)
= (Number of motor generator pole pairs) x (Number of revolutions) / 60 (Hz) ... (2)

  The rotation speed or frequency for switching the control mode can be arbitrarily changed.

  In the PHM control described below, the switching frequency of the switching element is less than that in the PWM control when the rotation speed of the motor generator 192 is in a low speed state including a stopped state. Therefore, depending on the magnitude of the AC power to be output, the distortion of the current waveform flowing in the inductance circuit of the motor generator 192 may increase, resulting in a problem in controllability. However, in the middle and high speed range where the inductance load of the motor generator 192 becomes large, if a specific harmonic component is deleted from the AC output to be output, even if the number of switching is reduced, the distortion of the current waveform flowing through the inductance circuit is reduced. it can. Therefore, there is an effect that the conduction noise due to the power loss of the switching element and the leakage current to the motor stray capacitance can be reduced. Therefore, such a drawback can be compensated by combining with the control by the PWM control method.

  For example, when the automobile starts running from a stopped state, the motor generator 192 needs to generate a large torque in the stopped state. In order to give the vehicle a high-class feel, smooth start and acceleration are desirable. When starting and accelerating the vehicle, it is desirable to reduce the distortion of the alternating current supplied to the motor generator 192 in order to realize smooth acceleration. Switching of the switching element included in the power switching circuit 144 by the PWM control method is desirable. Control the behavior.

  In the low-speed operation state of the motor generator 192, there is a limit to the AC current that can be supplied, and control is performed while suppressing the maximum generated torque. As the rotational speed of the motor generator 192 increases, the internal induced voltage increases and the amount of current supplied tends to decrease. For this reason, the output torque of the motor generator 192 tends to decrease as the rotational speed increases.

  The rotational speed of the motor generator for switching between control by the PWM method and PHM control is not particularly limited, but the middle / high speed region where the inductance load of the motor generator 192 is large is an operation region that is very suitable for control by the PHM method. In this region, the control of the PHM system has fewer switching times of the switching element than the control by the PWM system, and the effect of reducing the loss and the common mode noise by the common mode current is large. This driving region is a driving region that is easy to use in urban driving, and PHM control exhibits a great effect in a driving region closely related to daily life.

  In this embodiment, the mode controlled by the PWM control method (hereinafter referred to as PWM control mode) is used in a region where the rotational speed of the motor generator 192 is relatively low, while the PHM control mode described later is used in a region where the rotational speed is relatively high. use. In the PWM control mode, the power converter 140 performs control using the PWM signal as described above. That is, the microcomputer in the control circuit 172 calculates the voltage command values for the d and q axes of the motor generator 192 based on the input target torque value, and calculates the voltage command values for the U phase, V phase, and W phase. Convert to Then, a sine wave corresponding to the voltage command value of each phase is used as a fundamental wave, and this is compared with a triangular wave having a predetermined period as a carrier wave, and a pulse-like modulated wave having a pulse width determined based on the comparison result is Output to the circuit 174. By outputting a drive signal corresponding to the modulated wave from the driver circuit 174 to the IGBTs 328 and 330 corresponding to the upper and lower arms of each phase, the DC voltage output from the battery 136 is converted into a three-phase AC voltage, and the motor generator 192.

  The contents of the PHM will be described in detail later. The modulated wave generated by the control circuit 172 in the PHM control mode is output to the driver circuit 174. As a result, a drive signal corresponding to the modulated wave is output from the driver circuit 174 to the corresponding IGBTs 328 and 330 of each phase. As a result, the DC voltage output from the battery 136 is converted into a three-phase AC voltage and supplied to the motor generator 192.

  When DC power is converted into AC power using a switching element like the power converter 140, switching loss can be reduced by reducing the number of switching per unit time or per predetermined phase of AC output, and further common While it is possible to reduce common mode noise due to mode current, there is a possibility that torque pulsation increases due to the tendency that the converted AC output contains a lot of harmonic components, and the responsiveness of motor generator control may deteriorate. Therefore, in the present invention, as described above, the PWM control mode and the PHM control mode are switched according to the frequency of the AC output to be converted or the rotational speed of the motor generator related to this frequency, thereby lowering the lower harmonics. The PHM control method is applied to the motor generator rotation region that is not easily affected by the waves, that is, the medium-high speed rotation region, and the PWM control method is applied to the low-speed rotation region where torque pulsation is likely to occur. By doing in this way, increase of torque pulsation can be suppressed comparatively low, and switching loss and common mode noise can be reduced.

  As a control state of the motor generator that minimizes the number of times of switching, there is a control state by a rectangular wave in which the switching elements of each phase are turned on and off once for each electrical angle 2π of the motor. The control state by the rectangular wave is a control form of the PHM control system as the final state of the switching frequency per half cycle which decreases in accordance with the increase of the modulation degree in the converted AC output waveform in the above PHM control system. Can be understood as This point will be described in detail later.

  Next, in order to describe the PHM control method, first, PWM control and rectangular wave control will be described with reference to FIG. In the case of PWM control, the switching element is controlled based on the comparison of the magnitude of the carrier wave having a constant frequency and the AC waveform to be output, and the switching element is turned on and off. By using PWM control, an AC output with less pulsation can be supplied to the motor, and motor control with less torque pulsation becomes possible. On the other hand, since the number of times of switching per unit time or every cycle of the AC waveform is large, there is a disadvantage that common mode noise due to switching loss and common mode current is large. On the other hand, as an extreme example, when the switching element is controlled using one pulse of a rectangular wave, the switching loss can be reduced because the number of times of switching is small, and the common mode noise due to the common mode current can also be reduced. On the other hand, the AC waveform to be converted becomes a rectangular wave when the influence of the inductance load is ignored, and the sine wave includes harmonic components such as the fifth, seventh, eleventh,... Can see. When a rectangular wave is Fourier-expanded, harmonic components such as fifth order, seventh order, eleventh order,. This harmonic component causes current distortion that causes torque pulsation. As described above, the PWM control and the rectangular wave control are opposite to each other.

FIG. 5 shows an example of harmonic components generated in the AC output when it is assumed that conduction and interruption of the switching element are controlled in a rectangular wave shape. FIG. 5A is an example in which an alternating waveform that changes in a rectangular waveform is decomposed into a sine wave that is a fundamental wave and harmonic components such as fifth, seventh, eleventh,... The Fourier series expansion of the rectangular wave shown in FIG. 5 (a) is expressed as Equation (3).
f (ωt) = 4 / π × {sinωt + (sin3ωt) / 3 + (sin5ωt) / 5 + (sin7ωt) / 7 + ...} (3)

  Equation (3) is obtained from the sine wave of the fundamental wave represented by 4 / π · (sinωt) and the third, fifth, seventh,... It shows that the rectangular wave shown in (a) is formed. Thus, it turns out that it approximates a rectangular wave by synthesize | combining a higher order harmonic with respect to a fundamental wave.

  FIG. 5B shows a comparison of the amplitudes of the fundamental wave, the third harmonic, and the fifth harmonic. When the amplitude of the rectangular wave in FIG. 5A is 1, the amplitude of the fundamental wave is 1.27, the amplitude of the third harmonic is 0.42, and the amplitude of the fifth harmonic is 0.25. . Thus, it can be seen that the influence of the rectangular wave control becomes smaller because the amplitude becomes smaller as the order of the harmonics increases.

  From the viewpoint of torque pulsation that may occur when the switching element is turned on and off in a rectangular wave shape, while removing high-order harmonic components that have a large impact, On the other hand, by disregarding the influence and including these harmonic components, it is possible to realize a power conversion device that can reduce the number of times of switching, reduce the switching loss, and suppress the increase in torque pulsation. In the PHM control used in the present embodiment, attention is paid to the motor generator line voltage actually controlled by the inverter, and the motor generator AC current is removed by removing some harmonic components from the motor generator line voltage to be output. Are reduced to some extent according to the state of control. This reduces the effect of motor control torque pulsation, while reducing the number of switchings and reducing switching loss by making the motor generator AC current contain harmonic components in a range where there is no problem in use. I try to reduce it. Such a control method is described as a PHM control method in this specification as described above.

  Further, in the following embodiments, the PWM control method is used in the motor generator low rotation range where the influence of the harmonics in the PHM control method is large or the controllability is poor, that is, in the state where the AC output is at a low frequency. . Specifically, by switching between PWM control and PHM control according to the rotational speed of the motor, and using the PWM method in a region where the rotational speed is low, a motor that is desirable in each of the low-speed rotational region and the high-speed rotational region Control is performed. Alternatively, switching according to the frequency of the alternating current output, for example, the alternating voltage, which is about to output the PWM control and the PHM control, and using the PWM method in the low frequency region, the low frequency region and the high frequency region are controlled. In each case, desirable motor control is performed.

  FIG. 6 shows a motor control system by the control circuit 172 according to the embodiment of the present invention. A torque command T * as a target torque value is input to the control circuit 172 from a host control device. The torque command / current command converter 410 stores a torque-rotation speed map stored in advance based on the input torque command T * and rotation speed information based on the magnetic pole position signal θ detected by the rotating magnetic pole sensor 193. Are used to obtain a d-axis current command signal Id * and a q-axis current command signal Iq *. The d-axis current command signal Id * and the q-axis current command signal Iq * obtained by the torque command / current command converter 410 are output to the current controllers (ACR) 420 and 421, respectively.

  The current controllers (ACR) 420 and 421 include the d-axis current command signal Id * and the q-axis current command signal Iq * output from the torque command / current command converter 410 and the motor generator 192 detected by the current sensor 180. Phase current detection signals lu, lv, and lw are converted into Id and Iq current signals converted on the d and q axes by a magnetic pole position signal from a rotation sensor in a three-phase two-phase converter (not shown) on the control circuit 172. Based on this, the d-axis voltage command signal Vd * and the q-axis voltage command signal Vq * are respectively calculated so that the current flowing through the motor generator 192 follows the d-axis current command signal Id * and the q-axis current command signal Iq *. . The d-axis voltage command signal Vd * and the q-axis voltage command signal Vq * obtained by the current controller (ACR) 420 are output to the pulse modulator 430 for PHM control. On the other hand, the d-axis voltage command signal Vd * and the q-axis voltage command signal Vq * obtained by the current controller (ACR) 421 are output to the pulse modulator 440 for PWM control.

  The pulse modulator 430 for PHM control includes a voltage phase difference calculator 431, a modulation degree calculator 432, and a pulse generator 434. The d-axis voltage command signal Vd * and the q-axis voltage command signal Vq * output from the current controller 420 are input to the voltage phase difference calculator 431 and the modulation factor calculator 432 in the pulse modulator 430.

Voltage phase difference calculator 431 calculates the phase difference between the magnetic pole position of motor generator 192 and the voltage phase represented by d-axis voltage command signal Vd * and q-axis voltage command signal Vq *, that is, the voltage phase difference. Assuming that this voltage phase difference is δ, the voltage phase difference δ is expressed by Equation (4).
δ = arctan (-Vd * / Vq *) (4)

The voltage phase difference calculator 431 further calculates the voltage phase by adding the magnetic pole position represented by the magnetic pole position signal θ from the rotating magnetic pole sensor 193 to the voltage phase difference δ. Then, a voltage phase signal θv corresponding to the calculated voltage phase is output to the pulse generator 434. This voltage phase signal θv is expressed by equation (5), where the magnetic pole position represented by the magnetic pole position signal θ is θe.
θv = δ + θe + π (5)

The modulation factor calculator 432 calculates the modulation factor by normalizing the magnitude of the vector represented by the d-axis voltage command signal Vd * and the q-axis voltage command signal Vq * with the voltage of the battery 136, and according to the modulation factor. The modulation degree signal a is output to the pulse generator 434. In this embodiment, the modulation degree signal a is determined based on the battery voltage that is a DC voltage supplied to the power switching circuit 144 shown in FIG. 2, and the modulation degree a decreases as the battery voltage increases. Tend to be. Further, as the amplitude value of the command value increases, the degree of modulation a tends to increase. Specifically, when the battery voltage is Vdc, it is expressed by Expression (6). In equation (6), Vd represents the amplitude value of the d-axis voltage command signal Vd *, and Vq represents the amplitude value of the q-axis voltage command signal Vq *.
a = (√ (2/3)) (√ (Vd ^ 2 + Vq ^ 2)) / (Vdc / 2) ... (6)

  Based on the voltage phase signal θv from the voltage phase difference calculator 431 and the modulation degree signal a from the modulation degree calculator 432, the pulse generator 434 applies to the upper and lower arms of the U phase, V phase, and W phase, respectively. A pulse signal based on the corresponding six types of PHM control is generated. Then, the generated pulse signal is output to the switching unit 450, and is output from the switching unit 450 to the driver circuit 174, and a drive signal is output to each switching element. A method for generating a pulse signal based on PHM control (hereinafter referred to as a PHM pulse signal) will be described in detail later.

  On the other hand, the pulse modulator 440 for PWM control has a magnetic pole represented by the d-axis voltage command signal Vd * and the q-axis voltage command signal Vq * output from the current controller 421 and the magnetic pole position signal θ from the rotating magnetic pole sensor 193. Based on the position θe, six types of pulse signals (hereinafter referred to as PWM pulse signals) based on PWM control corresponding to the U-phase, V-phase, and W-phase upper and lower arms are generated by a known PWM method. . Then, the generated PWM pulse signal is output to the switch 450, supplied from the switch 450 to the drive circuit 174, and the drive signal is supplied from the drive circuit 174 to each switching element.

  The switch 450 selects either the PHM pulse signal output from the pulse modulator 430 for PHM control or the PWM pulse signal output from the pulse modulator 440 for PWM control. The selection of the pulse signal by the switch 450 is performed according to the rotational speed of the motor generator 192 as described above. That is, when the rotational speed of motor generator 192 is lower than a predetermined threshold set as a switching line, PWM control method is applied in power converter 140 by selecting a PWM pulse signal. . When the rotational speed of motor generator 192 is higher than the threshold value, the PHM control method is applied to power converter 140 by selecting the PHM pulse signal. The PHM pulse signal or PWM pulse signal thus selected by the switch 450 is output to the driver circuit 174 (not shown).

  As described above, the switch 450 selects either the PHM pulse signal output from the pulse modulator for PHM control 430 or the PWM pulse signal output from the pulse modulator for PWM control 440. The selection of the pulse signal by the switch 450 may be performed according to the AC output to be output from the control circuit 172 represented by the above-described equation (2), for example, the frequency of the AC voltage. That is, when the frequency to be output from the control circuit 172 to the motor generator 192 is lower than a predetermined threshold set as a switching line, the PWM control method is selected in the power converter 140 by selecting the PWM pulse signal. To be applied. In addition, when the frequency to be output from the control circuit 172 to the motor generator 192 is higher than the threshold value, the PHM control method is applied to the power converter 140 by selecting the PHM pulse signal. The PHM pulse signal or PWM pulse signal thus selected by the switch 450 is output to the driver circuit 174 (not shown).

  As described above, a PHM pulse signal or a PWM pulse signal is output as a modulated wave from the control circuit 172 to the driver circuit 174. In response to the modulated wave, a drive signal is output from the driver circuit 174 to the IGBTs 328 and 330 of the power switching circuit 144.

  Details of the pulse generator 434 in FIG. 6 will be described here. The pulse generator 434 is realized by a phase searcher 435 and a timer counter or a phase counter comparator 436, for example, as shown in FIG. The phase search unit 435 performs switching stored in advance based on the voltage phase signal θv from the voltage phase difference calculator 431, the modulation degree signal a from the modulation degree calculator 432, and the rotational speed ω information based on the magnetic pole position signal θ. A phase for which a switching pulse is to be output is searched for from the upper and lower arms of the U-phase, V-phase, and W-phase from the pulse phase information table, and information on the search result is output to the timer counter or phase counter comparator 436. The timer counter or phase counter comparator 436 generates PHM pulse signals as switching commands for the upper and lower arms of the U-phase, V-phase, and W-phase, respectively, based on the search result output from the phase searcher 435. Six types of PHM pulse signals for the upper and lower arms of each phase generated by the timer counter comparator 436 are output to the switch 450 as described above.

  FIG. 8 is a flowchart illustrating in detail the procedure of pulse generation by the phase searcher 435 and the timer counter comparator 436 in FIG. The phase searcher 435 takes in the modulation degree signal a as an input signal in step 801 and takes in the voltage phase signal θv as an input signal in step 802. In the subsequent step 803, the phase searcher 435 calculates a voltage phase range corresponding to the next control period in consideration of the control delay time and the rotation speed based on the input current voltage phase signal θv. Thereafter, in step 804, the phase searcher 435 performs a ROM search. In this ROM search, switching on and off phases are searched from a table stored in advance in a ROM (not shown) within the voltage phase range calculated in step 803 based on the input modulation degree signal a. .

  The phase searcher 435 outputs the information on the switching ON / OFF phase obtained by the ROM search in step 804 to the timer counter comparator 436 in step 805. The timer counter comparator 436 converts this phase information into time information in step 806, and generates a PHM pulse signal using a compare match function with the timer counter. In addition, the process of converting phase information into time information uses information of a rotational speed signal. Alternatively, the PHM pulse may be generated by using the phase matching information of switching on and off obtained by the ROM search in step 804 and using the compare match function with the phase counter in step 806.

  The timer counter comparator 436 outputs the PHM pulse signal generated at step 806 to the switch 450 at the next step 807. The processes in steps 801 to 807 described above are performed in the phase searcher 435 and the timer counter comparator 436, whereby a PHM pulse signal is generated in the pulse generator 434.

  Alternatively, pulse generation may be performed by executing the processing shown in the flowchart of FIG. 9 in the pulse generator 434 instead of the flowchart of FIG. 8. This process generates a switching phase for each control cycle of the current controller (ACR) without using a table retrieval method for retrieving a switching phase using a table stored in advance as shown in the flowchart of FIG. It is a method.

  The pulse generator 434 receives the modulation degree signal a in step 801 and receives the voltage phase signal θv in step 802. In the following step 820, the pulse generator 434 sets the on / off phase of switching based on the input modulation degree signal a and the voltage phase signal θv in consideration of the control delay time and the rotation speed. ACR) is determined every control cycle.

  Details of the switching phase determination processing in step 820 are shown in the flowchart of FIG. In step 821, the pulse generator 434 specifies a harmonic order to be deleted from the motor generator line voltage based on the rotation speed. In accordance with the harmonic order thus designated, the pulse generator 434 performs processing such as matrix calculation in the subsequent step 822, and outputs a pulse reference angle in step 823.

  The pulse generation process from steps 821 to 823 is calculated according to the determinants represented by the following equations (7) to (10).

  Here, as an example, the case of eliminating the third, fifth, and seventh order components is taken up.

  When the third, fifth, and seventh harmonic components are specified in step 821 as the harmonic orders to be deleted, the pulse generator 434 performs matrix calculation in the next step 822.

  Here, a row vector like Formula (7) is created with respect to the 3rd, 5th and 7th erasure orders.

  Each element in the right parenthesis of Expression (7) is k1 / 3, k2 / 5, k3 / 7. Any odd number can be selected for k1, k2, and k3. However, k1 = 3, 9, 15, k2 = 5, 15, 25, k3 = 7, 21, 35, etc. must not be selected. Under this condition, the third, fifth and seventh order components are completely eliminated.

  In more general terms, the value of each element of Equation (7) is determined by setting the harmonic order from which the denominator value is deleted and the numerator value being an arbitrary odd number excluding an odd multiple of the denominator. be able to. Here, in the example of Expression (7), the number of elements of the row vector is set to three because there are three types of deletion orders (third order, fifth order, and seventh order). Similarly, a row vector having N elements can be set for N types of erasure orders, and the value of each element can be determined.

  In addition, in the formula (7), by setting the numerator and denominator values of each element other than those described above, the spectrum can be shaped instead of deleting the harmonic component. Therefore, the numerator and denominator values of each element may be arbitrarily selected for the main purpose of spectrum shaping rather than elimination of harmonic components. In that case, the numerator and denominator values do not necessarily have to be integers, but the numerator value should not be an odd multiple of the denominator. Further, the values of the numerator and denominator need not be constants, and may be values that change according to time.

  As described above, when there are three elements whose values are determined by the combination of the denominator and the numerator, a vector of three columns can be set as in Expression (7). Similarly, a vector of N elements whose value is determined by a combination of a denominator and a numerator, that is, a vector of N columns can be set. Hereinafter, this N-column vector is referred to as a harmonic-based phase vector.

  When the harmonic compliant phase vector is a three-column vector as shown in Equation (7), the harmonic compliant phase vector is transposed to calculate Equation (8). As a result, pulse reference angles from S1 to S4 are obtained.

  The pulse reference angles S1 to S4 are parameters representing the center position of the voltage pulse, and are compared with a triangular wave carrier described later. As described above, when the pulse reference angle is four (S1 to S4), generally, the number of pulses per one cycle of the line voltage is 16.

  Further, when the harmonic compliant phase vector is four columns as in the equation (9) instead of the equation (7), the matrix operation equation (10) is applied.

  As a result, pulse reference angle outputs from S1 to S8 are obtained. At this time, the number of pulses per cycle of the line voltage is 32.

  The relationship between the number of harmonic components to be deleted from the line voltage of the motor generator and the number of pulses is generally as follows. That is, when there are two harmonic components to be deleted, the number of pulses per cycle of the line voltage is 8 pulses, and when there are 3 harmonic components to be deleted, the number of pulses per cycle of the line voltage Is 16 pulses, and when there are 4 harmonic components to be deleted, the number of pulses per cycle of the line voltage is 32 pulses, and when there are 5 harmonic components to be deleted, one cycle of the line voltage The number of hits is 64 pulses. Similarly, as the number of harmonic components to be deleted increases by one, the number of pulses per cycle of the line voltage doubles. Here, in the normal motor generator line voltage, high-order harmonics that are multiples of 3 cancel each other out, so it is not necessary to add them to the harmonic components to be deleted. However, in this PHM pulse generation calculation process, only the third harmonic is included in the harmonic component to be deleted for convenience.

  However, in the case of a pulse arrangement in which a positive pulse and a negative pulse are overlapped by a line voltage, the number of pulses may be different from the above.

  With the PHM pulse signal generated in the pulse generator 434 as described above, a pulse waveform is formed in each of three types of line voltages, that is, a UV line voltage, a VW line voltage, and a WU line voltage. The pulse waveforms of these line voltages are the same pulse waveform having a phase difference of 2π / 3. Therefore, only the UV line voltage will be described below as a representative of each line voltage.

  Here, the relationship between the reference phase θuvl of the voltage between the UV rays, the voltage phase signal θv, and the magnetic pole position θe is represented by the equation (11).

  θuvl = θv + π / 6 = θe + δ + 7π / 6 [rad] (11)

  The waveform of the UV line voltage represented by Equation (11) is line symmetric about the position of θuvl = π / 2, 3π / 2, and point-symmetrical about the position of θuvl = 0, π. Become. Therefore, the waveform of one cycle of UV voltage pulse (θuvl is from 0 to 2π) is symmetrical or up / down every π / 2 based on the pulse waveform between θuvl from 0 to π / 2. It can be expressed by arranging them symmetrically.

  One method for realizing this is to compare the center phase of the UV line voltage pulse in the range of 0 ≦ θuvl ≦ π / 2 with a 4-channel phase counter, and based on the comparison result, one period, that is, 0 ≦ θuvl. This is an algorithm for generating a UV line voltage pulse in a range of ≦ 2π. The conceptual diagram is shown in FIG.

  FIG. 11 shows an example in which there are four line voltage pulses in the range of 0 ≦ θuvl ≦ π / 2. In FIG. 11, pulse reference angles S1 to S4 represent the center phases of the four pulses.

  carr1 (θuvl), carr2 (θuvl), carr3 (θuvl), and carr4 (θuvl) represent each of the 4-channel phase counters. Each of these phase counters is a triangular wave having a period of 2π rad with respect to the reference phase θuvl. Further, carr1 (θuvl) and carr2 (θuvl) have a deviation of dθ in the amplitude direction, and the relationship between carr3 (θuvl) and carr4 (θuvl) is the same.

  dθ represents the width of the line voltage pulse. The amplitude of the fundamental wave changes linearly with respect to this pulse width dθ.

  The line voltage pulse is a pulse reference angle S1 that represents the center phase of each pulse in the range of 0 ≦ θuvl ≦ π / 2 and each phase counter carr1 (θuvl), carr2 (θuvl), carr3 (θuvl), carr4 (θuvl) Formed at each intersection with ~ S4. Thereby, a symmetrical pulse signal is generated every 90 degrees.

  More specifically, a pulse with a width dθ having a positive amplitude is generated at a point where carr1 (θuvl), carr2 (θuvl) and S1 to S4 coincide with each other. On the other hand, at the point where carr3 (θuvl), carr4 (θuvl) and S1 to S4 coincide with each other, a pulse having a negative amplitude and a width dθ is generated.

  FIG. 12 shows an example in which the waveform of the line voltage generated using the method described above is drawn for each modulation degree. In FIG. 12, k1 = 1, k2 = 1, and k3 = 3 are selected as the values of k1, k2, and k3 in Equation (7), respectively, and the modulation factor is changed from 0 to 1.0. An example of a voltage pulse waveform is shown. FIG. 12 shows that the pulse width increases almost in proportion to the increase in the modulation degree. The effective value of the voltage can be increased by increasing the pulse width in this way. However, for pulses near θuvl = 0, π, 2π, the pulse width does not change even when the modulation degree changes at a modulation degree of 0.4 or more. Such a phenomenon is caused by overlapping of a pulse having a positive amplitude and a pulse having a negative amplitude.

  As described above, in the above embodiment, each switching element performs a switching operation based on the phase of the AC output to be output by sending a drive signal from the driver circuit 174 to each switching element of the power switching circuit 144. . The number of switching times of the switching element in one cycle of the AC output tends to increase as the number of harmonics to be removed increases.

  From another viewpoint, when the voltage of the supplied DC power decreases, the degree of modulation increases, and the conduction period of each conducting switching operation tends to be long. Also, when driving a rotating electrical machine such as a motor, when the generated torque of the rotating electrical machine is increased, the degree of modulation increases, and as a result, the conduction period of each switching operation becomes longer and the generated torque of the rotating electrical machine decreases. The conduction period of each switching operation is shortened. If the conduction period increases and the interruption time becomes shorter, that is, if the switching interval is shortened to some extent, there is a possibility that the switching element cannot be safely interrupted. Control leading to the conduction period is performed. Conversely, even when the conduction period of each switching operation is shortened and the energization period is shortened, there is a possibility that the switching element cannot be energized safely. In this case, control that leads to the cutoff period is performed without energization.

  From another viewpoint, in a low frequency state where the influence of distortion of the output AC output is large, particularly in a state where the rotating electric machine is stopped or the rotational speed is very low, the PHM system control is not used. The power switching circuit 144 is controlled by a PWM method using a carrier wave of the same, and the power switching circuit 144 is controlled by switching to the PHM method while the rotation speed is increased. When the present invention is applied to a power conversion device for driving an automobile, the stage of starting and accelerating from a stopped state particularly reduces the influence of torque pulsation because it affects the sense of luxury of the car. Is desirable. For this reason, at least when the vehicle starts from a stopped state, the power switching circuit 144 is controlled by the PWM method, and after a certain acceleration, the control is switched to the PHM method. In this way, control with less torque pulsation can be realized at least at the time of starting, and it is possible to control with the PHM method with less switching loss at least in the state of shifting to constant speed driving which is normal operation. Control with less loss can be realized while suppressing the influence of

  The PHM pulse signal used in the present invention is characterized in that when the modulation degree is fixed as described above, a line voltage waveform is formed by a pulse train having the same pulse width except for exceptions. Note that the case where the pulse width of the line voltage is unequal to other pulse trains is an exception when a pulse having a positive amplitude and a pulse having a negative amplitude overlap as described above. In this case, if the portion where the pulses overlap is decomposed into a pulse having a positive amplitude and a pulse having a negative amplitude, the widths of the pulses are always equal throughout. That is, the degree of modulation changes with a change in pulse width.

  Here, the case where the pulse width of the line voltage is unequal to other pulse trains will be described in detail with reference to FIG. The upper part of FIG. 13 shows an enlarged line voltage pulse waveform at a modulation degree of 1.0 in FIG. 12 in the range of π / 2 ≦ θuvl ≦ 3π / 2. In this line voltage pulse waveform, two pulses near the center have different pulse widths from other pulses.

  The lower part of FIG. 13 shows a state where such a part having a different pulse width is disassembled. From this figure, in this part, a pulse having a positive amplitude and a pulse having a negative amplitude each having the same pulse width as other pulses are overlapped, and these pulses are combined to be different from others. It can be seen that a pulse having a pulse width is formed. That is, by decomposing the overlap of pulses in this way, it can be seen that the pulse waveform of the line voltage formed according to the PHM pulse signal is composed of pulses having a constant pulse width.

  FIG. 14 shows another example of the line voltage pulse waveform by the PHM pulse signal generated by the present invention. Here, k1 = 1, k2 = 1, and k3 = 5 are selected as the values of k1, k2, and k3 in Equation (7), respectively, and the line voltage when the modulation degree is changed from 0 to 1.27. An example of a pulse waveform is shown. In FIG. 14, when the degree of modulation is 1.17 or more, there is no gap between two symmetrically adjacent pulses at the positions of θuvl = π / 2 and 3π / 2. Therefore, it can be seen that the target harmonic component can be deleted when the modulation factor is less than 1.17, but the harmonic component cannot be effectively deleted when the modulation factor exceeds this value. As the degree of modulation is further increased, the gap between adjacent pulses disappears at other positions, and finally a rectangular line voltage pulse waveform is obtained at a degree of modulation of 1.27.

  Even in the main line voltage pulse waveform example, the pulse width is not constant, but in the same way as the principle explained in FIG. 13, a pulse having a positive amplitude and a pulse having a negative amplitude having the same pulse width overlap each other. Thus, it is the same that these pulses are combined to form a pulse having a different pulse width.

  FIG. 15 shows an example in which the line voltage pulse waveform shown in FIG. 14 is represented by a corresponding phase voltage pulse waveform. In FIG. 15, as in FIG. 14, it can be seen that the gap between two adjacent pulses disappears when the modulation degree becomes 1.17 or more. There is a phase difference of π / 6 between the phase voltage pulse waveform of FIG. 15 and the line voltage pulse waveform of FIG.

  As shown in FIG. 14, when increasing the modulation level and increasing the peak value of the output AC voltage, the control circuit 172 does not change the number of line voltage pulses in the range where the modulation level is less than 1.17. Increase the width of As a result, the number of conductions and the number of interruptions of the switching element are made the same, and the conduction width is controlled to be increased. On the other hand, when the peak value of the AC output to be output is increased by increasing the modulation degree, the period during which the line voltage pulse is 0, that is, the cutoff width of the switching element is narrowed accordingly, and the modulation degree is reduced. If it is 1.17 or more, the switching element cannot be shut off. In such a state, the control circuit 172 combines the portions where the line voltage pulses overlap with each other to reduce the number of pulses. Accordingly, the conduction state of the switching element is continued during the period, so that the number of conduction times and the number of interruptions of the switching element are reduced. The line voltage pulse waveform by the PHM pulse signal has such characteristics.

  Next, a method for converting a line voltage pulse into a phase voltage pulse will be described. FIG. 16 shows an example of a conversion table used in the conversion from the line voltage pulse to the phase voltage pulse. The modes 1 to 6 described in the leftmost column in this table are assigned numbers for each possible switching state. In modes 1 to 6, the relationship from the line voltage to the output voltage is determined on a one-to-one basis. Each of these modes corresponds to an active period in which energy is transferred between the DC side and the three-phase AC side. Note that the line voltages described in the table of FIG. 16 are obtained by normalizing patterns that can be taken as potential differences between different phases with the battery voltage Vdc.

  In FIG. 16, for example, in mode 1, Vuv → 1, Vvw → 0, and Vu → −1 are indicated, but this is Vu−Vv = Vdc, Vv−Vw = 0, and Vw−Vu = −Vdc. The case is shown normalized. According to the table of FIG. 16, the phase voltage at this time, that is, the phase terminal voltage (proportional to the gate voltage) is Vu → 1 (the U-arm upper arm is on and the lower arm is off), Vv → 0 (V-phase upper arm) Off, lower arm on), Vw → 0 (W-phase upper arm off, lower arm on). That is, the table of FIG. 16 shows a normalized case where Vu = Vdc, Vv = 0, and Vw = 0. Modes 2 to 6 are also based on the same concept as mode 1.

  FIG. 17 shows an example in which the line voltage pulse is converted into the phase voltage pulse in the mode in which the power switching circuit 144 is controlled in a rectangular wave state using the conversion table of FIG. In FIG. 17, the upper part shows the UV line voltage Vuv as a typical example of the line voltage, and the U phase terminal voltage Vu, the V phase terminal voltage Vv, and the W phase terminal voltage Vw are shown below. As shown in FIG. 17, in the rectangular wave control mode, the modes shown in the conversion table of FIG. In the rectangular wave control mode, there is no later-described three-phase short-circuit period.

  FIG. 18 shows a state in which the line voltage pulse waveform illustrated in FIG. 12 is converted into a phase voltage pulse according to the conversion table of FIG. In FIG. 18, the upper stage shows a UV line voltage pulse as a representative example of the line voltage, and the U phase terminal voltage Vu, the V phase terminal voltage Vv, and the W phase terminal voltage Vw are shown below.

  The upper part of FIG. 18 shows the number of the mode (the active period in which energy is transferred between the DC side and the three-phase AC side) and the period in which the three-phase is short-circuited. During the three-phase short-circuit period, either the three-phase upper arm is turned on or the three-phase lower arm is turned on, either switch depending on the switching loss or conduction loss situation. Select a mode.

  For example, when the UV line voltage Vuv is 1, the U-phase terminal voltage Vu is 1 and the V-phase terminal voltage Vv is 0 (modes 1 and 6). When the UV line voltage Vuv is 0, the U-phase terminal voltage Vu and the V-phase terminal voltage Vv are the same value, that is, Vu is 1 and Vv is 1 (mode 2, 3-phase short circuit), or Vu is 0 and Vv is 0 (mode 5, 3-phase short circuit). When the UV line voltage Vuv is −1, the U-phase terminal voltage Vu is 0 and the V-phase terminal voltage Vv is 1 (modes 3 and 4). Based on such a relationship, each pulse of the phase voltage, that is, the phase terminal voltage (gate voltage pulse) is generated.

  In FIG. 18, the pattern of the line voltage pulse and the phase terminal voltage pulse of each phase is a pattern that repeats quasi-periodically with π / 3 as the minimum unit with respect to the phase θuvl. That is, the pattern in which 1 and 0 of the U-phase terminal voltage in the period of 0 ≦ θuvl ≦ π / 3 are inverted is the same as the pattern of the W-phase terminal voltage of π / 3 ≦ θuvl ≦ 2π / 3. A pattern obtained by inverting 1 and 0 of the V-phase terminal voltage in the period of 0 ≦ θuvl ≦ π / 3 is the same as the pattern of the U-phase terminal voltage of π / 3 ≦ θuvl ≦ 2π / 3, and 0 ≦ The pattern obtained by inverting 1 and 0 of the W-phase terminal voltage during the period of θuvl ≦ π / 3 is the same as the pattern of the V-phase terminal voltage of π / 3 ≦ θuvl ≦ 2π / 3. Such a characteristic is particularly noticeable in a steady state where the rotational speed and output of the motor are constant.

  Here, the above modes 1 to 6 are set as the first period in which the upper arm IGBT 328 and the lower arm IGBT 330 are turned on in different phases and current is supplied from the battery 136 as a DC power source to the motor generator 192. Define. Further, the three-phase short-circuit period is defined as a second period in which either the upper arm IGBT 328 or the lower arm IGBT 330 is turned on and the torque is maintained with the energy accumulated in the motor generator 192 in all phases. . In the example shown in FIG. 18, it can be seen that the first period and the second period are alternately formed according to the electrical angle.

  Further, in FIG. 18, for example, in a period of 0 ≦ θuvl ≦ π / 3, modes 6 and 5 as the first period are alternately repeated with a three-phase short-circuit period as the second period interposed therebetween. As can be seen from FIG. 16, in mode 6, the lower arm IGBT 330 is turned on in the V phase, while in the other U phase and W phase, the side different from the V phase, that is, the upper arm IGBT 328 is turned on. doing. On the other hand, in mode 5, the upper arm IGBT 328 is turned on in the W phase, while in the other U phase and V phase, the side different from the W phase, that is, the lower arm IGBT 330 is turned on. That is, in the first period, one of the U phase, the V phase, and the W phase (the V phase in mode 6 and the W phase in mode 5) is selected, and the selected one phase is used for the upper arm. IGBT 328 or lower arm IGBT 330 is turned on, and for the other two phases (U phase and W phase in mode 6, U phase and V phase in mode 5), IGBT 328 for the arm on the side different from the selected one phase , 330 are turned on. Moreover, the 1 phase (V phase, W phase) selected for every 1st period is replaced.

  In the period other than 0 ≦ θuvl ≦ π / 3, similarly to the above, any one of the modes 1 to 6 as the first period is alternately repeated with the three-phase short-circuit period as the second period in between. That is, modes 1 and 6 are performed in the period of π / 3 ≦ θuvl ≦ 2π / 3, modes 2 and 1 are performed in the period of 2π / 3 ≦ θuvl ≦ π, and modes 3 and 2 in the period of π ≦ θuvl ≦ 4π / 3. The modes 4 and 3 are alternately repeated in the period of 4π / 3 ≦ θuvl ≦ 5π, and the modes 5 and 4 are alternately repeated in the period of 5π / 3 ≦ θuvl ≦ 2π. Accordingly, in the same manner as above, in the first period, any one of the U phase, the V phase, and the W phase is selected, and the IGBT 328 for the upper arm or the IGBT 330 for the lower arm is selected for the selected one phase. At the same time, the IGBTs 328 and 330 for the arm on the side different from the selected one phase are turned on for the other two phases. Moreover, the 1 phase selected for every 1st period is replaced.

  As described above, when the upper arm IGBT 328 of any two phases of the U phase, the V phase, and the W phase is in the conductive state, the other one-phase upper arm IGBT 328 is in the conductive state. It becomes a three-phase short-circuit period, and it will be in any mode by setting it as a cutoff state. That is, the control circuit 172 can control the three-phase short-circuit period by turning on and off the IGBT 328. Similarly, when the IGBT 330 for the lower arm of any two phases of the U phase, the V phase, and the W phase is in the conductive state, the IGBT 330 for the lower arm of the other one phase is similarly set in the conductive state by 3 It becomes a short circuit period, and it will be in either mode by setting it as a cutoff state. That is, the control circuit 172 can control the three-phase short-circuit period by turning on and off the IGBT 330.

  By the way, the electrical angle position forming the first period, that is, the period of modes 1 to 6 and the length of this period can be changed in accordance with a request command such as torque or rotational speed for the motor generator 192. . That is, as described above, the specific electrical angle position forming the first period is changed in order to change the order of the harmonics to be deleted in accordance with changes in the rotational speed and torque of the motor. Alternatively, the modulation factor is changed by changing the length of the first period, that is, the pulse width, in accordance with changes in the rotational speed or torque of the motor. Thereby, the waveform of the alternating current flowing through the motor, more specifically, the harmonic component of the alternating current is changed to a desired value, and the electric power supplied from the battery 136 to the motor generator 192 can be controlled by this change. . Note that only one of the specific electrical angle position and the length of the first period may be changed, or both may be changed simultaneously.

  Here, the pulse shape and voltage have the following relationship. The illustrated pulse width has an effect of changing the effective value of the voltage. When the pulse width of the line voltage is wide, the effective value of the voltage is large, and when it is narrow, the effective value of the voltage is small. When the number of harmonics to be deleted is small, the effective value of the voltage is high, so that the upper limit of the modulation degree approaches a rectangular wave. This effect is effective in the rotation region where the induced voltage of the rotating electrical machine (motor generator 192) is high, and a voltage higher than the line voltage when controlled by normal PWM can be supplied to the rotating electrical machine. That is, by changing the length of the first period for supplying power from the battery 136 that is a DC power source to the motor generator 192 and the specific electrical angle position forming the first period, the voltage is applied to the motor generator 192. By changing the effective value of the alternating voltage to be output, an output corresponding to the rotation state of the motor generator 192 can be obtained.

  Further, the pulse shape of the drive signal shown in FIG. 18 is asymmetrical about an arbitrary θuvl, that is, an electrical angle, for each of the U phase, the V phase, and the W phase. Furthermore, at least one of the on period and the off period of the pulse includes a period in which θuvl (electrical angle) continues for π / 3 or more. For example, the U phase has an on period of π / 6 or more around the vicinity of θuvl = π / 2 and an off period of π / 6 or more around the vicinity of θuvl = 3π / 2. Similarly, the V phase has an off period of π / 6 or more centered around θuvl = π / 6 and an on period of π / 6 or more centered around θuvl = 7π / 6. The W phase has an off period of π / 6 or more around θuvl = 5π / 6, and an on period of π / 6 or more around θuvl = 11π / 6. . As described above, the number of pulses per electrical angle 2π of each phase of the U phase, V phase, and W phase is sequentially determined according to the number of pulses of the line voltage, but the pulse intervals between the electrical angles 2π are not uniform. It is. It has such a pulse shape feature.

  As described above, according to the power conversion device of the present embodiment, when the PHM control mode is selected, the first period in which power is supplied from the DC power supply to the motor and all phases of the three-phase full bridge The second period during which the upper arm is turned on or the lower arm of all phases is turned on is alternately generated at a specific timing according to the electrical angle. Thereby, compared with the case where the PWM control mode is selected, the switching frequency may be 1/7 to 1/10 or less. Therefore, switching loss can be reduced.

  Next, how the harmonic component is deleted from the line voltage pulse waveform when the modulation degree is changed as illustrated in FIG. 14 will be described. FIG. 19 is a diagram illustrating the amplitudes of the fundamental wave and the harmonic component to be deleted in the line voltage pulse when the modulation degree is changed.

  FIG. 19 (a) shows an example of the fundamental wave and the amplitude of each harmonic in the line voltage pulse for which the third and fifth harmonics are to be deleted. According to this figure, it can be seen that the fifth harmonic appears without being completely deleted when the modulation degree is 1.2 or more. FIG. 19B shows an example of the fundamental wave and the amplitude of each harmonic in the line voltage pulse for which the third, fifth and seventh harmonics are to be deleted. According to this figure, it can be seen that the fifth and seventh harmonics are not completely deleted when the modulation degree is 1.17 or more.

  Examples of the line voltage pulse waveform and the phase voltage pulse waveform corresponding to FIG. 19A are shown in FIGS. Here, a row vector having 2 elements is set, k1 = 1 and k2 = 3 are selected as the values of k1 and k2 in each element (k1 / 3, k2 / 5), and the modulation degree is 0. 6 shows an example of a line voltage pulse waveform and a phase voltage waveform when changing from 1 to 1.27. FIG. 19B corresponds to the line voltage pulse waveform and the phase voltage pulse waveform shown in FIGS.

  From the above description, it can be seen that when the degree of modulation exceeds a certain value, the harmonics to be deleted begin to appear without being completely deleted. It can also be seen that the higher the number (number) of harmonics to be deleted, the more the harmonics cannot be deleted with a lower modulation degree.

  Next, a method of generating a PWM pulse signal in the pulse modulator 440 for PWM control shown in FIG. 6 will be described with reference to FIG. FIG. 22A shows waveforms of the voltage command signal in each phase of the U phase, the V phase, and the W phase and the triangular wave carrier used for generating the PWM pulse. The voltage command signal for each phase is a sine wave command signal whose phases are shifted by 2π / 3 from each other, and the amplitude changes according to the degree of modulation. The voltage command signal and the triangular wave carrier signal are compared for each of the U, V, and W phases, and the intersection of the two is set as the on / off timing of the pulse, whereby FIGS. 22 (b), (c), and (d) Voltage pulse waveforms for the U phase, V phase, and W phase as shown in FIG. Note that the number of pulses in these pulse waveforms is equal to the number of triangular wave pulses in the triangular wave carrier.

  FIG. 22 (e) shows the waveform of the voltage between UV rays. The number of pulses is equal to twice the number of triangular wave pulses in the triangular wave carrier, that is, twice the number of pulses in the voltage pulse waveform for each phase. The same applies to other line voltages, that is, the VW line voltage and the WU line voltage.

  FIG. 23 shows an example in which the waveform of the line voltage formed by the PWM pulse signal is drawn for each modulation degree. Here, an example of a line voltage pulse waveform when the modulation degree is changed from 0 to 1.27 is shown. In FIG. 23, when the degree of modulation is 1.17 or more, there is no gap between two adjacent pulses, and a total of one pulse. Such a pulse signal is called an overmodulated PWM pulse. Eventually, the line voltage pulse waveform is a rectangular wave at a modulation degree of 1.27.

  FIG. 24 shows an example in which the line voltage pulse waveform shown in FIG. 23 is represented by a corresponding phase voltage pulse waveform. 24, as in FIG. 23, it can be seen that the gap between two adjacent pulses disappears when the modulation degree is 1.17 or more. Note that there is a phase difference of π / 6 between the phase voltage pulse waveform of FIG. 24 and the line voltage pulse waveform of FIG.

  Here, the line voltage pulse waveform by the PHM pulse signal is compared with the line voltage pulse waveform by the PWM pulse signal. FIG. 25 (a) shows an example of a line voltage pulse waveform by a PHM pulse signal. This corresponds to a line voltage pulse waveform having a modulation degree of 0.4 in FIG. On the other hand, FIG. 25B shows an example of the line voltage pulse waveform by the PWM pulse signal. This corresponds to a line voltage pulse waveform having a modulation degree of 0.4 in FIG.

  25 (a) and FIG. 25 (b) are compared with respect to the number of pulses, the line voltage pulse waveform based on the PHM pulse signal shown in FIG. 25 (a) is based on the PWM pulse signal shown in FIG. 25 (b). It can be seen that the number of pulses is significantly smaller than the line voltage pulse waveform. Therefore, when the PHM pulse signal is used, the control responsiveness is lower than the case of the PWM signal because the number of generated line voltage pulses is small. However, the number of times of switching is greatly reduced as compared with the case of using the PWM signal. Can do. As a result, switching loss can be greatly reduced.

  Next, the difference in pulse shape between PWM control and PHM control will be described with reference to FIG. FIG. 26A shows a triangular wave carrier used for generating a PWM pulse signal, and a U-phase voltage, a V-phase voltage, and a UV line voltage generated by the PWM pulse signal. FIG. 26B shows the U-phase voltage, the V-phase voltage, and the UV line voltage generated by the PHM pulse signal. Comparing these figures, when the PWM pulse signal is used, the pulse width of each pulse of the UV line voltage is not constant, whereas when the PHM pulse signal is used, the pulse of each UV line voltage is It can be seen that the pulse width is constant. As mentioned above, the pulse width may not be constant, but this is due to the overlap of a pulse with a positive amplitude and a pulse with a negative amplitude. The same pulse width is obtained with this pulse. In addition, when the PWM pulse signal is used, the triangular wave carrier is constant regardless of the fluctuation of the motor rotation speed, so that the interval of each pulse of the UV line voltage is also constant regardless of the motor rotation speed. When the PHM pulse signal is used, it can be seen that the interval of each pulse of the UV line voltage changes according to the motor rotation speed.

  FIG. 27 shows the relationship between the motor rotation speed and the line voltage pulse waveform based on the PHM pulse signal. FIG. 27A shows an example of a line voltage pulse waveform by a PHM pulse signal at a predetermined motor rotation speed. This corresponds to a line voltage pulse waveform with a modulation degree of 0.4 in FIG. 12, and has 16 pulses per 2π electrical angle (reference phase θuvl of UV line voltage).

  FIG. 27B shows an example of the line voltage pulse waveform by the PHM pulse signal when the motor rotation speed of FIG. 27A is doubled. The length of the horizontal axis in FIG. 27B is equivalent to that in FIG. 27A with respect to the time axis. When comparing FIG. 27 (a) and FIG. 27 (b), the number of pulses per electrical angle 2π remains unchanged at 16 pulses, but the number of pulses within the same time is doubled in FIG. 27 (b). I understand that.

  FIG. 27 (c) shows an example of a line voltage pulse waveform by a PHM pulse signal when the motor rotation speed of FIG. 27 (a) is halved. Note that the length of the horizontal axis in FIG. 27C is also equivalent to that in FIG. 27A with respect to the time axis, as in FIG. When comparing FIG. 27A and FIG. 27C, since the number of pulses per electrical angle π is 8 in FIG. 27C, the number of pulses per electrical angle 2π is 16 pulses. It can be seen that the number of pulses in the same time is ½ times in FIG.

  As described above, when the PHM pulse signal is used, the number of line voltage pulses per unit time changes in proportion to the motor rotation speed. That is, considering the number of pulses per electrical angle 2π, this is constant regardless of the motor rotation speed. On the other hand, when the PWM pulse signal is used, as described in FIG. 26, the number of line voltage pulses is constant regardless of the motor rotation speed. That is, considering the number of pulses per electrical angle 2π, this decreases as the motor rotation speed increases.

  FIG. 28A shows the relationship between the number of line voltage pulses per electrical angle 2π (that is, per line voltage period) generated in the PHM control and PWM control, respectively, and the motor rotation speed. FIG. 28B shows the relationship between the number of phase voltage pulses per electrical angle 2π (that is, per phase voltage period) generated in the PHM control and PWM control, respectively, and the motor rotation speed. In FIGS. 28A and 28B, an 8-pole motor (pole pair number 4) is used, and the harmonic components to be deleted in the PHM control are set to the third, fifth, and seventh orders, and the sine wave PWM control is performed. An example in which the frequency of the triangular wave carrier used in FIG. As described above, the number of line voltage pulses and the number of phase voltage pulses per electrical angle 2π decrease as the motor rotation speed increases in the case of PWM control, whereas in the case of PHM control, the number of line voltage pulses and the phase voltage pulse decrease. It turns out that it is constant regardless. Note that the number of line voltage pulses in the PWM control can be obtained by Expression (12).

(Number of voltage pulses between lines)
= (Frequency of triangular wave carrier) / {(number of pole pairs) × (motor rotational speed) / 60} × 2
(12)

  In FIGS. 28A and 28B, the number of line voltage pulses per cycle of the line voltage when the number of harmonic components to be deleted in PHM control is three is 16, and the number of phase voltage pulses However, the number of line voltage pulses changes as described above according to the number of harmonic components to be deleted. That is, when there are two harmonic components to be deleted, 8 when there are four harmonic components to be deleted, 64 when there are five harmonic components to be deleted, and so on. As the number of harmonic components to be deleted increases by one, the number of pulses per cycle of the line voltage doubles.

  FIG. 29 shows a flowchart of motor control performed by the control circuit 172 according to the embodiment described above. In step 901, the control circuit 172 acquires the rotational speed information of the motor generator. The rotational speed information is obtained based on the magnetic pole position signal θ output from the rotating magnetic pole sensor 193.

  In step 902, the control circuit 172 determines whether the motor rotation speed is equal to or higher than a predetermined switching rotation speed based on the rotation speed information acquired in step 901. If the motor rotation speed is equal to or higher than the switching rotation speed, the process proceeds to step 904, and if it is less than the switching rotation speed, the process proceeds to step 903.

  In step 904, the control circuit 172 determines the order of the harmonics to be deleted in the PHM control. Here, as described above, harmonics such as the third order, the fifth order, and the seventh order can be determined as deletion targets. Note that the number of harmonics to be deleted may be changed according to the motor rotation speed. For example, when the motor rotation speed is relatively low, the third, fifth, and seventh harmonics are targeted for deletion, and when the motor rotation speed is relatively high, the third and fifth harmonics are targeted for deletion. Thus, by reducing the number of harmonics to be deleted as the motor rotation speed increases, the number of pulses of the PHM pulse signal is reduced in a high-speed rotation region that is not easily affected by torque pulsation due to the harmonics, and switching loss. Can be more effectively reduced.

  In step 905, the control circuit 172 performs PHM control in which the harmonics of the order determined in step 904 are to be deleted. At this time, a PHM pulse signal corresponding to the order of the harmonics to be deleted is generated by the pulse modulator 430 according to the generation method as described above, and the PHM pulse signal is selected by the switch 450, and from the control circuit 172 It is output to the driver circuit 174. After step 905 is executed, the control circuit 172 returns to step 901 and repeats the above processing.

  In step 906, the control circuit 172 performs rectangular wave control. As described above, the rectangular wave control can be considered as one form of the PHM control, that is, the one in which the degree of modulation is maximized in the PHM control, or the harmonic order to be deleted is not present. Although the harmonic wave cannot be deleted by the rectangular wave control, the number of times of switching can be minimized. Note that the pulse signal used for the rectangular wave control can be generated by the pulse modulator 430 as in the case of the PHM control. This pulse signal is selected by the switch 450 and output from the control circuit 172 to the driver circuit 174. After step 906 is executed, the control circuit 172 returns to step 901 and repeats the above processing.

In step 903, the control circuit 172 performs PWM control. At this time, the PWM pulse signal is generated in the pulse modulator 440 by the generation method as described above based on the comparison result between the predetermined triangular wave carrier and the voltage command signal, and the PWM pulse signal is selected by the switch 450. And output from the control circuit 172 to the driver circuit 174. After executing Step 903, the control circuit 172 returns to Step 901 and repeats the above processing.

  According to the form and the PHM control mode of the present embodiment described above, the following effects can be obtained by using the PHM control mode that achieves the above-described effects and further reduces the number of switching times of the switching element as compared with the PWM control mode. There is also an effect.

FIG. 30A shows the U-phase voltage pulse waveform extracted from (b) from each waveform of the PWM control shown in FIG. Here, assuming that each pulse width of the PWM phase voltage is Duty = 50% as shown in FIG. 30 (b), each pulse shape is approximated by a trapezoidal wave having rise and fall times as shown in FIG. 30 (c). In this case, the characteristics of the PWM phase voltage pulse waveform shown in FIG. 30A can be expressed by the characteristic values shown in the following equation (13), for example.
PWM carrier frequency: Fc = 10 (kHz)
Rise / fall time: τp = 0.2 (μs)
Pulse period: Tp = 1 / Fc (s)
Pulse width: αp = 0.5 Tp (s) (From Duty 50%, αp / Tp = 0.5)
(13)

FIG. 31A shows the U-phase voltage pulse waveform (3rd, 5th, and 7th harmonics deleted) extracted from each waveform of the PHM control shown in FIG. Here, assuming that the pulse width of the PHM phase voltage is 50% per electrical angle 2π as shown in FIG. 31 (b), each pulse shape has a rise and fall time as shown in FIG. 31 (c). The characteristics of the PHM phase voltage pulse waveform shown in FIG. 31A can be expressed by the characteristic values shown in the following equation (14), for example.
Rise / fall time: τh = 0.2 (μs)
Number of motor generator pole pairs: P = 4
Motor generator rotation speed: N = 2000 (r / min)
Number of PHM phase voltage pulses: n = 11 (pulse / 2π)
Pulse period: Th = 1 / (P × (N / 60) × n) (s)
Pulse width: αh = 0.5 Th (s) (From Duty 50%, αh / Th = 0.5)
(14)

  FIG. 32 is a graph showing the PWM control pulse width αp and the PHM control pulse width αh according to equations (13) and (14). In the case of PWM control, the value of αp is determined by the value of the carrier frequency Fc and is constant as long as the carrier frequency Fc does not change. On the other hand, in the case of PHM control, the number of pulses is smaller than that of PWM, and as shown in FIG. 28, the number of pulses per electrical angle 2π is constant regardless of the rotational speed of the motor generator. As shown in (), the number of pulses per unit time decreases and the value of αh increases as the speed decreases.

  FIG. 33A shows a voltage spectrum of the U-phase voltage in PWM control from the rise / fall time τp and the pulse width αp shown in FIGS. FIG. 33B shows the voltage spectrum of the U-phase voltage in the PHM control from the rise / fall time τh and the pulse width αh shown in FIGS. In FIGS. 33A and 33B, the voltage spectrum of the U-phase voltage is shown as a representative example of the phase voltage spectrum, but the voltage spectrum of the V-phase and the W-phase may be considered the same. In FIG. 33 (b), the phase voltage spectrum by the PWM control of FIG. 33 (a) is indicated by a dotted line for comparison. 33 (a) and 33 (b), it can be seen that the amplitude of the phase voltage spectrum at the same frequency is smaller in the PHM control than in the PWM control. This is because the PHM control can reduce the number of switching of the switching element while suppressing the torque ripple of the motor generator 192.

  FIG. 33 (b) shows the phase voltage spectrum by PHM control when the motor generator rotation speed is N = 2000 (r / min). As shown in the graph of FIG. Similarly, in the range of the motor generator rotation speed larger than αp, the phase voltage spectrum of the PHM control tends to be lower than the phase voltage spectrum of the PWM control.

  Here, in FIG. 2A, the leakage current 192i flowing in the stray capacitance (stray capacitor) 192nc between the neutral point 192n and the GND of the motor generator 192 is caused by the potential difference between the neutral point 192n and GND. This leakage current 192i flows via GND into a stray capacitor 192gc between motor generator 192 and GND, a stray capacitor 170gc between control unit 170 and GND, and a stray capacitor 200gc between power converter 200 and GND. That is, the stray capacitor 192nc between the neutral point 192n and GND can be regarded as a common mode noise source as shown in FIG.

  From the results of FIG. 33 and the equation (1), the voltage spectrum of the neutral point voltage Vn can be lowered at the same time by lowering the voltage spectrum of the U, V, W phase voltages. That is, in the PHM control, the number of switching times of the switching element is reduced as compared with the PWM control, so that not only the neutral point voltage spectrum can be reduced, but also the neutral point voltage fluctuation number itself can be reduced. Therefore, the common mode noise shown in FIG. 2B can be reduced.

-Modification-
The embodiment described above can be modified as follows.

(1) In each of the above embodiments, the PHM control including the rectangular wave control is performed if the motor rotation speed is equal to or higher than the predetermined switching rotation speed, and the PWM control is performed if the motor rotation speed is less than the switching rotation speed, whereby the power conversion device. In 140, the control mode is switched. However, such control mode switching is not limited to the mode described in each embodiment, and can be applied at any motor rotation speed. For example, when the motor rotation speed is 0 to 10,000 r / min, PWM control is performed in the range of 0 to 1,500 r / min, PHM control is performed in the range of 1,500 to 4,000 r / min, and in the range of 4,000 to 6,000 r / min. PWM control and PHM control can be performed in the range of 6,000 to 10,000 r / min, respectively. In this way, it is possible to realize even finer motor control using an optimal control mode according to the motor rotation speed.

(2) Next, another form of PHM control will be described. In normal PHM control, first, the harmonic order to be deleted is selected from the motor generator line voltage, but now the creation of a PHM pulse pattern is considered for the purpose of reducing the neutral point voltage fluctuation. Normally, as described above, when the third, fifth, and seventh harmonic components are specified in step 821 in FIG. 10 as the harmonic order to be deleted, the pulse generator 434 performs matrix calculation in the next step 822. For the third, fifth, and seventh order erasure orders, a row vector as shown in equation (7) is created. Each element in the right parenthesis of Expression (7) is k1 / 3, k2 / 5, k3 / 7. Any odd number can be selected for k1, k2, and k3, but k1 = 3, 9, 15, k2 = 5, 15, 25, k3 = 7, 21, 35, etc. must not be selected. Under this condition, the third, fifth and seventh order components are completely eliminated. What has been described above is the rule of the harmonic compliant phase vector for harmonic elimination.

  Here, consider a PHM pulse pattern in which the deleted harmonic order is only the third and fifth orders and the neutral point voltage fluctuation is reduced instead of the seventh order harmonic deletion. The third term k3 / 7 on the right side of the above equation (7) is obtained in order to eliminate the 7th harmonic, and the value of k3 is a function k3 = f (a) of the modulation degree signal a, and the value is an odd number. It is assumed that any value can be taken without being limited to an integer value such as.

  When the expression (8) is expanded, the following expression (15) is obtained.

Further, by substituting equation (7) and selecting, for example, k1 = 1, k2 = 1, k3 = f (a), equation (15) can be expressed as equation (16) below.

  Here, from the equation (16), the pulse reference angles S1 to S4 can be expressed as shown in FIG. In FIG. 35, the interval between the pulse reference angles S1 and S2 and the interval between S3 and S4 are πf (a) / 7, respectively. The interval between the midpoint of the pulse reference angles S1 and S2 and the midpoint of S3 and S4 is π / 5. The pulse reference angles S1 to S4 that satisfy these conditions are arranged for each π / 2 in terms of the UV line voltage, and the center interval between the combinations of adjacent pulse reference angles S1 to S4 is π / 3.

  In FIG. 35, two pulses are combined into one pulse at positions where the voltage phase between UV rays is π / 2 and 3π / 2. Each pulse width in the line voltage of the PHM control is equal as described above, and the pulse widths at the positions of π / 2 and 3π / 2 are two line voltage pulses. That is, at the positions of π / 2 and 3π / 2, the two pulses are continuously output without a gap so as not to overlap each other. Since the line voltage pulse width of the PHM control changes in proportion to the modulation degree, if the pulse reference angles S1 to S4 do not change, the two pulses at the positions of π / 2 and 3π / 2 will increase as the modulation degree increases. If they overlap each other and become smaller, the two pulses are separated from each other. However, if the value of f (a) is appropriately selected according to the degree of modulation, the two pulses can be combined into one pulse at π / 2 and 3π / 2 in the UV line voltage phase without overlapping each other. Is possible.

  f (a) is a function of the modulation factor a, and is set so as to decrease as the modulation factor a increases and increase as the modulation factor a decreases. As a result, as shown in FIG. 38, when the modulation degree a increases, the value of f (a) is decreased, and the position of the pulse reference angle S1 is moved away from π / 2 and 3π / 2 by the increase of the pulse width. To prevent overlapping of two pulses. Also, if the degree of modulation decreases, the value of f (a) is increased, and the position of the pulse reference angle S1 is brought closer to π / 2 and 3π / 2 by the decrease in the pulse width so that the two pulses do not move away. To. When three harmonics are to be deleted, the number of line voltage pulses is normally 16 pulses per cycle as described above. However, in the case of the PHM pulse pattern for reducing the neutral point voltage fluctuation as described above, the two pulses are combined into one pulse at the positions of π / 2 and 3π / 2. The number is 14 pulses per cycle.

  When determining the positions of the pulse reference angles S1 to S4 shown in FIG. 35, the values of the first term k1 / 3 and the second term k2 / 5 in the equation (7) are in accordance with the rules of the harmonic compliant phase vector. For example, as shown in Expression (16), an arbitrary odd number excluding k1 = 3, 9, 15, k2 = 5, 15, 25, etc. is selected. Therefore, the third harmonic and the fifth harmonic can be deleted by the PHM pulse pattern shown in FIG. On the other hand, the value of k3 / 7 in the third term on the right side in equation (7) is a function f (a) of the modulation degree a as in equation (16) without following the harmonic-based phase vector rule. The harmonics of the component cannot be deleted, but the neutral point voltage fluctuation can be reduced instead.

  When the pulse reference angles S1 to S4 are determined in this way, a pulse is obtained by triangular wave comparison as shown in FIG. Here, using the same method as described with reference to FIG. 11, using a four-channel phase counter, the pulse width is dθ at positions other than π / 2 and 3π / 2, and the phases are π / 2 and 3π. At the position of / 2, a line voltage pulse with a pulse width of 2dθ can be generated.

  36 shows the waveforms of the phase voltage pulse and the line voltage pulse in the PHM control of the neutral point voltage fluctuation suppression pattern in which the pulse train of FIG. 35 is expanded from the line voltage pulse to the phase voltage pulse in the same manner as FIG. It represents the fluctuation of the neutral point voltage according to the pulse waveform. On the other hand, FIG. 34 shows the waveform of the phase voltage pulse and the line voltage pulse in the PWM control, and the fluctuation of the neutral point voltage according to these pulse waveforms.

  Comparing the fluctuation of the neutral point voltage by the PWM control of FIG. 34 and the fluctuation of the neutral point voltage by the PHM control of the neutral point voltage fluctuation suppression pattern of FIG. 36, there are the following two differences. The first difference is that the number of phase voltage pulses per cycle is almost the same in both cases, but the number of fluctuations (fluctuation period) of the neutral point voltage is larger due to the PWM control shown in FIG. . The second difference is that when the magnitude of the fluctuation of the neutral point voltage generated according to each pulse of the U-phase, V-phase, and W-phase voltages is compared, ΔV1, ΔV2 (ΔV2 <ΔV1) in the PWM control shown in FIG. ), Two types of fluctuations having different sizes are mixed, but in the PHM control, there is no fluctuation of ΔV1, and all of them fall within the fluctuation of ΔV2. That is, in the case of PHM control of the neutral point voltage fluctuation suppression pattern shown in FIG. 36, the number of line voltage pulses increases from 8 to 14 as compared with the case of removing the fifth harmonic, but instead the neutral point voltage Fluctuation can be suppressed.

  In FIG. 36, in the period from the phase θ1 to π / 2 where the output of the UV line voltage pulse having a positive amplitude is started, both the U-phase voltage and the W-phase voltage are fixed to the positive voltage side and do not change. On the other hand, only the V-phase voltage has changed. That is, during this period, the U-phase and W-phase upper arm IGBTs 328 are connected in parallel to supply current to the stator winding of the motor generator 192, and within the conduction time, the V-phase lower arm The IGBT 330 repeats conduction and interruption operations a plurality of times, and current from the stator winding of the motor generator 192 flows through the IGBT 330. Further, during the period from the phase θ2 at which the output of the UV line voltage pulse having a positive amplitude ends to the phase θ3 at which the output of the UV line voltage pulse having a negative amplitude is started, the U phase voltage and the V phase voltage are output. Both are fixed to the positive voltage side and do not change, while only the W-phase voltage is changing, and during the period from 3π / 2 to the phase θ4 when the output of the UV line voltage pulse having a negative amplitude ends, Both the V-phase voltage and the W-phase voltage are fixed to the positive voltage side and do not change, while only the U-phase voltage changes. Similarly during these periods, the upper arm IGBT 328 is connected in parallel in the U phase and the V phase or the V phase and the W phase to supply current to the stator winding of the motor generator 192 and within the conduction time. The W-phase or U-phase lower arm IGBT 330 repeatedly conducts and shuts off a plurality of times, and the current from the stator winding of the motor generator 192 flows through the IGBT 330.

  In each of the above periods, the conduction time of the two-phase upper arm IGBT 328 in the parallel connection state is longer than the conduction time of the remaining one-phase lower arm IGBT 330. In the following description, the operation area of the switching element corresponding to these periods is referred to as a first operation area.

  In FIG. 36, during the period from π / 2 to θ2, both the V-phase voltage and the W-phase voltage are fixed to the negative voltage side and do not change, while only the U-phase voltage changes. That is, during this period, the lower-arm IGBT 330 is connected in parallel in the V-phase and the W-phase, and the current from the stator winding of the motor generator 192 flows through these IGBTs 330 and the U-phase is within the conduction time. The upper arm IGBT 328 repeats conduction and interruption several times, and supplies current to the stator winding of the motor generator 192 from the IGBT 328. In the period from θ3 to 3π / 2, both the U-phase voltage and the W-phase voltage are fixed to the negative voltage side and do not change, while only the V-phase voltage changes. In the period from θ4 to θ1, Both the U-phase voltage and the V-phase voltage are fixed to the negative voltage side and do not change, while only the W-phase voltage changes. Similarly, during these periods, the lower arm IGBT 330 is connected in parallel in the U phase and the W phase or the U phase and the V phase, and the current from the stator winding of the motor generator 192 flows through these IGBTs 330. The V-phase or W-phase upper arm IGBT 328 repeatedly conducts and shuts off a plurality of times within the conduction time, and supplies current from the IGBT 328 to the stator winding of the motor generator 192.

  In each of the above periods, the conduction time of the two-phase lower arm IGBT 330 in the parallel connection state is longer than the conduction time of the remaining one-phase upper arm IGBT 328. In the following description, the operation area of the switching element corresponding to these periods is referred to as a second operation area.

  In the first operating region, the first period described above is formed by setting the IGBT 330 for the lower arm, which is not connected in parallel, to the conductive state (the IGBT 328 for the upper arm is in the cut-off state). The above-described second period (three-phase short-circuit period) is formed by turning off the lower arm IGBT 330 (the upper arm IGBT 328 is conductive). Similarly, in the second operation region, the above-described first period is formed by setting the IGBT 328 for the upper arm, which is not connected in parallel, to the conductive state (the IGBT 330 for the lower arm is cut off). The above-described second period (three-phase short-circuit period) is formed by setting the arm IGBT 328 in a cut-off state (the lower arm IGBT 330 is in a conductive state). That is, the conduction time of the power switching circuit 144 is the conduction time of the IGBT 330 for the lower arm that is not connected in parallel (in the case of the first operation region) or the conduction time of the IGBT 328 for the upper arm (in the second operation region). Controlled).

  By forming the pulse waveform of each phase as described above, voltage changes in the U-phase, V-phase, and W-phase in the motor generator 192 can be suppressed as low as possible. As a result, the fluctuation of the neutral point voltage can be reduced.

  In addition, it can be seen that the above-described characteristics appear between the phase voltages even in the pulse waveform in the normal PHM control shown in FIG. However, in this case, the phase voltages of the two phases that are in the parallel connection state are not completely fixed, and change only a few times (one time) in either one of the phases. Even in this case, as in the case of FIG. 36, the change in voltage of the U phase, V phase, and W phase in the motor generator 192 can be suppressed as low as possible compared with the PWM control, thereby reducing the fluctuation of the neutral point voltage. It is possible to achieve the operational effects.

  The above description is merely an example, and the present invention is not limited to the configuration of each of the above embodiments.

43 power converter 110 electric vehicle 112 front wheel 114 front wheel axle 116 front wheel side differential gear (front wheel side DEF)
118 Transmission 120 Engine 122 Power split mechanism 136 Battery 138 DC connector 200 Power converter 140 Power converter 142 Power converter 144 Power switching circuit 150 Upper and lower arm series circuit 153 Collector electrode 154 Gate electrode 155 Emitter electrode 156 Diode 157 Positive terminal (P terminal)
158 Negative terminal (N terminal)
159 AC terminal 163 Collector electrode 164 Gate electrode 165 Emitter electrode 166 Diode 169 Connection point 170 Control unit 170 gc Stray capacitance between the control unit and GND (stray capacitor)
172 Control circuit 174 Driver circuit 186 AC power line 180 Current sensor 188 AC connector 192 Stray capacitance between the motor generator 192gc 192 and GND (stray capacitor)
192i Leakage current flowing between neutral point and GND (common mode current)
192n Neutral point 192nc Stray capacitance between neutral point and GND (stray capacitor)
193 Rotating magnetic pole sensor 194 Motor generator 200 Power converter 200gc Stray capacitance between the power converter and GND (stray capacitor)
195 Auxiliary motor 314 DC positive terminal 316 DC negative terminal 328 IGBT (switching element)
330 IGBT (switching element)
410 Torque command / current command converter 420 Current controller (ACR)
421 Current Controller (ACR)
430 Pulse modulator 431 for PHM control Voltage phase difference calculator 432 Modulation degree calculator 434 Pulse generator 435 Phase searcher 436 Timer counter or phase counter comparator 440 Pulse modulator 450 for PWM control 450 Switcher 500 Capacitor module

Claims (6)

An inverter circuit that receives DC power and generates AC power for driving the rotating electrical machine (or motor);
A smoothing capacitor for supplying DC power to the inverter circuit;
A control circuit for controlling the inverter circuit;
A driver circuit for driving the inverter circuit based on the output of the control circuit,
The inverter circuit includes a plurality of switching elements for configuring the upper arms of the U phase, the V phase, and the W phase, and a plurality of switching elements for configuring the lower arms of the U phase, the V phase, and the W phase, DC power from the smoothing capacitor is supplied to a series circuit configured by connecting a stator winding of the rotating electric machine (or motor) between the upper arm and the lower arm, Each of the U-phase, V-phase, and W-phase switching elements of the upper arm and the lower arm repeats conduction and interruption sequentially to generate AC power for driving the rotating electrical machine (or motor),
The control circuit supplies current to a stator winding of the rotating electrical machine (or motor) in a state where the upper arm is connected in parallel in a plurality of phases, and current from the stator winding passes through one lower arm. In the first operation region that flows, the conduction time of the upper arm connected in parallel is made longer than the conduction time of the lower arm, and the conduction time of the inverter circuit is controlled by the conduction time of the lower arm. In the second operating region in which current is supplied from the upper arm to the stator winding and the current from the stator winding flows through the lower arm connected in parallel, the lower arm connected in parallel from the upper arm conduction time. The conduction time of the inverter circuit is controlled by the conduction time of the upper arm by increasing the conduction time of the arm,
The power conversion device in which the driver circuit controls a switching element constituting an upper arm and a switching element constituting a lower arm based on a control signal from the control circuit. The power conversion device according to claim 1,
In the first operation area, the lower arm repeats conduction and shut-off multiple times within the conduction time of the upper arm,
In the second operation region, the upper arm repeats conduction and shut-off operations a plurality of times within the conduction time of the lower arm. In the power converter device according to claim 1 or 2,
In the control state in which the peak value of the AC output is increased, the control circuit increases the conduction width by setting the number of conductions and the number of interruptions of the switching element to be the same, and further increases the peak value of the AC output. In a state where the switching element has a narrow cut-off width and cannot be cut off, the conduction state of the switching element is continued to reduce the number of conductions and the number of interruptions. In the power converter device according to claim 1 or 2,
The control circuit conducts the switching element at a phase angle determined based on preset data within one cycle of the AC output, and controls the conduction width according to the peak value. In the power converter device in any one of Claims 1 thru | or 4,
The control circuit switches the switching element a predetermined number of times corresponding to a predetermined phase within one cycle of the AC output. In the power converter device in any one of Claims 1 thru | or 5,
The control circuit controls a three-phase short-circuit period by conduction and cutoff of the other one-phase upper arm when any two phases of the U-phase, V-phase, and W-phase are in a conductive state,
When the lower arm of any two phases of the U phase, the V phase, and the W phase is in a conductive state, the three-phase short-circuit period is controlled by the conduction and interruption of the lower arm of the other one phase.

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