JP3598909B2 - Synchronous motor control device, electric vehicle control device, and synchronous motor control method - Google Patents

Description

【００７９】
【数１５】

ただし、Ｌ０＝（Ｌｄ＋Ｌｑ）／２，Ｌ１＝（Ｌｄ−Ｌｑ）／２，ｋ１（θ），ｋ２（θ），ｋ３（θ），ｋ４（θ），ｋ５（θ），ｋ６（θ）はそれぞれθに関する関数となっている。
【００８０】
円筒形同期モータの場合には、Ｌ１＝０なので、β軸電流微分値ｐｉβはα軸電圧Ｖαには影響しないことがわかる。
【００８１】
Ｖ−Ｗ相短絡状態のとき、α軸電圧ＶαだけがＵ相電圧Ｖｕの状態により印加されていることになるが、β軸電流微分値ｐｉβはＶα＝０のときと変わらない。しかも、Ｖ−Ｗ相短絡状態なので、Ｖβ＝０となっているので、３相短絡状態のときのβ軸電流微分値ｐｉβと一致することを意味している。以上のことから、図１８が成り立つことがわかる。
【００８２】
また、Ｗ−Ｕ相短絡のときも同様に、β′軸電流微分値ｐｉβ′は３相短絡電流微分ベクトルｐｉｓのβ′軸成分と同じになる。従って、２相短絡状態の短絡軸の電流微分値（差分値）を検出すると、３相短絡電流微分ベクトルの位相γを図１８のベクトル図を計算することにより算出することができる。
【００８３】
今回の２相短絡モードＭｓｃ（ｎ）と前回の２相短絡モードＭｓｃ（ｎ−１）から、３相短絡電流微分ベクトルの位相γを求める場合、その短絡モードの組合わせにより、演算方法が異なる。
【００８４】
そのため、ステップ１１６では、図８のようなモードに分けた演算式を用いて３相短絡電流微分ベクトルの位相γを求めている。
【００８５】
ステップ１１７については、図４のステップ１０３と同様にして磁極位置θを得ることができる。
【００８６】
以上のように、本実施例を用いれば、比較的継続時間の長い２相短絡状態の電流の変化量すなわち差分値から３相短絡状態の電流微分ベクトルの方向を決定すなわち演算できるので、高精度の磁極位置検出を少ない電流の取り込みにより得られる効果がある。
【００８７】
また、この実施例の方式は微分回路を用いないので、ノイズに強く、比較的安価なコントローラで実現できる有利点も持っている。
【００８８】
図９の実施例は突極形同期モータ１６に本発明を適用したときの制御装置の構成図である。コントローラ４は先の実施例と同じく電気回路でもまたソフトウエアででも実現できる。図１の実施例に対して、図９の実施例は２相スイッチング演算部１８を用いること、ＰＷＭ信号発生部９からの検出用パルスＰ３とＰ４を使用すること，演算部１７の処理方法が異なる。
【００８９】
２相スイッチング演算部１８の処理内容について、図１０のタイムチャートを用いて説明する。
【００９０】
２相スイッチングとは３相のＰＷＭ信号のうち、１相のスイッチングを停止しながら３相スイッチングと同じ正弦波電流を流す手法をいう。
【００９１】
図１０において例えばＵ相のスイッチングを停止しながら３相スイッチングと同じ正弦波電流を流す手法を説明する。Ｕ相以外のＶ相またはＷ相も同じ考え方でスイッチングを停止する相とすることができる。Ｕ相電圧指令値Ｖｕｒを常に搬送波の最大値と同じ値になるように付加電圧Ｖ０を強制的に加算している。これにより、Ｕ相ＰＷＭ信号Ｐｕｐは常にｈｉｇｈ状態になるので、スイッチング素子Ｓｕｐがオン状態となっている。
【００９２】
Ｖ相電圧指令値Ｖｖｒ，Ｗ相電圧指令値Ｖｗｒには、通常の指令値に付加電圧Ｖ０をそれぞれ加算した値を演算し、それによりＰＷＭ信号Ｐｖｐ，Ｐｗｐを発生している。
【００９３】
すべての相に同一の電圧を加算しても線間電圧には影響しないので、同期モータ１６を流れる電流は付加電圧Ｖ０を加えないときの電流と同じになる。これが２相スイッチングであり、よく知られている方法である。この方法を用いると、図１０に示す１回あたりの３相短絡状態は図３の場合よりも長く継続していることがわかる。
【００９４】
ＰＷＭ信号発生部９から発生する検出用パルスＰ３，Ｐ４も図１０に示す。
【００９５】
検出用パルスＰ３は搬送波の最大値に同期して発生するようになっており、図９の電流検出部１０で各相の電流平均値ｉｕａ，ｉｖａを得るために用いている。
【００９６】
また、電流検出用パルスＰ４は延長された３相短絡状態の開始時と終了時に発生するようになっている。図９の電流検出部２７では、電流検出用パルスＰ４によりＵ相電流ｉｕ，Ｖ相電流ｉｖを入力している。
【００９７】
これらの電流値は演算部１７に入力され、図１１のフローチャートに示す処理を行って磁極位置θを演算する。
【００９８】
図１１の処理方法は次のようにして行われる。ステップ１２１において、３相短絡状態の開始時刻ｔ（ｎ−１）のＵ相電流ｉｕ（ｎ−１），Ｖ相電流ｉｖ（ｎ−１）と、終了時刻ｔ（ｎ）のＵ相電流ｉｕ（ｎ），Ｖ相電流ｉｖ（ｎ）を用いて、各相の電流差分値ｐｉｕ，ｐｉｖ，ｐｉｗを演算する。その処理方法は図７のステップ１１３と同じである。
【００９９】
次のステップ１２２では、電流差分値ｐｉｕ，ｐｉｖ，ｐｉｗを用いて、３相短絡電流の微分ベクトルの位相γを演算する。この処理は図４のステップ１０２と同様である。
【０１００】
以下の手法では、コントローラ４内でその時点で制御に用いている磁極位置をθ′、同期モータ１６の実際の磁極位置をθとする。また、コントローラ４内の磁極位置θ′により演算されたｄ軸電流，ｑ軸電流をｉｄ′，ｉｑ′、同期モータ１６の実際のｄ軸電流，ｑ軸電流をそれぞれｉｄ，ｉｑとして説明する。
【０１０１】
ステップ１２３では、磁極位置θ′と電流検出部１０から入力した電流平均値ｉｕａ，ｉｖａを用いて、ｄ軸電流ｉｄ′，ｑ軸電流ｉｑ′を算出する。
【０１０２】
ステップ１２４では、ｉｄ，ｉｑの代わりに、ｉｄ′，ｉｑ′を用いて、式 （１０）の演算を行い、磁極位置（ｄ軸）から３相短絡電流微分ベクトルまでの位相δを求める。
【０１０３】
モータ速度ωが所定値以上の場合には次の近似式により求めてもよい。
【０１０４】
【数１６】
ｔａｎ（δ）≒−Ｌｄ｛（Ｌｄ−Ｌｑ）ｉｄ＋Φ｝／｛Ｌｑ（Ｌｑ−Ｌｄ）ｉｑ｝…（１６）
ステップ１２５では、ステップ１２２で得られた位相γを用いて、磁極位置θを次式で求める。
【０１０５】
【数１７】
θ＝γ−δ …（１７）
この関係は図１７のベクトル図に示している。
【０１０６】
ステップ１２６では、ステップ１２５で求めた磁極位置θがステップ１２３のｉｄ′，ｉｑ′を求めるときの磁極位置θ′とほぼ一致しているかを判断する。一致していない場合には、再びステップ１２３からステップ１２５までの処理を行い、磁極位置θを算出する。
【０１０７】
実際の磁極位置θとコントローラ内の磁極位置θ′とが異なると、ｉｄ′，ｉｑ′がｉｄ，ｉｑと一致しないため、位相δには誤差が生じる。しかし、その誤差はステップ１２３からステップ１２５までの処理を行う毎に減少していき、コントローラ内の磁極位置θ′は真の磁極位置θに収束する。これをステップ１２６で判断し、ほぼ磁極位置θの演算が収束したとき、演算を終了する。
【０１０８】
また、この演算は２，３回で数度以内に収束することが見込まれるため、収束の判断を磁極位置θの演算結果でなく、演算回数で例えば２回の演算で終了するようにしてもよい。
【０１０９】
さらに、磁極位置を検出するサンプリング時間とモータ速度との関係によっては、ステップ１２６を省略して、数回のサンプリングで磁極位置を検出する方法を採用することもできる。
【０１１０】
このように、突極形同期モータの磁極位置を検出する場合には、誤差を含んだｄ軸電流ｉｄ′，ｑ軸電流ｉｑ′を用いて演算する必要があるが、これを収束できるようにしたアルゴリズムを設けた点に本実施例の特徴がある。そのため、３相短絡状態の電流の変化を利用して突極形同期モータのセンサレス制御システムを構築できる利点がある。
【０１１１】
本システムでは、２相スイッチング方式のように３相短絡時間を延長する方式を併用することにより、３相短絡期間における電流の変化幅を大きくできる。そのため、微分回路を用いることなく、３相短絡電流微分ベクトルを直接計測でき、ノイズに強い磁極位置検出方式を簡単なソフトウエア処理により実現できる。図１２は２相短絡状態から磁極位置を検出する突極形同期モータの実施例で、電気自動車に適用するための高信頼化システムの構成を示している。この実施例でコントローラ４は電気回路ばかりでなくソフトウエアによっても実現できる。先の実施例と同様、コントローラ４の各ブロックはソフトウエアの処理機能を表わしている。
【０１１２】
図５の円筒形同期モータの場合と比べて、図１２の異なる点は、演算部２０の処理内容が異なる点、突極形同期モータ１６で電気自動車のタイヤ２４，２５を駆動する機構になっている点、電気自動車の信頼性を向上する目的でモータ１６の磁極位置を機械的に直接に検出する磁極位置センサ２３を備えている点である。
【０１１３】
まず、演算部２０の処理内容を説明する。この処理内容のフローチャートを図１３に示す。
【０１１４】
ステップ１３１からステップ１３４までの処理は図７のステップ１１１からステップ１１４までの処理と同じである。
【０１１５】
ステップ１３５の突極性補正位相εは同期モータ１６の突極性の影響を考慮するために必要な補正量である。
【０１１６】
式（１５）で示したように、突極形同期モータ１６の場合、Ｌ１≠０なので、β軸電流微分値ｐｉβはα軸電圧Ｖαにより変化する。そのため、３相短絡電流微分ベクトルのβ軸成分とは異なる値になる。
【０１１７】
図１９はα軸電圧Ｖαにより発生するα軸電流微分値ｐｉα１，β軸電流微分値ｐｉβ１、及び、その合成である電流微分ベクトルｐｉ１を示している。
【０１１８】
電流微分ベクトルｐｉ１と一致した方向の軸をｘ軸、それに直交する軸をｙ軸とすると、電流微分ベクトルｐｉ１のｙ軸成分はα軸電圧Ｖαによらず常に０であることがわかる。そのため、ｐｉ１のｙ軸成分は３相短絡電流微分ベクトル
ｐｉｓのｙ軸成分と一致する。これを突極性補正位相εとよぶ。
【０１１９】
そこで、突極形同期モータの場合、β軸でなく、突極性補正位相εだけ進んだｙ軸の電流微分値（差分値）を検出する。
【０１２０】
実際には３つの２相短絡状態があるので、Ｖ−Ｗ相短絡，Ｗ−Ｕ相短絡，Ｕ−Ｖ相短絡の場合の突極性補正位相をそれぞれε１，ε２，ε３とし、その方向の軸をｙ′軸，ｙ″軸とする。
【０１２１】
突極性補正位相ε１，ε２，ε３は式（１４）および式（１５）からそれぞれ次式となる。
【０１２２】
【数１８】
ｔａｎ（ε１）＝−（Ｌ１ｓｉｎ２θ）／（Ｌ０−Ｌ１ｃｏｓ２θ） …（１８）
【０１２３】
【数１９】

【０１２４】
【数２０】

以上のことから、ステップ１３５では、２相短絡状態に応じて式（１８），式（１９）、および式（２０）のいずれかの演算を行い、突極性補正位相を求めている。
【０１２５】
これらの演算で用いる磁極位置θはコントローラ４での値であり、誤差を含んでいるが、図１１のように収束させながら正確な磁極位置を求めていくこともできる。
【０１２６】
ステップ１３６においては、電流差分値ｐｉｕ（ｎ），ｐｉｖ（ｎ）から図１３の表を用いて補正した短絡軸（ｙ軸，ｙ′軸，ｙ″軸のいずれか）の短絡電流差分値演算を行い、短絡軸の短絡電流差分値ｐｉｓｃを算出する。
【０１２７】
短絡軸とは既に説明したように、α軸電圧により電流微分値（差分値）に影響を受けない方向の軸をいう。
【０１２８】
次のステップ１３７では、図１４に示したように、今回と前回の２相短絡状態により計算するモードを変更し、図１４の演算式を用いて３相短絡電流微分ベクトルの位相γを得る。
【０１２９】
このときのベクトル図の一例を図２０に示すが、この関係を図１４の演算式により求めていることになる。
【０１３０】
ステップ１３８からステップ１４０までの処理は図１１のステップ１２３からステップ１２５までの処理と同じで、突極形同期モータ１６における磁極位置から電流微分ベクトルまでの位相を考慮したものである。
【０１３１】
以上のように、演算部２０を用いれば、突極形同期モータ１６に対しても、２相短絡状態の電流を検出するだけで磁極位置を検出することができる。
【０１３２】
図１２の電気自動車駆動システムでは、磁極位置センサ２３からの信号を磁極位置検出部２１に入力して、磁極位置θ１を検出している。
【０１３３】
磁極位置異常検出部２２では、磁極位置検出部２１からの磁極位置θ１と演算部２０からの磁極位置θを入力し、図１５に示す処理を行っている。
【０１３４】
ステップ１４１で入力した磁極位置θとθ１を比較し、その差が予め決められた正常範囲内にあるかをステップ１４２で判断する。
【０１３５】
正常と判断した場合には、ステップ１４３で磁極位置θ１を出力用磁極位置θ２にストアし、ステップ１４４でθ２を座標変換部８，１１に出力している。ステップ１４２において、２つの磁極位置を比較して正常範囲にないと判断したときには、ステップ１４５で電気自動車を一旦停止する処理を行う。
【０１３６】
ステップ１４６は同期モータ１６の回転が停止したか否かを判断する。停止したと判断した場合に、ステップ１４７で正常な磁極位置を用いて安全速度内で走行が行えるように処理する。安全速度内で走行とは自動車の車速の上限を時速４０ｋｍあるいは５０ｋｍに制限し、この速度以下で走行するように図示しない制御装置で制御し、この制御のもとで走行することである。
【０１３７】
このように、本実施例を用いれば、通常の磁極位置センサ２３を設けることで、磁極位置センサ２３の出力に基づき磁極位置を求めると共にこれとは別にモータ電流から演算処理により磁極位置を求めているので、信頼性の高い電気自動車を提供できる効果がある。
【０１３８】
特に、リラクタンストルクを用いてモータの小型化，電気自動車の軽量化を図るためには、本実施例は適している。
【０１３９】
図１６は電流センサだけで磁極位置を検出し、しかも、その特性に異常がないかを自己診断する機能を有した磁極位置センサレスの制御システムの実施例を示す。
【０１４０】
図１２の実施例に対して、自己診断部２６を設けた点が図１６の実施例の特徴である。
【０１４１】
演算部２０では、図１３の処理に加え、モータ速度ωを検出するための演算を行う。
【０１４２】
前述した実施例では、電流微分ベクトルについては、その位相γを検出することで磁極位置θを得る方式であるため、電流微分ベクトルの大きさに関する情報を無視している。
【０１４３】
そこで、式（６），式（７）から逆にモータ速度ωを求めることにする。つまり、
【０１４４】
【数２１】
ω＝（Ｌｄ ｐｉｄ＋Ｒ ｉｄ）／Ｌｑ ｉｑ …（２１）
【０１４５】
【数２２】
ω＝−（Ｌｑｐｉｑ＋Ｒ ｉｑ）／（Ｌｄ ｉｄ＋Φ） …（２２）
の一方、あるいは、両方の式を用いて、モータ速度ωを演算する。
【０１４６】
なお、式（２１），式（２２）において、抵抗Ｒを無視する簡易式を用いてもよい。このモータ速度ωを自己診断部２６に出力する。
【０１４７】
また、図１３のステップ１４０で得られた磁極位置θも自己診断部２６に出力する。
【０１４８】
自己診断部２６では、磁極位置θの変化状態とモータ速度ωを比較することにより、何らかの異常がないか否かを判断する。
【０１４９】
もし、異常があると判断した場合には、異常診断信号Ｓｅを出力し、センサレス制御システムを停止するようにシステム的な処理を行う。
【０１５０】
このように電流微分ベクトルの独立した変数を複数用いてモータ速度を推定することで、他のセンサを用いることなく、自己診断機能を持たせることができる。
【０１５１】
以上が、電流センサだけを用いて同期モータの磁極位置を検出する実施例である。
【０１５２】
同期モータの他に、リラクタンスモータでも突極性を利用して本発明を適用することができる。
【０１５３】
また、本実施例では、説明が複雑になるために省略したが、モータの回転子がサンプリング時間中に回転することによる影響を考慮して磁極位置を演算するようにしても、本実施例を適用できることはいうまでもない。
【０１５４】
ここでは電気自動車の例を述べたが、現在１２０度通電方式のインバータ制御を用いてセンサレス制御している磁石モータに対しても、本実施例を用いることにより、１８０度通電方式のインバータ制御によりトルク脈動と低騒音のセンサレスシステムを提供できる。
【０１５５】
図１から図２０を用いて説明した実施の形態は、モータ１が所定回転速度以上で回転している状態での制御に最適である。ここで所定回転速度以上とは例えば８００回転／分以上である。モータが停止状態を含む低速状態では後で説明する理由で磁極位置の検知を更に高精度に行う必要性がある場合が考えられる。モータの停止状態を含む低速状態の磁極位置を高精度に行える実施例を図２１を用いて説明する。図１と図２１との大きな相違点は、停止状態を含むモータの低速回転時の磁極位置を検知する演算部５２を設けたことである。なおコントローラ４は他の実施例と同様、電気回路ばかりでなくソフトウエアによっても実現できる。ソフトウエアで実現する場合、コントローラ４の各ブロックはソフトウエアの処理機能を表わしている。
【０１５６】
演算部１４で磁極位置を検出する処理は図１で説明した通りである。また電流指令値発生部６や電流制御部７，座標変換部８，ＰＷＭ信号発生部９，電流検出部１０，座標変換部１１，インバータ３の動作は図１と基本的に同じである。上で同期モータが３相短絡状態のときのモータ電流の変化量、または、変化方向に基づいてモータの磁極位置を演算する処理方法について説明したが再度説明する。
【０１５７】
この磁極位置の演算方法では、インバータからの印加電圧の影響を受けない３相短絡状態でのモータ電流の変化量または変化方向に基づいて演算し、磁極位置を求める。上記３相短絡状態でのモータ電流の変化量または変化方向を検出するため、図３で説明の通り、この３相短絡状態でタイミングを制御するためにパルスＰ１を発生する。その代表例として図３に示す搬送波の最高値あるいは最低値あるいはその両方でパルスＰ１を発生し、このパルスＰ１に同期して電流値あるいは電流値の微分値を検出する。すなわちＰＷＭ信号発生部９でパルスＰ１発生し、パルスＰ１をモータの３相短絡状態を表わす信号とし、このパルスＰ１をトリガとしてモータ電流の変化量を検出する。このときに検出される３相短絡時のモータ電流変化量のベクトル、すなわち、微分ベクトルの関係は先に説明した図１７の通りである。
【０１５８】
図１７において、検出すべき磁極位置は静止座標α軸と回転座標ｄ軸との間の位相θであり、先の式（１７）で表すことができる。
【０１５９】
【数１７】
θ＝γ−δ …（１７）
ここで、γは３相短絡時のモータ電流微分ベクトルｐＩｓのα軸に対する位相であり、δはモータ電流微分ベクトルｐＩｓのｄ軸に対する位相である。上記位相γについては、まず、図１に示す電流微分回路１２においてモータ電流ｉｕとｉｖを微分し、さらに電流微分値検出部１３において、モータ短絡状態の検出用パルスＰ１に同期して短絡時のモータ電流の微分値ｐｉｕ，ｐｉｖを取り込む。さらに、この短絡時のモータ電流の微分値ｐｉｕ，ｐｉｖを式（１）と式（２）に従いα−β軸の電流微分値ｐｉαとｐｉβを算出し、式（２３）に基づき位相γを算出する。
【０１６０】
【数１】
ｐｉα＝（√３／２）ｐｉｕ …（１）
【０１６１】
【数２】
ｐｉβ＝（１／√２）（ｐｉｕ＋２ｐｉｖ） …（２）
【０１６２】
【数２３】
γ＝ｔａｎ^−１（ｐｉβ／ｐｉα） …（２３）
なお、ここではｐｉα，ｐｉβを求めるのにｐｉｕ，ｐｉｖの２相分を用いているが、ｐｉｕ，ｐｉｖ，ｐｉｗの３相電流の微分値を用いても演算可能である。さらに、３相短絡時の電流微分値を演算するために微分回路を用いているが、微分回路の構成が不可能な場合には電流微分値の代わりに３相短絡区間での電流変化量を演算し、電流変化量を短絡時間で割ることによって求められる電流変化率を用いても実現可能である。加えて、短絡区間が非常に短く、３相短絡区間の電流変化率の演算が不可能な場合には、３相短絡区間より確実に短絡区間が長い２相短絡区間を用いても３相短絡区間の電流変化率を演算することは可能である。詳細は上で述べた通りである。
【０１６３】
以上のようにして、３相短絡時のモータ電流微分ベクトルｐＩｓのα軸に対する位相γは求められる。次にモータ電流微分ベクトルｐＩｓのｄ軸に対する位相δは以下のようにして求める。まず、回転座標ｄ−ｑ軸での同期モータの基本式は次の式で表すことができる。なおこれらの式は先に説明の通りである。
【０１６４】
【数４】
Ｖｄ＝（Ｒ＋ｐＬｄ）ｉｄ−ωＬｑ ｉｑ …（４）
【０１６５】
【数５】
Ｖｑ＝（Ｒ＋ｐＬｑ）ｉｑ＋ω（Ｌｄ ｉｄ＋Φ） …（５）
ここで、Ｖｄ，Ｖｑはｄ−ｑ軸電圧、Ｌｄ，Ｌｑはｄ−ｑ軸のインダクタンス、Ｒは巻線抵抗、ωはモータ角速度、φは界磁主磁束、ｐはｄ／ｄｔである。上記基本式において、３相短絡時はｄ−ｑ軸での印加電圧は０となるので、３相短絡状態での基本式は以下のようになる。
【０１６６】
【数６】
ｐｉｄ＝（ωＬｑ ｉｑ−Ｒ ｉｄ）／Ｌｄ …（６）
【０１６７】
【数７】
ｐｉｑ＝−｛ω（Ｌｄ ｉｄ＋Φ）＋Ｒ ｉｑ｝／Ｌｑ …（７）
静止座標α−β軸における電流微分ベクトルは、式（６）と式（７）で表される回転座標ｄ−ｑ軸の電流微分ベクトルとｄ−ｑ軸がモータ角速度ωで回転することにより発生する電流微分ベクトルの和で表される。よって、α−β軸で見たｄ−ｑ軸の電流微分値は以下のようになる。
【０１６８】
【数８】
ｐｉｄｓ＝｛ω（Ｌｑ−Ｌｄ）ｉｑ−Ｒ ｉｄ｝／Ｌｄ …（８）
【０１６９】
【数９】
ｐｉｑｓ＝−｛ω（Ｌｄ−Ｌｑ）ｉｄ＋Φ）＋Ｒ ｉｑ｝／Ｌｑ …（９）
従って、モータ電流微分ベクトルｐＩｓ のｄ軸に対する位相δは式（２４）で表される。
【０１７０】
【数２４】

よって、上記式（１７）と式（２３）と式（２４）により磁極位置を求めることができる。ここで、式（２４）に含まれるωは位相推定値の変化量より求められる角速度の推定値を用いることになる。また、角速度が十分大きく、Ｒ成分を無視できる領域であればωの影響はなくなる。
【０１７１】
以上が同期モータ３相短絡時のモータ電流の変化量、または、変化方向に基づいて同期モータの磁極位置を検出する位置検出方式の概要である。この方式は突極型の同期モータのみならず、円筒型の同期モータにも適用可能である。
【０１７２】
上記位置検出方式では、通常のＰＷＭ動作時に発生するモータ短絡状態の電流変化量を用いるため、特別な推定信号を印加することなく高速領域まで磁極位置の推定が可能となる。
【０１７３】
しかし、以下の動作範囲においては、この検出方式だけでは位置検出精度が低下する可能性がある。図２２にその動作範囲を示す。速度が零すなわちモータ１が回転を停止している状態はモータ１のトルクも零である。モータの回転停止状態例えばモータを起動しようとする状態ではモータ電流がまだ流れていないために、ＰＷＭ制御を行おうとしても短絡時の電流の変化量を検出することができない。さらに起動後であっても回転速度が低くトルクの小さいときは電流値が小さく、上記測定が難しい。磁極位置の検出精度が低下する。さらには速度ゼロ付近で電流が流れている領域においては、短絡時の電流の変化量により磁束の方向は検出できるものの誘起電圧の影響が非常に小さいために、検出した方向がＮ極か，Ｓ極か、すなわち極性を判別することができない。
【０１７４】
上記のような動作範囲例えば８００ｒｐｍ（回転／分）においては、本実施例は検出部５２を新たに設けることにより、磁極位置の検出制度を向上することができる。検出部５２を図２３により説明する。検出部５２は基本的には磁石位置の推定用信号を発生する信号発生部５４と極性判別部５６を有している。
【０１７５】
磁石位置の推定用信号を発生する信号発生部５４は、電流が流れていない領域あるいは電流値が小さい領域で磁極位置を検出する場合、例えばモータ１の起動時に、モータ短絡電流の変化量が検出できる程度の電流が流れるように位置検出用の電流指令値ｉｄｈｒを発生し、電流指令値発生部６に電流指令値ｉｄｈｒを与える。
【０１７６】
電流指令値発生部６の構成を図２４に示す。図１から図２０の実施例および図２１の制御系では、コントローラ４の上位の制御部より電流指令値発生部６にトルク指令Ｔｒが入力される。電流指令値発生部６は、トルク制御部６３を有しており、同期モータ１が指令通りのトルクを発生するようにｄ−ｑ軸の電流指令ｉｄｒ，ｉｑｒを演算する。通常は、トルク指令とモータ速度（図２４には図示せず）を入力として、現動作点において最高効率となるようなｉｄｒとｉｑｒをマップ検索等を使用して演算する。ここで通常とは例えば同期モータ１がある決められた回転速度である８００回転／分より早い速度で回転している状態である。
【０１７７】
上記で説明したように、同期モータの起動時などの８００回転／分以下の低速回転時に磁極位置を検出するための電流指令ｉｄｈｒが信号発生部５４から出力され、加算器６５に印加される。この電流指令ｉｄｈｒは加算器６５でｄ軸方向の電流指令ｉｄｒ１に加算され、加算結果がｄ軸電流指令ｉｄｒとして、電流指令値発生部６から電流制御部７に加えられる。本実施例では磁極位置を検出するための電流指令ｉｄｈｒはｄ軸方向の電流指令に印加している。これは、ｉｄｈｒの印加による余計なトルク発生を防ぐためである。すなわちｑ軸方向の電流が零ならば基本的にはｄ軸に電流を印加してもトルクの発生はない。しかし、ｑ軸方向にｉｄｈｒを印加した場合でも、もしくはｄ、ｑ軸双方に印加した場合でも磁極位置の検出は可能である。従ってトルクの発生が制御全体において許容できるものであれば、磁石位置の検出のためにｑ軸方向の電流指令に加算することができる。さらに測定に対応した短い時間のみ加えるようにしても良い。
【０１７８】
さらに、印加する位置推定用の電流指令ｉｄｈｒは交流信号でも直流信号でも位置検出は可能である。このうち電流指令ｉｄｈｒが交流信号である場合は、推定動作初期に生じる位置検出値とモータの磁極位置との間にずれによる発生トルクの平均値は零となる。従ってトルク変動の影響を抑えられる。
【０１７９】
以上のようにして、モータの速度が零で、電流が流れていないような動作点、例えば停止状態から回転を開始する起動時、においては、位置推定用の信号を印加することにより、モータ短絡時の電流変化量を用いた位置検出が可能となる。しかし、検出された位置は磁束方向ではあるが、それがＮ極方向であるか、もしくはＳ極方向であるかは不明である。それはモータの角速度ωが零であるために、誘起電圧の影響を受けないためである。
【０１８０】
この課題を解決するために図２３に示す実施例では、極性判別部５６を設けて、演算された位置がＮ極であるか、Ｓ極であるかすなわち磁極の極性を判別する。この極性判別部５６において用いられる判別方法は特に限定されるものではないが、モータの磁気飽和特性を用いる方法が有効である。以下に極性判別方法の一例として磁気飽和特性を用いる方法を説明する。
【０１８１】
同期モータの磁気特性は回転子が永久磁石による磁束を有しているため、図２５に示すように磁束軸であるｄ軸の電流ｉｄが零のときでも磁束が存在する。この磁気特性により、ｄ軸インダクタンスＬｄの特性は図２６のようになる。図２６からｄ軸の電流ｉｄの正負の違いによりインダクタンスＬｄの大きさが異なる領域、図２６中斜線の領域、が存在することがわかる。よって、図２６の斜線領域でｉｄが流れるようなバイアス成分をもつ交流信号をｄ軸に印加すれば、インダクタンスＬｄの大きさにより電流制御系の応答が変化するために、Ｎ極，Ｓ極の極性の違いがモータ電流の振幅の違いに現れ、モータ電流の振幅の大きさを計測することにより磁極の極性を判別することが可能となる。以上が極性判別方法の一例である。
【０１８２】
さらに、この極性判別方法の手順を図２３を用いて表すと次のようになる。まず、磁極位置を検出するための信号を発生する信号発生部５４より直流のバイアス成分を有する交流信号をｄ軸電流指令に印加する。次に検出されたｄ軸電流ｉｄを極性判別部５６に入力し、極性判別部５６においてｉｄの交流成分の振幅値を計測し、現在の位置設定値θ＾の極性を判別する。
【０１８３】
もし、判別された結果がＮ極であったならば、そのまま制御系の位置設定値θ＾を制御に用いる。それに対して判別された結果がＳ極であったならば、位置設定値θ＾に１８０°加えまたは減じ、位置設定値θ＾をＮ極に補正する。また、この極性判別方式では、モータに磁気飽和が生じる程度までｄ軸方向に電流を流すことになるので、位置設定値θ＾に少しでも誤差が生じていたならば、モータに一定方向のトルクが生じる。そこで、起動時の磁極位置検出動作は、このトルクによってモータの軸が回転しないようにモータの回転軸やロータの回転を機械的に一時的に阻止するロック機構を設けておき、回転軸の回転が阻止されている状態において起動するようにしておけば良い。例えば図２１でモータ１のロータ７０あるいは回転軸７２の回転を阻止するロック機構７４を設ける。このロック機構７４は通常のブレーキ機構と同じ機構であり、検出部５２からの信号で一時的に回転を阻止する。なおこの機構が働くのはモータの回転開始時であり、トルクが小さいので簡単な機構であっても目的を達することが可能である。回転を開始するための磁極位置や極性の検知が終わればこのロック機構は検出部５２からの信号で回転の阻止を解除する。
【０１８４】
以上が、極性判別方式の一例の説明である。
【０１８５】
次に、これまで説明してきた零速度における磁極位置の検出部によるモータ起動時の処理手順を図２７のフローチャートにより説明する。図２７で文字Ｓはステップすなわち手順を意味する。まず図２７のステップ３０において、モータの回転軸の回転がロック機構により阻止されている状態かを判定する。その結果、回転軸の回転が阻止されているならば、ステップ３１において磁石位置の検出用の信号ｉｄｈｒを極性判別部５６から出力し、電流指令値発生部６の電流指令ｉｄｒｌと加算演算するために加算器６５に印加する。
【０１８６】
以上の説明はモータの起動時すなわち回転開始時にモータ回転を阻止するロック機構がある場合であり、そのような機構がない場合はステップ３０は不要である。ロック機構がある場合でそのロック機構が動作していない場合は、ステップ３０でロック機構を動作させる指示を出すか、他の制御部からの指示で動作するのを待つ。そしてロック機構が動作した状態でステップ３１へ処理が移る。
【０１８７】
ステップ３２ではＰＷＭ制御中に発生する短絡電流を検出し、先で説明した短絡電流の変化量による磁極位置検出方法によって検出値θ＾の演算を行う。さらにステップ３３において、ステップ３２で得られた磁極位置の検出値θ＾がＮ極方向であるか、もしくはＳ極方向であるかを判定する。この判定の結果、Ｎ極であると判定された場合にはステップ３４において検出値θ＾が現在の磁極位置とされる。それに対して、ステップ３３でＳ極であると判定された場合には、ステップ３５において検出値θ＾は１８０°加算あるいは減算の補正がなされて、現在の磁極位置が求められる。次にステップ３６において、得られた磁極位置の検出値を用いてモータの駆動制御が開始される。
【０１８８】
以上が、モータ回転が零速度の磁極位置の検出方式を用いたモータ起動時の処理手順である。モータの短絡電流の変化量による磁極位置検出方式のみでは検出精度が低い場合は、上記処理を行うことで高精度に磁極位置を検出することが可能となる。なお本実施例では、磁極位置の検出動作の開始の条件としてモータの回転軸がロックされていることとしたが、上述の如く回転軸がロックされていなくても磁極位置の検出は可能である。
【０１８９】
モータの回転速度が零で、かつモータ電流が流れているような場合での演算部５２の動作を説明する。このような動作環境は、回転停止時において負荷トルクすなわちモータにより駆動される装置の駆動に必要なモータのトルクがモータの発生しているトルクより大きく、さらにノイズ等の影響によりコントローラ内の磁極位置の設定値が消去された（正確でなくなった）場合等である。このような場合、モータ電流が流れているので、先で説明したモータ短絡電流の変化量による磁極位置の検出が可能である。しかし、モータ速度が零なので、極性判別を行う必要がある。ここでの極性判別はモータの回転軸はロックされておらずモータ駆動中なので、先で説明した磁気飽和特性を利用した極性判別方式より次に説明する方式の方が望ましい場合が多い。
【０１９０】
ここで説明する方式は、モータトルクの発生方向とモータ軸の回転方向とから判別する方式である。この判別方式の動作すなわち処理内容を図２８のフローチャートを用いて説明する。まずステップ４０において、モータの短絡電流の変化量によって得られた検出値θ＾を磁極位置の設定値として所望のトルクを発生させる。その後、ステップ４１においてモータ軸の回転方向と発生させたトルクの方向とを参照し、両者の方向が一致しているならば、ステップ４２において、磁極位置の設定値を現在の磁極位置とする。それに対してステップ４１において、モータ軸の回転方向と発生させたトルクの方向とが一致していない場合には極性が反対であると判断し、ステップ４３において設定値を１８０°補正し現在の磁極位置とする。さらに、ステップ４４でトルク指令を増加させる。以上の動作を所定回数繰り返し、設定回数以上連続で、モータ軸の回転方向と発生させたトルクの方向とが一致したときに極性判別を終了させる。以上がトルクの発生方向とモータの回転軸の回転方向とから磁極位置の設定値の極性を判別する方式の処理手順である。この方式では特別な極性判別用の信号の印加は不要である。
【０１９１】
以上、モータの短絡電流の変化量を用いる磁極位置検出方式を用いた、モータ速度が零である場合の磁極位置検出方法について述べたが、本方式は誘起電圧が非常に小さいような極低速時にも適用できる。例えば８００回転／分以下の回転で効果が大きい。
【０１９２】
さらには、本方式は電気自動車やモータとエンジンの両方を有するハイブリッド車をはじめとする電気車の駆動装置にそのまま適用できる。電気自動車やハイブリッド車の駆動装置においては、位置検出用の信号を印加して検出を行う場合や磁気飽和特性を利用した極性判別方式を用いる場合には、ブレーキが掛けられた状態、もしくは動作レンジがパーキングレンジに入っている状態で行えば、不要なトルク発生による車両の振動や移動が避けられる。
【０１９３】
上述の実施例によれば、印加電圧の状態に影響されることなく、かつ安価なコントローラで通常のＰＷＭ制御を行いながら、同期モータの全動作範囲において磁極位置検出が可能となる。
【０１９４】
さらに図２１から図２８の実施例によれば、モータの停止時から正確な制御が可能である。
【０１９５】
さらに図１から図２８に記載の実施例において、モータ電流の変化量、または変化方向から同期モータの磁極位置を求めるので、位置検出器を設けることなく、通常のＰＷＭ制御を行いながら磁極位置を検出することができる。
【０１９６】
また、磁極検出器を備え、磁極検出器で検出した磁極位置と、モータ電流から求めた磁極位置とを比較することにより、通常のＰＷＭ制御を行いながら磁極位置検出器の異常を検知することができる。
【０１９７】
【発明の効果】
本発明によれば、制御性の優れた同期モータの駆動システムを低コストで提供できる。
【０１９８】
また、通常の磁極位置センサを用いる場合には、そのセンサの異常を検出できるので、信頼性の高い同期モータ駆動システムを提供できる。
【図面の簡単な説明】
【図１】電流微分回路を用いて円筒形同期モータの磁極位置を検出する本発明を適用したときの一実施例を示す構成図である。
【図２】図１のインバータ３の結線方法を示す構成図である。
【図３】搬送波信号と３相の電圧指令値、ＰＷＭ信号との関係を示すとともに、電流の取り込みタイミングを示したタイムチャートである。
【図４】図１の構成方法のときの磁極位置を検出するためのフローチャートである。
【図５】円筒形同期モータの２相短絡状態のときの電流を検出して磁極位置を演算するための実施例を示す構成図である。
【図６】３相のＰＷＭ信号と図５の電流の検出タイミングを示すタイムチャートである。
【図７】図５の構成方法のときの磁極位置を検出するためのフローチャートである。
【図８】図７の２相短絡電流差分値、及び、３相短絡電流微分ベクトルの位相を演算するための演算式の一覧表である。
【図９】３相短絡時間を延長しながら電流の差分を用いて突極形同期モータの磁極位置を検出する他の実施例を示す構成図である。
【図１０】３相のＰＷＭ信号と図９の電流の検出タイミングを示すタイムチャートである。
【図１１】図９の構成方法において、磁極位置を高精度に検出するためのフローチャートである。
【図１２】第１の磁極位置検出器を用いて突極形同期モータを制御する電気自動車において、２相短絡状態の電流で磁極位置を検出する第２の磁極位置検出器を有する他の実施例を示す構成図である。
【図１３】図１２の構成方法において、突極形同期モータの磁極位置を２相短絡状態の電流を用いて検出するためのフローチャートである。
【図１４】図１３の２相短絡電流差分値、及び、３相短絡電流微分ベクトルの位相を演算するための演算式の一覧表である。
【図１５】図１２の磁極位置の異常判断を行うためのフローチャートである。
【図１６】２相短絡状態の電流で磁極位置を検出する磁極位置検出器を有する突極形同期モータにおいて、磁極位置検出の故障を自己診断するための他の実施例を示す構成図である。
【図１７】同期モータの電流ベクトル，電流微分ベクトル，磁極位置（ｄ軸）の関係の一例を示すベクトル図である。
【図１８】図９の円筒形同期モータにおいて２相短絡時の電流微分ベクトルと３相短絡時の電流微分ベクトルの関係を示すベクトル図である。
【図１９】突極形同期モータのα軸に印加した電圧により発生する電流微分ベクトルの関係を示すベクトル図である。
【図２０】図１６の突極形同期モータにおいて２相短絡時の電流微分ベクトルと３相短絡時の電流微分ベクトルの関係を示すベクトル図である。
【図２１】本発明の他の実施例を示すモータ制御システムの構成図である。
【図２２】磁石位置の検出が低下する可能性がある領域を示す図である。
【図２３】演算部５２の構成を示す図である。
【図２４】電流指令値発生部６の構成を示す図である。
【図２５】同期モータの磁気特性を示す図である。
【図２６】ｄ軸インダクタンスの特性を示す図である。
【図２７】モータの起動時の磁極位置を検出する処理手順を示す図である。
【図２８】トルクの発生方向とモータの回転軸の回転方向とから極性を判別する処理手順を示す図である。
【符号の説明】
１…円筒形同期モータ、２…バッテリー、３…インバータ、４…コントローラ、５ａ，５ｂ…電流センサ、６…電流指令値発生部、７…電流制御部、８，１１…座標変換部、９…ＰＷＭ信号発生部、１０，２７…電流検出部、１２…電流微分回路、１３…電流微分検出部、１４，１５，１７，２０，５２…演算部、１６…突極形同期モータ、１８…２相スイッチング演算部、１９…磁極位置・電流演算部、２１…磁極位置検出部、２２…磁極位置異常検出部、２３…磁極位置センサ、２４，２５…タイヤ、２６…自己診断部、５４…信号発生部、５６…極性判別部、６５…加算器。７０…ロータ、７２…回転軸、７４…ロック機構。[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a control device and a control method for a synchronous motor or an electric vehicle control device using them. In the present invention, the synchronous motor includes a reluctance motor.
[0002]
[Prior art]
In order to control the speed and torque of the synchronous motor, it is necessary to detect or estimate the magnetic pole position. By performing current control or voltage control based on the detected magnetic pole position, the torque and speed of the synchronous motor can be controlled.
[0003]
Previously, the magnetic pole position was detected by a position detector. However, in recent years, a method of controlling a synchronous motor by estimating a magnetic pole position, which is different from a method of detecting a magnetic pole position with a magnetic pole position detector, that is, a magnetic pole position sensorless control method has been proposed.
[0004]
For example, IEICE Transactions Vol. 117-D, no. No. 1,1997, “Sensorless salient-pole brushless DC motor control based on speed electromotive force estimation” (Takeshita, Ichikawa et al.) Performs speed control while estimating speed electromotive force using a motor model. A method has been proposed.
[0005]
Japanese Patent Application Laid-Open No. Hei 8-205578 discloses that the saliency of a synchronous motor is detected from a correlation between a voltage vector applied to the synchronous motor by pulse width control (PWM control) and a ripple component of a motor current vector corresponding thereto. A method is described.
[0006]
The technique described in the above paper is a method of estimating a magnetic pole position from a difference between a current calculated by a control model and an actually flowing motor current, and has a feature that a control system can be configured only by control calculation of a controller.
[0007]
Further, the technique described in Japanese Patent Application Laid-Open No. Hei 8-205578 has an advantage that it is not necessary to add an additional signal for detection since a general PWM signal for controlling the voltage of the synchronous motor is used.
[0008]
[Problems to be solved by the invention]
In the method of estimating the magnetic pole position from the difference between the current calculated by the control model and the actual flowing motor current, it is necessary to solve the problem that if the synchronous motor loses synchronization for some reason, control becomes impossible. Must.
[0009]
On the other hand, the technique disclosed in Japanese Patent Application Laid-Open No. 8-205578 can always detect the magnetic pole position of the synchronous motor with saliency, so that the control cannot be disabled.
[0010]
However, in the method of detecting the magnetic pole position of the synchronous motor with saliency, it is necessary to detect the correlation between the state of the motor current and the applied voltage every time the PWM signal changes.
[0011]
That is, it is necessary to detect the state of the motor current at least six times in one cycle of the carrier wave and to grasp the state of the applied voltage. Therefore, there is a problem that the operation speed cannot keep up unless a high-performance controller is used.
[0012]
A first object of the present invention is to provide a low-cost synchronous motor control device.
A second object of the present invention is to provide a highly reliable control system.
[0013]
[Means for Solving the Problems]
One of the solving means of the present invention is to calculate or estimate the magnetic pole position of the synchronous motor based on a change amount or a change direction of the motor current when the synchronous motor is in a short circuit state, and based on the calculated magnetic pole position. And controlling the synchronous motor.
[0014]
Other solutions will be described in the description of the embodiments of the invention.
[0015]
BEST MODE FOR CARRYING OUT THE INVENTION
Hereinafter, an embodiment of the present invention will be described with reference to FIG.
[0016]
FIG. 1 is a configuration diagram of a motor control device that drives a cylindrical synchronous motor 1 with DC energy of a battery 2. The DC voltage of the battery 2 is converted into a three-phase AC voltage by the inverter 3 and applied to the cylindrical synchronous motor 1. The inverter 3 is controlled based on the output of the controller 4.
[0017]
The output of the controller 4 is determined based on the result of the following calculation. Here, the controller 4 is configured by a functional block diagram, but these can be realized not only by hardware but also by software. The current differentiating circuit 12, the current detecting unit 10, and the PWM signal generating unit 9 described below partially use an input / output circuit of a computer. For example, an input / output circuit is an analog / digital converter or a pulse output circuit, and by using these, all can be processed by a program.
[0018]
First, the current command value generator 6 determines a d-axis current command value idr and a q-axis current command value iqr for a torque command value τr to be generated by the motor. Note that a command is issued to the current command value generator 6 from a higher-level control device of the controller 4 or a control program for the torque command value τr.
[0019]
The d-axis indicates the direction of the magnetic pole position (magnetic flux), and the q-axis indicates the direction electrically orthogonal to the d-axis, forming a dq-axis coordinate system. When the rotor having the magnet rotates, the dq axis coordinate system also rotates. Therefore, the phase from the stationary coordinate system (α-β axis coordinate system) is set to θ. That is, the purpose of this embodiment is to detect the phase θ of the magnetic pole (hereinafter, referred to as the magnetic pole position θ) from the current.
[0020]
FIG. 17 is a vector diagram showing an example of the relationship between the coordinate system and the current. If the d-axis current and the q-axis current can be controlled according to the command values, the synchronous motor 1 can generate a torque that matches the torque command value τr. The torque command value τr may be instructed directly or may be instructed from a speed control arithmetic circuit (not shown). Further, signals indicating the values of the u-phase current iu and the v-phase current iv are sent from the current sensors 5a and 5b to the current detection unit 10, and the current detection unit 10 detects the signals at the timing of a detection pulse P1 described later. The detected current values are converted by the coordinate conversion unit 11 into d-axis current id and q-axis current iq in the dq axis coordinate system.
[0021]
In this embodiment, the currents detected by the current detection unit 10 are the two phase currents iu and iv of the U phase and the V phase, but the W phase current iw can be obtained from iu and iv. Detection is omitted. As a matter of course, all three-phase currents may be detected.
[0022]
The current controller 7 calculates the d-axis current command value idr and the d-axis current deviation of the d-axis current id, and calculates the q-axis current command value iqr and the q-axis current deviation of the q-axis current iq. A d-axis voltage command value Vdr and a q-axis voltage command value Vqr are obtained by proportional / integral control calculations.
[0023]
The coordinate conversion unit 8 that inputs the d-axis voltage command value Vdr and the q-axis voltage command value Vqr calculates the three-phase voltage command values Vur, Vvr, and Vwr of the stationary coordinate system based on the magnetic pole position θ. Output.
[0024]
The PWM signal generator 9 outputs three-phase PWM pulses Pup, Pvp, Pwp, Pun, Pvn, and Pwn to the inverter 3 by calculation.
[0025]
FIG. 2 shows the relationship between the connection method of the inverter 3 and the PWM pulse. For example, when Pup is high, the switching element Sup is turned on, and when Pup is low, the switching element Sup is turned off.
[0026]
In addition, the PWM pulses Pup and Pun basically have a relationship in which high and low are opposite to each other. However, in order to prevent a power supply short-circuit, a short-circuit prevention period is provided in which both of the PWM pulses are in a low state when the PWM pulse is inverted.
[0027]
The processing content of the PWM signal generator 9 will be described with reference to a timing chart shown in FIG. By comparing the waveforms of the voltage command values Vur, Vvr, Vwr of each phase with the triangular carrier wave, three-phase PWM pulses Pup, Pvp, Pwp can be obtained. Note that the above-described short-circuit prevention period is omitted for simplicity of description.
[0028]
That is, in FIG. 3, when the PWM pulses Pup, Pvp, and Pwp are high, the switching elements Sup, Svp, and Swp of the upper arm in FIG. 2 are turned on, and the switching elements Sun, Svn, and Swn of the lower arm are turned off. State. When the PWM pulses Pup, Pvp, Pwp are low, the switching elements Sun, Svn, Swn are turned on, and the switching elements Sup, Svp, Swp are turned off.
[0029]
As can be seen from FIG. 3, when the voltage command value of each phase is within a predetermined range including the minimum value and the maximum value of the carrier, there is a period in which the upper arm or the lower arm is in a three-phase short-circuit state. Here, if the processing is performed so that the detection pulse P1 is generated when the carrier wave has the maximum value and the minimum value, the detection pulse P1 is generated when the synchronous motor is in the three-phase short-circuit state. become.
[0030]
It is known that, when the current detection unit 10 detects the current of each phase when the pulse P1 is generated, the instantaneous value of the current is almost the average value of the current of the phase. The state where the phase winding of the synchronous motor is short-circuited is not only the moment of the minimum value and the maximum value of the carrier wave shown in FIG. The predetermined range is a range of the pulse having the smallest width among the PWM pulses Pup, Pvp, and Pwp, and a time between the pulse next to the pulse having the widest pulse. At time t1, the range of Pvp pulse width, at time t2, between Pup and the next Pup, at time t3, range of Pvp pulse width, at time t4, between Pup and next Pup, at time t5, Pwp pulse The range of the width is between Pup and the next Pup at time t6. The times t1 to t5 indicate the instants of the minimum value and the maximum value of the carrier. As described above, the short-circuit state of the phase winding occurs for a predetermined period including the moment of each of the minimum value and the maximum value, and this is repeated. A pulse P1 is created to capture the current in the winding in the short-circuit state of the phase winding. This pulse P1 may be generated within the above-mentioned predetermined period. The method of generating at the minimum value and the maximum value of the carrier as in this embodiment has advantages such as easy generation of a pulse, or almost at the center of a short-circuit state and less possibility of malfunction.
[0031]
Next, features of the embodiment of FIG. 1 will be described.
[0032]
Signals representing the U-phase current iu and the V-phase current iv are input to the current differentiating circuit 12, and the current differential values piu and piv, which are signals representing the differential (or pseudo-differential) values of these current values, are output.
[0033]
The values piu and piv are input to the detection unit 13, and are held and output when the detection pulse P1 is generated. That is, the current differential value piu or piv is detected at the timing of the pulse P1. That is, it is taken into the calculation unit 14.
[0034]
The calculation unit 14 calculates the magnetic pole position θ by performing the processing of the flowchart shown in FIG. 4 by the magnetic pole position calculation unit 14 that calculates the magnetic pole position.
[0035]
First, in step 101, the current differential values piu and piv at the time of three-phase short-circuit are input to the arithmetic unit.
[0036]
In step 102, an operation for obtaining the phase γ of the current differential vector pi when the three phases are short-circuited is performed.
[0037]
FIG. 17 shows the phase relationship of the current differential vector pi. The α-axis current differential value piα and the β-axis current differential value piβ can be obtained from the current differential values piu and piv when the three phases are short-circuited.
[0038]
When the U-phase axis coincides with the α-axis, it is obtained by the following equation.
[0039]
(Equation 1)
piα = (√3 / 2) piu (1)
[0040]
(Equation 2)
piβ = (1 / √2) (piu−2piv) (2)
Next, the phase γ is calculated from piα and piβ using the relationship shown in FIG.
[0041]
In step 103, the magnetic pole position θ is obtained by the following equation.
[0042]
[Equation 3]
θ = γ + π / 2 (3)
The feature of the present embodiment is that it has been newly found that the relationship between the magnetic pole position θ and the phase γ of the three-phase short-circuit current is approximately expressed by Expression (3). The reason will be described below. The basic equation of the synchronous motor can be expressed by the following equation in the dq axis coordinate system. Here, p = d / dt. Ω is the rotation speed of the motor.
[0043]
(Equation 4)
Vd = (R + pLd) id−ωLq iq (4)
[0044]
(Equation 5)
Vq = (R + pLq) iq + ω (Ld id + Φ) (5)
When the synchronous motor is brought into the three-phase short-circuit state, the applied voltage of the synchronous motor becomes Vd = Vq = 0, so the equation for the three-phase short-circuit state is as follows.
[0045]
(Equation 6)
pid = (ωLq iq−R id) / Ld (6)
[0046]
(Equation 7)
piq = − {ω (Ld id + Φ) + R iq} / Lq (7)
The current differential vector in the α-β axis coordinate system in the stationary coordinate system is the sum of the current differential vector in the dq axis coordinate system and the current differential vector generated when the dq axis coordinate system rotates at a speed ω. . Therefore, the d-axis current differential value pids and the q-axis current differential value piqs viewed in the α-β axis coordinate system are respectively
[0047]
(Equation 8)
pids = {ω (Lq−Ld) iq−R id} / Ld (8)
[0048]
(Equation 9)
piqs = − {ω (Ld−Lq) id + Φ) + R iq} / Lq (9)
It becomes. Accordingly, the phase δ of the three-phase short-circuit current differential vector with respect to the d-axis, that is, the magnetic pole position θ is obtained by the following equation.
[0049]
(Equation 10)
tan (δ) ≡piqs / pids
= −Ld [ω ｛(Ld−Lq) id + Φ｝ + R iq]
/ [Lq {ω (Lq-Ld) iq-Rid}] (10)
In the case of the present embodiment, since the condition is Ld = Lq because of the cylindrical synchronous motor,
[0050]
(Equation 11)
tan (δ) = Ld (ωΦ + R iq) / (Lq R id) (11)
It becomes. Here, if id <0, the phase δ is approximated by the following equation.
[0051]
(Equation 12)
δ ≒ −π / 2 (12)
For this reason, the content of the calculation in step 103 is expressed by equation (3).
[0052]
When the motor speed ω is low, the error of the equation (12) becomes large, so that it can be asymptotically obtained by the equation (11). This method will be described in another embodiment described later.
[0053]
As described above, the calculation unit 14 in FIG. 1 can obtain the magnetic pole position θ by a simple calculation. If the coordinate conversion of the coordinate conversion units 8 and 11 is performed using the magnetic pole position θ, it is possible to control the motor to generate a torque according to the required torque command value.
[0054]
Therefore, according to this embodiment, relatively simple calculation can be performed only with the current sensor without using a magnetic pole position sensor for directly measuring a mechanical rotation position such as a resolver or an encoder for the cylindrical synchronous motor. It has the characteristic that the magnetic pole position can be detected by using. Therefore, the control device is inexpensive.
[0055]
Further, even when the synchronous motor loses synchronism for some reason, the magnetic pole position can be detected, so that the motor does not fall into the uncontrolled state.
[0056]
Moreover, since the sensorless control system can be configured only with information obtained when performing the PWM control while performing the normal PWM control, noise and torque pulsation can be reduced as compared with the method of detecting the magnetic pole position by adding a detection additional signal. It has a feature that can be reduced.
[0057]
FIG. 5 shows a control device for another embodiment, that is, a cylindrical synchronous motor capable of detecting a magnetic pole position without using a current differentiating circuit. This embodiment can be realized not only by an electric circuit but also by a program as in FIG.
[0058]
The main differences from the embodiment shown in FIG. 1 are that the current differentiating circuit 12 is not used, the timing of current detection is changed by the detection pulse P2, It is different. It is important that this embodiment does not directly detect the three-phase short-circuit current.
[0059]
First, the detection pulse P2 for controlling the detection timing of the current detection unit 10 will be described with reference to FIG. FIG. 6 shows the same state as the PWM signal of FIG. 3, but the detection pulse P2 of FIG. 6 is different from the detection pulse P1 of FIG. 3 in the following point.
[0060]
Normally, in each phase of the 180-degree conducting three-phase inverter shown in FIG. 2, one of the upper-arm switching element and the lower-arm switching element is in an on state and the other is in an off state. Therefore, at least two of the three phases are always in a short-circuit state.
[0061]
FIG. 6 shows the section. For example, in a section from time t (n-2) to time t (n-1), the switching elements Svn and Swn of the V-phase and W-phase lower arms are turned on, and the V-phase and W-phase of the synchronous motor 1 are turned on. The phases are short-circuited.
[0062]
The section from time t (n-1) to time t (n) indicates that the upper arms of the U-phase and V-phase are in a short-circuit state.
[0063]
As described above, in the 180-degree conduction type inverter, there are two-phase two-phase short-circuit states in one cycle of the carrier wave.
[0064]
As shown in FIG. 6, the detection pulse P2 is generated when the mode in the two-phase short-circuit state is switched.
[0065]
The PWM signal generator 9 synchronizes with the change of the PWM signal in which the phase of the voltage command value having the second largest value among the three phase voltage command values, that is, the intermediate value (duration, ie, pulse width), occurs. To generate a detection pulse P2.
[0066]
The current detection unit 10 captures two-phase current values output by the current sensors 5a and 5b every time the detection pulse P2 is generated, for example, signals iu and iv indicating the values of the U-phase current iu and the V-phase current iv.
[0067]
The U-phase and V-phase currents obtained at this timing are input from the current detector 10 to the calculator 15, and the calculator 15 performs the processing shown in FIG. Here, the average value ua of the U-phase current and the average value iva of the V-phase current selected by the calculation are input from the calculation unit 15 to the coordinate conversion unit 11. The calculated magnetic pole position θ is input to the coordinate conversion units 8 and 11 from the calculation unit 15. The coordinate conversion units 8 and 11 in FIG. 5 perform the same operations as those in FIG.
[0068]
A flowchart of FIG. 7 showing the processing performed by the arithmetic unit 15 will be described. Using the U-phase current iu (n) and V-phase current iv (n) at time t (n) input from the current sensors 5a and 5b to the current detection unit 10 based on the detection pulse P2 in step 111, the U-phase The average value ua (n) and the V-phase current average value iva (n) are calculated in step 112. If the average of the U-phase current iu (n-1) at time t (n-1) and the U-phase current iu (n) at time t (n) is calculated, it is almost equal to the U-phase current iu at time t5 in FIG. Have the same value. Since the U-phase current when the detection pulse P1 is generated is almost the average value, the process of step 112 is performed.
[0069]
In the next step 113, a difference value, that is, a differential value of the current of each phase at time t (n-1) and time t (n) is calculated.
[0070]
Step 114 determines the two-phase short-circuit mode Msc indicating which phase is in the two-phase short-circuit state in the section from time t (n-1) to time t (n). In this case, it can be seen from FIG. 6 that the U-phase and the V-phase of the upper arm are determined. The two-phase short-circuit mode Msc (n-1) in the section from the previous time t (n-2) to the time t (n-1) is "V-W phase short-circuit".
[0071]
In step 115, the short-circuit current difference value is calculated using the arithmetic expression shown in the table of FIG.
[0072]
The short-circuit current difference value picc of the short-circuit axis will be described. In FIG. 18, the short-circuit axis refers to the β-axis when the VW phase is short-circuited, the β ′ axis when the WU phase is short-circuited, and the β ″ axis when the UV-phase short-circuited.
[0073]
For example, when converting a three-phase voltage to an α-β axis coordinate system (α axis coincides with U-phase axis), β-axis voltage Vβ is expressed by the following equation.
[0074]
(Equation 13)
Vβ = (Vv−Vw) / (√2) (13)
Here, in the case of the VW phase short-circuit state, since Vv = Vw, Vβ = 0. That is, it can be said that the β axis is in a short-circuit state, and this axis is referred to as a short-circuit axis. Similarly, in the case of the WU phase short-circuit, the β 'axis rotated by 120 degrees from the β-axis becomes the short-circuit axis, and in the case of the UV phase short-circuit, the β ″ axis rotated by 240 degrees from the β axis becomes the short-circuit axis. .
[0075]
In the case of a cylindrical synchronous motor, the short-circuit current difference value picc of this short-circuit axis matches the short-circuit axis component of the three-phase short-circuit current differential vector pi. FIG. 18 shows the relationship between the vector diagrams.
[0076]
The reason why the vector diagram of FIG. 18 is established will be described by expanding Expressions (4) and (5).
[0077]
The α-axis current differential value piα and the β-axis current differential value piβ are given by the following expressions from Expressions (4) and (5).
[0078]
[Equation 14]

[0079]
(Equation 15)

However, L0 = (Ld + Lq) / 2, L1 = (Ld−Lq) / 2, k1 (θ), k2 (θ), k3 (θ), k4 (θ), k5 (θ), k6 (θ) are Each is a function related to θ.
[0080]
In the case of the cylindrical synchronous motor, since L1 = 0, it can be seen that the β-axis current differential value piβ does not affect the α-axis voltage Vα.
[0081]
In the VW phase short-circuit state, only the α-axis voltage Vα is applied depending on the state of the U-phase voltage Vu, but the β-axis current differential value piβ is the same as when Vα = 0. Further, since Vβ = 0 since the VW phase is short-circuited, it means that the β-axis current differential value piβ in the three-phase short-circuit state coincides. From the above, it can be seen that FIG. 18 holds.
[0082]
Similarly, when the WU phase is short-circuited, the β′-axis current differential value piβ ′ is the same as the β′-axis component of the three-phase short-circuit current differential vector pi. Therefore, when the current differential value (difference value) of the short-circuit axis in the two-phase short-circuit state is detected, the phase γ of the three-phase short-circuit current differential vector can be calculated by calculating the vector diagram of FIG.
[0083]
When the phase γ of the three-phase short-circuit current differential vector is obtained from the current two-phase short-circuit mode Msc (n) and the previous two-phase short-circuit mode Msc (n−1), the calculation method differs depending on the combination of the short-circuit modes. .
[0084]
Therefore, in step 116, the phase γ of the three-phase short-circuit current differential vector is obtained by using the arithmetic expressions divided into the modes as shown in FIG.
[0085]
In step 117, the magnetic pole position θ can be obtained in the same manner as in step 103 in FIG.
[0086]
As described above, according to the present embodiment, the direction of the current differential vector in the three-phase short-circuit state can be determined or calculated from the amount of change in the current in the two-phase short-circuit state having a relatively long duration, that is, the difference value. Is obtained by taking in a small amount of current.
[0087]
Further, since the system of this embodiment does not use a differentiating circuit, it has an advantage that it is resistant to noise and can be realized by a relatively inexpensive controller.
[0088]
The embodiment of FIG. 9 is a configuration diagram of a control device when the present invention is applied to the salient-pole synchronous motor 16. The controller 4 can be realized by an electric circuit or software as in the previous embodiment. In contrast to the embodiment of FIG. 1, the embodiment of FIG. 9 uses the two-phase switching operation unit 18, uses the detection pulses P3 and P4 from the PWM signal generation unit 9, and uses a processing method of the operation unit 17. different.
[0089]
The processing contents of the two-phase switching operation unit 18 will be described with reference to the time chart of FIG.
[0090]
Two-phase switching refers to a method of flowing the same sine wave current as in three-phase switching while stopping one-phase switching among three-phase PWM signals.
[0091]
In FIG. 10, a method of flowing the same sine wave current as in the three-phase switching while stopping the U-phase switching will be described. The V-phase or W-phase other than the U-phase can be a phase for stopping the switching based on the same concept. The additional voltage V0 is forcibly added so that the U-phase voltage command value Vur always becomes the same value as the maximum value of the carrier wave. As a result, the U-phase PWM signal Pup is always in the high state, and the switching element Sup is in the on state.
[0092]
For the V-phase voltage command value Vvr and the W-phase voltage command value Vwr, values obtained by adding an additional voltage V0 to a normal command value are calculated, thereby generating PWM signals Pvp and Pwp.
[0093]
Even if the same voltage is added to all phases, the line voltage is not affected, so that the current flowing through the synchronous motor 16 is the same as the current when the additional voltage V0 is not applied. This is two-phase switching, a well-known method. When this method is used, it can be seen that the three-phase short-circuit state per operation shown in FIG. 10 is longer than in the case of FIG.
[0094]
FIG. 10 also shows detection pulses P3 and P4 generated from the PWM signal generator 9.
[0095]
The detection pulse P3 is generated in synchronization with the maximum value of the carrier wave, and is used by the current detection unit 10 in FIG. 9 to obtain the current average values iua and iva of each phase.
[0096]
The current detection pulse P4 is generated at the start and end of the extended three-phase short-circuit state. 9, the U-phase current iu and the V-phase current iv are input by the current detection pulse P4.
[0097]
These current values are input to the calculation unit 17, and the processing shown in the flowchart of FIG. 11 is performed to calculate the magnetic pole position θ.
[0098]
The processing method of FIG. 11 is performed as follows. In step 121, the U-phase current iu (n-1) and the V-phase current iv (n-1) at the start time t (n-1) of the three-phase short-circuit state and the U-phase current iu at the end time t (n) (N), The current difference value piu, piv, piw of each phase is calculated using the V-phase current iv (n). The processing method is the same as step 113 in FIG.
[0099]
In the next step 122, the phase γ of the differential vector of the three-phase short-circuit current is calculated using the current difference values piu, piv, piw. This process is the same as step 102 in FIG.
[0100]
In the following method, the magnetic pole position used for control in the controller 4 at that time is θ ′, and the actual magnetic pole position of the synchronous motor 16 is θ. The d-axis current and the q-axis current calculated based on the magnetic pole position θ ′ in the controller 4 are described as id ′ and iq ′, and the actual d-axis current and the q-axis current of the synchronous motor 16 are described as id and iq, respectively.
[0101]
In step 123, the d-axis current id 'and the q-axis current iq' are calculated using the magnetic pole position θ 'and the current average values iua and iva input from the current detector 10.
[0102]
In step 124, the operation of Expression (10) is performed using id 'and iq' instead of id and iq, and the phase δ from the magnetic pole position (d-axis) to the three-phase short-circuit current differential vector is obtained.
[0103]
When the motor speed ω is equal to or higher than a predetermined value, the motor speed ω may be obtained by the following approximate expression.
[0104]
(Equation 16)
tan (δ) ≒ −Ld ｛(Ld−Lq) id + Φ｝ / {Lq (Lq−Ld) iq} (16)
In step 125, using the phase γ obtained in step 122, the magnetic pole position θ is obtained by the following equation.
[0105]
[Equation 17]
θ = γ−δ (17)
This relationship is shown in the vector diagram of FIG.
[0106]
In step 126, it is determined whether or not the magnetic pole position θ obtained in step 125 substantially coincides with the magnetic pole position θ ′ when id ′ and iq ′ are obtained in step 123. If they do not match, the processing from step 123 to step 125 is performed again to calculate the magnetic pole position θ.
[0107]
If the actual magnetic pole position θ is different from the magnetic pole position θ ′ in the controller, id ′ and iq ′ do not match id and iq, so that an error occurs in the phase δ. However, the error decreases each time the processing from step 123 to step 125 is performed, and the magnetic pole position θ ′ in the controller converges to the true magnetic pole position θ. This is determined in step 126, and when the calculation of the magnetic pole position θ has almost converged, the calculation ends.
[0108]
In addition, since this calculation is expected to converge within a few degrees in a few times, the convergence determination may be completed not by the calculation result of the magnetic pole position θ but by, for example, two calculations. Good.
[0109]
Further, depending on the relationship between the sampling time for detecting the magnetic pole position and the motor speed, step 126 may be omitted and a method of detecting the magnetic pole position by sampling several times may be employed.
[0110]
As described above, when detecting the magnetic pole position of the salient-pole synchronous motor, it is necessary to calculate using the d-axis current id 'and the q-axis current iq' including errors. The feature of the present embodiment lies in that the algorithm described above is provided. Therefore, there is an advantage that a sensorless control system for a salient-pole synchronous motor can be constructed by utilizing a change in current in a three-phase short-circuit state.
[0111]
In the present system, by using together a method of extending the three-phase short-circuit time, such as a two-phase switching method, the range of change in current during the three-phase short-circuit period can be increased. Therefore, the three-phase short-circuit current differential vector can be directly measured without using a differentiating circuit, and a magnetic pole position detection method that is resistant to noise can be realized by simple software processing. FIG. 12 shows an embodiment of a salient pole type synchronous motor for detecting a magnetic pole position from a two-phase short-circuit state, and shows a configuration of a high reliability system applied to an electric vehicle. In this embodiment, the controller 4 can be realized not only by an electric circuit but also by software. As in the previous embodiment, each block of the controller 4 represents a processing function of software.
[0112]
12 is different from the case of the cylindrical synchronous motor of FIG. 5 in that the processing contents of the arithmetic unit 20 are different, and that the salient-pole synchronous motor 16 drives the tires 24 and 25 of the electric vehicle. And a magnetic pole position sensor 23 that mechanically directly detects the magnetic pole position of the motor 16 for the purpose of improving the reliability of the electric vehicle.
[0113]
First, the processing contents of the arithmetic unit 20 will be described. FIG. 13 shows a flowchart of this processing content.
[0114]
The processing from step 131 to step 134 is the same as the processing from step 111 to step 114 in FIG.
[0115]
The saliency correction phase ε in step 135 is a correction amount necessary for considering the effect of the saliency of the synchronous motor 16.
[0116]
As shown in equation (15), in the case of the salient pole type synchronous motor 16, since L1 ≠ 0, the β-axis current differential value piβ changes according to the α-axis voltage Vα. Therefore, the value becomes different from the β-axis component of the three-phase short-circuit current differential vector.
[0117]
FIG. 19 shows an α-axis current differential value piα1 generated by the α-axis voltage Vα, a β-axis current differential value piβ1, and a current differential vector pi1 that is a combination thereof.
[0118]
Assuming that the axis in the direction coincident with the current differential vector pi1 is the x-axis and the axis orthogonal thereto is the y-axis, the y-axis component of the current differential vector pi1 is always 0 regardless of the α-axis voltage Vα. Therefore, the y-axis component of pi1 is a three-phase short-circuit current differential vector
It matches the y-axis component of pis. This is called a saliency correction phase ε.
[0119]
Therefore, in the case of the salient-pole synchronous motor, the current differential value (difference value) of the y-axis advanced not by the β-axis but by the salient polarity correction phase ε is detected.
[0120]
Actually, since there are three two-phase short-circuit states, the saliency correction phases in the case of the VW-phase short-circuit, the WU-phase short-circuit, and the U-V-phase short-circuit are ε1, ε2, ε3, respectively, and the axis in that direction is Are the y ′ axis and the y ″ axis.
[0121]
The saliency correction phases ε1, ε2, ε3 are given by the following equations from equations (14) and (15).
[0122]
(Equation 18)
tan (ε1) = − (L1 sin2θ) / (L0−L1cos2θ) (18)
[0123]
[Equation 19]

[0124]
(Equation 20)

As described above, in step 135, any one of the equations (18), (19), and (20) is performed according to the two-phase short-circuit state, and the saliency correction phase is obtained.
[0125]
The magnetic pole position θ used in these calculations is a value in the controller 4 and includes an error. However, an accurate magnetic pole position can be obtained while converging as shown in FIG.
[0126]
In step 136, a short-circuit current difference value calculation of the short-circuit axis (any of the y-axis, y'-axis, and y "-axis) corrected from the current difference values piu (n), piv (n) using the table of FIG. To calculate the short-circuit current difference value picc of the short-circuit axis.
[0127]
As described above, the short-circuit axis refers to an axis in a direction that is not affected by the current differential value (difference value) due to the α-axis voltage.
[0128]
In the next step 137, as shown in FIG. 14, the mode to be calculated based on the current and previous two-phase short-circuit state is changed, and the phase γ of the three-phase short-circuit current differential vector is obtained using the arithmetic expression in FIG.
[0129]
An example of the vector diagram at this time is shown in FIG. 20, and this relationship is obtained by the operation formula in FIG.
[0130]
The processing from step 138 to step 140 is the same as the processing from step 123 to step 125 in FIG. 11, taking into account the phase from the magnetic pole position to the current differential vector in the salient-pole synchronous motor 16.
[0131]
As described above, if the arithmetic unit 20 is used, the magnetic pole position can be detected for the salient pole synchronous motor 16 only by detecting the current in the two-phase short-circuit state.
[0132]
In the electric vehicle drive system of FIG. 12, a signal from the magnetic pole position sensor 23 is input to the magnetic pole position detection unit 21 to detect the magnetic pole position θ1.
[0133]
The magnetic pole position abnormality detection unit 22 receives the magnetic pole position θ1 from the magnetic pole position detection unit 21 and the magnetic pole position θ from the calculation unit 20 and performs the processing shown in FIG.
[0134]
At step 141, the magnetic pole position θ input at step 141 is compared with θ1, and it is determined at step 142 whether the difference is within a predetermined normal range.
[0135]
If it is determined to be normal, the magnetic pole position θ1 is stored in the output magnetic pole position θ2 in step 143, and θ2 is output to the coordinate converters 8 and 11 in step 144. If it is determined in step 142 that the two magnetic pole positions are not in the normal range, the electric vehicle is temporarily stopped in step 145.
[0136]
Step 146 determines whether or not the rotation of the synchronous motor 16 has stopped. If it is determined that the vehicle has stopped, a process is performed in step 147 so that the vehicle can run within the safe speed using the normal magnetic pole position. Running within the safe speed means that the upper limit of the speed of the vehicle is limited to 40 km / h or 50 km / h, and the vehicle is controlled by a control device (not shown) so as to run at a speed lower than this speed, and runs under this control.
[0137]
As described above, according to this embodiment, by providing the normal magnetic pole position sensor 23, the magnetic pole position is obtained based on the output of the magnetic pole position sensor 23, and separately from the motor current, the magnetic pole position is obtained by the arithmetic processing. Therefore, there is an effect that a highly reliable electric vehicle can be provided.
[0138]
This embodiment is particularly suitable for reducing the size of the motor and the weight of the electric vehicle by using the reluctance torque.
[0139]
FIG. 16 shows an embodiment of a control system without a magnetic pole position sensor, which has a function of detecting the magnetic pole position only by the current sensor and self-diagnosing whether there is any abnormality in the characteristics.
[0140]
A feature of the embodiment of FIG. 16 is that a self-diagnosis unit 26 is provided in the embodiment of FIG.
[0141]
The calculation unit 20 performs a calculation for detecting the motor speed ω in addition to the processing in FIG.
[0142]
In the above-described embodiment, since the magnetic pole position θ is obtained by detecting the phase γ of the current differential vector, information on the magnitude of the current differential vector is ignored.
[0143]
Therefore, the motor speed ω is determined in reverse from the equations (6) and (7). That is,
[0144]
(Equation 21)
ω = (Ld pid + R id) / Lq iq (21)
[0145]
(Equation 22)
ω = − (Lqpiq + R iq) / (Ld id + Φ) (22)
The motor speed ω is calculated using one or both formulas.
[0146]
Note that in Expressions (21) and (22), a simplified expression that ignores the resistance R may be used. The motor speed ω is output to the self-diagnosis unit 26.
[0147]
Further, the magnetic pole position θ obtained in step 140 of FIG.
[0148]
The self-diagnosis unit 26 determines whether there is any abnormality by comparing the change state of the magnetic pole position θ with the motor speed ω.
[0149]
If it is determined that there is an abnormality, an abnormality diagnosis signal Se is output, and systematic processing is performed to stop the sensorless control system.
[0150]
By estimating the motor speed using a plurality of independent variables of the current differential vector in this manner, a self-diagnosis function can be provided without using another sensor.
[0151]
The above is the embodiment in which the magnetic pole position of the synchronous motor is detected using only the current sensor.
[0152]
In addition to the synchronous motor, the present invention can be applied to a reluctance motor using saliency.
[0153]
In the present embodiment, the description is omitted because the description is complicated.However, the magnetic pole position may be calculated in consideration of the influence of the rotation of the motor rotor during the sampling time. It goes without saying that it can be applied.
[0154]
Although the example of the electric vehicle has been described here, the present embodiment is also used for the magnet motor that is currently subjected to sensorless control using the 120-degree conduction type inverter control, by using the 180-degree conduction type inverter control. A sensorless system with low torque pulsation and low noise can be provided.
[0155]
The embodiment described with reference to FIGS. 1 to 20 is most suitable for control in a state where the motor 1 is rotating at a predetermined rotation speed or higher. Here, the predetermined rotation speed or more is, for example, 800 rotations / minute or more. In a low-speed state including a stopped state of the motor, it may be necessary to detect the magnetic pole position with higher accuracy for a reason to be described later. An embodiment in which the magnetic pole position in the low-speed state including the stop state of the motor can be performed with high accuracy will be described with reference to FIG. A major difference between FIG. 1 and FIG. 21 is that an arithmetic unit 52 for detecting the magnetic pole position at the time of low-speed rotation of the motor including the stopped state is provided. The controller 4 can be realized not only by an electric circuit but also by software, as in the other embodiments. When implemented by software, each block of the controller 4 represents a processing function of the software.
[0156]
The processing of detecting the magnetic pole position by the arithmetic unit 14 is as described with reference to FIG. The operations of the current command value generator 6, current controller 7, coordinate converter 8, PWM signal generator 9, current detector 10, coordinate converter 11, and inverter 3 are basically the same as those in FIG. The processing method of calculating the magnetic pole position of the motor based on the change amount or the change direction of the motor current when the synchronous motor is in the three-phase short-circuit state has been described above, but will be described again.
[0157]
In this magnetic pole position calculation method, the magnetic pole position is calculated based on the change amount or change direction of the motor current in a three-phase short-circuit state that is not affected by the voltage applied from the inverter. In order to detect the amount or direction of change of the motor current in the three-phase short-circuit state, a pulse P1 is generated to control the timing in the three-phase short-circuit state as described with reference to FIG. As a typical example, the pulse P1 is generated at the highest value and / or the lowest value of the carrier wave shown in FIG. 3, and a current value or a differential value of the current value is detected in synchronization with the pulse P1. That is, the pulse P1 is generated by the PWM signal generating section 9, and the pulse P1 is used as a signal indicating a three-phase short-circuit state of the motor, and the pulse P1 is used as a trigger to detect a change amount of the motor current. The relationship between the vector of the motor current change amount at the time of the three-phase short circuit detected at this time, that is, the differential vector is as shown in FIG. 17 described above.
[0158]
In FIG. 17, the magnetic pole position to be detected is the phase θ between the stationary coordinate α axis and the rotating coordinate d axis, and can be expressed by the above equation (17).
[0159]
[Equation 17]
θ = γ−δ (17)
Here, γ is the phase of the motor current differential vector pIs with respect to the α-axis when the three phases are short-circuited, and δ is the phase of the motor current differential vector pIs with respect to the d-axis. For the phase γ, first, the motor currents iu and iv are differentiated by the current differentiating circuit 12 shown in FIG. 1, and the current differential value detecting unit 13 detects the short-circuited state in synchronism with the motor short-circuit detection pulse P1. The differential values piu and piv of the motor current are fetched. Further, the differential values piu and piv of the motor current at the time of this short circuit are calculated as the current differential values piα and piβ of the α-β axis according to the equations (1) and (2), and the phase γ is calculated based on the equation (23). I do.
[0160]
(Equation 1)
piα = (√3 / 2) piu (1)
[0161]
(Equation 2)
piβ = (1 / √2) (piu + 2piv) (2)
[0162]
(Equation 23)
γ = tan ^-1 (Piβ / piα) (23)
Here, two phases of piu and piv are used to obtain piα and piβ, but the calculation can also be performed using differential values of three-phase currents of piu, piv, and piw. Furthermore, although a differentiating circuit is used to calculate the current differential value at the time of the three-phase short circuit, if the configuration of the differentiating circuit is not possible, instead of the current differential value, the amount of current change in the three-phase short circuit section is calculated. It can also be realized by using a current change rate obtained by calculating and dividing a current change amount by a short-circuit time. In addition, when the short-circuit section is very short and the calculation of the current change rate in the three-phase short-circuit section cannot be performed, the three-phase short-circuit section can be used even if the two-phase short-circuit section is surely longer than the three-phase short-circuit section. It is possible to calculate the current change rate in the section. Details are as described above.
[0163]
As described above, the phase γ of the motor current differential vector pIs at the time of the three-phase short circuit with respect to the α-axis is obtained. Next, the phase δ of the motor current differential vector pIs with respect to the d-axis is obtained as follows. First, the basic equation of the synchronous motor on the d-q rotation coordinates can be represented by the following equation. Note that these equations are as described above.
[0164]
(Equation 4)
Vd = (R + pLd) id−ωLq iq (4)
[0165]
(Equation 5)
Vq = (R + pLq) iq + ω (Ld id + Φ) (5)
Here, Vd and Vq are dq axis voltages, Ld and Lq are dq axis inductances, R is winding resistance, ω is motor angular velocity, φ is field magnetic flux, and p is d / dt. In the above basic formula, when a three-phase short circuit occurs, the applied voltage on the dq axis is 0, so the basic formula in a three-phase short circuit state is as follows.
[0166]
(Equation 6)
pid = (ωLq iq−R id) / Ld (6)
[0167]
(Equation 7)
piq = − {ω (Ld id + Φ) + R iq} / Lq (7)
The current differential vector on the stationary coordinate α-β axis is generated by rotating the current differential vector on the rotating coordinate dq axis and the dq axis expressed by the equations (6) and (7) at the motor angular velocity ω. The current differential vector is expressed as Therefore, the current differential value of the dq axis viewed on the α-β axis is as follows.
[0168]
(Equation 8)
pids = {ω (Lq−Ld) iq−R id} / Ld (8)
[0169]
(Equation 9)
piqs = − {ω (Ld−Lq) id + Φ) + R iq} / Lq (9)
Therefore, the phase δ of the motor current differential vector pIs with respect to the d-axis is represented by Expression (24).
[0170]
(Equation 24)

Therefore, the magnetic pole position can be obtained from the above equations (17), (23) and (24). Here, ω included in the equation (24) uses an estimated value of the angular velocity obtained from the change amount of the phase estimated value. Further, in a region where the angular velocity is sufficiently large and the R component can be ignored, the influence of ω is eliminated.
[0171]
The above is the outline of the position detection method for detecting the magnetic pole position of the synchronous motor based on the amount of change or the direction of change of the motor current when the synchronous motor is short-circuited in three phases. This method is applicable not only to salient pole type synchronous motors but also to cylindrical type synchronous motors.
[0172]
In the above-described position detection method, since the amount of current change in the motor short-circuit state that occurs during normal PWM operation is used, the magnetic pole position can be estimated up to a high-speed region without applying a special estimation signal.
[0173]
However, in the following operation range, there is a possibility that the position detection accuracy is reduced only by this detection method. FIG. 22 shows the operation range. When the speed is zero, that is, when the motor 1 stops rotating, the torque of the motor 1 is also zero. In a motor rotation stop state, for example, a state in which the motor is to be started, the motor current has not yet flowed, so that even if the PWM control is performed, the amount of change in the current at the time of short circuit cannot be detected. Furthermore, even after startup, when the rotation speed is low and the torque is small, the current value is small, and the above measurement is difficult. The detection accuracy of the magnetic pole position decreases. Further, in a region where the current flows near zero speed, the direction of the magnetic flux can be detected by the amount of change in the current at the time of the short circuit, but the effect of the induced voltage is very small. The polarity, that is, the polarity cannot be determined.
[0174]
In the above-described operating range, for example, 800 rpm (rotation / minute), the present embodiment can improve the detection accuracy of the magnetic pole position by newly providing the detection unit 52. The detection unit 52 will be described with reference to FIG. The detection unit 52 basically has a signal generation unit 54 that generates a magnet position estimation signal and a polarity determination unit 56.
[0175]
When detecting the magnetic pole position in an area where no current flows or an area where the current value is small, for example, when the motor 1 is started, the signal generation unit 54 that generates the magnet position estimation signal detects the change amount of the motor short-circuit current. A current command value idhr for position detection is generated so that a current as large as possible flows, and the current command value idhr is given to the current command value generating section 6.
[0176]
FIG. 24 shows the configuration of the current command value generator 6. In the embodiments of FIGS. 1 to 20 and the control system of FIG. 21, the torque command Tr is input to the current command value generator 6 from the upper control unit of the controller 4. The current command value generator 6 has a torque controller 63, and calculates current commands idr and iqr for the dq axes so that the synchronous motor 1 generates a torque as instructed. Normally, a torque command and a motor speed (not shown in FIG. 24) are input, and idr and iqr that maximize the efficiency at the current operating point are calculated using a map search or the like. Here, the normal state means, for example, a state where the synchronous motor 1 is rotating at a speed higher than a predetermined rotation speed of 800 rotations / minute.
[0177]
As described above, the current command idhr for detecting the magnetic pole position at the time of low-speed rotation of 800 revolutions / minute or less, such as when the synchronous motor is started, is output from the signal generator 54 and applied to the adder 65. The current command idhr is added to the current command idr1 in the d-axis direction by the adder 65, and the addition result is added from the current command value generator 6 to the current controller 7 as a d-axis current command idr. In this embodiment, the current command idhr for detecting the magnetic pole position is applied to the current command in the d-axis direction. This is to prevent unnecessary torque generation due to application of idhr. That is, if the current in the q-axis direction is zero, basically, no torque is generated even if a current is applied to the d-axis. However, the magnetic pole position can be detected even when idhr is applied in the q-axis direction or when both d and q axes are applied. Therefore, if the generation of torque is permissible in the entire control, it can be added to the current command in the q-axis direction for detecting the magnet position. Furthermore, only the short time corresponding to the measurement may be added.
[0178]
Further, the position can be detected by using an AC signal or a DC signal as the applied position estimation current command idhr. When the current command idhr is an AC signal, the average value of the torque generated due to the deviation between the position detection value generated at the initial stage of the estimation operation and the magnetic pole position of the motor is zero. Therefore, the influence of torque fluctuation can be suppressed.
[0179]
As described above, at the operating point where the motor speed is zero and no current is flowing, for example, at the time of start-up when starting rotation from a stopped state, the motor is short-circuited by applying a position estimation signal. Position detection using the current change amount at the time becomes possible. However, although the detected position is in the magnetic flux direction, it is unknown whether it is in the N-pole direction or the S-pole direction. This is because the motor is not affected by the induced voltage because the angular velocity ω is zero.
[0180]
In order to solve this problem, in the embodiment shown in FIG. 23, a polarity discriminating unit 56 is provided to discriminate whether the calculated position is the N pole or the S pole, that is, the polarity of the magnetic pole. The discrimination method used in the polarity discrimination unit 56 is not particularly limited, but a method using the magnetic saturation characteristics of the motor is effective. Hereinafter, a method using magnetic saturation characteristics will be described as an example of the polarity determination method.
[0181]
In the magnetic characteristics of the synchronous motor, since the rotor has a magnetic flux generated by the permanent magnet, a magnetic flux exists even when the current id of the d-axis, which is the magnetic flux axis, is zero as shown in FIG. Due to this magnetic characteristic, the characteristic of the d-axis inductance Ld is as shown in FIG. From FIG. 26, it can be seen that there is a region where the magnitude of the inductance Ld is different depending on the difference between the positive and negative of the current id on the d-axis, and a region hatched in FIG. Therefore, if an AC signal having a bias component such that id flows in the shaded area in FIG. 26 is applied to the d-axis, the response of the current control system changes depending on the magnitude of the inductance Ld. The difference in the polarity appears in the difference in the amplitude of the motor current, and the polarity of the magnetic pole can be determined by measuring the magnitude of the amplitude of the motor current. The above is an example of the polarity determination method.
[0182]
Further, the procedure of this polarity discrimination method is represented as follows using FIG. First, an AC signal having a DC bias component is applied to a d-axis current command from a signal generator 54 that generates a signal for detecting a magnetic pole position. Next, the detected d-axis current id is input to the polarity discriminating unit 56, and the polarity discriminating unit 56 measures the amplitude value of the AC component of id, and discriminates the polarity of the current position set value θ ＾.
[0183]
If the determined result is the N pole, the position set value θ ＾ of the control system is used for control as it is. On the other hand, if the determined result is the S pole, the position setting value θ ＾ is added or subtracted by 180 ° to correct the position setting value θ ＾ to the N pole. Also, in this polarity discrimination method, current flows in the d-axis direction to the extent that magnetic saturation occurs in the motor. Therefore, if there is any error in the position set value θ ＾, torque in a certain direction is applied to the motor. Occurs. Therefore, the magnetic pole position detecting operation at the time of starting is provided with a lock mechanism that mechanically temporarily prevents the rotation of the rotating shaft of the motor or the rotor so that the shaft of the motor is not rotated by this torque, and the rotation of the rotating shaft is provided. May be started in a state where is blocked. For example, in FIG. 21, a lock mechanism 74 for preventing the rotation of the rotor 70 or the rotating shaft 72 of the motor 1 is provided. This lock mechanism 74 is the same mechanism as a normal brake mechanism, and temporarily stops rotation by a signal from the detection unit 52. This mechanism works at the start of rotation of the motor. Since the torque is small, the purpose can be achieved even with a simple mechanism. When the detection of the magnetic pole position and the polarity for starting the rotation is completed, the lock mechanism releases the inhibition of the rotation by a signal from the detection unit 52.
[0184]
The above is an example of the polarity discrimination method.
[0185]
Next, the processing procedure at the time of starting the motor by the magnetic pole position detection unit at zero speed described above will be described with reference to the flowchart of FIG. In FIG. 27, the letter S means a step, that is, a procedure. First, in step 30 of FIG. 27, it is determined whether the rotation of the rotation shaft of the motor is blocked by the lock mechanism. As a result, if the rotation of the rotating shaft is prevented, in step 31 a signal idhr for detecting the magnet position is output from the polarity discriminating section 56 and added to the current command idrl of the current command value generating section 6 to calculate. Is applied to the adder 65.
[0186]
The above description is for the case where there is a lock mechanism for preventing the rotation of the motor when the motor is started, that is, when the motor starts rotating. If there is no such mechanism, step 30 is unnecessary. If there is a lock mechanism and the lock mechanism is not operating, an instruction to operate the lock mechanism is issued in step 30 or the operation is awaited by an instruction from another control unit. Then, the process proceeds to step 31 with the lock mechanism operated.
[0187]
In step 32, the short circuit current generated during the PWM control is detected, and the detection value θ ＾ is calculated by the magnetic pole position detection method based on the change amount of the short circuit current described above. Further, in step 33, it is determined whether the detected value θ ＾ of the magnetic pole position obtained in step 32 is in the N-pole direction or the S-pole direction. If the result of this determination is that it is an N pole, in step 34 the detected value θ 検 出 is made the current magnetic pole position. On the other hand, if it is determined in step 33 that the magnetic pole is the south pole, the detection value θ ＾ is corrected by adding or subtracting 180 ° in step 35, and the current magnetic pole position is obtained. Next, in step 36, the drive control of the motor is started using the obtained detected value of the magnetic pole position.
[0188]
The above is the processing procedure at the time of starting the motor using the detection method of the magnetic pole position where the motor rotation is zero speed. If the detection accuracy is low only by the magnetic pole position detection method based on the change amount of the short-circuit current of the motor, the above processing can be performed to detect the magnetic pole position with high accuracy. In this embodiment, the rotation axis of the motor is locked as a condition for starting the magnetic pole position detection operation. However, the magnetic pole position can be detected even when the rotation axis is not locked as described above. .
[0189]
The operation of the calculation unit 52 when the rotation speed of the motor is zero and the motor current is flowing will be described. In such an operation environment, when the rotation is stopped, the load torque, that is, the torque of the motor required for driving the device driven by the motor is larger than the torque generated by the motor, and further, the magnetic pole position in the controller is affected by noise and the like. Is deleted (it is no longer accurate). In such a case, since the motor current is flowing, the magnetic pole position can be detected based on the change amount of the motor short-circuit current described above. However, since the motor speed is zero, it is necessary to determine the polarity. In this case, since the rotation axis of the motor is not locked and the motor is being driven, the polarity determination method using the magnetic saturation characteristic described above is often more desirable than the method described below.
[0190]
The method described here is a method of determining from the direction in which the motor torque is generated and the rotation direction of the motor shaft. The operation of this determination method, that is, the processing content will be described with reference to the flowchart of FIG. First, in step 40, a desired torque is generated by using the detected value θ に よ っ て obtained by the change amount of the short-circuit current of the motor as a set value of the magnetic pole position. Thereafter, the rotational direction of the motor shaft and the direction of the generated torque are referred to in step 41, and if both directions match, in step 42, the set value of the magnetic pole position is set as the current magnetic pole position. On the other hand, in step 41, if the rotation direction of the motor shaft does not match the direction of the generated torque, it is determined that the polarities are opposite, and in step 43, the set value is corrected by 180 ° and the current magnetic pole is corrected. Position. Further, at step 44, the torque command is increased. The above operation is repeated a predetermined number of times, and when the rotation direction of the motor shaft coincides with the direction of the generated torque continuously for a set number of times or more, the polarity determination is terminated. The above is the processing procedure of the method of determining the polarity of the set value of the magnetic pole position from the direction in which the torque is generated and the rotation direction of the rotating shaft of the motor. In this method, it is not necessary to apply a special polarity discrimination signal.
[0191]
As described above, the magnetic pole position detection method using the magnetic pole position detection method using the change amount of the short-circuit current of the motor and the magnetic pole position detection method when the motor speed is zero have been described. Is also applicable. For example, the effect is large at a rotation of 800 revolutions / minute or less.
[0192]
Further, the present method can be directly applied to a drive device of an electric vehicle such as an electric vehicle or a hybrid vehicle having both a motor and an engine. In an electric vehicle or hybrid vehicle drive device, when a signal for position detection is applied to perform detection or when a polarity discrimination method using magnetic saturation characteristics is used, a brake is applied or an operating range is applied. If the vehicle is in the parking range, vibration and movement of the vehicle due to unnecessary torque generation can be avoided.
[0193]
According to the above-described embodiment, the magnetic pole position can be detected in the entire operation range of the synchronous motor without being affected by the state of the applied voltage and while performing ordinary PWM control with an inexpensive controller.
[0194]
Further, according to the embodiment of FIGS. 21 to 28, accurate control can be performed from the time when the motor is stopped.
[0195]
Further, in the embodiment shown in FIGS. 1 to 28, the magnetic pole position of the synchronous motor is obtained from the change amount or the change direction of the motor current, so that the magnetic pole position can be determined while performing the normal PWM control without providing a position detector. Can be detected.
[0196]
Further, a magnetic pole detector is provided, and by comparing the magnetic pole position detected by the magnetic pole detector with the magnetic pole position obtained from the motor current, it is possible to detect abnormality of the magnetic pole position detector while performing normal PWM control. it can.
[0197]
【The invention's effect】
ADVANTAGE OF THE INVENTION According to this invention, the drive system of the synchronous motor excellent in controllability can be provided at low cost.
[0198]
Further, when a normal magnetic pole position sensor is used, an abnormality of the sensor can be detected, so that a highly reliable synchronous motor drive system can be provided.
[Brief description of the drawings]
FIG. 1 is a configuration diagram showing an embodiment when the present invention for detecting a magnetic pole position of a cylindrical synchronous motor using a current differentiating circuit is applied.
FIG. 2 is a configuration diagram showing a method of connecting an inverter 3 of FIG. 1;
FIG. 3 is a time chart showing a relationship between a carrier signal, a three-phase voltage command value, and a PWM signal, and showing a timing at which a current is taken.
FIG. 4 is a flowchart for detecting a magnetic pole position in the configuration method of FIG. 1;
FIG. 5 is a block diagram showing an embodiment for calculating a magnetic pole position by detecting a current in a two-phase short-circuit state of the cylindrical synchronous motor.
6 is a time chart showing detection timings of a three-phase PWM signal and the current shown in FIG. 5;
FIG. 7 is a flowchart for detecting a magnetic pole position in the configuration method of FIG. 5;
8 is a list of arithmetic expressions for calculating the phase of a two-phase short-circuit current differential value and the three-phase short-circuit current differential vector of FIG. 7;
FIG. 9 is a configuration diagram showing another embodiment for detecting the magnetic pole position of the salient-pole synchronous motor by using the current difference while extending the three-phase short-circuit time.
10 is a time chart showing detection timings of a three-phase PWM signal and the current shown in FIG. 9;
FIG. 11 is a flowchart for detecting a magnetic pole position with high accuracy in the configuration method of FIG. 9;
FIG. 12 is an electric vehicle in which a salient-pole synchronous motor is controlled using a first magnetic pole position detector, in which another embodiment having a second magnetic pole position detector for detecting a magnetic pole position with a current in a two-phase short-circuit state; It is a block diagram showing an example.
FIG. 13 is a flowchart for detecting the magnetic pole position of the salient pole type synchronous motor using the current in a two-phase short-circuit state in the configuration method of FIG. 12;
14 is a list of arithmetic expressions for calculating the two-phase short-circuit current difference value and the phase of the three-phase short-circuit current differential vector in FIG.
FIG. 15 is a flowchart for performing abnormality determination of a magnetic pole position in FIG. 12;
FIG. 16 is a configuration diagram showing another embodiment for self-diagnosing a failure in magnetic pole position detection in a salient pole synchronous motor having a magnetic pole position detector that detects a magnetic pole position with a current in a two-phase short-circuit state. .
FIG. 17 is a vector diagram showing an example of a relationship among a current vector, a current differential vector, and a magnetic pole position (d-axis) of the synchronous motor.
18 is a vector diagram showing a relationship between a current differential vector at the time of two-phase short-circuit and a current differential vector at the time of three-phase short-circuit in the cylindrical synchronous motor of FIG. 9;
FIG. 19 is a vector diagram showing a relation of a current differential vector generated by a voltage applied to the α-axis of the salient-pole synchronous motor.
20 is a vector diagram showing a relationship between a current differential vector at the time of two-phase short-circuit and a current differential vector at the time of three-phase short-circuit in the salient-pole synchronous motor of FIG. 16;
FIG. 21 is a configuration diagram of a motor control system showing another embodiment of the present invention.
FIG. 22 is a diagram illustrating an area where detection of a magnet position may be reduced.
FIG. 23 is a diagram illustrating a configuration of a calculation unit 52.
FIG. 24 is a diagram showing a configuration of a current command value generator 6.
FIG. 25 is a diagram showing magnetic characteristics of the synchronous motor.
FIG. 26 is a diagram showing characteristics of d-axis inductance.
FIG. 27 is a diagram showing a processing procedure for detecting a magnetic pole position at the time of starting the motor.
FIG. 28 is a diagram illustrating a processing procedure for determining polarity from a torque generation direction and a rotation direction of a rotating shaft of a motor.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 1 ... Cylindrical synchronous motor, 2 ... Battery, 3 ... Inverter, 4 ... Controller, 5a, 5b ... Current sensor, 6 ... Current command value generation part, 7 ... Current control part, 8, 11 ... Coordinate conversion part, 9 ... PWM signal generation unit, 10, 27 current detection unit, 12 current differentiation circuit, 13 current differentiation detection unit, 14, 15, 17, 20, 52 ... calculation unit, 16 ... salient pole synchronous motor, 18 ... 2 Phase switching calculator, 19: magnetic pole position / current calculator, 21: magnetic pole position detector, 22: magnetic pole position abnormality detector, 23: magnetic pole position sensor, 24, 25 ... tire, 26: self-diagnosis unit, 54: signal Generation unit, 56: polarity discrimination unit, 65: adder. 70: rotor, 72: rotating shaft, 74: lock mechanism.

Claims (28)

A synchronous motor, an inverter that drives the synchronous motor, and a magnetic pole position of the synchronous motor that is detected based on a change amount of a motor current when the synchronous motor is in a short circuit state, and the synchronization is performed based on the detected magnetic pole position. A controller that outputs a control signal for controlling the motor, wherein the inverter controls the synchronous motor based on the control signal. A synchronous motor, an inverter that drives the synchronous motor, and a magnetic pole position of the synchronous motor that is detected based on a change direction of the motor current when the synchronous motor is in a short circuit state, and the synchronous motor is detected based on the detected magnetic pole position. A controller for generating a control signal for controlling, wherein the inverter controls the synchronous motor based on the control signal. A synchronous motor control device including a synchronous motor, an inverter that drives the synchronous motor, and a controller that generates a control signal, wherein the inverter drives the synchronous motor based on a control signal generated by the controller. ,
The controller detects a change direction of the motor current when the synchronous motor is in a short-circuit state, and sets a dq-axis coordinate system in which a magnetic flux direction is a d-axis and a direction orthogonal to the d-axis is a q-axis. The d-axis current and the q-axis current on the dq-axis coordinate system, and calculates the magnetic pole position of the synchronous motor based on the detected change direction, the d-axis current, and the q-axis current. A synchronous motor control device for generating the control signal according to a position. A three-phase synchronous motor, a controller that generates a PWM signal based on a three-phase voltage command value, and an inverter that drives the synchronous motor based on the PWM signal;
The controller detects a current in synchronization with a PWM signal of a phase that commands an intermediate value among the three-phase voltage command values, detects a magnetic pole position of the synchronous motor based on the current, and detects the magnetic pole position based on the detected magnetic pole position. A synchronous motor control device, wherein the three-phase voltage command value is determined based on the value. 4. The synchronous motor control device according to claim 1, wherein the short-circuit state of the synchronous motor is a state in which all phases are short-circuited. 5. 4. The synchronous motor control device according to claim 1, wherein the controller is configured to control the motor current when two phases generated when the inverter controls the synchronous motor by PWM control are in a short-circuit state. 5. A synchronous motor control device for detecting a change amount or a change direction of the motor. 3. The synchronous motor control device according to claim 1, wherein the controller is configured to control the motor current in a three-phase short-circuit state generated when the inverter controls the synchronous motor by PWM control. 4. A synchronous motor control device for detecting a change amount or a change direction. 4. The synchronous motor control device according to claim 1, wherein the controller is configured such that a change amount or a change direction of the motor current in the short-circuit state is a change in a motor current in a three-phase short-circuit state of the motor. 5. An amount or direction of change, wherein the amount of change or direction of change is obtained from the amount or direction of change of the motor current when the synchronous motor is in a short-circuit state of a plurality of different two phases. . 4. The synchronous motor control device according to claim 1, wherein the controller includes an extension unit that lengthens a three-phase short-circuit state. 5. 10. The synchronous motor control device according to claim 9, wherein the extension means performs a two-phase switching operation to lengthen a three-phase short-circuit state. 5. The synchronous motor control device according to claim 1, wherein the controller includes a first motor speed obtained from the detected change state of the magnetic pole position and a second motor speed obtained from a change amount of the motor current. 6. A synchronous motor control device for determining an abnormality by comparing the motor speed with the motor speed of the motor. A synchronous motor, a magnetic pole position detector for detecting a magnetic pole position of the synchronous motor, a controller for controlling the synchronous motor based on a magnetic pole position detected by the magnetic pole position detector, and driving the synchronous motor based on a signal from the controller The controller determines a magnetic pole position of the synchronous motor based on a change amount or a change direction of the motor current when the synchronous motor is in a short circuit state, and detects a magnetic pole position detected by the magnetic pole position detector. And a magnetic pole position obtained from a change amount or a change direction of the motor current to detect an abnormality of the magnetic pole position detector or the controller. 13. The synchronous motor control device according to claim 12, wherein the controller controls the synchronous motor based on a magnetic pole position obtained from a change amount or a change direction of the motor current when detecting an abnormality of the magnetic pole position detector. A synchronous motor control device characterized by the above-mentioned. A synchronous motor that drives a vehicle, a magnetic pole position detector that detects a magnetic pole position of the synchronous motor, a controller that controls the synchronous motor based on a magnetic pole position detected by the magnetic pole position detector, and a controller that controls the synchronous motor based on a signal from the controller. An inverter for driving a synchronous motor, the controller obtains a magnetic pole position of the synchronous motor based on a change amount or a change direction of a motor current when the synchronous motor is in a short-circuit state, and the magnetic pole position detector An electric vehicle control device that compares a detected magnetic pole position with a magnetic pole position obtained from a change amount or a change direction of a motor current to detect an abnormality of a magnetic pole position detector or a controller. 13. The electric vehicle control device according to claim 12, wherein the controller controls the synchronous motor based on a magnetic pole position obtained from a change amount or a change direction of the motor current when detecting an abnormality of the magnetic pole position detector. An electric vehicle control device characterized by the above-mentioned. 15. The electric vehicle control device according to claim 14, wherein the controller stops the vehicle when detecting the abnormality. 17. The electric vehicle control device according to claim 16, wherein after stopping the vehicle, the controller is configured to select one of a magnetic pole position obtained from a magnetic pole position detector and a magnetic pole position obtained from a motor current by a normal controller. An electric vehicle control device, wherein the vehicle is driven using normal magnetic pole position detection means. A first step of detecting a change direction of the motor current when the synchronous motor is in a short circuit state, and a second step of setting a dq-axis coordinate system in which a magnetic flux direction is a d-axis and a direction orthogonal to the d-axis is a q-axis. And a third step of detecting the d-axis current and the q-axis current on the set dq-axis coordinate system; and performing the synchronization based on the detected change direction, the d-axis current, and the q-axis current. A synchronous motor control method comprising: a fourth step of calculating a magnetic pole position of a motor; and a fifth step of controlling the synchronous motor based on the magnetic pole position. 19. The synchronous motor control method according to claim 18, wherein a difference between the magnetic pole position in the dq axis coordinate system set in the second step and the magnetic pole position calculated in the fourth step is within a predetermined value range. Wherein the synchronous motor is controlled based on the magnetic pole position calculated in the fourth step. A synchronous motor, a controller, and an inverter that drives the synchronous motor based on an output of the controller,
A controller configured to detect a magnetic pole position of the synchronous motor based on a change amount or a change direction of the motor current when the synchronous motor is in a short-circuit state; and to generate the output based on the detected magnetic pole position. And a control unit that performs
Furthermore, a second detector is provided for detecting the magnetic pole position at a low speed including the rotation stop state when the speed of the synchronous motor is low, and an output of the second detection unit is provided at a low speed including the rotation stop state when the synchronous motor speed is low. And controlling the motor based on the output of the first detection unit in a region where the rotation speed is higher than the low speed. The control device according to claim 20, wherein
The second detection unit generates a signal for detecting a current used to calculate a magnetic pole position, and determines whether the calculated magnetic pole position is in the N-pole direction or the S-pole direction. A control device for a synchronous motor, comprising: a polarity determination unit. The control device according to claim 21,
A PWM pulse is sent from the controller to the inverter so that a current flows through the motor based on an output of the signal generation unit, and a magnetic pole position is calculated by detecting a motor current based on an output of the signal generation unit, A synchronous motor control device for controlling a motor. The control device according to claim 21, wherein the control device has a current control unit that controls a torque of the motor,
Wherein the polarity discriminating unit applies a current command for polarity discrimination in the d-axis direction of the current control unit, and discriminates the polarity of a magnetic pole based on a difference in response characteristics of the synchronous motor. Control device. The control device according to claim 21,
The polarity determining unit may determine a polarity based on a torque of a synchronous motor generated by a magnetic pole position obtained based on the calculated magnetic pole position and a rotation direction of a rotation shaft of the synchronous motor. A synchronous motor control device. 21. The control device according to claim 20, further comprising: a mechanism for preventing rotation of the synchronous motor, wherein the rotation of the synchronous motor is prevented.
A control device for a synchronous motor, wherein a current based on an output signal of the second detection unit is supplied to the motor. A control device for an electric vehicle, comprising the control device for a synchronous motor according to at least one of claims 20 to 25 . The control device for an electric vehicle according to claim 26,
When the current supply to the motor based on the output of the second detection unit is started, the current supply is performed in a state in which the rotation of the drive wheels of the electric vehicle is stopped, including a state in which the brake of the vehicle is operating. Control device for electric vehicles. The control device for an electric vehicle according to claim 26,
A control device for an electric vehicle, characterized in that at the start of current supply to the motor based on the output of the second detection unit, the operation is performed in a state where the operating range of the electric vehicle is a parking range.

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