JP2000175485A - Synchronous motor control device and electric vehicle control device, and synchronous motor control method for the synchronous motor control device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To manufacture at low cost a high-performance control system of a synchronous motor, that does not affect the state of an impressed voltage and performs normal pulse width modulation(PWM) control with a low-cost controller. SOLUTION: This motor control device is provided with a synchronous motor 1, an inverter 3 and a controller 4. When the inverter 3 drives the synchronous motor 1 in a three-phase short-circuit, i.e., when its carrier signal reaches a maximum or minimum value, the change in the motor current is detected by the current differentiating and detecting part 13 of the controller 4. A computing part 14 computes a phase γfrom an α-axis of a stationary coordinate system to the differential vector of the three-phase short-circuit. A phase δ from a d-axis (magnetic pole position) to the differential vector of the three-phase short-circuit is inferred, using a d-axis current id and a q-axis current iq of a d-q axis coordinate system. Then, a magnetic pole position θ for the α-axis is calculated from the phases γ and δ. Based on this magnetic pole position θ, d-q axis current control system 11, 7, 8 are constituted to control the synchronous motor 1.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a synchronous motor control device and control method, or an electric vehicle control device using the same. In the present invention, the synchronous motor includes a reluctance motor.

[0002]

2. Description of the Related Art In order to control the speed and torque of a synchronous motor, it is necessary to detect or estimate a magnetic pole position. By performing current control or voltage control based on the detected magnetic pole position, the torque and speed of the synchronous motor can be controlled.

[0003] Previously, the magnetic pole position was detected by a position detector. However, in recent years, a method of controlling a synchronous motor by estimating a magnetic pole position, which is different from a method of detecting a magnetic pole position with a magnetic pole position detector, that is, a magnetic pole position sensorless control method has been proposed.

For example, the Transactions of the Institute of Electrical Engineers of Japan, Vol.
D, No. 1, 1997, "Sensorless salient-pole type brushless DC motor control based on speed electromotive force estimation" (Takeshita, Ichikawa et al.) Describes a method of estimating speed electromotive force using a motor model. A method for performing speed control has been proposed.

Japanese Unexamined Patent Publication No. Hei 8-205578 discloses that the saliency of a synchronous motor is determined from the correlation between the voltage vector applied to the synchronous motor by pulse width control (PWM control) and the ripple component of the motor current vector. Are described.

The technique described in the above paper is a method of estimating a magnetic pole position from a difference between a current calculated by a control model and an actually flowing motor current, and has a feature that a control system can be constituted only by control calculation of a controller. is there.

The technique described in Japanese Patent Application Laid-Open No. 8-205578 uses a general PWM signal for controlling the voltage of a synchronous motor, so that there is no need to add an additional signal for detection. There is.

[0008]

In the method of estimating the magnetic pole position from the difference between the current calculated by the control model and the motor current actually flowing, if the synchronous motor loses synchronization for some reason, the control becomes impossible. You have to solve the problem of becoming.

On the other hand, the technique described in Japanese Patent Application Laid-Open No. Hei 8-205578 can always detect the magnetic pole position of the synchronous motor with saliency, so that control cannot be disabled.

However, in the method of detecting the magnetic pole position of the synchronous motor with saliency, it is necessary to detect the correlation between the state of the motor current and the applied voltage every time the PWM signal changes.

That is, it is necessary to detect the state of the motor current at least six times in one cycle of the carrier wave and to grasp the state of the applied voltage. Therefore, there is a problem that the operation speed cannot keep up unless a high-performance controller is used.

A first object of the present invention is to provide a low-cost synchronous motor control device. A second object of the present invention is to provide a highly reliable control system.

[0013]

One of the solutions of the present invention is to calculate or estimate the magnetic pole position of the synchronous motor based on the amount or direction of change of the motor current when the synchronous motor is short-circuited. And controlling the synchronous motor based on the calculated magnetic pole position.

Another solution will be described in the description of the embodiments of the invention.

[0015]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS One embodiment of the present invention will be described below with reference to FIG.

FIG. 1 shows a cylindrical synchronous motor 1 with a battery 2
FIG. 2 is a configuration diagram of a control device for a motor driven by DC energy. The DC voltage of the battery 2 is converted into a three-phase AC voltage by the inverter 3 and applied to the cylindrical synchronous motor 1. The inverter 3 is controlled based on the output of the controller 4.

The output of the controller 4 is determined based on the result of the following operation. Here, the controller 4 is configured by a functional block diagram, but these can be realized not only by hardware but also by software. The current differentiating circuit 12 and the current detecting unit 10, which will be described below,
The PWM signal generator 9 partially uses an input / output circuit of a computer. For example, the input / output circuit is an analog / digital converter or a pulse output circuit, and by using these, all can be processed by a program.

First, the current command value generator 6 generates a d-axis current command value i for a torque command value τr to be generated by the motor.
The dr and q-axis current command values iqr are determined. Note that a command is issued to the current command value generator 6 from the control device or control program of the host 4 for the torque command value τr.

The d-axis indicates the direction of the magnetic pole position (magnetic flux), and the q-axis indicates the direction electrically orthogonal to the d-axis, forming a dq-axis coordinate system. When the rotor having the magnet rotates, d-
Since the q-axis coordinate system also rotates, the phase from the stationary coordinate system (α-β axis coordinate system) is set to θ. That is, the purpose of this embodiment is to detect the phase θ of the magnetic pole (hereinafter, referred to as the magnetic pole position θ) from the current.

FIG. 17 is a vector diagram showing an example of the relationship between the coordinate system and the current. If the d-axis current and the q-axis current can be controlled according to the command values, the synchronous motor 1 can generate a torque that matches the torque command value τr. The torque command value τr may be directly instructed, or may be instructed from a speed control arithmetic circuit (not shown). Further, signals indicating the values of the u-phase current iu and the v-phase current iv are sent from the current sensors 5a and 5b to the current detection unit 10, and the current detection unit 10 detects the signals at the timing of a detection pulse P1 described later. The detected current values are converted by the coordinate conversion unit 11 into d-axis current id and q-axis current iq in the dq axis coordinate system.

In this embodiment, the currents detected by the current detector 10 are two phase currents iu and iv of the U phase and the V phase, but the W phase current iw can be obtained from iu and iv. The detection of the phase current iw is omitted. As a matter of course, all three-phase currents may be detected.

In the current controller 7, the d-axis current command value idr
D-axis current deviation between d-axis current id and q-axis current command value iqr
And a q-axis current deviation of the q-axis current iq, and a d-axis voltage command value Vdr and a q-axis voltage command value Vqr are obtained by proportional / integral control calculation for each current deviation.

D-axis voltage command value Vdr, q-axis voltage command value V
The coordinate conversion unit 8 that inputs qr calculates three-phase voltage command values Vur, Vvr, and Vwr in the stationary coordinate system based on the magnetic pole position θ, and outputs the calculated values to the PWM signal generation unit 9.

According to the calculation in the PWM signal generator 9,
Three-phase PWM pulses Pup, Pvp, Pwp, Pun,
Pvn and Pwn are output to the inverter 3.

FIG. 2 shows the relationship between the connection method of the inverter 3 and the PWM pulse. For example, when Pup is high, the switching element Sup is on, and when Pup is low, the switching element Sup is off.

The PWM pulses Pup and Pun basically have a relationship between high and low. However, in order to prevent a power supply short circuit, when the PWM pulse is inverted,
In both cases, a short-circuit prevention period for providing a low state is provided.

The processing contents of the PWM signal generator 9 will be described with reference to a timing chart shown in FIG. By comparing the waveforms of the voltage command values Vur, Vvr, Vwr of each phase with the triangular carrier, the three-phase PWM pulses Pup,
Pvp and Pwp can be obtained. Note that the above-described short-circuit prevention period is omitted for simplicity of description.

That is, in FIG. 3, the PWM pulse Pu
When p, Pvp, and Pwp are high, the switching elements Sup, Svp, and Swp of the upper arm in FIG. 2 are turned on, and the switching elements Sun, Svn, and
Swn is turned off. PWM pulse Pu
When p, Pvp, and Pwp are low, the switching elements Sun, Svn, and Swn are turned on, and the switching elements Sup, Svp, and Swp are turned off.

As can be seen from FIG. 3, when the voltage command value of each phase is within a predetermined range including the minimum value and the maximum value of the carrier wave, the period during which the upper arm or the lower arm is in the three-phase short-circuit state There is. Here, if processing is performed such that the detection pulse P1 is generated when the carrier wave has the maximum value and the minimum value, the detection pulse P1 is generated when the synchronous motor is in a three-phase short-circuit state. become.

In the current detecting section 10, the pulse P
It is known that when the current of each phase is detected when 1 occurs, the instantaneous value of the current is substantially the average value of the current of the phase. The state where the phase winding of the synchronous motor is short-circuited is not only the moment of the minimum value and the maximum value of the carrier wave shown in FIG. The predetermined range is a range of the pulse having the narrowest width in the PWM pulses Pup, Pvp, and Pwp, and a time between the pulse next to the pulse having the widest pulse. At time t1, the range of Pvp pulse width, at time t2, between Pup and the next Pup, at time t3, range of Pvp pulse width, at time t4, between Pup and next Pup, and at time t5, Pwp pulse The range of the width is between Pup and the next Pup at time t6. The times t1 to t5 indicate the instants of the minimum value and the maximum value of the carrier. As described above, the short-circuit state of the phase winding occurs for a predetermined period including the moment of each minimum value and maximum value, and this is repeated. A pulse P1 is created to capture the current in the windings in the shorted state of the phase windings. This pulse P1 may be generated within the above-mentioned predetermined period. The method of generating at the minimum value and the maximum value of the carrier as in this embodiment has advantages such as easy generation of a pulse, or almost at the center of a short-circuit state and less possibility of malfunction.

Next, the features of the embodiment of FIG. 1 will be described.

Signals representing the U-phase current iu and the V-phase current iv are input to the current differentiating circuit 12, and the current values are differentiated.
(Or pseudo-differential) current differential value p which is a signal representing the value
iu and piv are output.

The values piu and piv are input to the detection unit 13, and are held and output when the detection pulse P1 is generated. That is, the current differential value piu or piv is detected at the timing of the pulse P1. That is, the operation unit 14
It is taken in.

The calculating section 14 calculates the magnetic pole position θ by performing the processing of the flowchart shown in FIG. 4 by the magnetic pole position calculating section 14 for calculating the magnetic pole position.

First, in step 101, the operation unit 14 receives the current differential values piu and piv at the time of short-circuiting three phases as inputs.
Take in.

In step 102, calculation is performed to find the phase γ of the current differential vector pi when the three phases are short-circuited.

FIG. 17 shows the phase relationship of the current differential vector pi. Current differential values piu, pi at the time of three-phase short circuit
From v, α-axis current differential value piα, β-axis current differential value piβ
Can be requested.

When the U-phase axis coincides with the α-axis, the following equation is obtained.

[0039]

Piα = (√3 / 2) piu (1)

[0040]

Piβ = (1 / √2) (piu−2piv) (2) Next, the phase γ is calculated from piα and piβ using the relationship shown in FIG.

In step 103, the magnetic pole position θ is obtained by the following equation.

[0042]

## EQU3 ## The present embodiment is a new finding that the relationship between the magnetic pole position θ and the phase γ of the three-phase short-circuit current is approximately expressed by Expression (3). It is a feature of. The reason will be described below. The basic equation of the synchronous motor can be expressed by the following equation in the dq axis coordinate system. Here, p = d / dt. Ω is the rotation speed of the motor.

[0043]

Vd = (R + pLd) id−ωLq iq (4)

[0044]

Vq = (R + pLq) iq + ω (Ld id + Φ) (5) When the synchronous motor is in a three-phase short-circuit state, the applied voltage of the synchronous motor is Vd = Vq = 0. It looks like this:

[0045]

Pid = (ωLq iq−R id) / Ld (6)

[0046]

## EQU00007 ## piq =-{. Omega. (Ld id + .PHI.) + R iq} / Lq (7) The current differential vector in the .alpha .-. Beta. Axis coordinate system of the stationary coordinate system is the current differential vector of the dq axis coordinate system and d-. This is the sum with the current differential vector generated when the q-axis coordinate system rotates at the speed ω. Therefore, the d-axis current differential value pids and the q-axis current differential value piqs viewed in the α-β axis coordinate system are respectively

[0047]

Pids = {ω (Lq−Ld) iq−R id} / Ld (8)

[0048]

[Mathematical formula-see original document] piqs =-{[omega] (Ld-Lq) id + [Phi]) + Riq} / Lq (9) Therefore, with respect to the d axis, that is, the magnetic pole position θ,
The phase δ of the three-phase short-circuit current differential vector is obtained by the following equation.

[0049]

Tan (δ) ≡piqs / pids = −Ld [ω ｛(Ld−Lq) id + Φ｝ + Riq] / [Lq ｛ω (Lq−Ld) iq−Rid}] (10) This embodiment In the case of the example, since it is a cylindrical synchronous motor, Ld = Lq
Is given,

[0050]

Tan (δ) = Ld (ωΦ + R iq) / (Lq R id) (11) Here, if id <0, the phase δ is approximated by the following equation.

[0051]

Δ ≒ −π / 2 (12) Therefore, the content of the calculation in step 103 is expressed by equation (3).

When the motor speed ω is low, the equation (12)
Can be obtained asymptotically by equation (11). This method will be described in another embodiment described later.

As described above, the calculation unit 14 in FIG. 1 can determine the magnetic pole position θ by a simple calculation. If the coordinate conversion of the coordinate conversion units 8 and 11 is performed using the magnetic pole position θ, it is possible to control the motor to generate a torque according to the required torque command value.

Therefore, according to the present embodiment, a magnetic pole position sensor such as a resolver or an encoder for directly measuring a mechanical rotational position is not used for a cylindrical synchronous motor, and only a current sensor is used. The magnetic pole position can be detected by easy calculation. Therefore, the control device is inexpensive.

Further, even when the synchronous motor loses synchronism for some reason, the magnetic pole position can be detected, so that the motor does not fall into the uncontrolled state.

Further, while performing the normal PWM control,
Since a sensorless control system can be configured using only information obtained when the PWM control is performed, it has a feature that noise and torque pulsation can be reduced as compared with the method of detecting the magnetic pole position by adding a detection additional signal.

FIG. 5 shows another embodiment, that is, a control device for a cylindrical synchronous motor capable of detecting a magnetic pole position without using a current differentiating circuit. This embodiment can be realized not only by an electric circuit but also by a program as in FIG.

The main differences from the embodiment shown in FIG. 1 are that the current differentiating circuit 12 is not used, that the current detection timing is changed by the detection pulse P2, and that the processing contents of the calculating unit 15 are the same as those of FIG. This is different from the section 14. It is important that this embodiment does not directly detect the three-phase short-circuit current.

First, the detection pulse P2 for controlling the detection timing of the current detector 10 will be described with reference to FIG. FIG. 6 shows the same state as the PWM signal of FIG. 3, but the detection pulse P2 of FIG. 6 is different from the detection pulse P1 of FIG. 3 in the following point.

In each phase of the 180-degree conducting three-phase inverter shown in FIG. 2, one of the upper-arm switching element and the lower-arm switching element is normally in an on state and the other is in an off state. Therefore, at least two of the three phases are always in a short-circuit state.

FIG. 6 shows the section. For example, in a section from time t (n-2) to time t (n-1), the switching elements Svn and Swn of the V-phase and W-phase lower arms are turned on, and the V-phase and W-phase of the synchronous motor 1 are turned on. The phase is short-circuited.

The section from time t (n-1) to time t (n) indicates that the upper arms of the U-phase and V-phase are in a short-circuit state.

As described above, in the 180-degree conduction type inverter, two modes of two-phase short-circuit exist during one cycle of the carrier wave.

As shown in FIG. 6, the detection pulse P2 is generated when the two-phase short-circuit mode is switched.

In the PWM signal generator 9, a change in the PWM signal in which a phase of the voltage command value having the second largest value among the three phase voltage command values, that is, an intermediate value (duration, ie, pulse width), is generated. Pulse P2 in synchronization with
Is performed.

The current detector 10 outputs two-phase current values output by the current sensors 5a and 5b each time the detection pulse P2 is generated, for example, signals iu and iv indicating the values of the U-phase current iu and the V-phase current iv. Take in.

The U-phase and V-phase currents obtained at this timing are input from the current detector 10 to the calculator 15,
Performs the processing shown in FIG. Here, the average value ua of the U-phase current and the average value i of the V-phase current selected by the calculation
va is input from the calculation unit 15 to the coordinate conversion unit 11. The calculated magnetic pole position θ is input from the calculation unit 15 to the coordinate conversion units 8 and 11, respectively. The coordinate transformation units 8 and 1 in FIG.
1 performs the same operations as those of FIG.

The flowchart of FIG. 7 showing the contents of the processing performed by the arithmetic unit 15 will be described. In step 111, the U-phase current iu (n) at time t (n) input from the current sensors 5a and 5b to the current detection unit 10 based on the detection pulse P2.
And V-phase current iv (n), U-phase average value iua
(n), the V-phase current average value iva (n) is calculated in step 112. If the average of the U-phase current iu (n-1) at time t (n-1) and the U-phase current iu (n) at time t (n) is calculated, it is almost equal to the U-phase current iu at time t5 in FIG. Have the same value. Since the U-phase current when the detection pulse P1 is generated is almost the average value, the process of step 112 is performed.

At the next step 113, time t (n-
A difference value, that is, a differential value, of the current of each phase between 1) and time t (n) is calculated.

Step 114 determines the two-phase short-circuit mode Msc indicating which phase is in the two-phase short-circuit state in the section from time t (n-1) to time t (n). in this case,
From FIG. 6, it can be seen that the upper arm has a U phase and a V phase.
This is determined in step 114, and the two-phase short-circuit mode Msc
(n) is "UV phase short circuit". Note that the previous time t
The two-phase short-circuit mode Msc (n-1) in the section from (n-2) to time t (n-1) is "V-W phase short-circuit".

In step 115, the short-circuit current difference value is calculated using the arithmetic expression shown in the table of FIG.

The short-circuit current difference value “pisc” of the short-circuit axis will be described. In FIG. 18, the short-circuit axis refers to the β-axis when the VW phase is short-circuited, the β ′ axis when the WU phase is short-circuited, and the β ″ axis when the UV-phase short-circuited.

For example, when converting a three-phase voltage to an α-β axis coordinate system (α axis coincides with U-phase axis), the β-axis voltage Vβ
Is represented by the following equation.

[0074]

Vβ = (Vv−Vw) / (√2) (13) Here, if the VW phase is short-circuited, Vβ = 0 because Vv = Vw. That is, it can be said that the β axis is in a short-circuit state, and this axis is referred to as a short-circuit axis. Similarly, WU
At the time of phase short-circuit, the β 'axis rotated 120 degrees from the β axis
In the case of the −V phase short-circuit, the β ″ axes rotated 240 degrees from the β-axis become the short-circuit axes.

In the case of a cylindrical synchronous motor, the short-circuit current difference value picc of the short-circuit axis is a three-phase short-circuit current differential vector pi.
s coincides with the short-circuit axis component. FIG. 18 shows the relationship between the vector diagrams.

The reason why the vector diagram of FIG. 18 is established will be described by expanding equations (4) and (5).

Α-axis current differential value piα, β-axis current differential value p
iβ is given by the following equation from equations (4) and (5).

[0078]

Piα = [(L0−L1cos2θ) Vα− (L1sin2θ) Vβ + k1 (θ) iα + k2 (θ) iβ + k3 (θ) φ] / (L0 ＾ 2−L1 ＾ 2) (14)

[0079]

Piβ = [− (L1 sin2θ) Vα + (L0 + L1cos2θ) Vβ + k4 (θ) iα + k5 (θ) iβ + k6 (θ) φ] / (L0 ＾ 2−L1 ＾ 2) (15) where L0 = (Ld + Lq) ) / 2, L1 = (Ld-L
q) / 2, k1 (θ), k2 (θ), k3 (θ), k4 (θ),
k5 (θ) and k6 (θ) are functions relating to θ.

In the case of the cylindrical synchronous motor, since L1 = 0, it can be seen that the β-axis current differential value piβ does not affect the α-axis voltage Vα.

In the V-W phase short-circuit state, only the α-axis voltage Vα is applied according to the state of the U-phase voltage Vu, but the β-axis current differential value piβ is the same as when Vα = 0. Moreover, since Vβ = 0 because of the VW phase short-circuit state, the β-axis current differential value p in the three-phase short-circuit state
It means that it matches iβ. From the above,
It can be seen that FIG. 18 holds.

Similarly, when the WU phase is short-circuited, β ′
The shaft current differential value piβ 'is a three-phase short-circuit current differential vector pi
It becomes the same as the β 'axis component of s. Therefore, when the current differential value (difference value) of the short-circuit axis in the two-phase short-circuit state is detected, the phase γ of the three-phase short-circuit current differential vector can be calculated by calculating the vector diagram of FIG.

When the phase γ of the three-phase short-circuit current differential vector is obtained from the current two-phase short-circuit mode Msc (n) and the previous two-phase short-circuit mode Msc (n−1), the calculation is performed by combining the short-circuit modes. The method is different.

For this reason, in step 116, the phase γ of the three-phase short-circuit current differential vector is obtained by using the arithmetic expressions divided into the modes as shown in FIG.

In step 117, the magnetic pole position θ can be obtained in the same manner as in step 103 in FIG.

As described above, according to the present embodiment, the direction of the current differential vector in the three-phase short-circuit state can be determined or calculated from the change amount of the current in the two-phase short-circuit state having a relatively long duration, that is, the difference value. In addition, there is an effect that highly accurate magnetic pole position detection can be obtained by taking in a small amount of current.

Further, since the system of this embodiment does not use a differentiating circuit, it has an advantage that it is resistant to noise and can be realized by a relatively inexpensive controller.

The embodiment of FIG. 9 is a block diagram of a control device when the present invention is applied to the salient-pole synchronous motor 16. The controller 4 can be realized by an electric circuit or software as in the previous embodiment. In contrast to the embodiment of FIG.
In the embodiment, the two-phase switching operation unit 18 is used,
The use of the detection pulses P3 and P4 from the PWM signal generator 9 and the processing method of the calculator 17 are different.

The processing contents of the two-phase switching operation section 18 will be described with reference to the time chart of FIG.

Two-phase switching refers to a method of flowing the same sine wave current as in three-phase switching while stopping one-phase switching among three-phase PWM signals.

Referring to FIG. 10, for example, a method of flowing the same sine wave current as in three-phase switching while stopping the U-phase switching will be described. A V-phase or a W-phase other than the U-phase can be a phase in which switching is stopped based on the same concept.
The additional voltage V0 is forcibly added so that the U-phase voltage command value Vur always becomes the same value as the maximum value of the carrier. As a result, the U-phase PWM signal Pup is always in the high state, and the switching element Sup is in the on state.

V-phase voltage command value Vvr, W-phase voltage command value V
wr is calculated as a value obtained by adding the additional voltage V0 to the normal command value, thereby obtaining the PWM signals Pvp and Pwp.
Has occurred.

Since the line voltage is not affected even if the same voltage is added to all the phases, the current flowing through the synchronous motor 16 is the same as the current when the additional voltage V0 is not applied. This is two-phase switching, a well-known method. When this method is used, it can be seen that the three-phase short-circuit state per operation shown in FIG. 10 is longer than in the case of FIG.

FIG. 10 also shows detection pulses P3 and P4 generated from the PWM signal generator 9.

The detection pulse P3 is generated in synchronization with the maximum value of the carrier wave.
Are used to obtain current average values iua and iva for each phase.

The current detection pulse P4 is generated at the start and end of the extended three-phase short-circuit state. In the current detection unit 27 of FIG.
4 inputs the U-phase current iu and the V-phase current iv.

[0097] These current values are input to the arithmetic unit 17,
The processing shown in the flowchart of FIG.
Is calculated.

The processing method shown in FIG. 11 is performed as follows. In step 121, the U-phase current iu (n-1) and the V-phase current iv (n) at the start time t (n-1) of the three-phase short-circuit state
−1) and the U-phase current iu (n) and the V-phase current iv (n) at the end time t (n), the current difference values piu, pi of each phase.
Calculate v and piw. The processing method is the same as step 113 in FIG.

In the next step 122, the current difference value pi
The phase γ of the differential vector of the three-phase short-circuit current is calculated using u, piv, and piw. This processing corresponds to step 1 in FIG.
Same as 02.

In the following method, the magnetic pole position used for control at that time in the controller 4 is θ ′, and the synchronous motor 1
6 is assumed to be θ. Controller 4
The d-axis current and the q-axis current calculated based on the magnetic pole position θ ′ are id ′ and iq ′, and the actual d-axis current of the synchronous motor 16 is
The description will be made assuming that the q-axis currents are id and iq, respectively.

In step 123, the d-axis current id 'and the q-axis current iq' are calculated using the magnetic pole position θ 'and the current average values iua and iva input from the current detector 10.

In step 124, the expression (10) is calculated by using id 'and iq' instead of id and iq, and the phase δ from the magnetic pole position (d-axis) to the three-phase short-circuit current differential vector is calculated. Ask.

When the motor speed ω is equal to or higher than a predetermined value, it may be obtained by the following approximate expression.

[0104]

Tan (δ) ≒ −Ld {(Ld−Lq) id + Φ} / {Lq (Lq−Ld) iq} (16) In step 125, the phase γ obtained in step 122 is obtained.
Is used to determine the magnetic pole position θ by the following equation.

[0105]

(17) θ = γ−δ (17) This relationship is shown in the vector diagram of FIG.

In step 126, it is determined whether or not the magnetic pole position θ obtained in step 125 substantially coincides with the magnetic pole position θ ′ used for obtaining id ′ and iq ′ in step 123. If they do not match, the processing from step 123 to step 125 is performed again to calculate the magnetic pole position θ.

If the actual magnetic pole position θ is different from the magnetic pole position θ ′ in the controller, id ′ and iq ′ do not match id and iq, so that an error occurs in the phase δ. However, the error decreases each time the processing from step 123 to step 125 is performed, and the magnetic pole position θ ′ in the controller converges to the true magnetic pole position θ. Step 12
When it is determined in step 6 that the calculation of the magnetic pole position θ has almost converged, the calculation is terminated.

Since this operation is expected to converge within a few degrees within a few times, the convergence is determined by the magnetic pole position θ.
Instead of the calculation result, the calculation may be terminated by the number of calculations, for example, two.

Further, depending on the relationship between the sampling time for detecting the magnetic pole position and the motor speed, step 126 is executed.
May be omitted, and a method of detecting the magnetic pole position by sampling several times may be adopted.

As described above, when detecting the magnetic pole position of the salient-pole synchronous motor, the d-axis current id ', q
Although the calculation needs to be performed using the axis current iq ', the present embodiment is characterized in that an algorithm is provided which can converge the calculation. Therefore, there is an advantage that a sensorless control system for a salient-pole synchronous motor can be constructed by utilizing a change in current in a three-phase short-circuit state.

In the present system, by using together a method of extending the three-phase short-circuit time, such as a two-phase switching method, the width of change in current during the three-phase short-circuit period can be increased. Therefore, a three-phase short-circuit current differential vector can be directly measured without using a differentiating circuit, and a magnetic pole position detection method that is resistant to noise can be realized by simple software processing.
FIG. 12 shows an embodiment of a salient pole synchronous motor for detecting a magnetic pole position from a two-phase short-circuit state, and shows a configuration of a highly reliable system applied to an electric vehicle. In this embodiment, the controller 4 can be realized not only by an electric circuit but also by software. As in the previous embodiment, each block of the controller 4 represents a software processing function.

Compared to the case of the cylindrical synchronous motor shown in FIG.
12 is different from the processing contents of the arithmetic unit 20 in that the salient-pole synchronous motor 16 uses the tires 24,
25, and a magnetic pole position sensor 23 that directly detects the magnetic pole position of the motor 16 mechanically for the purpose of improving the reliability of the electric vehicle.

First, the processing contents of the arithmetic unit 20 will be described.
FIG. 13 shows a flowchart of this processing content.

The processing from step 131 to step 134 is the same as the processing from step 111 to step 114 in FIG.

The saliency correction phase ε in step 135 is a correction amount necessary for considering the effect of the saliency of the synchronous motor 16.

As shown in equation (15), in the case of the salient-pole synchronous motor 16, since L1 ≠ 0, the β-axis current differential value pi
β changes depending on the α-axis voltage Vα. Therefore, the value becomes different from the β-axis component of the three-phase short-circuit current differential vector.

FIG. 19 shows an α-axis current differential value piα1 generated by the α-axis voltage Vα, a β-axis current differential value piβ1, and
The current differential vector pi1 that is the composite is shown.

Assuming that the axis in the direction coinciding with the current differential vector pi1 is the x-axis and the axis orthogonal thereto is the y-axis, the y-axis component of the current differential vector pi1 is always 0 regardless of the α-axis voltage Vα. . Therefore, the y-axis component of pi1 matches the y-axis component of the three-phase short-circuit current differential vector pi. This is called a saliency correction phase ε.

Therefore, in the case of the salient pole type synchronous motor, the current differential value (difference value) of the y-axis advanced not by the β-axis but by the salient polarity correction phase ε is detected.

Since there are actually three two-phase short-circuit conditions,
The saliency correction phases in the case of VW phase short-circuit, WU phase short-circuit, and UV phase short-circuit are ε1, ε2, ε3, respectively, and the axes in that direction are the y ′ axis and the y ″ axis.

The saliency correction phases ε1, ε2, ε3 are given by the formula (1)
4) and Expression (15) are as follows.

[0122]

Tan (ε1) = − (L1 sin2θ) / (L0−L1cos2θ) (18)

[0123]

Tan (ε2) = − {L1 sin (2θ−4π / 3)} / {L0−L1cos (2θ−4π / 3)} (19)

[0124]

Tan (ε3) = − {L1sin (2θ−2π / 3)} / {L0−L1cos (2θ−2π / 3)} (20) From the above, in step 135, the two-phase short-circuit state occurs. Is calculated according to the equation (18), the equation (19), or the equation (20) to obtain the saliency correction phase.

The magnetic pole position θ used in these calculations is a value in the controller 4 and includes an error, but it is also possible to obtain an accurate magnetic pole position while converging as shown in FIG.

In step 136, the current difference value p
A short-circuit current difference value of the short-circuit axis (any of the y-axis, y'-axis, and y "-axis) corrected from iu (n) and piv (n) using the table of FIG. Difference value picc
Is calculated.

As described above, the short-circuit axis is an axis in a direction that is not affected by the current differential value (difference value) by the α-axis voltage.

In the next step 137, as shown in FIG. 14, the mode of calculation based on the current and previous two-phase short-circuit state is changed, and the phase γ of the three-phase short-circuit current differential vector is calculated using the arithmetic expression of FIG. Get.

An example of the vector diagram at this time is shown in FIG. 20, and this relationship is obtained by the operation formula in FIG.

The processing from step 138 to step 140 is the same as the processing from step 123 to step 125 in FIG. 11, and takes into account the phase from the magnetic pole position to the current differential vector in the salient-pole synchronous motor 16.

As described above, if the arithmetic unit 20 is used, the magnetic pole position can be detected for the salient-pole synchronous motor 16 only by detecting the current in the two-phase short-circuit state.

In the electric vehicle drive system shown in FIG. 12, a signal from the magnetic pole position sensor 23 is input to the magnetic pole position detecting section 21 to detect the magnetic pole position θ1.

The magnetic pole position abnormality detecting section 22 receives the magnetic pole position θ1 from the magnetic pole position detecting section 21 and the magnetic pole position θ from the calculating section 20, and performs the processing shown in FIG.

The magnetic pole positions θ and θ input in step 141
In step 142, it is determined whether the difference is within a predetermined normal range.

If it is determined that it is normal, step 143 is executed.
Stores the magnetic pole position θ1 in the output magnetic pole position θ2, and outputs the θ2 to the coordinate conversion units 8 and 11 in step 144. If it is determined in step 142 that the two magnetic pole positions are not in the normal range by comparing the two magnetic pole positions, step 145 is performed.
To temporarily stop the electric vehicle.

At step 146, it is determined whether or not the rotation of the synchronous motor 16 has stopped. If it is determined that the vehicle has stopped, a process is performed in step 147 so that the vehicle can travel within the safe speed using the normal magnetic pole position. Running within the safe speed means that the upper limit of the vehicle speed is 40 km / h or 50 km / h
Is controlled by a control device (not shown) so as to run at a speed lower than this speed, and the vehicle runs under this control.

As described above, according to this embodiment, by providing the normal magnetic pole position sensor 23, the magnetic pole position sensor 2 can be used.
Since the magnetic pole position is obtained based on the output of No. 3 and the magnetic pole position is obtained separately from the motor current by arithmetic processing, a highly reliable electric vehicle can be provided.

This embodiment is particularly suitable for reducing the size of the motor and the weight of the electric vehicle by using the reluctance torque.

FIG. 16 shows an embodiment of a magnetic pole position sensorless control system having a function of detecting a magnetic pole position only by a current sensor and performing a self-diagnosis on whether there is any abnormality in its characteristics.

In contrast to the embodiment shown in FIG.
The feature of the embodiment shown in FIG.

The arithmetic section 20 performs an arithmetic operation for detecting the motor speed ω in addition to the processing shown in FIG.

In the above-described embodiment, since the magnetic pole position θ is obtained by detecting the phase γ of the current differential vector, information on the magnitude of the current differential vector is ignored.

Therefore, the motor speed ω is determined in reverse from the equations (6) and (7). That is,

[0144]

Ω = (Ld pid + R id) / Lq iq (21)

[0145]

Ω = − (Lqpiq + R iq) / (Ld id + Φ) (22) The motor speed ω is calculated using one or both of the expressions.

It should be noted that in equations (21) and (22),
A simple formula that ignores the resistance R may be used. The motor speed ω is output to the self-diagnosis unit 26.

The magnetic pole position θ obtained in step 140 of FIG.

The self-diagnosis unit 26 determines whether there is any abnormality by comparing the change state of the magnetic pole position θ with the motor speed ω.

If it is determined that there is an abnormality, an abnormality diagnosis signal Se is output, and systematic processing is performed so as to stop the sensorless control system.

As described above, by estimating the motor speed using a plurality of independent variables of the current differential vector, a self-diagnosis function can be provided without using another sensor.

The above is the embodiment in which the magnetic pole position of the synchronous motor is detected using only the current sensor.

In addition to the synchronous motor, the present invention can be applied to a reluctance motor utilizing saliency.

In this embodiment, the description is omitted for the sake of simplicity. However, even if the magnetic pole position is calculated in consideration of the influence of the rotation of the motor rotor during the sampling time, the present invention is not limited to this. It goes without saying that the embodiment can be applied.

Although the example of the electric vehicle has been described here, the present embodiment is also used for a magnet motor that is currently subjected to sensorless control using 120-degree conduction type inverter control. Inverter control can provide a sensorless system with low torque pulsation and low noise.

The embodiment described with reference to FIGS. 1 to 20 is most suitable for control in a state where the motor 1 is rotating at a predetermined rotation speed or higher. Here, the predetermined rotation speed or more is, for example, 800 rotations / minute or more. In a low-speed state including a stopped state of the motor, it may be necessary to detect the magnetic pole position with higher accuracy for a reason to be described later. An embodiment in which the magnetic pole position in the low-speed state including the stop state of the motor can be performed with high accuracy will be described with reference to FIG. A major difference between FIG. 1 and FIG. 21 is that a calculation unit 52 for detecting the magnetic pole position at the time of low-speed rotation of the motor including the stopped state is provided. The controller 4 can be realized not only by an electric circuit but also by software as in the other embodiments. When implemented by software, each block of the controller 4 represents a processing function of the software.

The processing for detecting the magnetic pole position in the arithmetic unit 14 is as described with reference to FIG. The operations of the current command value generator 6, current controller 7, coordinate converter 8, PWM signal generator 9, current detector 10, coordinate converter 11, and inverter 3 are basically the same as those in FIG. The processing method of calculating the magnetic pole position of the motor based on the change amount or the change direction of the motor current when the synchronous motor is in the three-phase short-circuit state has been described, but will be described again.

In this method of calculating the magnetic pole position, the magnetic pole position is calculated based on the change amount or the change direction of the motor current in the three-phase short-circuit state which is not affected by the voltage applied from the inverter. In order to detect the amount or direction of change of the motor current in the three-phase short-circuit state, as described in FIG.
The pulse P1 is generated to control the timing in the three-phase short-circuit state. As a typical example, a pulse P1 is generated at the highest value and / or the lowest value of the carrier wave shown in FIG. 3, and a current value or a differential value of the current value is detected in synchronization with the pulse P1. That is, a pulse P1 is generated by the PWM signal generator 9, and the pulse P1 is used as a signal indicating a three-phase short-circuit state of the motor, and the pulse P1 is used as a trigger to detect a change amount of the motor current. The relationship between the vector of the motor current change amount at the time of the three-phase short circuit detected at this time, that is, the differential vector is as shown in FIG. 17 described above.

In FIG. 17, the magnetic pole position to be detected is the phase θ between the stationary coordinate α axis and the rotating coordinate d axis, and can be expressed by the above equation (17).

[0159]

(17) where γ is the motor current differential vector pI at the time of three-phase short circuit.
s is the phase with respect to the α-axis, and δ is the phase of the motor current differential vector pIs with respect to the d-axis. As for the phase γ, first, the motor currents iu and iv are differentiated by the current differentiating circuit 12 shown in FIG. 1, and the current differential value detecting unit 13 detects the short-circuited state in synchronism with the motor short-circuit detection pulse P1. The differential values piu and piv of the motor current are captured. Further, the differential value pi of the motor current at the time of this short circuit
For u and piv, the current differential values piα and piβ on the α-β axis are calculated according to the equations (1) and (2), and the phase γ is calculated based on the equation (23).

[0160]

Piα = (√3 / 2) piu (1)

[0161]

Piβ = (1 / √2) (piu + 2piv) (2)

[0162]

Γ = tan ⁻¹ (piβ / piα) (23) Here, piu, p is used to obtain piα, piβ.
iv, two phases of iv, piu, piv, piw
The calculation can also be performed using the differential value of the three-phase current. Furthermore, although a differentiating circuit is used to calculate the current differential value at the time of the three-phase short circuit, if the configuration of the differentiating circuit is not possible, instead of the current differential value, the amount of current change in the three-phase short circuit section is calculated. It can also be realized by using a current change rate obtained by calculating and dividing a current change amount by a short-circuit time. In addition, if the short-circuit interval is very short and it is not possible to calculate the current change rate in the three-phase short-circuit interval, the three-phase short-circuit is used even if a two-phase short-circuit interval is used that is longer than the three-phase short-circuit interval. It is possible to calculate the current change rate in the section. Details are as described above.

As described above, the phase γ of the motor current differential vector pIs with respect to the α-axis when the three phases are short-circuited is obtained. Next, the phase δ of the motor current differential vector pIs with respect to the d-axis is obtained as follows. First, the rotation coordinate d−
The basic equation of the synchronous motor on the q-axis can be expressed by the following equation. Note that these equations are as described above.

[0164]

Vd = (R + pLd) id−ωLq iq (4)

[0165]

Vq = (R + pLq) iq + ω (Ld id + Φ) (5) where Vd and Vq are dq axis voltages, and Ld and Lq are d−
The q-axis inductance, R is the winding resistance, ω is the motor angular velocity, φ is the field main magnetic flux, and p is d / dt. In the above basic formula, when a three-phase short circuit occurs, the applied voltage on the dq axis is 0, so the basic formula in the three-phase short circuit state is as follows.

[0166]

Pid = (ωLq iq−R id) / Ld (6)

[0167]

[Mathematical formula-see original document] piq =-{[omega] (Ld id + [Phi]) + R iq} / Lq (7) The current differential vector on the stationary coordinate [alpha]-[beta] axis is the rotational coordinate d represented by Expressions (6) and (7) It is represented by the sum of a current differential vector on the −q axis and a current differential vector generated when the dq axis rotates at the motor angular velocity ω. Therefore, the current differential value on the dq axis viewed on the α-β axis is as follows.

[0168]

Pids = {ω (Lq−Ld) iq−R id} / Ld (8)

[0169]

Piqs = − {ω (Ld−Lq) id + Φ) + R iq} / Lq (9) Accordingly, the phase δ of the motor current differential vector pIs with respect to the d-axis is represented by Expression (24).

[0170]

Δ = tan ⁻¹ (piqs / pids) = tan ⁻¹ [−Ld ｛ω ((Ld−Lq) id + φ) + Riq} / {Lq (ω (Lq−Ld) iq−Rid)}] (24) Therefore, the magnetic pole position can be obtained from the above equations (17), (23) and (24). Here, equation (24)
Is used as the estimated value of the angular velocity obtained from the change amount of the phase estimated value. Further, in a region where the angular velocity is sufficiently large and the R component can be ignored, the influence of ω is eliminated.

The above is an outline of the position detection method for detecting the magnetic pole position of the synchronous motor based on the amount of change or the direction of change of the motor current when the synchronous motor is short-circuited in three phases. This method is applicable not only to a salient pole type synchronous motor but also to a cylindrical type synchronous motor.

In the above-described position detection method, since the amount of current change in the motor short-circuit state generated during normal PWM operation is used, the magnetic pole position can be estimated up to a high-speed region without applying a special estimation signal.

However, in the following operation range, there is a possibility that the position detection accuracy is reduced only by this detection method. FIG. 22 shows the operation range. When the speed is zero, that is, when the motor 1 stops rotating, the torque of the motor 1 is also zero. In a motor rotation stop state, for example, in a state in which the motor is to be started, the motor current has not yet flowed, so that even if an attempt is made to perform the PWM control, the amount of change in the current at the time of short circuit cannot be detected. Furthermore, even after startup, when the rotation speed is low and the torque is small, the current value is small, and the above measurement is difficult. The detection accuracy of the magnetic pole position decreases. Furthermore, in the area where current flows near zero speed,
Although the direction of the magnetic flux can be detected based on the amount of change in the current at the time of the short circuit, the influence of the induced voltage is very small.

In the above operating range, for example, at 800 rpm (rotation / minute), the present embodiment can improve the detection accuracy of the magnetic pole position by newly providing the detecting section 52. The detection unit 52 will be described with reference to FIG. Detector 5
2 basically has a signal generator 54 for generating a magnet position estimation signal and a polarity discriminator 56.

The signal generator 54 for generating the magnet position estimation signal detects a magnetic pole position in a region where no current flows or a region where the current value is small. A current command value idhr for position detection is generated so that a current of an amount that can be detected flows, and the current command value idhr is given to the current command value generator 6.

FIG. 24 shows the configuration of the current command value generator 6. In the embodiments of FIGS. 1 to 20 and the control system of FIG. 21, the torque command Tr is input to the current command value generator 6 from the upper control unit of the controller 4. The current command value generator 6 includes a torque controller 63, and calculates current commands idr and iqr for the dq axes so that the synchronous motor 1 generates a torque as instructed. Normally, a torque command and a motor speed (not shown in FIG. 24) are input, and idr and iqr that maximize the efficiency at the current operating point are calculated using a map search or the like. Here, “normal” means, for example, that the synchronous motor 1 has a predetermined rotation speed of 800 rotations / sec.
It is rotating at a speed faster than a minute.

As described above, the current command idhr for detecting the magnetic pole position at the time of low-speed rotation of 800 revolutions / minute or less, such as when the synchronous motor is started, is output from the signal generator 54 and applied to the adder 65. Is done. This current command id
The hr is added to the d-axis direction current command idr1 by the adder 65, and the addition result is added from the current command value generator 6 to the current controller 7 as a d-axis current command idr. In this embodiment, the current command idhr for detecting the magnetic pole position is applied to the current command in the d-axis direction. This is to prevent unnecessary torque generation due to application of idhr. That is, if the current in the q-axis direction is zero, basically no torque is generated even if a current is applied to the d-axis. However, the magnetic pole position can be detected even when idhr is applied in the q-axis direction or when both d and q axes are applied. Therefore, if the generation of torque is permissible in the entire control, it can be added to the current command in the q-axis direction for detecting the magnet position. Further, it may be added only for a short time corresponding to the measurement.

Further, a current command i for position estimation to be applied is
dhr can detect the position using either an AC signal or a DC signal. When the current command idhr is an AC signal, the average value of the torque generated due to the difference between the position detection value generated at the initial stage of the estimation operation and the magnetic pole position of the motor is zero.
Therefore, the influence of torque fluctuation can be suppressed.

As described above, when the motor speed is zero,
At an operating point where no current is flowing, for example, at the time of starting to start rotation from a stopped state, by applying a signal for position estimation, it is possible to detect a position using a current change amount when a motor is short-circuited. Become. However, although the detected position is in the magnetic flux direction, it is unknown whether it is in the N-pole direction or the S-pole direction. This is because the motor is not affected by the induced voltage because the angular velocity ω of the motor is zero.

In order to solve this problem, in the embodiment shown in FIG. 23, a polarity discriminator 56 is provided to discriminate whether the calculated position is the N pole or the S pole, that is, the polarity of the magnetic pole. The discrimination method used in the polarity discrimination unit 56 is not particularly limited, but a method using the magnetic saturation characteristics of the motor is effective. Hereinafter, a method using magnetic saturation characteristics will be described as an example of a polarity determination method.

As for the magnetic characteristics of the synchronous motor, since the rotor has a magnetic flux due to the permanent magnet, a magnetic flux exists even when the current id of the d-axis which is the magnetic flux axis is zero as shown in FIG. Due to this magnetic characteristic, the characteristic of the d-axis inductance Ld is as shown in FIG. From FIG. 26, it can be seen that there is a region where the magnitude of the inductance Ld differs due to the difference between the positive and negative of the current id on the d-axis, and a region indicated by oblique lines in FIG.
Therefore, if an AC signal having a bias component such that id flows in the shaded region in FIG. 26 is applied to the d-axis, the response of the current control system changes depending on the magnitude of the inductance Ld. The difference in the polarity appears in the difference in the amplitude of the motor current, and the polarity of the magnetic pole can be determined by measuring the magnitude of the amplitude of the motor current. The above is an example of the polarity determination method.

Further, the procedure of this polarity discrimination method is shown in FIG.
Is expressed as follows. First, an AC signal having a DC bias component is applied to a d-axis current command from a signal generator 54 that generates a signal for detecting a magnetic pole position. Next, the detected d-axis current id is input to the polarity discrimination unit 56, and the polarity discrimination unit 56 measures the amplitude value of the AC component of the id, and discriminates the polarity of the current position set value θ ＾.

If the determined result is the N pole, the position set value θ ＾ of the control system is used for control as it is.
On the other hand, if the determined result is the south pole, the position set value θ ＾ is added or subtracted by 180 ° to obtain the position set value θ.
＾ is corrected to the N pole. In this polarity discrimination method, current flows in the d-axis direction until magnetic saturation occurs in the motor. Therefore, if there is any error in the position set value θ ＾, torque in a certain direction is applied to the motor. Occurs. Therefore, the magnetic pole position detection operation at the time of starting is provided with a lock mechanism that mechanically temporarily prevents rotation of the rotation shaft of the motor or the rotor so that the shaft of the motor is not rotated by this torque, and the rotation of the rotation shaft is provided. May be started in a state where is blocked. For example, in FIG.
A lock mechanism 74 for preventing the rotation of the rotor 70 or the rotating shaft 72 is provided. The lock mechanism 74 is the same mechanism as a normal brake mechanism, and temporarily stops rotation by a signal from the detection unit 52. This mechanism works when the motor starts to rotate, and since the torque is small, the purpose can be achieved even with a simple mechanism. When the detection of the magnetic pole position and the polarity for starting the rotation is completed, the lock mechanism releases the inhibition of the rotation by a signal from the detection unit 52.

The above is an explanation of an example of the polarity discrimination method.

Next, the processing procedure at the time of starting the motor by the magnetic pole position detection unit at zero speed described above will be described with reference to the flowchart of FIG. The letter S in FIG.
Means a step or procedure. First, in step 30 of FIG. 27, it is determined whether or not the rotation of the rotating shaft of the motor is blocked by the lock mechanism. As a result, if the rotation of the rotating shaft is prevented, the signal idhr for detecting the magnet position is supplied to the polarity discriminator 56 in step 31.
And is applied to an adder 65 for performing an addition operation with the current command idrl of the current command value generating section 6.

The above description is for the case where there is a lock mechanism that prevents the rotation of the motor when the motor is started, that is, when the motor starts rotating. If there is no such mechanism, step 30 is unnecessary. If there is a lock mechanism and the lock mechanism is not operating, an instruction to operate the lock mechanism is issued in step 30 or the operation is awaited by an instruction from another control unit. Then, in the state where the lock mechanism operates, step 3
The processing shifts to 1.

In step 32, the short-circuit current generated during the PWM control is detected, and the detection value θ ＾ is calculated by the above-described magnetic pole position detection method based on the variation of the short-circuit current.
Furthermore, in step 33, the detected value θ ＾ of the magnetic pole position obtained in step 32 is in the N-pole direction, or
It is determined whether the direction is polar. As a result of this determination, when it is determined that the number is the north pole, the detected value θ is determined in step 34.
＾ is the current magnetic pole position. On the other hand, if it is determined in step 33 that the pole is an S pole,
, The detection value θ ＾ is corrected by 180 ° addition or subtraction, and the current magnetic pole position is obtained. Next, in step 36, the drive control of the motor is started using the obtained detected value of the magnetic pole position.

The above is the processing procedure at the time of starting the motor using the detection method of the magnetic pole position where the motor rotation is zero speed. When the detection accuracy is low only by the magnetic pole position detection method based on the change amount of the short-circuit current of the motor, the magnetic pole position can be detected with high accuracy by performing the above processing. In this embodiment, the rotation axis of the motor is locked as a condition for starting the magnetic pole position detection operation. However, the magnetic pole position can be detected even when the rotation axis is not locked as described above. .

The operation of the arithmetic unit 52 when the rotation speed of the motor is zero and the motor current is flowing will be described. In such an operating environment, when the rotation is stopped, the load torque, that is, the torque of the motor required for driving the device driven by the motor is larger than the torque generated by the motor, and the magnetic pole position in the controller is affected by noise and the like. Is deleted (it is no longer accurate). In such a case, since the motor current is flowing, the magnetic pole position can be detected based on the change amount of the motor short-circuit current described above. However, since the motor speed is zero, it is necessary to determine the polarity. In this case, since the rotation axis of the motor is not locked and the motor is being driven, the polarity determination method using the magnetic saturation characteristic described above is often more desirable than the polarity determination method described below.

The method described here is a method of determining from the direction in which the motor torque is generated and the rotation direction of the motor shaft. The operation of this determination method, that is, the processing content will be described with reference to the flowchart of FIG. First, in step 40, a desired torque is generated using the detected value θ ＾ obtained from the amount of change in the short-circuit current of the motor as the set value of the magnetic pole position. Thereafter, the rotational direction of the motor shaft and the direction of the generated torque are referred to in step 41, and if both directions match, in step 42, the set value of the magnetic pole position is set as the current magnetic pole position. On the other hand, if the rotation direction of the motor shaft does not match the direction of the generated torque in step 41, it is determined that the polarities are opposite. In step 43, the set value is corrected by 180 ° and the current magnetic pole is corrected. Position. Further, at step 44, the torque command is increased. The above operation is repeated a predetermined number of times,
When the rotation direction of the motor shaft and the direction of the generated torque coincide with each other continuously for a set number of times or more, the polarity determination is terminated. The above is the processing procedure of the method of determining the polarity of the set value of the magnetic pole position from the torque generation direction and the rotation direction of the rotating shaft of the motor. In this method, it is not necessary to apply a special polarity discrimination signal.

The magnetic pole position detecting method using the magnetic pole position detecting method using the amount of change in the short-circuit current of the motor has been described above. However, in this method, the induced voltage is very small. Applicable even at extremely low speeds. For example, the effect is large at a rotation of 800 rotations / minute or less.

Further, the present method can be applied to drive devices for electric vehicles such as electric vehicles and hybrid vehicles having both a motor and an engine. In an electric vehicle or a hybrid vehicle drive device, when a position detection signal is applied to perform detection or when a polarity determination method using magnetic saturation characteristics is used, a brake is applied or an operating range is applied. If the vehicle is in the parking range, vibration and movement of the vehicle due to unnecessary torque generation can be avoided.

According to the above-described embodiment, the magnetic pole position can be detected in the entire operation range of the synchronous motor without being affected by the state of the applied voltage and while performing ordinary PWM control with an inexpensive controller.

Further, according to the embodiments shown in FIGS. 21 to 28, accurate control can be performed even when the motor is stopped.

Further, in the embodiment shown in FIGS. 1 to 28, the magnetic pole position of the synchronous motor is obtained from the change amount or the change direction of the motor current, so that ordinary PWM control can be performed without providing a position detector. The magnetic pole position can be detected.

Further, a magnetic pole detector is provided, and by comparing the magnetic pole position detected by the magnetic pole detector with the magnetic pole position obtained from the motor current, abnormality of the magnetic pole position detector is detected while performing normal PWM control. can do.

[0197]

According to the present invention, a synchronous motor drive system having excellent controllability can be provided at low cost.

When a normal magnetic pole position sensor is used, an abnormality of the sensor can be detected, so that a highly reliable synchronous motor drive system can be provided.

[Brief description of the drawings]

FIG. 1 is a configuration diagram showing an embodiment when the present invention for detecting a magnetic pole position of a cylindrical synchronous motor using a current differentiating circuit is applied.

FIG. 2 is a configuration diagram showing a method of connecting an inverter 3 of FIG. 1;

FIG. 3 is a time chart showing a relationship between a carrier signal, a three-phase voltage command value, and a PWM signal, and showing a timing at which current is taken.

FIG. 4 is a flowchart for detecting a magnetic pole position in the configuration method of FIG. 1;

FIG. 5 is a configuration diagram showing an embodiment for calculating a magnetic pole position by detecting a current when the cylindrical synchronous motor is in a two-phase short-circuit state.

6 is a time chart showing detection timings of a three-phase PWM signal and the current shown in FIG. 5;

FIG. 7 is a flowchart for detecting a magnetic pole position in the configuration method of FIG. 5;

FIG. 8 is a list of arithmetic expressions for calculating the two-phase short-circuit current difference value and the phase of the three-phase short-circuit current differential vector of FIG. 7;

FIG. 9 is a configuration diagram showing another embodiment for detecting the magnetic pole position of the salient-pole synchronous motor using the difference in current while extending the three-phase short-circuit time.

10 is a time chart showing detection timings of a three-phase PWM signal and the current shown in FIG. 9;

11 is a flowchart for detecting a magnetic pole position with high accuracy in the configuration method of FIG. 9;

FIG. 12 is an electric vehicle in which a salient-pole synchronous motor is controlled by using a first magnetic pole position detector, in which another embodiment having a second magnetic pole position detector for detecting a magnetic pole position by a current in a two-phase short-circuit state; It is a block diagram showing an example.

FIG. 13 is a flowchart for detecting the magnetic pole position of the salient-pole synchronous motor using the two-phase short-circuited current in the configuration method of FIG. 12;

14 is a list of arithmetic expressions for calculating the two-phase short-circuit current difference value and the phase of the three-phase short-circuit current differential vector in FIG.

FIG. 15 is a flowchart for performing abnormality determination of a magnetic pole position in FIG. 12;

FIG. 16 is a configuration diagram showing another embodiment for self-diagnosing a failure in magnetic pole position detection in a salient pole synchronous motor having a magnetic pole position detector that detects a magnetic pole position with a current in a two-phase short-circuit state. .

FIG. 17 is a vector diagram showing an example of a relationship among a current vector, a current differential vector, and a magnetic pole position (d-axis) of the synchronous motor.

18 is a vector diagram showing a relationship between a current differential vector at the time of two-phase short-circuit and a current differential vector at the time of three-phase short-circuit in the cylindrical synchronous motor of FIG. 9;

FIG. 19 is a vector diagram showing a relation of a current differential vector generated by a voltage applied to the α-axis of the salient-pole synchronous motor.

20 is a vector diagram showing a relationship between a current differential vector at the time of two-phase short-circuit and a current differential vector at the time of three-phase short-circuit in the salient-pole synchronous motor of FIG. 16;

FIG. 21 is a configuration diagram of a motor control system showing another embodiment of the present invention.

FIG. 22 is a diagram illustrating an area where detection of a magnet position may be reduced.

FIG. 23 is a diagram illustrating a configuration of a calculation unit 52.

FIG. 24 is a diagram showing a configuration of a current command value generator 6.

FIG. 25 is a diagram showing magnetic characteristics of the synchronous motor.

FIG. 26 is a diagram showing characteristics of d-axis inductance.

FIG. 27 is a diagram showing a processing procedure for detecting a magnetic pole position at the time of starting the motor.

FIG. 28 is a diagram illustrating a processing procedure for determining polarity from a torque generation direction and a rotation direction of a rotating shaft of a motor.

[Description of Signs] 1 ... Cylindrical synchronous motor, 2 ... Battery, 3 ... Inverter, 4 ... Controller, 5a, 5b ... Current sensor, 6 ...
Current command value generation unit, 7: current control unit, 8, 11: coordinate conversion unit, 9: PWM signal generation unit, 10, 27: current detection unit, 12: current differentiation circuit, 13: current differentiation detection unit, 1
4, 15, 17, 20, 52 arithmetic section, 16 salient pole synchronous motor, 18 two-phase switching arithmetic section, 19 magnetic pole position / current arithmetic section, 21 magnetic pole position detecting section, 22 magnetic pole position abnormality Detector, 23 ... Magnetic pole position sensor, 24, 25 ...
Tire, 26 ... Self-diagnosis unit, 54 ... Signal generation unit, 56 ...
Polarity discriminator, 65 ... adder. 70: rotor, 72: rotating shaft, 74: lock mechanism.

──────────────────────────────────────────────────続 き Continued on the front page (51) Int.Cl. ⁷ Identification symbol FI Theme coat ゛ (Reference) H02P 7/63 303 H02P 7/00 501

Claims (28)

[Claims] A synchronous motor; an inverter for driving the synchronous motor; and a magnetic pole position of the synchronous motor detected based on an amount of change in motor current when the synchronous motor is in a short-circuit state. A controller that outputs a control signal for controlling the synchronous motor based on
The synchronous motor control device, wherein the inverter controls the synchronous motor based on the control signal. 2. A synchronous motor, an inverter for driving the synchronous motor, and a magnetic pole position of the synchronous motor detected based on a change direction of a motor current when the synchronous motor is short-circuited. And a controller for generating a control signal for controlling the synchronous motor according to (1), wherein the inverter controls the synchronous motor based on the control signal. 3. A synchronous motor having a synchronous motor, an inverter for driving the synchronous motor, and a controller for generating a control signal, wherein the inverter drives the synchronous motor based on a control signal generated by the controller. A controller, wherein the controller detects a change direction of a motor current when the synchronous motor is in a short-circuit state, and sets a magnetic flux direction to d-axis, d
A dq-axis coordinate system having a direction orthogonal to the axis as the q-axis is set, and the d-axis current and the q-axis current on the set dq-axis coordinate system are detected. A synchronous motor control device, wherein a magnetic pole position of the synchronous motor is calculated based on a current and a q-axis current, and the control signal is generated based on the magnetic pole position. 4. A three-phase synchronous motor; a controller for generating a PWM signal based on a three-phase voltage command value;
An inverter that drives the synchronous motor by an M signal, wherein the controller detects a current in synchronization with a PWM signal of a phase that commands an intermediate value among the three-phase voltage command values, and uses the current to detect the current. A synchronous motor control device comprising: detecting a magnetic pole position of a synchronous motor; and determining the three-phase voltage command value based on the detected magnetic pole position. 5. The synchronous motor control device according to claim 1, wherein the short-circuit state of the synchronous motor is a state in which all phases are short-circuited. 6. The synchronous motor control device according to claim 1, wherein the controller is configured such that two phases generated when the inverter controls the synchronous motor by PWM control are in a short-circuit state. A synchronous motor control device for detecting a change amount or a change direction of the motor current at the time. 7. The synchronous motor control device according to claim 1, wherein the controller is configured to control a short-circuit state in a three-phase state that occurs when the inverter controls the synchronous motor by PWM control. Wherein the amount or direction of change of the motor current is detected. 8. The synchronous motor control device according to claim 1, wherein the controller determines that a change amount or a change direction of the motor current in the short-circuit state is a three-phase short-circuit state of the motor. The change amount or the change direction of the motor current, and the change amount or the change direction is obtained from the change amount or the change direction of the motor current when the synchronous motor is in a short-circuit state of a plurality of different two phases. Synchronous motor control device. 9. The synchronous motor control device according to claim 1, wherein said controller comprises:
A synchronous motor control device comprising an extension means for lengthening a short-circuit state of a phase. 10. The synchronous motor control device according to claim 9, wherein said extension means performs a two-phase switching operation to lengthen a three-phase short-circuit state. 11. The synchronous motor control device according to claim 1, wherein the controller includes a first motor speed obtained from the detected change state of the magnetic pole position and a change amount of the motor current. A synchronous motor control device for determining an abnormality by comparing the second motor speed with a second motor speed obtained by the second control device. 12. A synchronous motor, a magnetic pole position detector for detecting a magnetic pole position of the synchronous motor, a controller for controlling the synchronous motor based on the magnetic pole position detected by the magnetic pole position detector, and a signal based on a signal from the controller. An inverter for driving the synchronous motor, wherein the controller obtains a magnetic pole position of the synchronous motor based on a change amount or a change direction of a motor current when the synchronous motor is in a short-circuit state, and the magnetic pole position detector The magnetic pole position detected in
A synchronous motor control device for detecting an abnormality of a magnetic pole position detector or a controller by comparing magnetic pole positions obtained from a change amount or a change direction of a motor current. 13. The synchronous motor control device according to claim 12, wherein when the controller detects an abnormality in the magnetic pole position detector, the controller performs the synchronization based on a magnetic pole position obtained from a change amount or a change direction of the motor current. A synchronous motor control device for controlling a motor. 14. A synchronous motor for driving a vehicle, a magnetic pole position detector for detecting a magnetic pole position of the synchronous motor, a controller for controlling the synchronous motor based on a magnetic pole position detected by the magnetic pole position detector, and the controller An inverter that drives the synchronous motor based on the signal of the controller, the controller obtains a magnetic pole position of the synchronous motor based on a change amount or a change direction of the motor current when the synchronous motor is in a short-circuit state, An electric vehicle control device that compares a magnetic pole position detected by a magnetic pole position detector with a magnetic pole position obtained from a change amount or a change direction of a motor current to detect an abnormality in the magnetic pole position detector or the controller. 15. The electric vehicle control device according to claim 12, wherein when the controller detects an abnormality of the magnetic pole position detector, the controller controls the synchronization based on a magnetic pole position obtained from a change amount or a change direction of the motor current. An electric vehicle control device for controlling a motor. 16. The electric vehicle control device according to claim 14, wherein the controller detects the abnormality when the controller detects the abnormality.
An electric vehicle control device for stopping the vehicle. 17. The electric vehicle control device according to claim 16, wherein the controller is configured to stop the vehicle and then obtain a magnetic pole position obtained from a magnetic pole position detector or a magnetic pole position obtained from a motor current by a normal controller. An electric vehicle control device, wherein the vehicle is driven by using any normal magnetic pole position detecting means. 18. A first step for detecting a change direction of a motor current when a synchronous motor is in a short-circuit state, and a dq-axis coordinate system in which a magnetic flux direction is a d-axis and a direction orthogonal to the d-axis is a q-axis. A second step of setting the d-axis current and the q-axis current on the set dq-axis coordinate system, and the detected change direction, the d-axis current, and the q-axis current Calculating the magnetic pole position of the synchronous motor based on
And a fifth step of controlling the synchronous motor based on the magnetic pole position. 19. The synchronous motor control method according to claim 18, wherein a difference between the magnetic pole position of the dq axis coordinate system set in said second step and the magnetic pole position calculated in said fourth step is obtained. A synchronous motor control method comprising: controlling the synchronous motor based on the magnetic pole position calculated in the fourth step when the synchronous motor is within a predetermined value range. 20. A synchronous motor, comprising: a controller; and an inverter for driving the synchronous motor based on an output of the controller, wherein the controller is configured such that a change amount or a change direction of a motor current when the synchronous motor is in a short circuit state. A first detecting unit that detects the magnetic pole position of the synchronous motor based on the control signal; and a control unit that generates the output based on the detected magnetic pole position. A second detection unit for detecting the magnetic pole position at a low speed including a rotation stop state when the speed of the synchronous motor is low, including controlling the motor based on the output of the second detection unit; A synchronous motor control device for controlling a motor in a large area based on an output of a first detection unit. 21. The control device according to claim 20, wherein the second detection unit generates a signal for detecting a current used to calculate a magnetic pole position;
A polarity discriminating unit for discriminating whether the calculated magnetic pole position is in the N-pole direction or the S-pole direction. 22. The control device according to claim 21, wherein a P is transmitted from the controller to the inverter so that a current flows through the motor based on an output of the signal generating means.
A control device for a synchronous motor, wherein a WM pulse is sent, a current of the motor is detected based on an output of the signal generating means, a magnetic pole position is calculated, and the motor is controlled. 23. The control device according to claim 21, wherein
The control device has a current control unit that controls the torque of the motor, the polarity determination unit applies a current command for polarity determination in the d-axis direction of the current control unit, and determines a response characteristic of the synchronous motor. A control device for a synchronous motor, wherein a polarity of a magnetic pole is determined based on a difference. 24. The control device according to claim 21, wherein said polarity discriminating section comprises: a synchronous motor torque generated by a magnetic pole position obtained based on the calculated magnetic pole position; A controller for determining the polarity based on the rotation direction of the synchronous motor. 25. The control device according to claim 20, wherein
A synchronous motor, wherein a mechanism for preventing rotation of the synchronous motor is provided, and in a state where the rotation of the synchronous motor is prevented, a current based on an output signal of the second detection unit is supplied to the motor. Control device. 26. A control device for an electric vehicle, comprising a control device for a synchronous motor according to at least one of claims 20 to 26. 27. The electric vehicle control device according to claim 26, wherein the start of current supply to the motor based on the output of the second detection unit includes a state in which the brake of the vehicle is operating. A control device for an electric vehicle, wherein the control is performed in a state where rotation of a drive wheel of the electric vehicle is blocked. 28. The electric vehicle control device according to claim 26, wherein when the current supply to the motor based on the output of the second detection unit is started, the operating range of the electric vehicle is a parking range. A control device for an electric vehicle, which is performed in