JP2011101554A - Controller converter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To properly reduce the losses due to the actuation of a converter that carries out voltage conversion. <P>SOLUTION: A carrier generating unit 112 generates a carrier signal, used for controlling a converter having a switching element and a reactor and includes an amplitude-calculating unit 112C, a loss-calculating unit 112D and a frequency-determining unit 112E. The amplitude-calculating unit 112C calculates the amplitude of the ripple component of a current flowing through the reactor as a function of a carrier frequency, by using the inductance L of the reactor and an input voltage VL and an output voltage VH to and from the converter. The loss-calculating unit 112D calculates each energy loss generated in the reactor and the energy loss generated by the switching element as a function of carrier frequency, by using the ripple component amplitude calculated by the amplitude-calculating unit 112C. The frequency where the total of the calculated carrier frequency functions becomes a minimum, is selected by a determining unit 112E. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to control of a converter that performs voltage conversion between a DC power source and an electric load device.

  In recent years, electric vehicles using a motor as one drive source are becoming widespread. Usually, an electric vehicle is equipped with a battery and obtains driving force by rotating a motor with electric power supplied from the battery. Usually, an electric vehicle includes a converter that performs voltage conversion between a battery and a motor. By driving the converter at a switching frequency calculated using a pulse width modulation (hereinafter referred to as “PWM (Pulse Width Modulation)”) method, the converter boosts the voltage of the battery and outputs it to the motor.

  A technique for controlling the switching frequency of such a converter is disclosed in, for example, Japanese Patent No. 3692993 (Patent Document 1). The drive device disclosed in Patent Document 1 controls a converter including two switching elements and a reactor connected to a connection point between the switching elements. This drive device stores, for each current flowing through the reactor, an optimum carrier frequency that minimizes the converter loss, assuming that the converter loss is uniquely determined from the carrier frequency when the current flowing through the reactor is constant. The optimal carrier frequency corresponding to the target value or detected value of the current flowing through the reactor is set using the map. By driving the converter using the optimum carrier frequency set in this way, the converter can be efficiently driven at the optimum carrier frequency at which the converter loss is minimized.

Japanese Patent No. 3692993 JP 2007-325351 A JP 2008-178166 A JP 2008-131689 A

  In Patent Document 1, the carrier frequency is set on the assumption that the loss of the converter when the current flowing through the reactor is constant is uniquely determined from the carrier frequency.

  However, in reality, even if the DC component of the current flowing through the reactor is constant, the loss caused by the operation of the converter is not uniquely determined from the carrier frequency, and the converter input voltage, output voltage, reactor inductance It fluctuates by etc. The current flowing through the reactor includes a ripple component due to switching of the converter, and this ripple component is a major factor that causes loss, but the amplitude of this ripple component is not uniquely determined only by the carrier frequency. It is. Therefore, it is not sufficient to make the loss generated in the converter uniquely determined from the carrier frequency, and there is room for improvement. In particular, in an electric vehicle in which the input voltage of the converter, the output voltage, the inductance of the reactor, and the like frequently change according to the traveling state, improvement of this point is eagerly desired.

  The present invention has been made to solve the above-described problems, and an object of the present invention is to appropriately reduce the loss caused by the operation of the converter that performs voltage conversion between the DC power supply and the electric load device. It is to provide a control device that can.

  A control device according to the present invention controls a converter that performs voltage conversion between a DC power source and an electric load device. The converter includes two switching elements connected in series to each other, and a reactor provided between the connection point of the two switching elements and the positive electrode of the DC power supply. The current flowing through the reactor includes a ripple component due to switching of the two switching elements in addition to the DC component. The control device includes a signal generation unit that generates a signal for controlling the two switching elements based on a result of comparing the carrier wave and the command value, and a carrier generation unit that generates a carrier wave. The carrier generator generates energy generated by the operation of the converter based on the first voltage input from the DC power source to the converter, the second voltage output from the converter to the electric load device, the inductance of the reactor, and the current flowing through the reactor. A carrier wave having a frequency with a minimum loss is generated.

  Preferably, the carrier generation unit includes a first calculation unit that obtains a first relationship between the frequency of the carrier wave and the amplitude of the ripple component based on the first voltage, the second voltage, and the inductance, the current flowing through the reactor, and the first Based on the relationship, a second calculation unit that obtains a second relationship between the frequency of the carrier wave and the energy loss, and a frequency that minimizes the energy loss using the second relationship are selected, and the selected frequency is And a setting unit for setting the frequency.

  Preferably, the setting unit obtains a frequency range in which the amplitude of the ripple component is equal to or less than a predetermined value based on the first relationship, and sets the frequency of the carrier wave within the obtained frequency range.

  Preferably, the setting unit obtains a frequency range in which a peak value of a current flowing through the reactor and a peak value of a current flowing through the reactor is equal to or less than a predetermined value based on the first relationship, and calculates a carrier frequency within the obtained frequency range. Set.

  Preferably, the electric load device is a device for generating a driving force of the vehicle by an electric motor.

  Preferably, the setting unit determines whether or not the load on the motor changes suddenly, and when the load on the motor changes suddenly, the frequency that minimizes energy loss is set to the frequency of the carrier wave, and the load on the motor does not change suddenly The frequency at which the peak value of the current flowing through the reactor is not more than a predetermined value is set as the frequency of the carrier wave.

  Preferably, a 2nd calculation part calculates | requires several 2nd relationship respectively corresponding to several components in which energy loss arises. The setting unit selects a frequency that minimizes the total value of energy losses generated in the plurality of components using the plurality of second relationships.

  Preferably, a component for connecting the DC power source and the converter and a first capacitor are provided between the DC power source and the converter. A second capacitor is provided between the converter and the electric load device. The second calculation unit obtains a plurality of second relationships respectively corresponding to at least one of the reactor, the two switching elements, the DC power supply, the component for connection, the first capacitor, and the second capacitor.

  Preferably, the control device further includes a third calculation unit that calculates an inductance in accordance with a current flowing through the reactor.

  ADVANTAGE OF THE INVENTION According to this invention, the loss which arises by operation | movement of the converter which performs voltage conversion between DC power supply and an electric load apparatus can be reduced appropriately.

It is a circuit diagram of the motor drive device to which a control device is applied. It is a functional block diagram of a control device. It is the figure which showed the correspondence of carrier signal CR, switching signal Sq2, and current IL. It is a figure which shows the flow of electric current IL at the time of an upper arm ON. It is a figure which shows the flow of electric current IL at the time of a lower arm ON. It is a functional block diagram (the 1) of a carrier production | generation part. 6 is a map showing a correspondence relationship between current IL and inductance L. It is a figure which shows the correspondence of carrier frequency fc and ripple amplitude IL_pp. It is the figure which showed roughly the method of calculating | requiring the reactor loss function Loss1 (fc). It is the figure which showed roughly the method of calculating | requiring IPM loss function Loss2 (fc). It is a figure which shows the correspondence of carrier frequency fc and total loss Loss_total. It is a figure which shows the wiring state of electric power cable PC in an electric vehicle. It is a figure which shows the some map which defined the loss of the converter and its connection component for every component. It is a figure which shows the range of the carrier frequency fc which can suppress a ripple amplitude IL_pp below predetermined value IL_ (alpha). It is a figure which shows the selection range of carrier frequency fc. It is a figure which shows the relationship between peak value IL_peak of electric current IL, and ripple amplitude IL_pp. It is a functional block diagram (the 2) of a carrier production | generation part. It is FIG. (1) explaining the priority of loss minimization and component protection. It is FIG. (2) explaining the priority of loss minimization and component protection. FIG. 6 is a diagram (part 3) illustrating the priority order between loss minimization and component protection;

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In the drawings, the same or corresponding parts are denoted by the same reference numerals and description thereof will not be repeated.

[Embodiment 1]
FIG. 1 is a circuit diagram of a motor drive device to which a control device according to Embodiment 1 of the present invention is applied. Referring to FIG. 1, motor drive device 100 includes DC power supply B, converter 10, inverter 20, positive lines PL1 and PL2, negative line NL, current sensor 52, voltage sensors 54 and 56, A filter capacitor C1, a smoothing capacitor C2, and a control device 30 are provided.

  This motor drive device 100 is mounted on an electric vehicle such as a hybrid vehicle or an electric vehicle. The AC motor M is mechanically coupled to drive wheels (not shown) and generates torque for driving the vehicle. Alternatively, AC motor M may be mechanically connected to an engine (not shown), operate as a generator that generates electric power using engine power, and be incorporated in a hybrid vehicle as an electric motor that starts the engine. .

  Converter 10 includes a reactor L1, switching elements Q1 and Q2, and diodes D1 and D2 (hereinafter, Q1, Q2, D1, and D2 are also collectively referred to as “IPM (Intelligent Power Module)”). Reactor L1 has one end connected to a positive line PL1 connected to the positive terminal of DC power supply B, and the other end connected to an intermediate point between switching element Q1 and switching element Q2, that is, the emitter and switching element of switching element Q1. Connected to the connection point with the collector of Q2. Switching elements Q1, Q2 are connected in series between positive electrode line PL2 and negative electrode line NL. The collector of switching element Q1 is connected to positive line PL2, and the emitter of switching element Q2 is connected to negative line NL. Further, diodes D1 and D2 for passing a current from the emitter side to the collector side are connected between the collector and emitter of switching elements Q1 and Q2, respectively.

  For example, IGBTs (Insulated Gate Bipolar Transistors) or power MOS (Metal Oxide Semiconductor) transistors can be used as the switching elements Q3 to Q8 described later included in the switching elements Q1 and Q2 and the inverter 20.

  Inverter 20 includes a U-phase arm 22, a V-phase arm 24, and a W-phase arm 26. U-phase arm 22, V-phase arm 24, and W-phase arm 26 are connected in parallel between positive electrode line PL2 and negative electrode line NL. U-phase arm 22 includes switching elements Q3 and Q4 connected in series. V-phase arm 24 includes switching elements Q5 and Q6 connected in series. W-phase arm 26 includes switching elements Q7 and Q8 connected in series. Further, diodes D3 to D8 that flow current from the emitter side to the collector side are connected between the collector and emitter of switching elements Q3 to Q8, respectively. And the intermediate point of each phase arm is connected to each phase coil of AC motor M, respectively.

  DC power supply B is a rechargeable power storage device, for example, a secondary battery such as nickel metal hydride or lithium ion. Note that a large capacity capacitor may be used as the DC power source B instead of the secondary battery.

  Converter 10 boosts the voltage between positive electrode line PL2 and negative electrode line NL (hereinafter also referred to as “system voltage”) to a voltage equal to or higher than the output voltage of DC power supply B based on signal PWC from control device 30. Signal PWC includes a switching signal Sq1 for controlling the on-duty of switching element Q1, and a switching signal Sq2 for controlling the on-duty of switching element Q2. Switching signals Sq1 and Sq2 are related so that switching elements Q1 and Q2 are in opposite states (that is, Q2 is off when Q1 is on and Q2 is on when Q1 is off). When the system voltage is lower than the target voltage, a signal Sq2 that increases the on-duty of the switching element Q2 is supplied to the converter 10, thereby allowing a current to flow from the positive line PL1 to the positive line PL2, thereby increasing the system voltage. Can do. On the other hand, when the system voltage is higher than the target voltage, a signal Sq1 that increases the on-duty of the switching element Q1 is supplied to the converter 10, so that a current can flow from the positive line PL2 to the positive line PL1, thereby reducing the system voltage. Can be made.

  Inverter 20 converts DC power supplied from positive line PL2 and negative line NL into three-phase alternating current based on signal PWI from control device 30 and outputs it to alternating current motor M to drive alternating current motor M. As a result, AC motor M is driven so as to generate torque specified by torque command value TR1. Inverter 20 also converts the three-phase AC power generated by AC motor M to DC based on signal PWI during braking of a hybrid vehicle or electric vehicle on which motor drive device 100 is mounted, and generates positive line PL2 and negative line Output to line NL.

  Current sensor 52 detects current IL flowing through reactor L <b> 1 of converter 10 and outputs the detected value to control device 30. Current sensor 52 detects a current flowing from DC power supply B to reactor L1 as a positive value, and detects a current flowing from reactor L1 to DC power supply B as a negative value.

  Filter capacitor C1 is connected between positive electrode line PL1 and negative electrode line NL. Voltage sensor 54 detects voltage VL across filter capacitor C <b> 1 as an input voltage of converter 10, and outputs the detected value to control device 30.

  Smoothing capacitor C2 is connected between positive electrode line PL2 and negative electrode line NL. Smoothing capacitor C <b> 2 smoothes the DC voltage from converter 10, and supplies the smoothed DC voltage to inverter 20.

  Voltage sensor 56 detects voltage VH across smoothing capacitor C <b> 2 as the output voltage of converter 10, and outputs the detected value to control device 30.

  The rotation angle sensor 58 detects the rotation angle θ of the rotor of the AC motor M and outputs the detected value to the control device 30.

  Control device 30 generates a PWM signal for driving converter 10 using the pulse width modulation method, and outputs the generated PWM signal to converter 10 as signal PWC.

  Further, control device 30 generates a PWM signal for driving AC motor M based on torque command value TR1 and motor rotation speed MRN1 of AC motor M received from an external ECU (Electronic Control Unit) (not shown), The generated PWM signal is output to inverter 20 as signal PWI.

  FIG. 2 is a functional block diagram of a portion related to control of converter 10 of control device 30 shown in FIG. Referring to FIG. 2, control device 30 includes voltage command generation unit 102, subtraction units 104 and 108, voltage control calculation unit 106, current control calculation unit 110, drive signal generation unit 111, and carrier generation unit. 112 and a sample / hold (hereinafter referred to as “S / H”) circuit 116.

  Voltage command generation unit 102 generates a voltage command value VHcom indicating a target value of voltage VH that is the output voltage of converter 10. For example, voltage command generation unit 102 generates voltage command value VHcom based on the power of AC motor M calculated from torque command value TR1 of AC motor M and motor rotational speed MRN1.

  Subtraction unit 104 subtracts the detected value of voltage VH from voltage command value VHcom, and outputs the calculation result to voltage control calculation unit 106 as deviation ΔVH.

  When the deviation ΔVH is not “0”, the voltage control calculation unit 106 executes a feedback control calculation (for example, calculation by proportional integration control) for bringing the voltage VH close to the voltage command value VHcom. Then, the voltage control calculation unit 106 outputs the calculated control amount as the current command value IR.

  The carrier generation unit 112 generates a carrier signal CR composed of a triangular wave for generating a PWM signal in the drive signal generation unit 111 described later, and the generated carrier signal CR is used as the drive signal generation unit 111 and the S / H circuit 116. Output to. The function of the carrier generation unit 112 will be described in detail later.

  S / H circuit 116 samples current IL at the timing of peaks and valleys of carrier signal CR received from carrier generation section 112.

  The subtracting unit 108 subtracts the detected value of the current IL sampled / held by the S / H circuit 116 from the current command value IR output from the voltage control calculating unit 106 and sends the calculation result to the current control calculating unit 110. Output.

  The current control calculation unit 110 receives a value obtained by subtracting the detected value of the current IL from the current command value IR from the subtraction unit 108, and performs feedback control (hereinafter referred to as “current feedback control”) to bring the current IL close to the current command value IR. Is executed by, for example, proportional-integral control. Then, the current control calculation unit 110 outputs the calculated control amount to the drive signal generation unit 111 as the duty command value d. High control responsiveness of the converter 10 is realized by the current feedback control by the current control calculation unit 110. That is, the speed at which voltage VH follows voltage command value VHcom is improved.

  The drive signal generation unit 111 compares the duty command value d received from the current control calculation unit 110 with the carrier signal CR received from the carrier generation unit 112, and a signal PWC (switching signal) whose logic state changes according to the comparison result. Sq1, Sq2) are generated. Then, drive signal generation unit 111 outputs the generated signal PWC to switching elements Q1 and Q2 of converter 10.

  FIG. 3 is a diagram illustrating a correspondence relationship between the carrier signal CR, the switching signal Sq2, and the current IL. In FIG. 3, the current IL is a positive value. In the following description, the ratio of turning on the lower arm of the switching element Q1 (upper arm) and the switching element Q2 (lower arm) is “duty (0 ≦ duty ≦ 1)”.

  The magnitude of the energy loss that occurs in the components (reactor L1, IPM, etc.) constituting the converter 10 is determined by the current IL flowing through them and the carrier frequency fc, but as shown in FIG. In addition, a ripple component generated with IPM switching is included. If the direct current component of the current IL and the carrier frequency fc are constant, the larger the amplitude of the ripple component (hereinafter also referred to as “ripple amplitude IL_pp”), the greater the energy loss that occurs in the components constituting the converter 10.

  Here, when the inductance of the reactor L1 is L, the ripple amplitude IL_pp is expressed by the following equation (1).

IL_pp = (VL / L) × (1 / fc) × (VH−VL) / VH (1)
Hereinafter, the principle that the ripple amplitude IL_pp is expressed by the equation (1) will be described.

  First, the ripple amplitude IL_p1 when the upper arm is on (when Q1 is on) is derived. FIG. 4 shows the flow of current IL when the upper arm is on. When the upper arm is on, the current IL flows through the positive line PL2 via the diode D1 as indicated by the arrow a in FIG. The voltage equation in this case is the following equation (2a). When this equation (2a) is modified, the slope dIL / dt of the current IL is represented by equation (2b).

VL-L (dIL / dt) -VH = 0 (2a)
dIL / dt = (VL−VH) / L (2b)
On the other hand, the ripple amplitude IL_p1 can be expressed by the following equation (2c) using the slope dIL / dt of the current IL, the carrier period Tc, and the duty. Therefore, the above equation (2b) is substituted into this equation (2c). Then, the ripple amplitude IL_p1 can be expressed by the following equation (2d).

IL_p1 = (dIL / dt) × Tc × (1-duty) (2c)
IL_p1 = (VL−VH) / L × Tc × (1-duty) (2d)
Next, the ripple amplitude IL_p2 when the lower arm is on (when Q2 is on) is derived. FIG. 5 shows the flow of current IL when the lower arm is on. When the lower arm is on, the current IL flows through the negative electrode line NL via the switching element Q2, as indicated by the arrow b in FIG. The voltage equation in this case is the following equation (3a). When this equation (3a) is modified, the slope dIL / dt of the current IL is represented by equation (3b).

VL-L (dIL / dt) = 0 (3a)
dIL / dt = VL / L (3b)
On the other hand, the ripple amplitude IL_p2 can be expressed by the following formula (3c) using the slope dIL / dt of the current IL, the carrier period Tc, and the duty. Therefore, the above formula (3b) is substituted into this formula (3c). Then, the ripple amplitude IL_p2 can be expressed by the following equation (3d).

IL_p2 = (dIL / dt) × Tc × duty (3c)
IL_p2 = (VL / L) × Tc × duty (3d)
Here, the duty is determined by the voltages VL and VH, and duty = 1−VL / VH. Therefore, when this equation is substituted into the equations (2d) and (3d), the following equations (2e) and (3e) are obtained, respectively. It becomes.

IL_p1 = − (VH−VL) / L × Tc × (VL / VH) (2e)
IL_p2 = (VL / L) × (VL / L) × Tc × (1-VL / VH)
= (VH−VL) / L × Tc × (VL / VH) (3e)
Comparing equation (2e) and equation (3e), the absolute value of IL_p1 and the absolute value of IL_p2 (magnitude of amplitude) are the same, which is redefined as ripple amplitude IL_pp, and the carrier period Tc is When replaced with 1 / carrier frequency fc, the above-described equation (1) is obtained. Thus, it can be explained in principle that the ripple amplitude IL_pp is expressed by the equation (1).

  By the way, as is clear from the equation (1), the ripple amplitude IL_pp is determined from four parameters of the voltages VL and VH, the inductance L, and the carrier frequency fc. That is, the ripple amplitude IL_pp varies greatly depending on the inductance L and the voltages VL and VH, not the value obtained only from the carrier frequency fc.

  For example, even when the carrier frequency fc and the inductance L are the same value, at the time of double boosting (when VH = 2 × VL), the ripple amplitude IL_pp = VL / (2 × L × fc) from the equation (1) On the other hand, at the time of triple boosting (when VH = 3 × VL), the ripple amplitude IL_pp = 2 × VL / (3 × L × fc) from Equation (1), and the ripple amplitude IL_pp at the triple boosting Is about 1.3 times as much as that at the time of double boosting. Further, the inductance L varies according to the magnitude of the current IL. In an electric vehicle such as a hybrid vehicle or an electric vehicle, the voltages VL and VH and the current IL (inductance L) frequently change according to the driving state, etc., and therefore the fluctuation of the ripple amplitude IL_pp due to these changes is also taken into consideration. In addition, it is necessary to reduce the loss generated in the converter 10.

  In consideration of these points, the carrier generation unit 112 obtains the ripple amplitude IL_pp as a function of the carrier frequency fc using the inductance L and the voltages VL and VH, and further uses the obtained ripple amplitude IL_pp to determine the converter 10. The energy loss that occurs in each component that constitutes is determined for each component as a function of the carrier frequency fc, and the carrier frequency fc that minimizes the total value is selected. This is the most characteristic point of the present embodiment.

  FIG. 6 shows a functional block diagram of the carrier generation unit 112. The carrier generation unit 112 includes an acquisition unit 112A, an inductance calculation unit 112B, an amplitude calculation unit 112C, a loss calculation unit 112D, a frequency determination unit 112E, and a generation unit 112F.

  Acquisition unit 112A performs sampling of current IL and voltages VL and VH at the timing of peaks and valleys of carrier signal CR. Each value is not limited to the detected value, and may be a command value.

  The inductance calculation unit 112B calculates the inductance L corresponding to the current IL acquired by the acquisition unit 112A using the map shown in FIG. FIG. 7 is a map showing the correspondence relationship between the current IL and the inductance L stored in advance.

  The amplitude calculation unit 112C substitutes the inductance L calculated by the inductance calculation unit 112B and the voltages VL and VH acquired by the acquisition unit 112A into the above equation (1) and transforms the ripple amplitude IL_pp into the following equation ( It is obtained as a function of the carrier frequency fc shown in 4) (hereinafter also referred to as “ripple amplitude function IL_pp (fc)”).

IL_pp (fc) = k × (1 / fc) (4)
Here, k is a constant determined by (VL / L) × (VH−VL) / VH. The constant k is obtained using the inductance L from the inductance calculation unit 112B and the voltages VL and VH acquired by the acquisition unit 112A.

  FIG. 8 shows a correspondence relationship between the carrier frequency fc and the ripple amplitude IL_pp. As is clear from the equation (4), if the constant k is determined by L, VL, and VH, the ripple amplitude IL_pp is inversely proportional to the carrier frequency fc, where the constant k is a proportional constant.

  Returning to FIG. 6, the loss calculation unit 112 </ b> D obtains the energy loss generated in each component constituting the converter 10 as a function of the carrier frequency fc for each component using the ripple amplitude function IL_pp (fc).

  In the following description, the case where the loss of reactor L1 and IPM is calculated | required among each components which comprise the converter 10 is demonstrated exemplarily.

  The loss calculation unit 112D includes a reactor loss map M1 and an IPM loss map M2 obtained in advance through experiments or the like.

  Reactor loss map M1 is a map in which loss Loss1 of reactor L1 is determined using carrier frequency fc, ripple amplitude IL_pp, and current IL (DC component) as parameters. In reactor loss map M1, reactor loss Loss1 is set to a larger value as carrier frequency fc is lower, is set to a larger value as ripple amplitude IL_pp is larger, and is set to a larger value as current IL is larger. This is a result of reflecting the effects of iron loss and copper loss in reactor L1.

  The IPM loss map M2 is a map that defines the loss Loss2 of the IPM using the carrier frequency fc, the ripple amplitude IL_pp, and the current IL (DC component) as parameters. In the IPM loss map M2, the IPM loss Loss2 is set to a larger value as the carrier frequency fc is higher, is set to a larger value as the ripple amplitude IL_pp is larger, and is set to a larger value as the current IL is larger. This is a result reflecting the influence of switching loss and copper loss in IPM.

  The loss calculation unit 112D obtains a reactor loss function Loss1 (fc) that uses the reactor loss Loss1 as a function of the carrier frequency fc using the reactor loss map M1, the ripple amplitude function IL_pp (fc), and the current IL.

  Similarly, the loss calculation unit 112D uses the IPM loss map M2, the ripple amplitude function IL_pp (fc), and the current IL to obtain an IPM loss function Loss2 (fc) having the IPM loss Loss2 as a function of the carrier frequency fc.

  FIG. 9 is a diagram schematically showing a method for obtaining the reactor loss function Loss1 (fc). Since current IL is specified by the value acquired by acquisition unit 112A, reactor loss map M1 is a map in which reactor loss Loss1 is determined using carrier frequency fc and ripple amplitude IL_pp as parameters, as shown in FIG. Furthermore, since the ripple amplitude IL_pp is k × (1 / fc) and is a function of the carrier frequency fc, if the carrier frequency fc is uniquely determined, the ripple amplitude IL_pp is also uniquely determined, and the reactor loss map M1 is used. Reactor loss Loss1 is also uniquely determined. The loss calculation unit 112D obtains the reactor loss function Loss1 (fc) using this relationship. For example, the loss calculator 112D sequentially obtains the value of the reactor loss Loss1 for a plurality of sampling points of the carrier frequency fc, and stores the combination of each sampling point and the reactor loss Loss1 for each sampling point.

  As shown in FIG. 9, in the reactor loss function Loss1 (fc), the reactor loss Loss1 tends to decrease as the carrier frequency fc increases. That is, in reactor L1, the higher the carrier frequency fc, the lower the loss (iron loss) due to the carrier frequency fc, and the smaller the ripple amplitude IL_pp and the loss due to the ripple amplitude IL_pp (copper loss and iron loss). Also decreases. Therefore, the loss of reactor L1 tends to decrease as the carrier frequency fc increases.

  FIG. 10 is a diagram schematically showing a method for obtaining the IPM loss function Loss2 (fc). Since the current IL is specified by the value acquired by the acquisition unit 112A, the IPM loss map M2 is a map that defines the IPM loss Loss2 using the carrier frequency fc and the ripple amplitude IL_pp as parameters, as shown in FIG. Further, since the ripple amplitude IL_pp is k × (1 / fc) and is a function of the carrier frequency fc, if the carrier frequency fc is uniquely determined, the ripple amplitude IL_pp is also uniquely determined, and further, the IPM loss map M2 is used. The IPM loss Loss2 is also uniquely determined. The loss calculation unit 112D obtains the IPM loss function Loss2 (fc) using this relationship. For example, the loss calculator 112D sequentially obtains the value of the IPM loss Loss2 for a plurality of sampling points of the carrier frequency fc, and stores the combination of each sampling point and the IPM loss Loss2 for each sampling point.

  As shown in FIG. 10, in the IPM loss function Loss2 (fc), the reactor loss Loss2 tends to increase as the carrier frequency fc increases in a range where the carrier frequency fc exceeds a predetermined value. That is, in the IPM, as the carrier frequency fc increases, the ripple amplitude IL_pp decreases and the loss (copper loss) due to the ripple amplitude IL_pp decreases, but the loss due to the carrier frequency fc (loss due to switching) decreases. To increase. Therefore, the loss of IPM tends to increase as the carrier frequency fc increases in the range where the carrier frequency fc exceeds a predetermined value.

  The curve shapes of the reactor loss function Loss1 (fc) and the IPM loss function Loss2 (fc) shown in FIGS. 9 and 10 are merely examples, and are not limited to these.

  Returning to FIG. 6, the frequency determination unit 112E adds the reactor loss function Loss1 (fc) and the IPM loss function Loss2 (fc) to obtain the total loss Loss_total of the converter 10. That is, the total loss Loss_total = Loss1 (fc) + Loss2 (fc). As described above, the total loss Loss_total is a function of only the carrier frequency fc, but since the ripple amplitude IL_pp and the current IL are already taken into consideration at the stage of deriving Loss1 (fc) and Loss2 (fc), the total loss Loss_total is also calculated. In addition to the carrier frequency fc, the value takes into account the ripple amplitude IL_pp and the current IL.

  Then, as shown in FIG. 11, the frequency determination unit 112E selects a carrier frequency fc that minimizes the total loss Loss_total. Specifically, the frequency determination unit 112E obtains the total loss Loss_total for each carrier frequency fc, and then selects the carrier frequency fc corresponding to the smallest total loss Loss_total.

  The generation unit 112F generates a carrier signal CR having the carrier frequency fc selected by the frequency determination unit 112E and outputs the carrier signal CR to the drive signal generation unit 111.

  As described above, in the control 30 according to the embodiment of the present invention, the ripple amplitude IL_pp, which is a major factor causing loss in the converter 10, varies not only with the carrier frequency fc but also with the inductance L and the voltages VL and VH. Considering this point, the ripple amplitude IL_pp is obtained as a function of the carrier frequency fc using the inductance L and the voltages VL and VH. Then, using the obtained ripple amplitude IL_pp, the loss generated in the converter 10 is obtained as a function of the carrier frequency fc for each component constituting the converter 10, and the optimum carrier frequency fc that minimizes the total value is selected. . Therefore, the loss can be reduced more accurately than in the case where the loss is calculated assuming that the ripple amplitude IL_pp is uniquely determined from the carrier frequency fc. In particular, in an electric vehicle such as a hybrid vehicle or an electric vehicle in which the inductance L and the voltages VL and VH change frequently, the control according to the embodiment of the present invention is very effective for reducing the loss.

[Embodiment 2]
In the first embodiment described above, the components considered so that the loss is minimized are the reactor L1 and the IPM that are the components of the converter 10.

  However, the current flowing through the reactor L1 is not only the reactor L1 and the IPM, but also the DC power source B, the positive lines PL1 and PL2, the negative line NL, the filter capacitor C1, the smoothing capacitor C2, or a bus bar used as a connection terminal thereof (see FIG. (Not shown). Therefore, in order to more appropriately reduce the loss caused by the operation of the converter 10, it is necessary to consider the loss caused by these components. For example, when converter 10 is mounted at the front of the vehicle and DC power supply B is mounted at the rear of the vehicle, power cable PC including positive line PL1 and negative line NL becomes very long as shown in FIG. The loss caused by the power cable PC cannot be ignored.

  Therefore, the second embodiment is characterized in that the carrier frequency fc that minimizes the loss is selected in consideration of the loss of these components. That is, as shown in FIG. 13, the loss calculation unit 112D adds the loss Loss3 of the power cable PC, the loss Loss4 of the DC power supply B, the filter in addition to the reactor loss map M1 and the IPM loss map M2 described in the first embodiment. The loss maps M3 to M7, which respectively define the loss Loss5 of the capacitor C1, the loss Loss6 of the smoothing capacitor C2, and the loss Loss7 of the bus bar, are provided in advance. Each is determined as a function of the carrier frequency fc. Then, the frequency determination unit 112E selects the carrier frequency fc that minimizes the total value thereof.

  Thus, efficiency can be improved more appropriately by considering the loss of many components that cause a loss due to the operation of the converter 10 and selecting the carrier frequency fc that minimizes the total value thereof.

  As parameters of the loss maps M3 to M7, for example, as shown in FIG. 13, similarly to the reactor loss map M1 and the IPM loss map M2, the carrier frequency fc, the ripple amplitude IL_pp, and the current IL (DC component) are used. That's fine. The parameters may be added or deleted in consideration of the loss characteristics of each component. For example, since the loss caused by the filter capacitor C1 and the smoothing capacitor C2 is not significantly affected by the current IL (DC component) and the carrier frequency fc, only the ripple amplitude IL_pp may be used as a parameter for the loss maps M5 and M6.

[Embodiment 3]
In the first and second embodiments described above, the method of obtaining the ripple amplitude function IL_pp (fc) and selecting the carrier frequency fc that minimizes the total loss using the obtained ripple amplitude function IL_pp (fc) has been described. At this time, in Embodiments 1 and 2, there is no particular restriction on the magnitude of the ripple amplitude IL_pp.

  However, if the ripple amplitude IL_pp is large, the life of the DC power supply B may be shortened, or a high frequency sound in the audible region may be generated from the DC power supply B. That is, in order to extend the life of the DC power supply B or reduce noise generated by the DC power supply B, it is necessary to provide a restriction on the magnitude of the ripple amplitude IL_pp.

  Therefore, in the third embodiment, the range of the carrier frequency fc that can suppress the ripple amplitude IL_pp below the predetermined value IL_α is defined using the ripple amplitude function IL_pp (fc), and the carrier frequency fc is selected within the defined range. (That is, providing a restriction on the selection range of the carrier frequency fc).

  From the above equation (1), in order to make the ripple amplitude IL_pp equal to or smaller than the predetermined value IL_α, the following equation (5) needs to be established. When the equation (5) is modified, the following equation (6) is obtained.

(VL / L) × (VH−VL) / VH × (1 / fc) ≦ IL_α (5)
fc ≧ (VL / L) × (VH−VL) / VH × (1 / IL_α) (6)
That is, the value on the right side of Equation (6) is the carrier frequency fc when the ripple amplitude IL_pp becomes the predetermined value IL_α, and when this is defined as the frequency fc_α, the ripple amplitude IL_pp is set to the predetermined value as shown in FIG. In order to make IL_α or less, the carrier frequency fc needs to be made fc_α or more.

  Therefore, in the third embodiment, as shown in FIG. 15, the frequency determination unit 112E sets the range of the carrier frequency fc that can suppress the ripple amplitude IL_pp below the predetermined value IL_α to a range above the frequency fc_α. The carrier frequency fc that minimizes the total loss Loss_total is selected.

  As a result, it is possible to reduce the loss while extending the life of the DC power supply B and reducing noise generated in the DC power supply B. Furthermore, since the search range for the carrier frequency fc that minimizes the total loss Loss_total is limited, the search time for the carrier frequency fc can be shortened.

  In the motor drive device 100 shown in FIG. 1, the ripple amplitude IL_pp is applied to the DC power supply B after being attenuated by the filter capacitor C1. Therefore, the predetermined value IL_α is determined in consideration of not only the ripple amplitude that affects the life of the DC power supply B and noise but also the attenuation in the filter capacitor C1.

[Embodiment 4]
In the first to third embodiments described above, the method of minimizing the loss of the components connected to the converter 10 and the converter 10 has been described. However, in an electric vehicle such as a hybrid vehicle or an electric vehicle, the load of the AC motor M is The driver changes suddenly according to the driving situation, for example, when the driver requests sudden acceleration or tire slip / grip occurs.

  For example, when slip / grip occurs, as shown in FIG. 16, the current IL is rapidly decreased by the grip and then the current IL is rapidly decreased by the grip. From the viewpoint of protecting the IPM by setting the current flowing through the IPM to be equal to or less than the allowable current value, it is desirable to control the peak value of the current IL to be within a predetermined range. If the peak value of the current IL is defined as IL_peak, the peak value IL_peak is a value obtained by adding half of the ripple amplitude IL_pp to the current IL (DC component) as shown in FIG.

  Therefore, in the fourth embodiment, the range of the carrier frequency fc that can suppress the peak value of the current IL to be equal to or less than the predetermined value IL_β is defined using the ripple amplitude function IL_pp (fc), and the carrier frequency fc is within the defined range. It is characterized by protecting the IPM by selecting.

  FIG. 17 shows a functional block diagram of carrier generation section 112 according to Embodiment 4. The carrier generation unit 112 according to Embodiment 4 includes an acquisition unit 112A, an inductance calculation unit 112B, a frequency determination unit 112G, and a generation unit 112F. Since acquisition unit 112A, inductance calculation unit 112B, and generation unit 112F other than frequency determination unit 112G have been described in the first embodiment, detailed description thereof will not be repeated here.

  The frequency determination unit 112G selects a carrier frequency fc that can suppress the peak value of the current IL to be equal to or less than the predetermined value IL_β, and outputs the selected carrier frequency fc to the generation unit 112F.

  Hereinafter, a method of selecting the frequency fc_β performed by the frequency determination unit 112G will be described in detail. As described above, the peak value IL_peak is a value obtained by adding half of the ripple amplitude IL_pp to the current IL (DC component), and is represented by the following equation (7).

IL_peak = IL + IL_pp / 2 (7)
From the above equation (7), in order to make the peak value IL_peak less than or equal to the predetermined value IL_β, it is necessary to establish the following equation (8).

IL + {(VL / L) × (VH−VL) / VH × (1 / fc)} / 2 ≦ IL_β
... (8)
fc ≧ (VL / L) × (VH−VL) / VH × {1 / (IL_β−IL)} / 2
... (9)
That is, the value on the right side of the equation (9) is the carrier frequency fc when the peak value IL_peak becomes the predetermined value IL_β. If this is defined as fc_β, the carrier value can be less than or equal to the predetermined value IL_β. The frequency fc needs to be greater than or equal to fc_β.

  Therefore, the frequency determination unit 112G sets the range of the carrier frequency fc that can suppress the peak value IL_peak to be equal to or lower than the predetermined value IL_β as a range equal to or higher than fc_β, and selects the carrier frequency fc within this range. Thereby, the peak value IL_peak can be suppressed below the predetermined value IL_β, and the IPM can be protected.

  Thus, in the fourth embodiment, carrier frequency fc that can suppress peak value IL_peak to a predetermined value IL_β or less is obtained in advance, and converter 10 is controlled at the obtained carrier frequency fc. Therefore, the peak value IL_peak can be reliably suppressed to be equal to or less than the predetermined value IL_β.

  Further, if the carrier frequency fc that minimizes the total loss Loss_total is selected within the range of fc_β or more as described in the first to third embodiments, the loss can be reduced while the peak value IL_peak is suppressed to a predetermined value IL_β or less. Can also be achieved.

  The carrier frequency fc has an upper limit frequency that is determined by the upper limit value of the IPM heat generation and the switching speed. Therefore, actually, the carrier frequency fc is selected in the range of fc_β and the upper limit frequency.

  Further, for example, when the peak value IL_peak is predicted from the currents IL and duty, and the final means is provided to limit the duty so that the predicted value of the peak value IL_peak is equal to or less than the predetermined value IL_β, the peak value is more reliably obtained. IL_peak can be suppressed to a predetermined value IL_β or less.

[Embodiment 5]
Of the above-described first to fourth embodiments, the first to third embodiments have described a technique for minimizing loss, and the fourth embodiment has described a technique for protecting a component.

  On the other hand, the fifth embodiment is characterized in that the carrier frequency fc is controlled by switching between giving priority to loss minimization or giving priority to component protection in accordance with the traveling state of the vehicle.

  For example, as shown in FIG. 18, when current IL is smaller than a threshold value, loss minimization is prioritized and carrier frequency fc that minimizes loss is selected, slip / grip occurs, and current IL is When it becomes larger than the threshold value, the carrier frequency fc that suppresses the peak value IL_peak is selected with priority given to component protection.

  Since slip / grip occurs only rarely under normal driving conditions, priority is given to minimizing loss when the load is stable, and priority is given to parts protection during sudden changes in load such as slip / grip. Parts can be protected without substantially reducing fuel consumption.

  Further, for example, considering that the load is unlikely to change suddenly in the region where the torque of the AC motor M is low and the rotation speed is low, the control device 30 has a low torque and low rotation speed region as shown in FIG. In this case, priority may be given to minimizing loss and priority is given to component protection in other areas.

  Further, as shown in FIG. 20, the control device 30 determines whether or not a tire slip grip has occurred (whether or not the load has suddenly changed) based on a change in the rotational speed of the AC motor M (step ( (Hereinafter abbreviated as “S”) 10) When there is no slip / grip (NO at S10), priority is given to the loss minimization (S20), and when slip / grip occurs (YES at S10). Priority may be given to component protection (S30). As described above, since the slip / grip occurs only rarely in a normal running state, the components can be protected also in this case without substantially reducing the fuel consumption.

  The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

  10 converter, 20 inverter, 22 U-phase arm, 24 V-phase arm, 26 W-phase arm, 30 control device, 52 current sensor, 54, 56 voltage sensor, 58 rotation angle sensor, 100 motor drive device, 102 voltage command generator 104, 108 Subtraction unit, 106 Voltage control calculation unit, 110 Current control calculation unit, 111 Drive signal generation unit, 112B Inductance calculation unit, 112 Carrier generation unit, 112A acquisition unit, 112E, 112G Frequency determination unit, 112C Amplitude calculation unit , 112D loss calculation unit, 112F generation unit, 116 S / H circuit, B DC power supply, C1 filter capacitor, C2 smoothing capacitor, CR carrier signal, D1-D8 diode, Q1-Q8 switching element, L1 reactor, M AC motor, M1 reactor Loss map, M2 IPM loss map, M3-M7 loss map, NL negative line, PC power cable, PL1, PL2 positive line.

Claims (9)

A control device for a converter that performs voltage conversion between a DC power source and an electric load device,
The converter includes two switching elements connected in series with each other, and a reactor provided between a connection point of the two switching elements and a positive electrode of the DC power source,
The current flowing through the reactor includes a ripple component due to switching of the two switching elements in addition to a DC component,
The controller is
A signal generation unit that generates a signal for controlling the two switching elements based on a result of comparing the carrier wave and the command value;
A carrier generation unit for generating the carrier wave,
The carrier generation unit is based on a first voltage input from the DC power source to the converter, a second voltage output from the converter to the electric load device, an inductance of the reactor, and a current flowing through the reactor. A control device for a converter that generates the carrier wave having a frequency at which energy loss caused by the operation of the converter is minimized. The carrier generator is
A first calculation unit that obtains a first relationship between the frequency of the carrier wave and the amplitude of the ripple component based on the first voltage, the second voltage, and the inductance;
A second calculation unit for obtaining a second relationship between the frequency of the carrier wave and the energy loss based on the current flowing through the reactor and the first relationship;
2. The converter control device according to claim 1, further comprising: a setting unit that selects a frequency that minimizes the energy loss using the second relationship, and sets the selected frequency to the frequency of the carrier wave.   The setting unit obtains a frequency range in which an amplitude of the ripple component is equal to or less than a predetermined value based on the first relationship, and sets the frequency of the carrier wave within the obtained frequency range. The control device of the converter described in 1.   The setting unit obtains a frequency range in which a peak value of the current flowing through the reactor and a peak value of the current flowing through the reactor is equal to or less than a predetermined value based on the first relationship, and the carrier wave within the obtained frequency range The control device of the converter according to claim 2, wherein the frequency is set.   The converter control device according to claim 2, wherein the electric load device is a device for generating a driving force of a vehicle by an electric motor.   The setting unit determines whether or not the load of the motor changes suddenly, and when the load of the motor changes suddenly, sets the frequency at which the energy loss is minimum to the frequency of the carrier wave, and the load of the motor is The converter control device according to claim 5, wherein if it does not change suddenly, a frequency at which a peak value of a current flowing through the reactor is equal to or lower than a predetermined value is set as the frequency of the carrier wave. The second calculation unit obtains a plurality of the second relationships respectively corresponding to a plurality of parts in which the energy loss occurs.
3. The converter control device according to claim 2, wherein the setting unit selects a frequency at which a total value of energy losses generated in the plurality of components is minimized using the plurality of second relationships. Between the DC power supply and the converter, a component for connecting the DC power supply and the converter, and a first capacitor are provided,
A second capacitor is provided between the converter and the electric load device,
The second calculation unit includes a plurality of second units respectively corresponding to at least one of the reactor, the two switching elements, the DC power supply, the component for connection, the first capacitor, and the second capacitor. The converter control device according to claim 7, wherein a relationship is obtained.   The said control apparatus is a control apparatus of the converter of Claim 2 which further contains the 3rd calculation part which calculates the said inductance according to the electric current which flows through the said reactor.

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