JP4623065B2 - Voltage conversion apparatus and voltage conversion method - Google Patents

Description

The present invention relates to a voltage converter that converts a DC voltage from a DC power source into an output voltage, a voltage conversion method that converts a DC voltage into an output voltage, and a computer that controls voltage conversion that converts a DC voltage into an output voltage. The present invention relates to a computer-readable recording medium on which a program for recording is recorded.

  Recently, hybrid vehicles and electric vehicles have attracted a great deal of attention as environmentally friendly vehicles. Some hybrid vehicles have been put into practical use.

  This hybrid vehicle is a vehicle that uses a DC power source, an inverter, and a motor driven by the inverter as a power source in addition to a conventional engine. In other words, a power source is obtained by driving the engine, a DC voltage from a DC power source is converted into an AC voltage by an inverter, and a motor is rotated by the converted AC voltage to obtain a power source. An electric vehicle is a vehicle that uses a DC power source, an inverter, and a motor driven by the inverter as a power source.

  In such a hybrid vehicle or an electric vehicle, it is also considered that a DC voltage from a DC power source is boosted by a boost converter so that the boosted DC voltage is supplied to an inverter that drives a motor (for example, JP, 2001-275367, A, etc.).

  That is, the hybrid vehicle or the electric vehicle is equipped with the motor drive device shown in FIG. Referring to FIG. 15, motor drive device 300 includes DC power supply B, system relays SR <b> 1 and SR <b> 2, capacitors C <b> 1 and C <b> 2, bidirectional converter 310, voltage sensor 320, and inverter 330.

  The DC power source B outputs a DC voltage. When system relays SR1 and SR2 are turned on by a control device (not shown), DC voltage from DC power supply B is supplied to capacitor C1. Capacitor C1 smoothes the DC voltage supplied from DC power supply B via system relays SR1 and SR2, and supplies the smoothed DC voltage to bidirectional converter 310.

  Bidirectional converter 310 includes a reactor 311, NPN transistors 312 and 313, and diodes 314 and 315. Reactor 311 has one end connected to the power supply line of DC power supply B, and the other end connected to an intermediate point between NPN transistor 312 and NPN transistor 313, that is, between the emitter of NPN transistor 312 and the collector of NPN transistor 313. The NPN transistors 312 and 313 are connected in series between the power supply line and the earth line. The collector of the NPN transistor 312 is connected to the power supply line, and the emitter of the NPN transistor 313 is connected to the ground line. In addition, diodes 314 and 315 for passing a current from the emitter side to the collector side are connected between the collector and emitter of each NPN transistor 312 and 313.

  In bidirectional converter 310, NPN transistors 312 and 313 are turned on / off by a control device (not shown), the DC voltage supplied from capacitor C1 is boosted, and the output voltage is supplied to capacitor C2. Bidirectional converter 310 also steps down the DC voltage generated by AC motor M1 and converted by inverter 330 during regenerative braking of a hybrid vehicle or electric vehicle equipped with motor drive device 300, and supplies the voltage to capacitor C1. .

  Capacitor C <b> 2 smoothes the DC voltage supplied from bidirectional converter 310 and supplies the smoothed DC voltage to inverter 330. The voltage sensor 320 detects the voltage on both sides of the capacitor C2, that is, the output voltage Vm of the bidirectional converter 310.

  When a DC voltage is supplied from the capacitor C2, the inverter 330 converts the DC voltage into an AC voltage based on control from a control device (not shown) and drives the AC motor M1. Thus, AC motor M1 is driven so as to generate torque specified by the torque command value. Further, the inverter 330 converts the AC voltage generated by the AC motor M1 into a DC voltage based on the control from the control device at the time of regenerative braking of the hybrid vehicle or the electric vehicle on which the motor driving device 300 is mounted. DC voltage is supplied to bidirectional converter 310 via capacitor C2.

In the motor driving device 300, when the DC voltage output from the DC power source B is boosted and the output voltage Vm is supplied to the inverter 330, feedback control is performed so that the output voltage Vm detected by the voltage sensor 320 becomes the voltage command Vdccom. Is done.
JP 2001-275367 A

  However, in the range where the reactor current flowing from the DC power source B to the bidirectional converter 310 is smaller than the ripple current of the reactor current, the bidirectional current is affected by the dead time when both the NPN transistors 312 and 313 of the bidirectional converter 310 are turned off. There is a problem that the output voltage Vm of the converter 310 rapidly rises and falls.

  That is, referring to FIG. 16, when bidirectional converter 310 boosts the DC voltage to output voltage Vm, the control device (not shown) outputs signal PWU to bidirectional converter 310. The signal PWU has a duty ratio of 50%. In this case, NPN transistors 312 and 313 of bidirectional converter 310 receive signals PWUU and PWUD at their gate terminals, respectively.

  Then, regions DT1 to DT4 are regions in which both signals PWUU and PWUD are at L (logic low) level, and are called dead time. When the reactor current is positive, the direct current flows to the positive bus of the inverter 330 via the direct current power source B, the reactor 311, and the upper arm (NPN transistor 312) of the bidirectional converter 310. The NPN transistors 312 and 313 are turned off during the dead time DT1, but since the direct current flows to the positive bus via the diode 314, the NPN transistor 312 is substantially turned on during the dead time DT1. It will be. The same applies to the dead times DT2 to DT4.

  As a result, the substantial on-duty DT1 of the NPN transistor 312 is longer than the on-duty DT in the signal PWU, and the NPN transistor 312 is turned on / off according to the signal PWU1.

  Further, when the reactor current is negative, the direct current flows in the direction of the negative and negative buses of the DC power supply B, the lower arm (NPN transistor 313) of the bidirectional converter 310, the reactor 311, the positive bus and the DC power supply B. Then, as described above, since the NPN transistors 312 and 313 have the dead times DT1 to DT4, the direct current flows through the diode 315 during the dead times DT1 to DT4.

  As a result, the on-duty DT2 of the NPN transistor 313 is longer than the on-duty DT in the signal PWU, and the NPN transistor 313 is turned on / off according to the signal PWU2.

  Further, when the reactor current is zero, the reactor current does not flow regardless of whether the NPN transistors 312 and 313 are turned on or off, and therefore the reactor current does not flow during the dead times DT1 to DT4. As a result, the substantial on-duty DT0 of the NPN transistors 312 and 313 is the same as the on-duty DT in the signal PWU, and the NPN transistors 312 and 313 are turned on / off according to the signal PWU0.

  Since the ripple current Ir is superimposed on the reactor current IL, the time change of the reactor current IL near zero is, for example, as shown in FIG. Referring to FIG. 17, it is assumed that reactor current IL flows between timing t1 and timing t5. The reactor current IL is positive between timing t1 and timing t2, and between timing t4 and timing t5. The reactor current IL is zero between timing t2 and timing t3, and between timing t3 and timing t4. The reactor current IL is negative.

  The influence of the ripple current Ir on the reactor current IL becomes significant when the reactor current IL approaches zero. As described above, when the reactor current IL is positive or negative, the NPN transistor 312 or the NPN transistor 313 is substantially not affected. Since the typical on-duty DT1 or DT2 becomes longer due to the influence of the dead times DT1 to DT4 of the NPN transistors 312, 313, the fluctuation of the output voltage Vm due to the ripple current flowing during the dead times DT1 to DT4 increases. As a result, as shown in FIG. 18, the output voltage Vm of the bidirectional converter 310 rises and falls rapidly around the voltage command Vdccom between timing t6 and timing t7.

  Then, output voltage Vm of bidirectional converter 310 cannot be held at voltage command Vdccom.

  Thus, when the reactor current approaches zero, there arises a problem that the output voltage obtained by boosting the DC voltage from the DC power supply rapidly rises and falls around the voltage command due to the influence of the dead time and ripple current of the NPN transistor.

  Accordingly, the present invention has been made to solve such a problem, and an object of the present invention is to provide a voltage conversion device that reduces the influence of the dead time of a transistor and converts a DC voltage into an output voltage. .

  Another object of the present invention is to provide a voltage conversion method for converting a DC voltage into an output voltage by reducing the influence of the dead time of the transistor.

  Furthermore, another object of the present invention is to provide a computer-readable recording medium storing a program for causing a computer to execute control of voltage conversion for converting a DC voltage into an output voltage by reducing the influence of a transistor dead time. Is to provide.

  A voltage converter according to the present invention is a voltage converter that converts a DC voltage from a DC power source into the output voltage so that the output voltage becomes a command voltage. The voltage converter includes a voltage converter, a detection unit, and a control unit. Prepare. The voltage converter includes a switching element and outputs an output voltage by changing the voltage level of the DC voltage. The detecting means detects a reactor current flowing from the DC power source to the voltage converter. The control means compares the absolute value of the reactor current detected by the detection means with the peak-to-peak value of the ripple current, which is the difference between the maximum value and the minimum value of the reactor current, and according to the comparison result, The voltage converter is controlled by changing a carrier frequency for turning on / off a switching element included in the voltage converter.

  Preferably, when the absolute value of the reactor current is equal to or less than the peak-to-peak value of the ripple current, the control means reduces the carrier frequency and controls the voltage converter.

  Preferably, the control means further controls the voltage converter while keeping the duty ratio of the switching element constant.

  Preferably, when the absolute value of the reactor current is larger than the peak-to-peak value of the ripple current, the control means sets the carrier frequency to the optimum carrier frequency and controls the voltage converter.

  Preferably, the voltage conversion device further includes first and second voltage sensors. The first voltage sensor detects a DC voltage from a DC power source. The second voltage sensor detects the output voltage of the voltage converter. Then, the control means calculates the peak-to-peak value of the ripple current using the detected DC voltage and output voltage, and compares it with the peak-to-peak value of the ripple current calculated as the absolute value of the reactor current. .

  Preferably, the voltage converter includes two switching elements for the upper arm and the lower arm. The control means adjusts the detected DC voltage, output voltage, on-time of the switching element for the upper arm, on-time of the switching element for the lower arm, and the carrier period of the signal that controls the on / off of the two switching elements. Based on this, the peak-to-peak value of the ripple current is calculated.

  In addition, according to the present invention, a computer-readable recording medium recording a program for causing a computer to execute voltage conversion control for converting a DC voltage from a DC power source into an output voltage so that the output voltage becomes a command voltage Is a ripple current that is a difference between a first step of detecting a reactor current flowing from a DC power source to a voltage converter that performs voltage conversion, and a detected reactor current absolute value and a maximum value and a minimum value of the reactor current. A second step of comparing with a peak-to-peak value of the first and a third step of controlling the voltage converter by changing a carrier frequency for turning on / off a switching element included in the voltage converter according to the comparison result A computer-readable recording medium recording a program for causing a computer to execute steps.

  Preferably, in the third step, when the absolute value of the reactor current is equal to or less than the peak-to-peak value of the ripple current, the carrier voltage is lowered to control the voltage converter.

  Preferably, the third step further controls the voltage converter by setting the carrier frequency to the optimum carrier frequency when the absolute value of the reactor current is larger than the peak-to-peak value of the ripple current.

  Preferably, the program executes a fourth step of detecting a DC voltage and an output voltage from the DC power supply, and a fifth step of calculating a peak-to-peak value of the ripple current using the detected DC voltage and output voltage. These steps are further executed by the computer.

  Furthermore, according to the present invention, the voltage conversion method is a voltage conversion method for converting a DC voltage from a DC power source into the output voltage so that the output voltage becomes a command voltage, and the voltage conversion method performs voltage conversion from the DC power source. The first step of detecting the reactor current flowing to the converter and comparing the absolute value of the detected reactor current with the peak-to-peak value of the ripple current, which is the difference between the maximum value and the minimum value of the reactor current A second step includes a third step of controlling the voltage converter by changing a carrier frequency for turning on / off a switching element included in the voltage converter according to the comparison result.

  Preferably, in the third step, when the absolute value of the reactor current is equal to or less than the peak-to-peak value of the ripple current, the carrier frequency is lowered to control the voltage converter.

  Preferably, in the third step, when the absolute value of the reactor current is larger than the peak-to-peak value of the ripple current, the voltage converter is controlled by setting the carrier frequency to the optimum carrier frequency.

  Preferably, in the voltage conversion method, a fourth step of detecting a DC voltage and an output voltage from the DC power supply, and a peak-to-peak value of the ripple current is calculated using the detected DC voltage and output voltage. A fifth step.

  In this invention, the absolute value of the reactor current flowing from the DC power supply to the voltage converter is compared with the peak-to-peak value of the ripple current of the reactor current, and the absolute value of the reactor current is the peak-to-peak value of the ripple current. When the following is true, the carrier frequency for turning on / off the switching element included in the voltage converter is lowered, and the switching element is turned on / off by the lowered carrier frequency.

  Therefore, according to this invention, the influence by the dead time of a switching element can be reduced.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In the drawings, the same reference numerals indicate the same or corresponding parts.

  Referring to FIG. 1, motor drive device 100 including a voltage conversion device according to an embodiment of the present invention includes a DC power supply B, voltage sensors 10, 13, current sensors 11, 24, and system relays SR1, SR2. And capacitors C1 and C2, a boost converter 12, an inverter 14, and a control device 30.

  AC motor M1 is a drive motor for generating torque for driving drive wheels of a hybrid vehicle or an electric vehicle. Alternatively, this motor has the function of a generator driven by an engine, and operates as an electric motor for the engine, for example, can be incorporated into a hybrid vehicle so that the engine can be started. Also good.

  Boost converter 12 includes a reactor L1, NPN transistors Q1, Q2, and diodes D1, D2. Reactor L1 has one end connected to the power supply line of DC power supply B, and the other end connected to an intermediate point between NPN transistor Q1 and NPN transistor Q2, that is, between the emitter of NPN transistor Q1 and the collector of NPN transistor Q2. The NPN transistors Q1 and Q2 are connected in series between the power supply line and the earth line. The collector of NPN transistor Q1 is connected to the power supply line, and the emitter of NPN transistor Q2 is connected to the ground line. In addition, diodes D1 and D2 that flow current from the emitter side to the collector side are connected between the collectors and emitters of the NPN transistors Q1 and Q2.

  Inverter 14 includes a U-phase arm 15, a V-phase arm 16, and a W-phase arm 17. U-phase arm 15, V-phase arm 16, and W-phase arm 17 are provided in parallel between the power supply line and the earth line.

  The U-phase arm 15 includes NPN transistors Q3 and Q4 connected in series, the V-phase arm 16 includes NPN transistors Q5 and Q6 connected in series, and the W-phase arm 17 includes NPN transistors Q7 and Q7 connected in series. Consists of Q8. Further, diodes D3 to D8 that flow current from the emitter side to the collector side are connected between the collectors and emitters of the NPN transistors Q3 to Q8, respectively.

  An intermediate point of each phase arm is connected to each phase end of each phase coil of AC motor M1. In other words, AC motor M1 is a three-phase permanent magnet motor, and is configured such that one end of three coils of U, V, and W phases is commonly connected to the middle point, and the other end of the U-phase coil is NPN transistor Q3. The other end of the V-phase coil is connected to the intermediate point of NPN transistors Q5 and Q6, and the other end of the W-phase coil is connected to the intermediate point of NPN transistors Q7 and Q8, respectively.

  The DC power source B is composed of a secondary battery such as nickel metal hydride or lithium ion. Voltage sensor 10 detects DC voltage Vb output from DC power supply B, and outputs the detected DC voltage Vb to control device 30. System relays SR1 and SR2 are turned on / off by signal SE from control device 30. More specifically, system relays SR1 and SR2 are turned on by H (logic high) level signal SE from control device 30 and turned off by L level signal SE from control device 30.

  Capacitor C1 smoothes the DC voltage supplied from DC power supply B, and supplies the smoothed DC voltage to boost converter 12. Current sensor 11 detects reactor current IL flowing through reactor L1 of boost converter 12, and outputs the detected reactor current IL to control device 30.

  Boost converter 12 boosts the DC voltage supplied from capacitor C1 and supplies the boosted voltage to capacitor C2. More specifically, when boosting converter 12 receives signal PWU from control device 30, boosting converter 12 boosts the DC voltage according to the period during which NPN transistor Q2 is turned on by signal PWU, and supplies the boosted voltage to capacitor C2.

  When boost converter 12 receives signal PWD from control device 30, boost converter 12 steps down the DC voltage supplied from inverter 14 via capacitor C 2 and charges DC power supply B. However, it goes without saying that boost converter 12 may be applied to a circuit configuration that performs only a boost function.

  Capacitor C <b> 2 smoothes the DC voltage from boost converter 12 and supplies the smoothed DC voltage to inverter 14. The voltage sensor 13 detects the voltage across the capacitor C2, that is, the output voltage Vm of the boost converter 12 (corresponding to the input voltage to the inverter 14; the same applies hereinafter), and the detected output voltage Vm is controlled by the control device 30. Output to.

  When the DC voltage is supplied from the capacitor C2, the inverter 14 converts the DC voltage into an AC voltage based on the signal PWMI from the control device 30 and drives the AC motor M1. As a result, AC motor M1 is driven so as to generate torque specified by torque command value TR. Further, the inverter 14 converts the AC voltage generated by the AC motor M1 into a DC voltage based on the signal PWMC from the control device 30 during regenerative braking of the hybrid vehicle or electric vehicle on which the motor drive device 100 is mounted, The converted DC voltage is supplied to boost converter 12 via capacitor C2. Note that regenerative braking here refers to braking with regenerative power generation when the driver driving a hybrid vehicle or electric vehicle performs foot braking, or turning off the accelerator pedal while driving, although the foot brake is not operated. This includes decelerating the vehicle (or stopping acceleration) while generating regenerative power.

  Current sensor 24 detects motor current MCRT flowing through AC motor M <b> 1 and outputs the detected motor current MCRT to control device 30.

  The control device 30 includes a torque command value TR and a motor rotational speed MRN input from an ECU (Electrical Control Unit) provided outside, a DC voltage Vb from the voltage sensor 10, an output voltage Vm from the voltage sensor 13, and a current. Based on the motor current MCRT from the sensor 24, a signal PWU for driving the boost converter 12 and a signal PWMI for driving the inverter 14 are generated by a method described later, and the generated signal PWU and signal PWMI are respectively generated. Output to boost converter 12 and inverter 14.

  The signal PWU is a signal for driving the boost converter 12 when the boost converter 12 converts the DC voltage from the capacitor C1 into the output voltage Vm. Then, when boost converter 12 converts DC voltage Vb to output voltage Vm, control device 30 performs feedback control on output voltage Vm, and controls boost converter 12 so that output voltage Vm becomes commanded voltage command Vdccom. A signal PWU for driving is generated. A method for generating the signal PWU will be described later.

  Control device 30 also adjusts the carrier frequency for turning on / off NPN transistors Q1, Q2 of boost converter 12. A method for adjusting the carrier frequency will be described later.

  Further, when control device 30 receives a signal indicating that the hybrid vehicle or the electric vehicle has entered the regenerative braking mode from an external ECU, signal PWMC for converting the AC voltage generated by AC motor M1 into a DC voltage. Is output to the inverter 14. In this case, the NPN transistors Q3 to Q8 of the inverter 14 are switching-controlled by the signal PWMC. Thereby, the inverter 14 converts the AC voltage generated by the AC motor M <b> 1 into a DC voltage and supplies it to the boost converter 12.

  Further, when receiving a signal indicating that the hybrid vehicle or the electric vehicle has entered the regenerative braking mode from the external ECU, the control device 30 generates a signal PWD for stepping down the DC voltage supplied from the inverter 14, The generated signal PWD is output to boost converter 12. As a result, the AC voltage generated by AC motor M1 is converted into a DC voltage, stepped down, and supplied to DC power supply B.

  Furthermore, control device 30 generates signal SE for turning on / off system relays SR1, SR2 and outputs the signal SE to system relays SR1, SR2.

  FIG. 2 is a functional block diagram of the control device 30. Referring to FIG. 2, control device 30 includes motor torque control means 301 and voltage conversion control means 302. The motor torque control means 301 is obtained by calculating a torque command value TR (obtained by calculating a torque command to be applied to the motor in consideration of the degree of depression of the accelerator pedal in the vehicle, and the operating state of the engine in a hybrid vehicle), DC Based on DC voltage Vb output from power supply B, motor current MCRT, motor rotation speed MRN, and output voltage Vm of boost converter 12, when AC motor M1 is driven, NPN transistors Q1, Q2 of boost converter 12 are driven by a method described later. Is generated, and signal PWMI for turning on / off NPN transistors Q3-Q8 of inverter 14 is generated, and the generated signal PWU and signal PWMI are supplied to boost converter 12 and inverter 14, respectively. Output.

  Further, the motor torque control means 301 calculates the peak-to-peak value Ir of the ripple current using the DC voltage Vb and the output voltage Vm by a method described later. Then, the motor torque control means 301 determines whether or not the absolute value of the reactor current IL received from the current sensor 11 is smaller than the peak-to-peak value Ir of the ripple current. When the absolute value of the reactor current IL is larger than the peak-to-peak value Ir of the ripple current, the motor torque control means 301 determines the carrier frequency fc of the signal PWU for turning on / off the NPN transistors Q1 and Q2 as the optimum carrier frequency fcopt. When the absolute value of the reactor current IL is equal to or less than the peak-to-peak value Ir of the ripple current (except IL = 0), the carrier frequency fc of the signal PWU is set to the optimum carrier frequency fcopt by a method described later. Lower carrier frequency fcdec.

  When the signal RGE indicating that the hybrid vehicle or the electric vehicle has entered the regenerative braking mode is received from an external ECU during regenerative braking, the voltage conversion control means 302 converts the AC voltage generated by the AC motor M1 into a DC voltage. Signal PWMC is generated and output to inverter 14.

  Further, when regenerative braking, voltage conversion control means 302 receives signal RGE from an external ECU, generates signal PWD for stepping down the DC voltage supplied from inverter 14 and outputs the signal to step-up converter 12. Thus, the boost converter 12 can also lower the DC voltage by the signal PWD for stepping down the DC voltage, and thus has a bidirectional converter function.

  FIG. 3 is a functional block diagram of the motor torque control means 301. Referring to FIG. 3, motor torque control means 301 includes motor control phase voltage calculation unit 40, inverter PWM signal conversion unit 42, inverter input voltage command calculation unit 50, feedback voltage command calculation unit 52, A duty ratio conversion unit 54 and a frequency adjustment unit 56 are included.

  Motor control phase voltage calculation unit 40 receives output voltage Vm of boost converter 12, that is, an input voltage to inverter 14 from voltage sensor 13, and receives motor current MCRT flowing in each phase of AC motor M <b> 1 from current sensor 24. The torque command value TR is received from the external ECU. The motor control phase voltage calculation unit 40 calculates the voltage to be applied to the coils of each phase of the AC motor M1 based on these input signals, and the calculated result is the inverter PWM signal conversion unit 42. To supply.

  Based on the calculation result received from the motor control phase voltage calculation unit 40, the inverter PWM signal conversion unit 42 generates a signal PWMI that actually turns on / off each of the NPN transistors Q3 to Q8 of the inverter 14, and generates the signal PWMI. The signal PWMI is output to the NPN transistors Q3 to Q8 of the inverter 14.

  Thereby, each NPN transistor Q3-Q8 is switching-controlled, and controls the electric current sent through each phase of AC motor M1 so that AC motor M1 may output the commanded torque. In this way, the motor drive current is controlled, and a motor torque corresponding to the torque command value TR is output.

  On the other hand, inverter input voltage command calculation unit 50 calculates an optimum value (target value) of the inverter input voltage based on torque command value TR and motor rotational speed MRN, that is, voltage command Vdccom, and calculates the calculated voltage command Vdccom. Output to the feedback voltage command calculation unit 52.

  Feedback voltage command calculation unit 52 calculates feedback voltage command Vdccom_fb based on output voltage Vm of boost converter 12 from voltage sensor 13 and voltage command Vdccom from inverter input voltage command calculation unit 50. Feedback voltage command Vdccom_fb is output to duty ratio converter 54.

  The duty ratio conversion unit 54 outputs the voltage from the voltage sensor 13 based on the battery voltage Vb from the voltage sensor 10, the output voltage Vm from the voltage sensor 13, and the feedback voltage command Vdccom_fb from the feedback voltage command calculation unit 52. The duty ratio DR for setting the voltage Vm to the feedback voltage command Vdccom_fb from the feedback voltage command calculation unit 52 is calculated, and the calculated duty ratio DR is output to the frequency adjustment unit 56. Then, duty ratio converter 54 generates a signal PWU for turning on / off NPN transistors Q1 and Q2 of boost converter 12 based on calculated duty ratio DR and carrier frequency fc from frequency adjusting unit 56. The generated signal PWU is output to NPN transistors Q1 and Q2 of boost converter 12.

  Note that increasing the on-duty of the NPN transistor Q2 on the lower side of the boost converter 12 increases the power storage in the reactor L1, so that a higher voltage output can be obtained. On the other hand, increasing the on-duty of the upper NPN transistor Q1 reduces the voltage of the power supply line. Therefore, by controlling the duty ratio of the NPN transistors Q1 and Q2, the voltage of the power supply line can be controlled to an arbitrary voltage equal to or higher than the output voltage of the DC power supply B.

  The frequency adjustment unit 56 uses the DC voltage Vb from the voltage sensor 10 and the output voltage Vm from the voltage sensor 13 to calculate the peak-to-peak value Ir of the ripple current by a method described later. The received reactor current IL is compared with the peak-to-peak value Ir of the calculated ripple current. Then, when the absolute value of the reactor current IL is larger than the peak-to-peak value Ir of the ripple current, the frequency adjusting unit 56 sets the carrier frequency fc to the optimum carrier frequency fcopt corresponding to each reactor current IL, and the reactor When the absolute value of the current IL is equal to or less than the peak-to-peak value Ir of the ripple current, the carrier frequency fc is set to a carrier frequency fcdec lower than the optimum carrier frequency fcopt.

  The optimum carrier frequency fcopt will be described with reference to FIG. 4 and FIG. FIG. 4 shows the relationship (solid line) between the carrier frequency fc for each reactor current IL flowing through the reactor L1 and the loss of the reactor L1, and the carrier frequency fc for each reactor current IL and the losses (switching loss) of the NPN transistors Q1 and Q2. The relationship (broken line) is shown.

  As shown by the solid line in FIG. 4, the loss of the reactor L1 increases as the reactor current IL increases or the carrier frequency fc decreases. On the other hand, as indicated by the broken line in FIG. 4, the loss of the NPN transistors Q1 and Q2 increases as the reactor current IL increases or the carrier frequency fc increases.

  Considering the loss of boost converter 12 as the sum of the loss of reactor L1 and the losses of NPN transistors Q1 and Q2, the loss characteristic of boost converter 12 for each reactor current IL is the characteristic indicated by the broken line in FIG. Therefore, the optimum carrier frequency fcopt that minimizes the loss of boost converter 12 exists for each reactor current IL. Then, boost converter 12 can be driven efficiently by setting optimum carrier frequency fcopt corresponding to each reactor current IL.

  The solid line in FIG. 5 represents a map showing the relationship between the reactor current IL and the optimum carrier frequency fcopt. Therefore, the frequency adjustment unit 56 holds a map indicating the relationship between the reactor current IL and the optimum carrier frequency fcopt, and when the absolute value of the reactor current IL is larger than the peak-to-peak value Ir of the ripple current, The optimum carrier frequency fcopt corresponding to the reactor current IL received from the current sensor 11 is determined with reference to the map. Then, the frequency adjustment unit 56 outputs the determined optimum carrier frequency fcopt to the duty ratio conversion unit 54.

  When reactor current IL approaches zero and the absolute value of reactor current IL becomes equal to or less than the peak-to-peak value Ir of ripple current, output voltage Vm of boost converter 12 becomes voltage due to the dead time of NPN transistors Q1 and Q2. It fluctuates up and down around the command Vdcccom. In order to reduce the influence of the dead time of the NPN transistors Q1 and Q2, the frequency adjustment unit 56 sets the carrier frequency fc to the optimum carrier when the absolute value of the reactor current IL is equal to or less than the peak-to-peak value Ir of the ripple current. The carrier frequency is changed to a carrier frequency fcdec lower than the frequency fcopt. In this case, the duty ratio DR is kept constant.

  Lowering the frequency while maintaining the duty ratio DR is equivalent to increasing the period, so the on-duty of the NPN transistors Q1 and Q2 is also increased. As a result, the ratio of the dead time to the on-duty of the NPN transistors Q1 and Q2 is reduced, and the influence of the dead time is reduced.

  That is, referring to FIG. 6, when carrier frequency fc of NPN transistors Q 1 and Q 2 is set to optimum carrier frequency fcopt, boost converter 12 receives signal PWUop, and NPN transistors Q 1 and Q 2 It is assumed that it is turned on / off by PWUUop and PWUDop. In this case, the on-duty of the NPN transistors Q1 and Q2 is TLOp, and the dead time of the NPN transistors Q1 and Q2 is TLd.

  When carrier frequency fc is changed to carrier frequency fcdec lower than optimum carrier frequency fcop, boost converter 12 receives signal PWUD, and NPN transistors Q1 and Q2 are turned on / off by signals PWUud and PWUDd, respectively. The

  The period T2 of the signal PWUd is longer than the period T1 of the signal PWUop. As a result, the on-duty TLOd of the NPN transistors Q1 and Q2 becomes longer as the cycle T2 becomes longer than the cycle T1.

  Then, the ratio TLd / TLOD of the dead time TLd to the on-duty TLOd is smaller than the ratio TLd / TLOP of the dead time TLd to the on-duty TLOp. That is, by reducing the carrier frequency fc from the optimum carrier frequency fcopt to the carrier frequency fcdec, the influence of the dead time of the NPN transistors Q1 and Q2 can be reduced.

  Next, a method for determining the carrier frequency fcdec will be described. Voltage fluctuation width ρ due to dead time of output voltage Vm of boost converter 12 is expressed by the following equation.

  However, D is the on-duty of the NPN transistors Q1 and Q2, Td is the dead time of the NPN transistors Q1 and Q2, and K is a constant. The constant K has a different value depending on each mode of the hybrid vehicle or the electric vehicle on which the motor driving device 100 is mounted. For example, when the hybrid vehicle or electric vehicle is in the power running mode, K = −1, when in the regeneration mode, K = 1, and when the reactor current IL is zero, K = 0.

  Then, when the carrier frequency fc is obtained from the equation (1), the following equation is obtained.

  Equation (2) represents the maximum frequency fcmax of the carrier frequency that can eliminate the influence of the dead time even if the dead time occurs in the NPN transistors Q1 and Q2. Therefore, by substituting the on-duty D, the dead time Td, the constant K, and the voltage fluctuation width ρ into the equation (2), the maximum frequency fcmax of the carrier frequency that can eliminate the influence of the dead time can be calculated.

  For example, by substituting constant K = 1, dead time Td = 5 μsec, on-duty D = 0.4, and voltage fluctuation rate ρ = 0.4 into equation (2), carrier frequency fc that can eliminate the influence of dead time The maximum frequency fcmax is 15238 Hz. Therefore, if the carrier frequency is set to 15 kHz or less, the influence of the dead time of NPN transistors Q1 and Q2 can be reduced.

  FIG. 7 shows the relationship between the on-duty D and the maximum frequency fcmax of the carrier frequency fc calculated using Equation (2) for each voltage fluctuation width ρ. In FIG. 7, the horizontal axis represents the on-duty D, and the vertical axis represents the maximum frequency fcmax of the carrier frequency fc. In FIG. 7, the constant K is fixed to “1”, and the dead time Td is fixed to 5 μsec.

  Referring to FIG. 7, curve k1 shows the case where voltage fluctuation width ρ is 0.6, curve k2 shows the case where voltage fluctuation width ρ is 0.4, and curve k3 shows voltage fluctuation width ρ. The case of 0.2 is shown.

  As the on-duty D increases, the maximum frequency fcmax of the carrier frequency fc that can eliminate the influence of the dead time increases. Further, at the same on-duty, the maximum frequency fcmax of the carrier frequency fc that can eliminate the influence of the dead time increases as the voltage fluctuation width ρ increases. The voltage fluctuation width ρ indicates an error of the output voltage Vm with respect to the voltage command Vdccom. Therefore, when the output voltage Vm may deviate greatly from the voltage command Vdccom, the influence of the dead time is set even if the carrier frequency fc is set high. Can be removed. In addition, the calculation example mentioned above shows the point A on the curve k2.

  Equation (2) shows the maximum carrier frequency that can eliminate the influence of dead time, but any carrier frequency can be selected as long as the carrier frequency is lower than the carrier frequency shown by Equation (2). There is no lower limit for the carrier frequency to be selected.

  If the carrier frequency is continuously lowered, the on-duty of the NPN transistor Q2 constituting the lower arm of the boost converter 12 becomes sufficiently long. Then, during the period when the NPN transistor Q2 is turned on, an overcurrent flows through a closed circuit including the DC power source B, the positive bus, the reactor L1, the NPN transistor Q2, and the negative bus, and the NPN transistor Q2 is damaged. Therefore, the lower limit value fcmin of the carrier frequency fc is set so that the current flowing through the NPN transistor Q2 does not exceed the maximum current value allowed for the NPN transistor Q2.

  Therefore, the frequency adjustment unit 56 holds the range of the maximum frequency fcmax and the minimum frequency fcmin of the carrier frequency described above for each on-duty D, and the absolute value of the reactor current IL is the peak-to-peak of the ripple current. When it is determined that the peak value is equal to or less than the Ir value, the carrier frequency fcdec is selected from the carrier frequency range fcmin to fcmax corresponding to the on-duty set at that time, and is output to the duty ratio converter 54.

Next, a method for obtaining the ripple current Ir of the reactor current IL will be described.
Referring to FIG. 8, reactor current IL periodically changes up and down when ripple current Ir is superimposed. The maximum values of the reactor current IL are In (K-1), In (K), In (K + 1),..., And the minimum values of the reactor current IL are Ip (K-1), Ip (K), Ip ( K + 1),. The time when the NPN transistor Q1 of the boost converter 12 is turned on is Tp, and the time when the NPN transistor Q2 is turned on is Tn. The sum of time Tp and time Tn corresponds to one cycle of signal PWU.

  Then, the currents In (K) and Ip (K) and the voltage Vc (K) across the capacitor C2 are expressed by the following equations.

  In Expression (3), R represents the resistance of the circuit from the DC power supply B to the boost converter 12, L represents the inductance of the reactor L1, and C represents the capacitance of the capacitor C2.

  In Equation (3), when K → ∞ and In (K) = In (K + 1), Ip (K) = Ip (K + 1) and Vc (K) = Vc (K + 1) = Vm, Equation (3) The peak-to-peak value Ir of the ripple current is expressed by the following equation.

  Therefore, the resistance R of the circuit from the DC power supply B to the boost converter 12, the DC voltage Vb output from the DC power supply B, the capacitance C of the capacitor C2, the inductance L of the reactor L1, the output voltage Vm across the capacitor C2, and the time Tp , Tn is substituted into the equation (4), the peak-to-peak value Ir of the ripple current can be obtained.

  Among these values, the resistance R, the capacitance C, and the inductance L are known values because they are fixed values for the wiring, the capacitor C2, and the reactor L1, and the DC voltage Vb is detected by the voltage sensor 10, and the output voltage Vm is detected by the voltage sensor 13. Further, the times Tp and Tn can be obtained from the carrier frequency fc (that is, the carrier period) of the signal PWU and the duty ratio DR.

  Therefore, the frequency adjustment unit 56 calculates the times Tp and Tn based on the optimum carrier frequency fcopt corresponding to the reactor current IL from the current sensor 11 and the duty ratio DR from the duty ratio conversion unit 54. The frequency adjusting unit 56 holds the resistance R of the circuit from the DC power supply B to the boost converter 12, the capacitance C of the capacitor C2, and the inductance L of the reactor L1, and the held resistance R, capacitance C, and inductance L. Then, the calculated times Tp and Tn, the DC voltage Vb of the voltage sensor 10 and the output voltage Vm from the voltage sensor 13 are substituted into the equation (4) to calculate the peak-to-peak value Ir of the ripple current. .

  As a result, the peak-to-peak value Ir of the ripple current that is the comparison target of the reactor current IL can be obtained.

  As described above, the frequency adjusting unit 56 includes the circuit resistance R from the DC power supply B to the boost converter 12, the DC voltage Vb output from the DC power supply B, the capacitance C of the capacitor C2, the inductance L of the reactor L1, and the capacitor C2. The peak-to-peak value Ir of the ripple current is calculated based on the output voltage Vm at both ends and the times Tp and Tn. Then, the frequency adjusting unit 56 determines whether or not the absolute value of the reactor current IL is larger than the peak-to-peak value Ir of the ripple current, and the absolute value of the reactor current IL is determined to be the peak-to-peak of the ripple current. When the value Ir is greater than the value Ir, the carrier frequency of the NPN transistors Q1 and Q2 is set to the optimum carrier frequency fcopt, and when the absolute value of the reactor current IL is equal to or less than the peak-to-peak value Ir of the ripple current, The carrier frequency of Q2 is set to a carrier frequency fcdec (fcmin to fcmax) lower than the optimum carrier frequency fcopt.

  That is, as shown in FIGS. 9 and 10, when the reactor current IL is larger than the peak-to-peak value Ir of the ripple current (IL> Ir), or the reactor current IL is the peak-to-peak value of the ripple current. When smaller than −Ir, the frequency adjustment unit 56 sets the optimum carrier frequency fcopt corresponding to the reactor current IL detected by the current sensor 11 as the carrier frequency fc of the NPN transistors Q1 and Q2. As shown in FIG. 11, when the reactor current IL is in the range of −Ir ≦ IL ≦ Ir (except IL = 0), the frequency adjustment unit 56 uses the carrier frequency fc of the NPN transistors Q1 and Q2. Is set to a carrier frequency fcdec (fcmin to fcmax) lower than the optimum carrier frequency fcopt.

  With reference to FIG. 12, the operation of adjusting the carrier frequency in the frequency adjusting unit 56 will be described. When the operation of adjusting the carrier frequency is started, the frequency adjusting unit 56 determines the optimum carrier frequency fcopt corresponding to the reactor current IL received from the current sensor 11 with reference to the map shown by the solid line in FIG. Then, the frequency adjustment unit 56 outputs the determined optimum carrier frequency fcopt to the duty ratio conversion unit 54 (step S10).

  Then, the duty ratio conversion unit 54 generates the output voltage Vm based on the feedback voltage command Vdccom_fb from the feedback voltage command calculation unit 52, the DC voltage Vb from the voltage sensor 10, and the output voltage Vm from the voltage sensor 13. The duty ratio DR for setting the feedback voltage command Vdccom_fb is calculated, and the calculated duty ratio DR is output to the frequency adjustment unit 56. Then, duty ratio conversion unit 54 generates signal PWU based on the calculated duty ratio DR and optimum carrier frequency fcopt from frequency adjustment unit 56 and outputs the signal PWU to boost converter 12.

  As a result, NPN transistors Q1 and Q2 of boost converter 12 are switching-controlled by signal PWU having an optimum carrier frequency fcopt corresponding to each reactor current IL, and output voltage Vm is set to voltage command Vdccom to output DC voltage Vb. Convert to voltage Vm.

  Thereafter, the frequency adjustment unit 56 calculates the times Tp and Tn based on the optimum carrier frequency fcopt output to the duty ratio conversion unit 54 in step S10 and the duty ratio DR from the duty ratio conversion unit 54. Further, the frequency adjusting unit 56 substitutes the resistance R, the capacitance C, the inductance L, the calculated times Tp and Tn, the DC voltage Vb from the voltage sensor 10 and the output voltage Vm from the voltage sensor 13 into the equation (4). The peak-to-peak value Ir of the ripple current is calculated.

  Then, the frequency adjustment unit 56 determines whether or not the reactor current IL is larger than the peak-to-peak value Ir of the ripple current, or whether or not the reactor current IL is smaller than the peak-to-peak value −Ir of the ripple current. Is determined (step S11). When the frequency adjustment unit 56 determines in step S11 that the reactor current IL is larger than the peak-to-peak value Ir of the ripple current, or the reactor current IL is larger than the peak-to-peak value -Ir of the ripple current. When it is determined that the value is smaller, steps S10 and S11 are repeatedly executed.

  On the other hand, when it is determined in step S11 that the reactor current IL is included in the range of −Ir to Ir (excluding IL = 0), the frequency adjustment unit 56 uses the above-described method to determine the optimum carrier frequency fcopt. The carrier frequency fcdec that is lower than the frequency and included in the range of fcmin to fcmax is selected and output to the duty ratio converter 54 (step S12).

  Then, the duty ratio conversion unit 54 generates the output voltage Vm based on the feedback voltage command Vdccom_fb from the feedback voltage command calculation unit 52, the DC voltage Vb from the voltage sensor 10, and the output voltage Vm from the voltage sensor 13. The duty ratio DR for setting the feedback voltage command Vdccom_fb is calculated, and the calculated duty ratio DR is output to the frequency adjustment unit 56. Then, duty ratio conversion unit 54 generates signal PWUd based on the calculated duty ratio DR and carrier frequency fcdec from frequency adjustment unit 56 and outputs the signal PWUd to boost converter 12.

  Thereby, NPN transistors Q1 and Q2 of boost converter 12 are switching-controlled by signal PWUd having carrier frequency fcdec lower than optimum carrier frequency fcopt, and boost converter 12 reduces the influence of dead time of NPN transistors Q1 and Q2. Then, the DC voltage Vb is converted into the output voltage Vm. Thus, the operation for adjusting the carrier frequency is completed.

  The routine for adjusting the carrier frequency shown in FIG. 12 is repeatedly executed every predetermined time, for example, every 20 μsec.

  As described above, according to the present invention, it is determined whether or not the reactor current IL is included in the range of −Ir to Ir (except IL = 0) at regular time intervals, and the reactor current IL is −Ir to When included in the range of Ir, the booster converter 12 is controlled by setting the carrier frequency fc of the NPN transistors Q1 and Q2 to a carrier frequency fcdec (: fcmin to fcmax) lower than the optimum carrier frequency fcopt, and the NPN transistors Q1 and Q2 Reduce the effect of dead time.

  With reference to FIG. 1 again, the operation in the motor drive device 100 will be described. When torque command value TR is input from an external ECU, control device 30 generates H-level signal SE for turning on system relays SR1 and SR2, and outputs the signal to system relays SR1 and SR2. DC power supply B outputs DC voltage Vb, and system relays SR1 and SR2 supply DC voltage Vb to capacitor C1. Capacitor C1 smoothes the supplied DC voltage Vb and supplies the smoothed DC voltage Vb to boost converter 12.

  Then, control device 30 selects optimal carrier frequency fcopt corresponding to reactor current IL received from current sensor 11, and boost converter 12 and inverter so that AC motor M1 generates torque specified by torque command value TR. 14 is generated and output to boost converter 12 and inverter 14, respectively.

  Then, NPN transistors Q1 and Q2 of boost converter 12 are turned on / off in response to signal PWU from control device 30, converts DC voltage Vb into output voltage Vm, and supplies the same to capacitor C2. The voltage sensor 13 detects an output voltage Vm that is a voltage across the capacitor C <b> 2 and outputs the detected output voltage Vm to the control device 30.

  Capacitor C <b> 2 smoothes the DC voltage supplied from boost converter 12 and supplies it to inverter 14. NPN transistors Q3 to Q8 of inverter 14 are turned on / off in accordance with signal PWMI from control device 30, inverter 14 converts a DC voltage into an AC voltage, and AC motor M1 converts the torque specified by torque command value TR. A predetermined alternating current is passed through each of the U-phase, V-phase, and W-phase of AC motor M1 so as to be generated. Thereby, AC motor M1 generates torque specified by torque command value TR.

  Then, according to the flowchart shown in FIG. 12, the control device 30 executes a routine for adjusting the carrier frequency at regular intervals, and the reactor current IL from the current sensor 11 falls within a range of −Ir ≦ IL ≦ Ir (however, (Except IL = 0), the carrier frequency fc of the signal PWU is set to a carrier frequency fcdec lower than the optimum carrier frequency fcopt, and the influence of the dead time of the NPN transistors Q1 and Q2 is reduced.

  When the hybrid vehicle or electric vehicle on which the motor driving device 100 is mounted enters the regenerative braking mode, the control device 30 receives a signal indicating that the regenerative braking mode has been entered from the external ECU, and receives the signal PWMC and the signal PWD. And output to inverter 14 and boost converter 12, respectively.

  AC motor M <b> 1 generates AC voltage and supplies the generated AC voltage to inverter 14. Then, inverter 14 converts an AC voltage into a DC voltage in accordance with signal PWMC from control device 30, and supplies the converted DC voltage to boost converter 12 via capacitor C2.

  Boost converter 12 steps down the DC voltage in accordance with signal PWD from control device 30 and supplies it to DC power supply B to charge DC power supply B.

  Thus, in motor drive device 100, when reactor current IL is in a range of −Ir ≦ IL ≦ Ir (except IL = 0), carrier frequency fc of signal PWU is greater than optimal carrier frequency fcopt. Is set to a lower carrier frequency fcdec, and the influence of the dead time of the NPN transistors Q1 and Q2 is reduced.

  In the present invention, the voltage sensors 10 and 13, the current sensor 11, the boost converter 12, the feedback voltage command calculation unit 52, the duty ratio conversion unit 54 and the frequency adjustment unit 56 of the control device 30 are “voltage conversion devices”. Constitute.

  In the present invention, feedback voltage command calculation unit 52, duty ratio conversion unit 54, and frequency adjustment unit 56 constitute “control means” for controlling boost converter 12 as a voltage converter.

  Furthermore, the voltage conversion method according to the present invention is a voltage conversion method for adjusting the carrier frequency of the signal PWU according to the flowchart shown in FIG. 12 and converting the DC voltage into the output voltage Vm.

  Further, feedback control in feedback voltage command calculation unit 52, duty ratio conversion unit 54, and frequency adjustment unit 56 is actually performed by a CPU (Central Processing Unit), and the CPU includes the steps of the flowchart shown in FIG. Voltage conversion for reading a program from a ROM (Read Only Memory), converting the DC voltage to the output voltage Vm while executing the read program and adjusting the carrier frequency for switching the NPN transistors Q1 and Q2 according to the flowchart shown in FIG. To control. Therefore, the ROM corresponds to a computer (CPU) readable recording medium in which a program including each step of the flowchart shown in FIG. 12 is recorded.

  Further, the motor drive device provided with the voltage conversion device according to the embodiment of the present invention may be a motor drive device 100A shown in FIG. Referring to FIG. 13, motor drive device 100A is obtained by adding current sensor 28 and inverter 31 to motor drive device 100 and replacing control device 30 of motor drive device 100 with control device 30A. This is the same as the motor drive device 100.

  Capacitor C2 receives output voltage Vm from step-up converter 12 via nodes N1 and N2, smoothes the received output voltage Vm and supplies it not only to inverter 14 but also to inverter 31. Current sensor 24 detects motor current MCRT1 and outputs it to control device 30A. Further, inverter 14 converts the DC voltage from capacitor C2 into an AC voltage based on signal PWMI1 from control device 30A to drive AC motor M1, and uses the AC voltage generated by AC motor M1 based on signal PWMC1. Convert to DC voltage.

  The inverter 31 has the same configuration as the inverter 14. The inverter 31 converts the DC voltage from the capacitor C2 into an AC voltage based on the signal PWMI2 from the control device 30A to drive the AC motor M2, and the AC voltage generated by the AC motor M2 based on the signal PWMC2. Is converted to a DC voltage. Current sensor 28 detects motor current MCRT2 flowing in each phase of AC motor M2, and outputs the detected current to control device 30A.

  Control device 30A receives DC voltage Vb output from DC power supply B from voltage sensor 10, receives reactor current IL from current sensor 11, receives motor currents MCRT1 and MCRT2 from current sensors 24 and 28, respectively, and boost converter 12 Output voltage Vm (that is, input voltage to inverters 14 and 31) is received from voltage sensor 13, and torque command values TR1 and TR2 and motor rotational speeds MRN1 and MRN2 are received from an external ECU. Then, control device 30A controls inverter 14 when inverter 14 drives AC motor M1 by the above-described method based on DC voltage Vb, output voltage Vm, motor current MCRT1, torque command value TR1, and motor rotational speed MRN1. A signal PWMI1 for switching control of NPN transistors Q3 to Q8 is generated, and the generated signal PWMI1 is output to inverter 14.

  Further, control device 30A controls inverter 31 when inverter 31 drives AC motor M2 by the above-described method based on DC voltage Vb, output voltage Vm, motor current MCRT2, torque command value TR2, and motor rotational speed MRN2. A signal PWMI2 for switching control of NPN transistors Q3 to Q8 is generated, and the generated signal PWMI2 is output to inverter 31.

  Furthermore, when inverter 14 (or 31) drives AC motor M1 (or M2), control device 30A provides DC voltage Vb, output voltage Vm, reactor current IL, motor current MCRT1 (or MCRT2), torque command value TR1. Based on (or TR2) and motor rotational speed MRN1 (or MRN2), signal PWU for switching control of NPN transistors Q1 and Q2 of boost converter 12 is set to optimum carrier frequency fcopt by the above-described method, and boost converter 12 is set. Output to.

  Furthermore, when reactor current IL is in a range of −Ir ≦ IL ≦ Ir (except IL = 0), control device 30A sets carrier frequency fc of signal PWU to a carrier frequency fcdec lower than optimal carrier frequency fcopt. To reduce the influence of the dead time of the NPN transistors Q1 and Q2.

  Further, control device 30A generates signal PWMC1 for converting the AC voltage generated by AC motor M1 during regenerative braking into a DC voltage, or signal PWMC2 for converting the AC voltage generated by AC motor M2 into a DC voltage. Then, the generated signal PWMC1 or signal PWMC2 is output to the inverter 14 or the inverter 31, respectively. In this case, control device 30 </ b> A generates signal PWD for controlling boost converter 12 so as to step down the DC voltage from inverter 14 or 31 and charge DC power supply B, and outputs the signal to boost converter 12.

  Further, control device 30A generates signal SE for turning on / off system relays SR1, SR2 and outputs the signal SE to system relays SR1, SR2.

  The overall operation in the motor drive device 100A will be described. When the entire operation is started and torque command values TR1 and TR2 are input from an external ECU, control device 30A generates H-level signal SE for turning on system relays SR1 and SR2, and system relay SR1. , SR2. DC power supply B outputs DC voltage Vb, and system relays SR1 and SR2 supply DC voltage Vb to capacitor C1. Capacitor C1 smoothes the supplied DC voltage Vb and supplies the smoothed DC voltage Vb to boost converter 12.

  Then, control device 30A selects optimal carrier frequency fcopt corresponding to reactor current IL received from current sensor 11, and AC motors M1 and M2 generate torques specified by torque command values TR1 and TR2, respectively. Signals PWU and signals PWMI1 and PWMI2 for controlling boost converter 12 and inverter 14 are generated and output to boost converter 12 and inverters 14 and 31, respectively.

  NPN transistors Q1 and Q2 of boost converter 12 are turned on / off in response to signal PWU from control device 30A, convert DC voltage Vb into output voltage Vm, and supply the same to capacitor C2. The voltage sensor 13 detects an output voltage Vm that is a voltage across the capacitor C <b> 2 and outputs the detected output voltage Vm to the control device 30.

  Capacitor C <b> 2 smoothes the DC voltage supplied from boost converter 12 and supplies it to inverters 14 and 31. NPN transistors Q3 to Q8 of inverter 14 are turned on / off in accordance with signal PWMI1 from control device 30A. Inverter 14 converts a DC voltage into an AC voltage, and AC motor M1 converts the torque specified by torque command value TR1. A predetermined alternating current is passed through each of the U-phase, V-phase, and W-phase of the AC motor M1 so as to be generated. Thereby, AC motor M1 generates a torque specified by torque command value TR1.

  Further, NPN transistors Q3 to Q8 of inverter 31 are turned on / off in accordance with signal PWMI2 from control device 30A, and inverter 31 converts a DC voltage into an AC voltage and converts the torque specified by torque command value TR2 to an AC motor. A predetermined alternating current is passed through each of the U phase, V phase, and W phase of the AC motor M2 so that M2 is generated. Thereby, AC motor M2 generates torque specified by torque command value TR2.

  Then, according to the flowchart shown in FIG. 12, the control device 30A executes a routine for adjusting the carrier frequency at regular intervals, and the reactor current IL from the current sensor 11 falls within a range of −Ir ≦ IL ≦ Ir (however, (Except IL = 0), the carrier frequency fc of the signal PWU is set to a carrier frequency fcdec lower than the optimum carrier frequency fcopt, and the influence of the dead time of the NPN transistors Q1 and Q2 is reduced.

  Further, at the time of regenerative braking of a hybrid vehicle or an electric vehicle on which motor drive device 100A is mounted, control device 30A receives signal RGE from the external ECU and generates signals PWMC1 and PWMC2 according to the received signal RGE. Output to inverters 14 and 31, respectively, generate signal PWD and output to boost converter 12.

  Then, inverter 14 converts the AC voltage generated by AC motor M1 into a DC voltage according to signal PWMC1, and supplies the converted DC voltage to boost converter 12 via capacitor C2. Inverter 31 also converts the AC voltage generated by AC motor M2 into a DC voltage according to signal PWMC2, and supplies the converted DC voltage to boost converter 12 via capacitor C2.

  Boost converter 12 receives the DC voltage from capacitor C2 via nodes N1 and N2, steps down the received DC voltage with signal PWD, and supplies the reduced DC voltage to DC power supply B. As a result, the power generated by the AC motor M1 or M2 is charged to the DC power source B.

  Thus, even when there are two AC motors, when the reactor current IL is in the range of −Ir ≦ IL ≦ Ir (except IL = 0), the carrier frequency fc of the signal PWU is the optimum carrier frequency fcopt. Is set to a lower carrier frequency fcdec, and the influence of the dead time of the NPN transistors Q1 and Q2 is reduced.

  In the above description, the DC current flowing from DC power supply B to boost converter 12 is referred to as reactor current IL. This DC current is synonymous with the DC current output from DC power supply B.

  In the above description, the reactor current IL is detected by the current sensor 11. However, in the present invention, the reactor current IL is applied to the output power of the AC motor M1 (or M2) and the output voltage Vm of the boost converter 12. You may make it ask | require based on. Alternatively, the output power of AC motor M1 (or M2) may be calculated from the torque of AC motor M1 (or M2), and reactor current IL may be obtained based on the calculated output power.

  Further, in the above description, the frequency adjustment unit 56 includes the circuit resistance R from the DC power source B to the boost converter 12, the DC voltage Vb output from the DC power source B, the capacitance C of the capacitor C2, the inductance L of the reactor L1, and the capacitor. Although it has been described that the peak-to-peak value Ir of the ripple current is calculated based on the output voltage Vm at both ends of C2 and the times Tp and Tn, in the present invention, the peak-to-peak value Ir of the ripple current is R is calculated in advance based on R, capacitance C, inductance L, DC voltage Vb, output voltage Vm, and times Tp and Tn, and the peak-to-peak value Ir of the calculated ripple current is held in the frequency adjustment unit 56. You may comprise.

  Further, in the above, the optimum carrier frequency fcopt of the signal PWU for turning on / off the NPN transistors Q1 and Q2 of the boost converter 12 is represented by the reactor current IL detected by the current sensor 11 and the map shown by the solid line in FIG. In the present invention, the optimum carrier frequency fcopt may be selected according to the flowchart shown in FIG.

  Referring to FIG. 14, when the operation of selecting optimum carrier frequency fcopt is started, frequency adjustment unit 56 performs torque command value TR (or TR1, TR2), motor rotational speed MRN (or MRN1, MRN2), and The battery voltage Vb is received (step S100), and based on the received torque command value TR and the motor rotation speed MRN, the output request BP * of the DC power source B, which is the power necessary for driving the AC motor M1, is calculated ( Step S102). Note that the output request BP * of the DC power supply B is calculated as power converted from the power from the AC motor M1.

  After that, the frequency adjustment unit 56 divides the calculated output request BP * by the DC voltage Vb from the voltage sensor 10 to calculate the target reactor current IL * that flows through the reactor L1 of the boost converter 12 (step S104). Based on the target reactor current IL *, the optimum carrier frequency fcopt, which is the optimum carrier frequency for efficiently switching the NPN transistors Q1 and Q2 of the boost converter 12, is set (step S106). Then, using this optimum carrier frequency fcopt, boost converter 12 is controlled so that target reactor current IL * flows to reactor L1 (step S108), and the routine for selecting the optimum carrier frequency ends.

  When the carrier frequency fc is set to the optimum carrier frequency fcopt according to the flowchart shown in FIG. 14, the operation for selecting the optimum carrier frequency in step S10 in FIG. 12 is performed according to the flowchart shown in FIG.

  The AC motors M1 and M2 are AC motors that can function as generators during regenerative braking as described above. Preferably, the AC motor generates power by the rotational force of the crankshaft of the engine when the engine is driven. It may function as a machine. Thus, as an aspect in which the AC motor functions as a generator when the engine is driven, the AC motor M1 functions as a motor that generates torque for driving the drive wheels, and the AC motor M2 rotates the crankshaft of the engine. This is a case of functioning as a generator that generates power by force.

  Alternatively, a hybrid system in which AC motors M1, M2 and the output shaft of the engine are connected to a planetary gear may be configured.

  Further, AC motors M1 and M2 may be provided corresponding to different drive wheels of the vehicle.

Further, other AC motors may be appropriately added in parallel to the AC motors M1 and M2.
According to the embodiment of the present invention, the voltage converter that changes the voltage level of the DC voltage by the switching element and outputs the output voltage, the detection means that detects the reactor current flowing from the DC power supply to the voltage converter, and the reactor current Control means for controlling the voltage converter by lowering the carrier frequency for turning on / off the switching element when the absolute value of the ripple current is less than the peak-to-peak value of the ripple current. The impact can be reduced.

  The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

It is a schematic block diagram of the motor drive device provided with the voltage converter by embodiment of this invention. It is a functional block diagram of the control apparatus shown in FIG. It is a functional block diagram for demonstrating the function of the motor torque control means shown in FIG. It is a relationship diagram of loss and carrier frequency. It is a relationship figure of the loss of a boost converter, and carrier frequency. It is a timing chart of a signal. It is a relationship diagram between the on-duty of a transistor and a carrier frequency. It is a waveform diagram of a ripple current. It is a wave form diagram of a reactor current and a ripple current. It is a wave form diagram of a reactor current and a ripple current. It is a wave form diagram of a reactor current and a ripple current. It is a flowchart for demonstrating the operation | movement which adjusts a carrier frequency. It is another schematic block diagram of the motor drive device provided with the voltage converter by embodiment of this invention. It is a flowchart for demonstrating the operation | movement which calculates | requires the optimal carrier frequency. It is a schematic block diagram of the conventional motor drive device. It is a timing chart of the control signal of the bidirectional converter shown in FIG. It is a wave form diagram of a reactor current. FIG. 5 is a waveform diagram of an output voltage for explaining an influence due to a dead time of a transistor.

Explanation of symbols

  10, 13, 320 Voltage sensor, 11, 24, 28 Current sensor, 12 Boost converter, 14, 31, 330 Inverter, 15 U-phase arm, 16 V-phase arm, 17 W-phase arm, 30, 30A Control device, 40 motor Control phase voltage calculation unit, 42 PWM signal conversion unit, 50 inverter input voltage command calculation unit, 52 feedback voltage command calculation unit, 54 duty ratio conversion unit, 56 frequency adjustment unit, 100, 100A, 300 motor drive device, 301 motor Torque control means, 302 voltage conversion control means, 310 bidirectional converter, 311, L1 reactor, 312, 313, Q1-Q8 NPN transistor, 314, 315, D1-D8 diode, C1, C2 capacitor, M1, M2 AC motor, SR1, SR2 system Relay.

Claims (7)

A voltage converter for converting a DC voltage from a DC power source into the output voltage so that the output voltage becomes a command voltage,
A voltage converter that includes a switching element and outputs an output voltage by changing a voltage level of the DC voltage;
Detecting means for detecting a reactor current flowing from the DC power source to the voltage converter;
The absolute value of the detected reactor current is compared with the peak-to-peak value of the ripple current, which is the difference between the maximum value and the minimum value of the reactor current, and the switching element is turned on according to the comparison result. Control means for controlling the voltage converter by changing the carrier frequency to be turned off ,
When the absolute value of the reactor current is larger than the peak-to-peak value of the ripple current, the control means sets the carrier frequency to an optimum carrier frequency for efficiently switching the switching element, When the absolute value of the reactor current is equal to or less than the peak-to-peak value of the ripple current, the carrier frequency is set to a carrier frequency lower than the optimum carrier frequency while maintaining the duty ratio of the switching element. there are, and to set such that the following maximum frequency of the carrier frequency that can remove the influence of dead time, the voltage conversion device. The control means is set such that, for each on-duty of the switching element, the maximum frequency of the carrier frequency and the current flowing through the switching element do not exceed the maximum current value allowed for the switching element. Holds the minimum carrier frequency,
When the absolute value of the reactor current is equal to or less than the peak-to-peak value of the ripple current, the maximum frequency of the carrier frequency corresponding to the on-duty set at that time and the minimum frequency of the carrier frequency The voltage converter according to claim 1 , wherein the carrier frequency is set within a range . A first voltage sensor for detecting a DC voltage from the DC power supply;
A second voltage sensor for detecting an output voltage of the voltage converter,
The control means calculates the peak-to-peak value of the ripple current using the detected DC voltage and output voltage, and calculates the absolute value of the reactor current from the calculated peak-to-peak value of the ripple current. The voltage converter of Claim 1 or Claim 2 compared with. The voltage converter includes two switching elements for an upper arm and a lower arm,
The control means controls the detected DC voltage, the output voltage, the on-time of the upper arm switching element, the on-time of the lower arm switching element, and the on / off of the two switching elements. The voltage converter according to claim 3 , wherein a peak-to-peak value of the ripple current is calculated based on a carrier period of a signal to be transmitted. A voltage conversion method for converting a DC voltage from a DC power source into the output voltage so that the output voltage becomes a command voltage,
A first step of detecting a reactor current flowing from the DC power source to a voltage converter that performs the voltage conversion;
A second step of comparing the absolute value of the detected reactor current with a peak-to-peak value of a ripple current, which is a difference between a maximum value and a minimum value of the reactor current;
Depending on the comparison result, it is seen including a third step of switching elements included in said voltage converter by changing the on / off the carrier frequency for controlling said voltage converter,
In the third step, when the absolute value of the reactor current is larger than the peak-to-peak value of the ripple current, the carrier frequency is set to an optimum carrier frequency for efficiently switching the switching element. When the absolute value of the reactor current is equal to or less than the peak-to-peak value of the ripple current, the carrier frequency is set to a carrier lower than the optimum carrier frequency while maintaining the duty ratio of the switching element. A voltage conversion method in which the frequency is set to be equal to or lower than the maximum carrier frequency that can eliminate the influence of dead time . The third step is set so that the maximum frequency of the carrier frequency and the current flowing through the switching element do not exceed the maximum current value allowed for the switching element for each on-duty of the switching element. Holding the minimum frequency of the carrier frequency,
When the absolute value of the reactor current is equal to or less than the peak-to-peak value of the ripple current, the maximum frequency of the carrier frequency corresponding to the on-duty set at that time and the minimum frequency of the carrier frequency The voltage conversion method according to claim 5 , wherein the carrier frequency is set within a range . A fourth step of detecting a DC voltage from the DC power supply and the output voltage;
The voltage conversion method according to claim 5 , further comprising a fifth step of calculating a peak-to-peak value of the ripple current using the detected DC voltage and output voltage.

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