JP2013090401A - Rotating electrical machine control system - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a rotating electrical machine control system that effectively prevents the occurrence of an overcurrent and an overvoltage even if an error is superimposed in a current value detected by a current sensor.SOLUTION: A rotating electrical machine control system 10 comprises: a motor generator MG2 as a rotating electrical machine; a DC/DC converter 14 including a reactor 20; smoothing capacitors C1, C2 connected to the DC/DC converter 14; and a control section 18 for controlling an inverter in a sinusoidal PWM control mode, an overmodulation control mode or a rectangular control mode. The control section 18 has a voltage drop control section 30 for reducing an input voltage VH to the inverter 16 and switching the mode of controlling the inverter 16 from the sinusoidal PWM control mode to the overmodulation control mode or the rectangular control mode when the frequency of power fluctuations of the motor generator MG2 matches a frequency in a resonance frequency range of an LC resonance circuit during the execution of the sinusoidal PWM control mode.

Description

  The present invention relates to a rotating electrical machine control system, and more particularly to a rotating electrical machine control system including a driving circuit that drives the rotating electrical machine and a control unit that switches a control method of the driving circuit according to a modulation rate.

  As a method of controlling a rotating electrical machine used as a motor or a generator, it is known to use a sine wave PWM (Pulse Width Modulation) control mode, an overmodulation control mode, and a rectangular wave control mode. The overmodulation control mode may be referred to as an overmodulation PWM control mode.

  For example, Patent Document 1 discloses an electric motor drive control device, in which a target operating point of a motor is included in a predetermined resonance region including a motor operating point when resonance occurs in a boost converter. It describes that the boost converter is controlled so that the voltage becomes higher than the voltage on the battery side, and that the inverter is controlled using a sine wave PWM control system. In this configuration, a resonant circuit is configured by the reactor of the boost converter and the smoothing capacitor connected to the boost converter, and when the operating point of the motor enters a predetermined region, resonance of voltage and current occurs, and the boost converter and The purpose is to prevent an excessive voltage or current from being applied to the smoothing capacitor.

JP 2009-225633 A

  However, when the inverter is controlled using the sine wave PWM control method as described above, there is a possibility that an offset error may be superimposed on the detected current value of the current sensor of the motor that is a rotating electrical machine due to product variation of the sensor. . In this case, since control is performed with the detection value of the current sensor being positive, the d-axis current and the q-axis current, which are actual currents, fluctuate in electrical primary, and power fluctuations that fluctuate in electrical primary occur in the motor. Furthermore, since the LC resonance circuit is configured by the reactor of the boost converter, which is a voltage conversion unit, and the smoothing capacitor, LC resonance occurs. In this case, when the frequency region of the LC resonance and the frequency of the power fluctuation coincide with each other, the high-voltage side voltage VH that is the boosted voltage greatly fluctuates, and the battery current IB that is the battery output current also fluctuates greatly. In particular, when the rotating electrical machine is controlled by sine wave PWM control, the control V is highly followable, and thus the fluctuations in the voltage VH and the current IB become significant. In addition, when the capacity of the smoothing capacitor is small or when the internal resistance of the battery is small, fluctuations in the voltage VH and the current IB become more remarkable. Thus, when both the voltage VH and the current IB are increased and an overcurrent and an overvoltage are generated, an effective measure is required to prevent exceeding a preset design component protection area value. For example, effective measures are required to prevent an overcurrent or overvoltage that exceeds a component protection threshold of a power control unit (PCU) including an inverter and a smoothing capacitor. For this reason, it is desired to realize a rotating electrical machine control system that can effectively prevent the occurrence of overvoltage and overcurrent related to the high-voltage side voltage VH that is the output voltage of the boost converter and the battery current IB.

  An object of the present invention is to effectively prevent the occurrence of an overcurrent and an overvoltage even when an error is superimposed on a detected current value of a current sensor in a rotating electrical machine control system.

  A rotating electrical machine control system according to the present invention includes a rotating electrical machine, a voltage converter connected to a DC power source and including a reactor, a drive circuit connected to the rotating electrical machine, and a smoothing capacitor connected to the voltage converter. When the modulation factor, which is the ratio of the effective value of the line voltage that is the necessary applied voltage of the rotating electrical machine to the input voltage of the driving circuit, is equal to or less than a preset first ratio, the driving is performed by a sine wave PWM control method. And a control unit that controls the drive circuit with an overmodulation control method or a rectangular wave control method when the modulation rate exceeds the first ratio, and the control unit controls the sine wave PWM control. When executing the method, when the frequency of the resonance frequency region of the resonance circuit including the reactor and the smoothing capacitor matches the frequency of the output fluctuation of the rotating electrical machine, the drive circuit A rotating electrical machine control system comprising: a voltage reduction control unit that reduces a force voltage and switches a control method of the drive circuit from the sine wave PWM control method to the overmodulation control method or the rectangular wave control method. .

  In the rotating electrical machine control system according to the present invention, it is preferable that the voltage drop control unit is configured such that the rotational speed of the rotating electrical machine is based on a frequency in the resonance frequency region when the sine wave PWM control method is executed. When the frequency in the resonance frequency region and the frequency of the output fluctuation of the rotating electrical machine coincide with each other when the frequency is within a specific rotational speed region set in advance, the input voltage of the drive circuit is reduced, The drive circuit control method is switched from the sine wave PWM control method to the overmodulation control method or the rectangular wave control method.

  In the rotating electrical machine control system according to the present invention, it is preferable that the voltage drop control unit is configured such that the rotational speed of the rotating electrical machine is based on a frequency in the resonance frequency region when the sine wave PWM control method is executed. Ripple components are removed from the actual current value that is the actual current value when the absolute value of the output of the rotating electrical machine is larger than the preset predetermined output even when it is within the preset specific rotation speed range. The input voltage of the drive circuit is reduced only when the absolute value of the difference between the filtered current value that has been filtered and the actual current value is greater than a predetermined current value that is set in advance. The control method is switched from the sine wave PWM control method to the overmodulation control method or the rectangular wave control method.

  According to the rotating electrical machine control system of the present invention, the frequency of the resonance frequency region of the resonant circuit including the reactor and the smoothing capacitor coincides with the frequency of the output fluctuation of the rotating electrical machine when the sine wave PWM control method is executed. Sometimes, the input voltage of the drive circuit is lowered, and the control method of the drive circuit is switched from the sine wave PWM control method to the overmodulation control method or the rectangular wave control method. For this reason, even when an error is superimposed on the detected current value of the current sensor that detects the current about the rotating electrical machine, the fluctuation of the input current to the voltage conversion unit in the resonance frequency region can be suppressed to a small level. Generation of overvoltage can be effectively prevented.

It is a circuit diagram which shows an example of the rotary electric machine control system of embodiment of this invention. It is a block diagram which shows the structure of the control part of FIG. In an embodiment of the present invention, it is a figure showing the relation between the number of rotations of a rotary electric machine, and torque for explaining change of the control method of a rotary electric machine. It is a figure which shows a mode that the sine wave PWM control part of FIG. 2 controls an inverter by a sine wave PWM control system. It is a figure which shows the relationship between the elapsed time in the case of accelerating rotation of a motor generator with constant acceleration, and a battery electric current value in the rotary electric machine control system of a comparative example. 3 is a flowchart illustrating an example of a method for determining a VH upper limit voltage that is an input voltage of an inverter used in the modulation degree control switching unit of FIG. 2. It is a flowchart which shows the subroutine which determines ON (= 1) or OFF (= 0) of the electric current fluctuation determination flag by S10 of FIG. It is a flowchart which shows the subroutine which determines ON (= 1) or OFF (= 0) of LC resonance zone determination flag by S10 of FIG.

  Embodiments according to the present invention will be described below in detail with reference to the drawings. In the following, a motor generator mounted on a vehicle will be described as a rotating electrical machine, but a rotating electrical machine used for purposes other than mounting on a vehicle may be used. Further, the rotating electrical machine may be used for an electric vehicle or a fuel cell vehicle mounted on a vehicle that simply functions as a motor.

  In the following, the same or corresponding elements in all drawings are denoted by the same reference numerals, and redundant description is omitted. In the description in the text, the symbols described before are used as necessary.

  FIG. 1 is a circuit diagram showing an example of a rotating electrical machine control system according to an embodiment of the present invention. The rotating electrical machine control system 10 is used by being mounted on a hybrid vehicle that uses one or both of an engine (not shown) and a motor generator MG2 that is a rotating electrical machine mainly used as a traveling motor as a main drive source. Such a rotating electrical machine control system 10 includes a motor generator MG2, a battery 12 that is a DC power source and a power storage unit, a DC / DC converter 14 that is a voltage conversion unit connected to the battery 12, and a DC / DC converter 14 Are provided with a smoothing capacitor C1 and a second smoothing capacitor C2, an inverter 16 serving as a drive circuit connected between the step-up side of the DC / DC converter 14 and the motor generator MG2, and a control unit 18.

  Note that the hybrid vehicle includes another motor generator MG1 (not shown) that is mainly driven by an engine and is used as a generator, and another inverter that drives another motor generator MG1. The DC converter 14 may be connected in parallel to the inverter 16 on the boost side, and another motor generator MG1 may be connected to another inverter. Although the control part 18 of this embodiment can control another inverter and another motor generator MG1 similarly, in the following description, the case where the inverter 16 and the motor generator MG2 are controlled will be described as a representative.

  Motor generator MG2 is a U-phase, V-phase, and W-phase three-phase rotating electric machine that functions as a motor when electric power is supplied from battery 12 and functions as a generator during braking of the vehicle. The generated electric power is supplied to the battery 12 via the inverter 16.

  The DC / DC converter 14 includes a reactor 20 and two switching elements Sa connected in series with each other. One end of the reactor 20 is connected between the two switching elements Sa, and the other end of the reactor 20 is connected to the positive electrode side of the battery 12 via the system relay SR and the fuse F. The switching element Sa is, for example, a transistor or an IGBT. In the illustrated example, the system relay SR includes a first relay R1 and a second relay R2 connected in parallel to the first relay R1 and having a resistor Rw connected in series. The control unit 18 enables the battery 12 and the DC / DC converter 14 to be electrically connected by controlling one of the first relay R1 and the second relay R2 to be turned on and the other to be turned on.

  A diode Da is connected in antiparallel to each switching element Sa of the DC / DC converter 14, and a negative electrode side of the battery 12 is connected to the switching element Sa on one side (the lower side in FIG. 1) of the two switching elements Sa. A smoothing capacitor C <b> 1 is connected between the other end of the reactor 20 and the negative electrode side of the battery 12. A second smoothing capacitor C <b> 2 is connected between both ends of the two switching elements Sa and the inverter 16.

  In such a DC / DC converter 14, switching of the switching element Sa is controlled by the control unit 18, and the high voltage side voltage VH obtained by boosting the low voltage side voltage VL that is the output side voltage of the battery 12 is supplied to the inverter 16. The high voltage side voltage VH supplied from the inverter 16 side is stepped down and supplied to the battery 12 side. For such control of the DC / DC converter 14, the rotating electrical machine control system 10 uses the low-voltage sensor 22 that detects the low-voltage side voltage VL of the DC / DC converter 14 and the high-voltage side voltage VH of the DC / DC converter 14. And a high-pressure sensor 24 for detection.

  The inverter 16 includes three arms Au, Av, Aw corresponding to the U-phase, V-phase, and W-phase connected in parallel with each other, and each phase arm Au, Av, Aw is connected to each other in series with a transistor, IGBT. The two switching elements Sw are included. A diode Di is connected to each switching element Sw in antiparallel. Switching of each switching element Sw is controlled by the control unit 18 to convert a DC voltage supplied from the high-voltage side of the DC / DC converter 14 into a three-phase AC voltage and output it to the motor generator MG2. When the vehicle is braked, the three-phase AC voltage output from the motor generator MG2 to the inverter 16 is converted into a DC voltage by the inverter 16, and the voltage is stepped down by the DC / DC converter 14 before being supplied to the battery 12. To charge.

  The rotating electrical machine control system 10 further includes a current sensor 28 that detects a current flowing through a power line that connects the stator coils 26u, 26v, and 26w of each phase of the motor generator MG2 and the inverter 16. The current sensor 28 will be described in detail later.

  Moreover, the control part 18 can be comprised by the vehicle-mounted computer, for example. The control unit 18 can be configured by one computer, but can also be configured by connecting a plurality of computers with cables or the like. For example, the control unit 18 can be a motor control unit that controls the operation of the motor generator MG2.

  FIG. 2 shows a part of the control unit 18 that performs motor control divided into functions. That is, the control unit 18 includes a sine wave PWM control unit 50, an overmodulation control unit 52, a rectangular wave control unit 54, a modulation degree control switching unit 56, a phase control switching unit 58, and a voltage drop control unit 30. including.

  Sine wave PWM control unit 50 controls motor generator MG2 by a sine wave PWM control method. Overmodulation control unit 52 controls motor generator MG2 by the overmodulation control method. The rectangular wave control unit 54 controls the motor generator MG2 by a rectangular wave control method.

In addition, the modulation degree control switching unit 56 performs switching of the control method among the sine wave PWM control mode, the overmodulation control mode, and the rectangular wave control mode, that is, the control mode, according to the modulation degree E that is the modulation rate. “Modulation E” is a line that is a necessary voltage applied to motor generator MG2 determined from a torque command value or the like with respect to high-voltage side voltage VH of DC / DC converter 14 that is a system voltage and an input voltage of inverter 16. This is the ratio (J / VH) of the effective value J of the inter-voltage. The effective value J of the line voltage of the motor generator MG2 is calculated using the d-axis voltage command value Vd ^* and the q-axis voltage command value Vq ^* as follows: J = {(Vd ^* ) ² + (Vq ^* ) ² } ^1/2 Given in. Therefore, the modulation degree E is obtained by the modulation degree E = [{(Vd ^* ) ² + (Vq ^* ) ² } ^1/2 ] / VH. When the modulation degree E is 0.61 or less which is a preset first ratio, the inverter 16 is controlled by the sine-wave PWM control method, and the modulation degree E exceeds 0.61 and is a preset second ratio. The modulation degree control switching unit 56 controls the inverter 16 so that the inverter 16 is controlled by the overmodulation control method when it is less than 0.78, and the inverter 16 is controlled by the rectangular wave control method when the modulation degree E is 0.78 or more. Switch. Note that in the general sine wave PWM control system, the maximum value of the modulation degree E can be increased only to 0.61. However, in the case of sinusoidal PWM control by the two-phase modulation method or the third harmonic superposition control, the maximum value of the modulation degree E can be increased to 0.70. In this case, the first ratio is 0.70. Thus, the first ratio is not limited to one value.

  The phase control switching unit 58 switches the control method for controlling the motor generator MG2, that is, the control mode from the rectangular wave control mode to the overmodulation control mode, and the operating point of the motor generator MG2 on the dq plane. Switch the control mode from. Here, the dq plane is for defining the operating point of the motor generator MG2 by the d axis and the q axis orthogonal to each other. For example, the d axis is a d axis current axis (Id axis) or a d axis voltage axis ( Vd axis), and q axis is Id axis or q axis current axis (Iq axis) or q axis voltage axis (Vd axis) perpendicular to Vd axis. Control unit 18 defines a maximum efficiency characteristic line obtained by connecting a current set of d-axis current and q-axis current that can be operated at maximum efficiency when current control is performed on motor generator MG2 on the dq plane. .

  Phase control switching portion 58 is set such that the operating point of motor generator MG2 is set on the retard side in the clockwise direction from the maximum efficiency characteristic line on the maximum voltage circle that is a circle centered on origin O on the dq plane. A function of switching from the rectangular wave control mode to the overmodulation control mode when the switching line is exceeded.

  Here, the sine wave PWM control method and the overmodulation control method are current feedback control, and are controls for outputting a PWM signal to the motor generator MG2 by comparing the voltage command value with a carrier wave. On the other hand, the rectangular wave control method is a control for outputting a one-pulse switching waveform to the motor generator MG2 in accordance with the electrical angle. The voltage amplitude is fixed to the maximum value, and the torque is feedback controlled by controlling the phase.

  FIG. 3 is a diagram for explaining how the control mode is selected according to the operating point of motor generator MG2. This figure shows the rotational speed of the motor generator MG2 on the horizontal axis, the torque on the vertical axis, its maximum torque characteristic line, and which control mode is used in the operation region indicated inside the maximum torque characteristic line. FIG. As shown in FIG. 3, a sine wave PWM control mode operation region is set on the low speed side, a rectangular wave control mode operation region on the high speed side, and an overmodulation control mode operation region in the middle.

  Next, switching between these three control modes will be described. As shown in FIG. 3, the control mode is switched according to the state of the operating point of motor generator MG2 given by the rotational speed and torque. As the speed and torque are gradually increased, the control mode is switched from the sine wave PWM control mode to the overmodulation control mode and from the overmodulation control mode to the rectangular wave control mode. For example, if the operating point of motor generator MG2 is at point α in FIG. 3, the sine wave PWM control mode is executed, and if it is at point β in FIG. 3, the overmodulation control mode is executed, and at point γ in FIG. In this case, the rectangular wave control mode is executed. In this case, the control mode can be switched according to the modulation degree E. That is, the sine wave PWM control mode is used when the modulation degree E is equal to or less than the first ratio such as 0.61, and overmodulation is performed until the modulation degree E exceeds the first ratio and is less than the second ratio such as 0.78. Using the control mode, the control mode is switched so that the rectangular wave control mode is used when the modulation degree E is equal to or greater than the second ratio.

  The degree of modulation E can also be used when switching the control mode in the opposite direction, but switching from the rectangular wave control mode to the overmodulation control mode has a constant voltage command amplitude in the rectangular wave control mode. This can be done by determining the switching timing based on the phase of the actual current with respect to the current command.

FIG. 4 is a diagram illustrating how the sine wave PWM control unit 50 of FIG. 2 controls the inverter 16 by the sine wave PWM control method. In FIG. 2, the sine wave PWM control unit 50 acquires a torque command value T ^* and a rotational angular velocity command value ω ^* from another control unit (not shown). These command values are calculated by estimating the user's required torque and required vehicle speed from an accelerator pedal operation amount, a brake pedal operation amount, etc. of a vehicle (not shown). The sine wave PWM control unit 50 includes a current command generation unit 34, a subtractor unit 36, a PI control unit 32, a 3-phase / 2-phase conversion unit 38, and a 2-phase / 3-phase conversion unit 40.

The current command generator 34 compares the actual rotational angular velocity ω of the motor generator MG2 with the rotational angular velocity command value ω ^*, and uses the previously created table or the like to determine the torque command value T ^* as the d-axis current command value Id ^*. And q-axis current command value Iq ^* .

The subtractor unit 36 subtracts the actual d-axis current value Id from the d-axis current command value Id ^* and calculates the d-axis current deviation δId, and the actual subtractor 36 from the q-axis current command value Iq ^*. And an Iq subtractor having a function of calculating a q-axis current deviation δIq by subtracting the q-axis current value Iq.

  The actual d-axis current value Id and the actual q-axis current value Iq in the motor generator MG2 are determined by the function of the three-phase / two-phase converter 38, and the rotation angle θ of the rotor of the motor generator MG2 and 3 of the motor generator MG2. It is calculated based on the detection values Iu, Iv, and Iw of the current sensor 28 that detects the current of the phase. The electrical angle of the rotor is detected by a rotation angle sensor MR such as a resolver. Current values Iu, Iv, Iw connect corresponding phase arms Au, Av, Aw (FIG. 1) of inverter 16 and corresponding phase stator coils 26u, 26v, 26w (FIG. 1) of motor generator MG2. It is obtained by detecting the current flowing through the power line. Since one end of each phase of the stator coils 26u, 26v, 26w of the motor generator MG2 is connected at a neutral point, the current values Iu for the remaining one phase can be calculated by detecting the currents Iv, Iw for two phases. It is. FIG. 1 shows that two of a V-phase current value Iv and a W-phase current value Iw are detected. The remaining U-phase current value Iu is obtained by Iu = − (Iv + Iw). It is also possible to provide current sensors 28 for three phases and directly detect the current values Iu, Iv, Iw for the three phases. The current sensor 28 has an accuracy capable of detecting the resonance frequency of the LC resonance circuit including the reactor 20 (FIG. 1) and the smoothing capacitors C1 and C2 (FIG. 1). Also, the high voltage sensor 24 for detecting the high voltage VH shown in FIG. 1 has an accuracy capable of detecting the resonance frequency of the LC resonance circuit.

The PI control unit 32 performs proportional-integral control on the d-axis current deviation δId and the q-axis current deviation δIq under a predetermined feedback gain G to obtain a control deviation corresponding to these, and d corresponding to the control deviation It has a function of calculating the shaft voltage command value Vd ^* and the q-axis voltage command value Vq ^* . Current subtraction in the PWM control mode is performed by the subtractor unit 36 and the PI control unit 32.

The two-phase / three-phase conversion unit 40 is obtained from the rotation angle θ of the rotor based on the d-axis voltage command value Vd ^* and the q-axis voltage command value Vq ^* input from the PI control unit 32. It has a function of calculating a U-phase voltage Vu, a V-phase voltage Vv, and a W-phase voltage Vw from a predicted angle that is predicted to be located after five control cycles.

  The calculated phase voltages Vu, Vv, Vw are converted into PWM signals by a PWM signal generator (not shown), and the PWM signals are output to a gate circuit (not shown). The gate circuit controls on / off of the switching element Sw by selecting the switching element Sw (FIG. 1) of the inverter 16 to which the control signal is applied.

  The current flowing through the power line connecting each phase arm of inverter 16 and each phase stator coil of motor generator MG2 is fed back to subtractor section 36 via three-phase / two-phase conversion section 38 as described above. . In this way, current feedback in the PWM control mode is performed.

  The above is the basic configuration of current feedback control. As shown in FIGS. 1 and 2, in the present embodiment, the control unit 18 further includes a voltage drop control unit 30. That is, when considering a comparative example that does not have the voltage drop control unit 30 with the above basic configuration, in the comparative example as well, the reactor 20 of the DC / DC converter 14 and the smoothing capacitors C1 and C2 Since the LC resonance circuit is configured by the above, if any frequency in the resonance frequency region matches the frequency of the power fluctuation that is the output fluctuation of the motor generator MG2, an overcurrent or an overvoltage may occur.

  FIG. 5 is a diagram illustrating a relationship between an elapsed time t when accelerating the rotation of the motor generator MG2 with a constant acceleration and a battery current value IB that is an output current value of the battery 12 in the rotating electrical machine control system of the comparative example. is there. The horizontal axis in FIG. 5 is the elapsed time t when accelerating the rotation of the motor generator MG2 with a constant acceleration, and can be replaced with the rotation speed N of the motor generator MG2. The vertical axis in FIG. 5 is the battery current value IB, and shows the change in IB when the rotational speed N is increased. In motor generator MG2, power fluctuation caused by the offset error of current sensor 28 occurs, and furthermore, LC resonance circuit is constituted by reactor 20 of DC / DC converter 14 and each of smoothing capacitors C1 and C2. Will occur.

  In this case, when the frequency region of the LC resonance and the frequency of the power fluctuation coincide with each other, a large fluctuation of the battery current IB based on the LC resonance occurs in the resonance rotational speed band indicated by the arrow L in FIG. In addition, large fluctuations in the high-voltage side voltage VH based on LC resonance also occur in the same resonance rotational speed band. That is, the fluctuation of the battery current IB and the fluctuation of the high-voltage side voltage VH due to the power fluctuation are expanded by the LC resonance phenomenon. As described above, when any frequency in the resonance frequency region matches the frequency of the power fluctuation, which is the output fluctuation of the motor generator MG2, an offset error deviating from the zero point is superimposed on the detection value of the current sensor 28. In addition, the fluctuation of the power fluctuation increases due to the LC resonance. As the fluctuation increases, the battery current IB that is the output current of the battery 12 and the high-voltage side voltage VH of the inverter that is the system voltage largely fluctuate, and an overcurrent or an overvoltage may occur.

  Thus, when the fluctuation of the battery current IB increases and the fluctuation of the high-voltage side voltage VH of the DC / DC converter 14, that is, the fluctuation of the system voltage of the inverter 16, the driving signal by the inverter 16 fluctuates, and the motor generator MG2 Control becomes unstable. Further, when the fluctuation of the battery current IB becomes large and exceeds the PCU protection threshold value X shown in FIG. 5, the performance of the components of the power control unit (PCU) including the inverter 16 and the smoothing capacitors C1 and C2 may be reduced. There is sex.

  In this embodiment, in order to solve such a problem in the comparative example, the control unit 18 includes a voltage drop control unit 30 as shown in FIGS. When the sinusoidal PWM control method is executed, the voltage drop control unit 30, as described above, when any frequency in the resonance frequency region of the LC resonance circuit coincides with the power fluctuation frequency of the motor generator MG2. The input voltage VH of the inverter, that is, the VH upper limit voltage is lowered to a preset predetermined voltage value or by a preset predetermined voltage. The inverter control method is modulated from the sine wave PWM control method by increasing the modulation degree E expressed by the ratio (J / VH) of the effective value J of the line voltage of the motor generator MG2 to the input voltage VH. The overmodulation control method or the rectangular wave control method is switched according to the degree E.

More preferably, the voltage drop control unit 30 specifies that the number of revolutions per predetermined time of the motor generator MG2, for example, the number of revolutions Arpm (= min ⁻¹ ) is preset based on the frequency in the resonance frequency region. When the current fluctuation determination flag is turned on (that is, becomes 1) and the absolute value | PM | of the power of the motor generator MG2 exceeds the predetermined power Pa (kw) (| PM) It is assumed that the frequency in the resonance frequency region and the power fluctuation frequency of motor generator MG2 coincide with |> Pa). In this case, the voltage drop control unit 30 reduces the input voltage VH of the inverter 16 and switches the control method of the inverter 16 from the sine wave PWM control method to the overmodulation control method or the rectangular wave control method.

  Here, the “current fluctuation determination flag” is the absolute difference between the actual current value IBa and the filtered current value IBf obtained by applying a filter to remove the ripple component to the actual current value IBa that is the actual battery current value IB. When the value (| IBa−IBf |) is larger than a predetermined current value Ic (A) set in advance (| IBa−IBf |> Ic), it is turned on (that is, becomes 1), otherwise it is turned off. (That is, 0). For this purpose, the detection value of the battery current sensor 33 (FIG. 1) that detects the battery current IB that is the output current of the battery 12 (FIG. 1) is input to the control unit 18.

Further, the voltage drop control unit 30 based on the angular velocity command value ω ^* and the torque command value T ^* in FIG. 4, the rotational speed Arpm (= min ⁻¹ ) of the motor generator MG2 and the absolute value of power | PM | And can be calculated. However, the detection value of the rotation angle sensor MR of the motor generator MG2 or the detection value of the rotation speed sensor that detects the rotation speed of the motor generator MG2 by using it instead of the rotation angle sensor MR is set to the rotation speed Arpm (= min ^{− 1} ) and the absolute value of power | PM |.

Further, when the absolute value | PM | of the power of the motor generator MG2 exceeds a predetermined power Pa (kw) (| PM |> Pa), and the rotational speed per predetermined time of the motor generator MG2 is the specific rotational speed. When in the region, the LC resonance band determination flag is turned on (that is, 1), and is otherwise turned off (that is, 0). For example, when it is between revolutions per minute of the motor generator MG2 Arpm (= min ^-1) is the specific rotation speed range T1rpm (= min ^-1) and T2rpm (= min ^-1), i.e. T1 < When | A | <T2, the LC resonance band determination flag is turned ON.

  Then, the voltage drop control unit 30 sets the LC resonance band in which the VH upper limit voltage is lower than the normal-time voltage used during normal operation when the current fluctuation determination flag is turned ON and the LC resonance band determination flag is turned ON. Determine the voltage. Modulation degree control switching unit 56 calculates modulation degree E using the determined VH upper limit voltage, and determines the control method of inverter 16 based on modulation degree E. Therefore, when the VH upper limit voltage is the LC resonance band voltage lower than the normal voltage, the modulation degree E is increased, and the inverter control method is changed from the sine wave PWM control method to the overmodulation control method or the rectangular wave control. Switch to the method.

  Next, an example of a method for determining the VH upper limit voltage that is the input voltage of the inverter 16 used in the modulation degree control switching unit 56 of FIG. 2 will be described using the flowcharts of FIGS. FIG. 7 is a flowchart showing a subroutine for determining whether the current fluctuation determination flag is ON (= 1) or OFF (= 0) in S10 of FIG. FIG. 8 is a flowchart showing a subroutine for determining whether the LC resonance band determination flag is ON (= 1) or OFF (= 0) in S10 of FIG.

  When determining the VH upper limit voltage, first, in step S10 of FIG. 6 (in the following description, “step S” is simply referred to as S), the current fluctuation determination flag is turned ON, and the LC resonance band determination flag is set. Whether or not is turned on is determined. For this purpose, the subroutine for determining the current fluctuation determination flag in FIG. 7 and the subroutine for determining the LC resonance band determination flag in FIG. 8 are executed.

  First, in S20 of FIG. 7, the absolute value (| IBa−IBf |) of the difference between the actual current value IBa, which is the actual battery 12 current value IB, and the filtered current value IBf is a predetermined current value Ic set in advance. It is determined whether it is larger than (A). If the determination result in S20 is affirmative, that is, if (| IBa-IBf |)> Ic, the current fluctuation determination flag is turned on (that is, becomes 1) (S22), and the determination result in S20 is negative. In this case, that is, if (| IBa−IBf |) ≦ Ic, the current fluctuation determination flag is turned off (that is, becomes 0) (S24).

Further, in S30 of FIG. 8, when the absolute value | PM | of the power of the motor generator MG2 exceeds the predetermined power Pa (kw) (| PM |> Pa) and the predetermined time of the motor generator MG2. The number of revolutions Arpm per minute (= min ⁻¹ ) is within a specific revolution number range (ie, between T1 rpm (= min ⁻¹ ) and T2 rpm (= min ⁻¹ )), that is, T1 <| A | < It is determined whether or not T2. When the determination result of S30 in FIG. 8 is affirmative, the LC resonance band determination flag is turned on (that is, becomes 1) (S32), and when the determination result of S30 in FIG. The determination flag is turned off (that is, 0) (S34). Note that the execution order of the subroutines of FIGS. 7 and 8 may be reversed.

  Next, when the determination result in S10 of FIG. 6 is affirmative, the VH upper limit voltage is determined to be the LC resonance band voltage VHL lower than the normal time voltage VHn (S12). On the other hand, if the determination result of S10 in FIG. 6 is negative, the VH upper limit voltage is determined to be the normal time voltage VHn (S14). Modulation degree control switching unit 56 shown in FIG. 2 calculates modulation degree E using the determined VH upper limit voltage, and determines the control method of motor generator MG2 according to the modulation degree E. Therefore, when the LC resonance band voltage VHL is determined as the VH upper limit voltage, the modulation degree E increases and the control method of the motor generator MG2 is changed from the sine wave PWM control method to the overmodulation control method or the rectangular wave control. The system is switched to the system, and the control response of the motor generator MG2 is deteriorated.

  Each function of the control unit 18 described above can be realized by executing software, but part of each function may be realized by hardware.

  According to the present embodiment, at the time of executing the sine wave PWM control system, any frequency in the resonance frequency region of the LC resonance circuit including the reactor 20 and the smoothing capacitors C1 and C2 and the power of the motor generator MG2 When the frequency of fluctuation coincides, the VH upper limit voltage that is the input voltage of the inverter 16 is reduced, and the control method of the inverter 16 is switched from the sine wave PWM control method to the overmodulation control method or the rectangular wave control method. The responsiveness of is worsened. For this reason, even when an offset error is superimposed on the detected current value of the current sensor 28 that detects the current for the motor generator MG2, the fluctuation of the input current to the inverter 16 in the resonance frequency region can be suppressed to a small level. Generation of current and overvoltage can be effectively prevented.

  In the present embodiment, the voltage drop control unit 30 determines the frequency of the resonance frequency region when the rotation number of the motor generator MG2 is within a specific rotation number region set in advance based on the frequency of the resonance frequency region. Assuming that the frequency of the power fluctuation of motor generator MG2 coincides, inverter input voltage VH is reduced. However, the present invention is not limited to such a configuration. For example, the rotational speed per unit time of an output shaft (not shown) or a wheel (not shown) connected to the motor generator MG2 so that power can be transmitted is within another specific rotational speed range that is preset based on the frequency in the resonance frequency range. In some cases, the inverter input voltage VH can also be reduced assuming that the frequency in the resonance frequency region matches the frequency of power fluctuations in the motor generator MG2.

  Further, in the present embodiment, voltage drop control unit 30 has a rotational speed of motor generator MG2 within a specific rotational speed range set in advance based on the frequency in the resonance frequency range (T1 <| A | <T2 However, the absolute value | PM | of the power of the motor generator MG2 is greater than the predetermined power Pa (kw) (| PM |> Pa), and the difference between the actual current value IBa and the filtered current value IBf The inverter input voltage VH is reduced only when the absolute value | IBa−IBf | of the current is larger than the predetermined current value Ic (| IBa−IBf |> Ic). That is, when | PM |> Pa is not satisfied, the power fluctuation is small, and when | IBa−IBf |> Ic is not satisfied, the fluctuation of the current ripple is small. Therefore, the control method is not the overmodulation control method or the rectangular wave control method. However, problems of overcurrent and overvoltage are unlikely to occur, and the control responsiveness can be increased by sine wave PWM control. For this reason, it is possible to effectively prevent the responsiveness of the control from being lowered excessively, and it is possible to improve the performance of equipment such as a vehicle equipped with the motor generator MG2.

  Unlike the present embodiment, the voltage drop control unit 30 performs the current fluctuation determination in FIG. 7 or determines whether or not the absolute value of the power of the motor generator MG2 in FIG. 8 is greater than a predetermined power. Without doing this, the inverter input voltage VH can also be lowered only by determining whether or not the rotational speed A of the motor generator MG2 is within the specific rotational speed region. In S12 and S14 of FIG. 6, the LC resonance band voltage and the normal voltage can be set so that a plurality of values are selected depending on whether or not a preset condition is satisfied. . For example, the storage unit included in the control unit 18 may store a plurality of LC resonance band voltages and normal voltage values.

  In the above description, the case where the voltage conversion unit is the DC / DC converter 14 having the step-up / step-down function has been described. However, the voltage conversion unit has only a function of boosting the voltage from the battery 12 side to the inverter 16 side. A step-up converter may be used, or a step-down converter having only a function of stepping down the voltage from the inverter 16 side to the battery 12 side.

  The rotating electrical machine control system according to the present invention can be used for controlling rotating electrical machines mounted on fuel cell vehicles, hybrid vehicles, and the like.

  DESCRIPTION OF SYMBOLS 10 Rotating electrical machine control system, 12 Battery, 14 DC / DC converter, 16 Inverter, 18 Control part, 20 Reactor, 22 Low voltage sensor, 24 High voltage sensor, 26u, 26v, 26w Stator coil, 28 Current sensor, 30 Voltage drop control part , 32 PI control unit, 33 battery current sensor, 34 current command generation unit, 36 subtractor unit, 38 3 phase / 2 phase conversion unit, 40 2 phase / 3 phase conversion unit, 50 sine wave PWM control unit, 52 overmodulation Control unit, 54 rectangular wave control unit, 56 modulation degree control switching unit, 58 phase control switching unit.

Claims (3)

Rotating electrical machinery,
A voltage converter connected to a DC power source and including a reactor;
A drive circuit connected to the rotating electrical machine;
A smoothing capacitor connected to the voltage converter;
When the modulation factor, which is the ratio of the effective value of the line voltage that is the required applied voltage of the rotating electrical machine to the input voltage of the drive circuit, is equal to or less than a preset first ratio, the drive circuit is controlled by a sine wave PWM control method. A control unit that controls the drive circuit in an overmodulation control method or a rectangular wave control method when the modulation rate exceeds the first ratio,
The controller is
When the sinusoidal PWM control method is executed, when the frequency in the resonance frequency region of the resonance circuit including the reactor and the smoothing capacitor matches the output fluctuation frequency of the rotating electrical machine, the input of the drive circuit A rotating electrical machine control system comprising: a voltage reduction control unit that reduces a voltage and switches a control method of the drive circuit from the sine wave PWM control method to the overmodulation control method or the rectangular wave control method. In the rotating electrical machine control system according to claim 1,
The voltage drop control unit, when executing the sine wave PWM control method, when the rotation speed of the rotating electrical machine is within a specific rotation speed range preset based on the frequency of the resonance frequency range, Assuming that the frequency in the resonance frequency region and the output fluctuation frequency of the rotating electrical machine coincide with each other, the input voltage of the drive circuit is reduced, and the control method of the drive circuit is changed from the sine wave PWM control method to the excess. A rotating electrical machine control system that switches to a modulation control system or the rectangular wave control system. In the rotating electrical machine control system according to claim 2,
The voltage drop control unit rotates at the time of execution of the sine wave PWM control method even when the rotation speed of the rotating electrical machine is within a specific rotation speed range set in advance based on the frequency of the resonance frequency range. When the absolute value of the output of the electric machine is larger than a predetermined output set in advance, and the filtered current value obtained by applying a filter that removes the ripple component to the actual current value that is the actual current value and the actual current value The input voltage of the drive circuit is lowered only when the absolute value of the difference between the two is larger than a predetermined current value set in advance, and the control method of the drive circuit is changed from the sine wave PWM control method to the overmodulation control method. Alternatively, the rotating electrical machine control system is switched to the rectangular wave control system.

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