JPS6333399B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a speed control device for an induction motor that can control the speed of the induction motor with sufficient accuracy without using a speed detector.

When an induction motor is driven using a variable frequency power supply device (hereinafter referred to as an inverter device), the rotational speed N of the motor is given by the following equation.

N=60/ _p0 ( ₁ -s)...(1) Here, _P0 : number of pole pairs, ₁ : drive frequency, s: slip frequency.

Since the slip frequency s in equation (1) changes in proportion to the torque, when the drive frequency ₁ is a constant value, the rotational speed N changes in accordance with the torque. Therefore, accurate speed control cannot be performed simply by keeping the drive frequency ₁ constant. On the other hand, high accuracy can be obtained by attaching a speed detector to the shaft end of the motor and controlling the drive frequency ₁ according to the deviation between the speed command and the signal from the speed detector. However, a speed detector is required. However, there are disadvantages such as the installation is troublesome, the structure around the motor is complicated, and a long distance cable is required for transmitting the speed signal.

Therefore, the slip frequency can be determined without using a speed detector.
s _is detected and the slip frequency is controlled so that the slip frequency s becomes a _{predetermined} value _.
A method has been proposed in which the slip frequency s is calculated from the ratio of the currents I ₁ and I _n . This is described, for example, in Japanese Unexamined Patent Publication No. 11413/1983.

However, the above-mentioned literature only describes that torque etc. can be stably controlled by calculating the slip frequency and controlling this slip frequency to a predetermined value, but there is nothing about accurately controlling the speed. Not disclosed.

An object of the present invention is to provide a speed control device for an induction motor that can perform speed control with high accuracy without using a speed detector.

The feature of the present invention is that the torque current component of the primary current is detected based on the phase command signal of the primary current supplied to the induction motor from the frequency converter and the output voltage of the frequency converter, and based on this torque current component. The primary frequency is modified so that it increases or decreases according to the increase or decrease in load.

Another feature of the present invention is that the torque current component of the primary current is detected based on the phase command signal of the primary current supplied to the induction motor from the frequency converter and the output voltage of the frequency converter. Based on this, the slip frequency of the induction motor is calculated by calculation, and the primary frequency is corrected based on this calculated slip frequency value, so that the primary frequency is increased or decreased according to the increase or decrease in the load.

FIG. 1 shows an embodiment of the present invention.

In Figure 1, 1 is a diode rectifier that rectifies the commercial AC voltage, 2 is a smoothing capacitor for smoothing the output voltage of rectifier 1, 3 is a voltage-type PWM inverter using a GTO thyristor, 4 is an induction motor, and 5 is a Speed command circuit, 6 is a current detector that detects the primary current of the motor, 7 is a current component detector that detects the torque current component I ₁ β of the primary current and the excitation current current component I ₁ α, respectively, 8 is I _{1 A} slip frequency calculation circuit that detects a signal proportional to the slip frequency (hereinafter referred to as the slip frequency signal) based on the β detection signal and the I ₁ α detection signal, 9 adds the slip frequency signal and the speed command signal, and a PWM inverter. an adder that outputs a frequency command signal that commands the output frequency of 1;
0 is a two-phase oscillator that generates a two-phase sine wave signal with a frequency proportional to the output signal voltage of adder 9, 11 is a magnetic flux command circuit that commands the amount of magnetic flux of the motor, and 12 is a multiplier for multiplying the magnetic flux command signal and the frequency command signal. 13 and 14 are multipliers that multiply the output signals of the oscillator 10 and the multiplier 12, 15 is a phase number converter that converts a two-phase signal into a three-phase signal, and 16 is a multiplier that outputs a voltage command signal.
17 is an oscillator that generates a carrier signal that controls the on/off frequency of the GTO thyristor of PWM inverter 3, and 17 compares the output signals of the phase number converter 15 and oscillator 16 and outputs a pulse width modulation signal (PWM signal). The comparator 18 is a gate amplifier that outputs a gate signal for controlling on/off the GTO thyristor of the PWM inverter 3.

FIG. 5 shows a detailed circuit diagram of the current component detector 7. 71 is an adder that adds or subtracts i _U , i _V , and i _W at a predetermined ratio; 72 is a subtracter that subtracts i _V from i _W ;
This converts the three-phase current signal i _U ~i _W into the two-phase current signal i _a ,
Convert to i _b . Multipliers 73 to 76 multiply the output signals a and b of the two-phase oscillator 10 by the two-phase current signals i _a and i _b , and adders and subtracters 77 and 78 add and subtract the output signals of the respective multipliers.

Next, its operation will be explained. As is well known, the voltage-type pulse width modulation inverter 3 can vary the output voltage by changing the on/off time of the GTO thyristor. Specifically, phase number converter 1
The voltage command signal (sine wave) from 5 and the carrier signal (triangular wave) from oscillator 16 are compared in comparator 17, and the GTO thyristor of inverter 3 is controlled on/off according to the PWM signal that is the output signal. and controls the output voltage so that it is proportional to the voltage command signal.

Next, the principle of the present invention will be explained first. The exciting current component I ₁ α and the torque current component I ₁ β of the primary current of the induction motor 4 are given by the following equations.

i ₁ α=1+PT ₂ /M′・φ ₂ ′ ……(2) I ₁ β=2πs・T ₂ /M′・φ ₂ ′ ……(3) Here, P: d/dt (operator) T ₂ : Secondary time constant M': Mutual inductance between primary and secondary φ ₂ ': Secondary interlinkage flux s: Slip frequency R ₂ ': Secondary resistance That is, the slip frequency s can be calculated according to the following formula.

s=1/2πT ₂ I ₁ β/[I ₁ α/(1+PT ₂ )] …(4) Or, s=M′/2πT ₂・I ₁ β/φ ₂ ′ …(5) On the other hand, induction The rotational frequency r of the electric motor 4 and the driving frequency ₁ have the following relationship: r = ₁ - s (for a two-pole machine) ... (6) Therefore, the driving frequency ₁ should be increased by the slip frequency s in advance. , ₁ ′= ₁ + s ……(7) If ₁ ′ is set as a new driving frequency, the relationship r= ₁ ′−s= ₁ ……(8) holds true. That is, the rotational frequency r is
Corresponds to the original drive frequency ₁ .

From this, it is possible to change the speed command signal to the original drive frequency.
₁ , and the slip frequency s
By adding corresponding signals and controlling the output frequency ₁ ' of the inverter 3 accordingly, the rotational speed r can be made to match the speed command with high precision.

The above is the principle. Next, the operation of the embodiment shown in FIG. 1 will be explained.

The current component detector 7 detects the torque current component I ₁ β and the excitation current component I ₁ α according to the following relationship. first,
The three-phase primary current signal i _U detected by the current detector 6,
i _V and i _W are converted into two-phase signals i _a and i _b by adders and subtracters 71 and 72 shown in FIG. The content of the calculation is as shown below, and the phase relationship of the vectors is shown in FIG.

where, k: proportional constant The two-phase signals i _a , i _b obtained in this way and the sine wave output signals a, b of the two-phase oscillator 10 are multiplied by the multipliers 73 to 7
Multiply by 6, excitation current component I ₁ α and torque current component
Detection signals i ₁ α and i ₁ β of I ₁ β are obtained. The contents of the calculation are as follows, and the phase relationship of the vectors is shown in FIG.

i ₁ α=-b・i _a +a・i _b =k′Isinθ i ₁ β=a・i _a +b・i _b =k′・Icosθ ……(10) Here, k′: constant of proportionality I: 1 Amplitude of the secondary current θ: Delay phase angle of signals i _a and i _b with respect to signals a and b Since the signals a and b of the two-phase oscillator 10 and the primary voltage of the motor 4 match in phase as described below,
By calculating equation (10), the torque current component of the primary current I ₁ β
and the excitation current component I ₁ α are detected.

The detailed configuration diagram of the slip frequency calculation circuit 8 is shown in the sixth figure.
As shown in the figure. Consists of a first-order delay circuit 81 and a divider 82, which performs calculations corresponding to equation (4) to determine the slip frequency s.
A slip frequency calculation signal proportional to is obtained.
This slip frequency calculation signal and the speed command signal from the speed command circuit 5 are added by an adder 9 to form a frequency command signal ₁ . This relationship is as described in equations (7) and (8).

A two-phase oscillator 10 generates two-phase sine wave signals a and b having a frequency proportional to the frequency command signal ₁ .
The two-phase oscillator 10 is, for example, a well-known integral type sawtooth wave oscillator and a function generator, and its output signals a and b are sine wave signals with constant amplitudes as shown in the following equation.

a=sin (2π ₁ t) b=cos (2π ₁ t) ...(11) Multipliers 13 and 14 multiply the two-phase signal a or b by the voltage command signal from multiplier 12, as shown in the following equation. Outputs signals c and d.

c=Asin (2π ₁ t) d=Acos (2π ₁ t) ...(12) Here, A: Voltage command signal voltage The phase number converter 15 converts the two-phase signals c and d into three-phase signals by a well-known method. Convert to The output voltage of the PWM inverter 3 is controlled to be proportional to this three-phase signal as described above, but at this time, the phase relationship between the output voltage E _U - E _W and the primary current I _U - I _W is 4-phase. The result will be as shown in the figure. At this time, the exciting current component I ₁ α and the torque current component I ₁ β of the primary current are given by the following equation,
A proportional relationship holds true with the detection signals i ₁ α and i ₁ β in equation (10).

I ₁ α=Isinθ I ₁ β=Icosθ (13) Control is performed as described above, and the slip frequency is calculated by detecting the torque current component of the primary current. The primary frequency is controlled by modifying the speed command signal.
Since the primary frequency is increased or decreased in response to an increase or decrease in the load, the rotational speed of the motor can be accurately controlled in proportion to the speed command signal. Furthermore, since the speed detector can be omitted, a great effect can be obtained especially in cases where its installation is difficult.

Here, in the embodiment shown in FIG. 1, the amount of magnetic flux can be arbitrarily controlled by the functions of the magnetic flux command circuit 11 and the multiplier 12. However, when operating under conditions where the amount of magnetic flux is constant, the excitation current component I ₁ α is constant, so as is clear from equation (4), s = k″i ₁ β
(k″: proportionality constant). Therefore, in this case, the slip frequency calculation circuit 8 in FIG. 1 can be omitted. Also, the slip frequency s is
Detection is also possible according to equation (5). Therefore,
Instead of the slip frequency calculation circuit 8, the torque current component I ₁ β is divided by the magnetic flux command signal to calculate the slip frequency s.
The same effect can be obtained by calculating and adding to the adder 9.

As explained above, the present invention detects the torque current component from the voltage phase command signal and the primary current, modifies the speed command signal based on this torque current component, and controls the primary frequency. Therefore, since it is not affected by voltage waveform distortion and speed changes due to load changes are corrected, highly accurate speed control can be performed.

[Brief explanation of the drawing]

FIG. 1 is a configuration diagram showing one embodiment of the present invention, and FIG.
Figures 4 to 4 are explanatory diagrams of the operation of Figure 1, and Figure 5 is an illustration of the operation of Figure 1.
FIG. 6 is a detailed circuit diagram of the current component detector shown in the figure, and FIG. 6 is a detailed circuit diagram of the slip frequency calculation circuit shown in FIG. 1... Diode rectifier, 2... Smoothing capacitor, 3... PWM inverter, 4... Induction motor, 5... Speed command circuit, 6... Current detector, 7
... Current component detector, 8 ... Slip frequency calculation circuit, 9 ... Adder, 10 ... Two-phase oscillator, 11 ...
...Magnetic flux command circuit, 12... Multiplier, 13, 14...
... Multiplier, 15 ... Phase number converter, 16 ... Oscillator, 17 ... Comparator, 18 ... Gate amplifier.

Claims (1)

[Scope of Claims] 1. In a speed control device for an induction motor in which a primary frequency is controlled by a frequency converter according to a speed command value, a primary current supplied from the frequency converter to the induction motor and the frequency converter The torque current component of the primary current is detected based on the phase command signal of the output voltage of the controller, and the primary frequency is corrected based on the torque current component, so that the primary frequency is increased or decreased in accordance with an increase or decrease in load. Features a speed control device for induction motors. 2. In a speed control device for an induction motor in which the primary frequency is controlled by a frequency converter according to a speed command value, the phase of the primary current supplied from the frequency converter to the induction motor and the output voltage of the frequency converter The torque current component of the primary current is detected by a command signal, the slip frequency of the induction motor is calculated based on this torque current component, and the primary frequency is corrected based on the calculated slip frequency value to increase or decrease the load. 1. A speed control device for an induction motor, characterized in that the primary frequency is increased or decreased depending on the speed of the induction motor.