JPS6355314B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Field of Industrial Application] The present invention relates to a speed control device for an induction motor that can control a squirrel cage induction motor with high response.

[Conventional technology]

In recent years, a control method called vector control, which controls the speed of squirrel cage induction motors with high response in the same way as DC motors, has been attracting attention. As is well known, this method separates the primary current of an induction motor into an excitation current component and a torque effect current component, and controls each component independently.This method prevents interference between the components, which was previously This eliminates the drawbacks of the control method described above, such as fluctuations in magnetic flux that occur in response to changes in torque, resulting in torque pulsations, and enables fast-response control.

However, conventionally, a method has been known in which the air gap magnetic flux of an induction motor is detected by a Hall generator, and this is used as a phase reference for the primary current of the motor to independently control the torque action component and the excitation component of the primary current. However, this method requires a Hall element to be installed inside the motor to detect the magnetic flux, which has the drawback of reducing the reliability and accuracy of the detector.

In order to solve this problem, a method that eliminates the magnetic flux detector has been announced. That is, this is a method that improves the slip frequency control method and controls the magnitude, phase, and slip frequency of the primary current based on the torque action current command and excitation current command of the primary current.

FIG. 1 is a block diagram showing one row of conventional devices.

The device includes an induction motor 101, an inverter 102 that supplies primary current to the motor, and a speed detector 10.
3. Speed command circuit 104, speed deviation amplifier 10
5. Slip frequency calculation circuit 106 that outputs a slip frequency command, adder 107 that outputs a speed detection signal and a slip frequency command signal, excitation current command circuit 108, torque action current command from amplifier 105 and excitation from command circuit 108 A current command calculation circuit 109 that outputs a primary current command signal (sine wave signal) based on the current command and the frequency command from the adder 107; a current detector 110 that detects the instantaneous value of the primary current; A current deviation amplifier 1 that amplifies the deviation of the primary current command signal and detection signal and controls the output voltage of the inverter 102 accordingly.
It consists of 11 parts.

In the device shown in FIG. 1, the primary current is controlled to be proportional to the primary current command signal according to the operation of the amplifier 111, etc., and the torque action component and excitation component of the primary current are controlled by the amplifier 105 and the command circuit 10, respectively.
8 output signals. Since the torque action current is controlled according to the speed deviation as described above, the rotational speed of the electric motor is proportional to the speed command.

[Problem that the invention seeks to solve]

However, in this device, the torque action component of the primary current and the slip frequency are controlled according to a fixed relationship, so if the secondary resistance of the motor changes due to temperature, the arithmetic circuit 10
The slip frequency command value output from 6 will deviate from the optimal value required by the motor at that time. As a result, each component of the primary current cannot be controlled independently, resulting in disadvantageous characteristics such as pulsation and response delay in the generated torque and fluctuations in the motor voltage (magnetic flux) when the current changes.

An object of the present invention is to provide a speed control device for an induction motor whose control characteristics do not change regardless of changes in secondary resistance due to temperature changes.

[Means for solving problems]

The present invention determines the torque current component of the primary current using the primary current supplied from the frequency converter to the induction motor and the phase command signal of the output voltage of the frequency converter. The torque current command is obtained by comparing the speed detection value of the rotational speed of the induction motor with the speed command value, and the output frequency of the frequency converter (primary frequency of the induction motor) is determined by the deviation between this torque current command and the torque current component. Control.

[Effect]

The primary frequency is controlled according to the deviation between the torque current command and the torque current component. When the torque current component is smaller than the torque current command, the primary frequency changes in an upward direction and the slip frequency increases.
Accordingly, the secondary current of the induction motor, which is proportional to the slip frequency, increases, and the torque current component of the primary current also increases. As a result, the torque current component is controlled to match the torque current command.
When the torque current component is larger than the torque current command, the operation completely opposite to the above is performed. When the secondary resistance changes, the secondary current changes in inverse proportion to the change in the secondary resistance. However, since the slip frequency is controlled as described above, the secondary current is controlled according to the torque current command. Therefore, high-response speed control can be performed without being affected by changes in secondary resistance.

〔Example〕

Hereinafter, the present invention will be explained based on examples.

FIG. 2 is a block diagram showing one embodiment of the present invention.

In FIG. 2, the devices include a rectifier 1, a smoothing capacitor 2, a PWM inverter 3, an induction motor 4, a speed command circuit 5, a speed detector 6, a speed deviation amplifier 7, a current detector 8, a current component detector 9, and an amplifier 1.
0, adder 11, two-phase oscillator 12, excitation current command circuit 13, deviation amplifier 14, multipliers 15, 16,
17, phase number converter 18, oscillator 19, comparator 2
0 and a gate amplifier 21. Commercial power is rectified by a rectifier 1, smoothed by a smoothing capacitor 2, and applied to a PWM inverter 3. The PWM inverter 3 performs predetermined control and supplies AC power to the induction motor 4. When controlling the PWM inverter 3, vector control is performed, and control is performed by both systems of excitation current component and torque action current component.

Control of the torque acting current component is performed based on the primary current detection value, motor speed, and speed command.
That is, the amplifier 7 outputs a signal corresponding to the speed deviation based on the output of the speed command circuit 5 that issues the speed command S and the output of the speed detector 6 connected to the motor shaft. Next, the deviation between the output of the amplifier 7 and the torque acting current component (effective component) from the current component detector 9 is amplified by the amplifier 10. The output signal is further added to the output signal of the speed detector 6 in an adder 11 and applied to a two-phase oscillator 12. The two-phase oscillator 12 generates a two-phase sine wave signal (phase command signal of the output voltage of the PWM inverter 3) with a frequency proportional to the output signal voltage of the adder 11, and sends it to the current component detector 9 and multipliers 16 and 17. Output.

On the other hand, the amplifier 1 detects the deviation between the excitation current component detected by the current component detector 9, that is, the reactive current component, and the excitation current command from the excitation current command circuit 13.
4, the output signal is multiplied by the output signal of adder 11 in multiplier 15, and the multiplied value is output to multipliers 16 and 17.

Both outputs of the multipliers 16 and 17 are applied to a phase number converter 18, where the two-phase signal is converted into a sine wave of a three-phase signal. This three-phase signal is sent to the comparator 20,
Turn on/off GTO thyristor of PWM inverter 3
A comparison is made with the output of an oscillator 19 that generates a carrier signal for controlling the off frequency, and a pulse width modulation signal (PWM signal) is generated based on the comparison result.
is applied to the gate amplifier 21 to control the PWM inverter 3.

Next, the operation of the embodiment shown in FIG. 2 will be explained. As is well known, the voltage-type pulse width modulation inverter device changes the on/off period of the thyristors that make up the inverter.
The output voltage of the inverter can be made variable, and the voltage command signal (sine wave) from the phase number converter 18 and the carrier wave signal (triangular wave) from the oscillator 19 are sent to the comparator 2.
0, and by controlling the GTO thyristor of the inverter 3 on and off according to the PWM signal that is its output signal, the output voltage of the inverter is controlled to be proportional to the voltage command signal.

Next, the basic characteristics of the electric motor will be explained. If we ignore the leakage impedance of the motor primary winding for simplicity, the exciting current component of the primary current I ₁ 〓,
The following relationship holds true between the torque action current component I ₁ 〓 and the generated torque τe.

I ₁ 〓=1+PT ₂ /M・φ ₂ ...(1) I ₁ 〓=2πfs/R ₂・φ ₂ ...(2) τe=PM/M+l ₂・I ₁ 〓・φ ₂ ...(3) Here, P: d/dt (operator) T ₂ : Secondary time constant M: Mutual inductance between primary and secondary φ ₂ : Secondary interlinkage flux _s : Slip frequency R ₂ : Secondary resistance p: Pole Logarithm l ₂ : Secondary leakage inductance As mentioned above, torque τe is proportional to I ₁ 〓 and φ ₂ ,
φ ₂ has a steady proportional relationship with I ₁ 〓, and furthermore,
I ₁ 〓 is proportional to _s and φ ₂ .

Therefore, by controlling I ₁ 〓, I ₁ 〓, φ ₂
and τe can be arbitrarily controlled.Next, a method for controlling I ₁ 〓 and I ₁ 〓, which is the content of the present invention, will be explained.

The current component detector 9 detects the torque action current component I ₁ 〓 and the exciting current component I ₁ 〓 of the primary current according to the relationship of the following equation. First, the three-phase primary current signals (i _u , i _v , i _w ) detected by the current detector 8 are converted into two-phase signals (i〓, i _b ).
Convert to The content of the calculation is as shown below, and the phase relationship of the vectors is shown in FIG.

Here, k: proportionality constant Next, the sine wave signal (a, b) of the two-phase oscillator is multiplied by the signals i _a , i _b , and the detection signals of I ₁ 〓, I ₁ 〓 are obtained as i ₁ 〓, i ₁ 〓 The calculation contents are as follows, and the phase relationship of the vectors is shown in FIG.

i ₁ 〓=-b・i〓+a・i _b =k′ Isinθ i ₁ 〓=−a・i〓+b・i _b =k′ Icosθ……(5) Here, k′: Constant of proportionality I: 1 Amplitude of the secondary current θ: Delay phase angle of signals i〓, i _b with respect to signals a and b Since the two-phase oscillator signal and the primary voltage of the motor 4 match in phase as described later, the calculation of equation (5) The effective component (torque action current component) and reactive component (excitation current component) of the primary current are detected.

The signal corresponding to the speed deviation from the amplifier 7 and the above-mentioned signal I ₁ are matched in the amplifier 10,
That deviation is amplified. In the adder 11, the output signals of the amplifier 10 and the speed detector 6 are added, and 2
Phase oscillator 12 generates a two-phase sinusoidal signal with a frequency proportional to the output signal of adder 11. At this time, the signal of the speed detector 6 is a signal proportional to the rotation frequency r, and the output signal of the amplifier 10 is a signal proportional to the slip frequency.
It corresponds to a signal proportional to _a , and the two-phase oscillator 1
2 output signal frequency, i.e. the primary frequency of the motor
_M becomes as follows.

_M = _r + _a ... (6) The two-phase oscillator 12 is composed of a well-known integral type sawtooth wave oscillator and a function generator, and its output signals a and b have amplitudes as shown in the following equation. It is a constant sinusoidal signal.

a=sin (2π _M t) b=cos (2π _M t) ...(7) On the other hand, the excitation current command from the command circuit 13 and the above-mentioned signal i ₁ are compared in the amplifier 14,
That deviation is amplified.

The output signals of the amplifier 14 and the adder 11 are multiplied in the multiplier 15, and a signal instructing the output voltage amplitude of the PWM inverter 3 is extracted. The voltage amplitude is basically the output signal of the adder 11, that is, _M
, but is additionally controlled by the output signal of amplifier 14.

In multipliers 16 and 17, the output signals of two-phase oscillator 12 and multiplier 15 are multiplied to obtain signals (c, d) shown in the following equation.

c=Asin(2π _M t) d=Acos(2π _M t) ...(8) Here, A: Output signal voltage of the multiplier 15 Next, in the phase number converter 18, the two-phase signal c is converted by a well-known method. , d into three-phase signals. PWM
The output voltage of the inverter 3 is controlled to be proportional to this three-phase signal as described above, but at this time, the phase relationship between the output voltage E _U ~ _EW and the primary current E _U ~ _EW is the 5th one. The result will be as shown in the figure. In this case, the reactive component I ₁ 〓 and the effective component I ₁ 〓 of the primary current are given by the following formula, (5)
A proportional relationship holds true with i ₁ 〓 and i ₁ 〓 in the equation.

I ₁ 〓=Isinθ I ₁ 〓=Icosθ ...(9) Since the circuit operates as described above, the motor's 2
The secondary flux linkage φ ₂ and the torque τ _e are controlled as follows. First, regarding φ ₂ , the output voltage of the inverter 3 is controlled according to the deviation between the command signal of the command circuit 13 and i ₁ 〓, and as a result, φ ₂ which is in a proportional relationship with i ₁ 〓 is controlled to a predetermined value.

As a method of controlling the amount of magnetic flux, a method has been known in which the output voltage of the inverter is controlled by detecting the motor voltage. However, with this method, when the rotation speed of the motor is low, the magnetic flux decreases due to the influence of primary resistance etc. It is difficult to control the amount to a predetermined value. However, in this method, even when the rotational speed is low, i ₁ 〓 is detected continuously, so the above-mentioned problem does not occur.

On the other hand, regarding the torque τe, the slip frequency _a is controlled according to the deviation between the torque command from the amplifier 7 and i ₁ 〓 under certain conditions for φ ₂ as described above. According to the relation I ₁ 〓 and τe
is controlled to be proportional to the torque command.

The rotational speed of the electric motor is then controlled to be proportional to the speed command.

Here, even if the secondary resistance R ₂ of the motor changes depending on the temperature, I ₁ 〓
Since it is controlled so that it becomes a predetermined value, the influence of the change can be prevented. Furthermore, since control of I ₁ 〓 can be performed with a quick response by a control loop including the current component detector 9 and the amplifier 10, it is possible to prevent undesirable torque fluctuations caused by various causes.

In the embodiment described above, the multiplier 15 is used to cause the output voltage of the inverter 3 to vary according to _M , but if the acceleration/deceleration rate of the motor is relatively slow, the multiplier 15 is unnecessary. This is because the output voltage of the inverter 3 is controlled by the output signal of the amplifier 14, and if the gain of the amplifier 14 is sufficient, the output voltage can be controlled with sufficient accuracy only by this output signal. be.

The same applies to adder 11. That is, the adder 11 is used to change the output frequency of the inverter in accordance with the output signal of the speed detector 6, but if the acceleration/deceleration rate is relatively slow, the adder 11 is unnecessary. . This is because the output frequency of the inverter 3 is controlled by the output signal of the amplifier 10, and if the gain of the amplifier 10 is sufficient, the frequency can be controlled only by this output signal.

FIG. 6 is a block diagram showing a second embodiment of the present invention.

The embodiment shown in FIG. 6 is obtained by removing the command circuit 13, amplifier 14, and multiplier 15 from the embodiment shown in FIG. 2, that is, removing the system for the reactive component of the primary current. Such a configuration can be applied when fluctuations in the DC input voltage of the PWM inverter 3 and a decrease in torque during low-speed rotation do not pose particular problems. In this case, the circuit for detecting i ₁ 〓 of the current component detector 9 becomes unnecessary.

FIG. 7 is a block diagram showing a third embodiment of the present invention. This embodiment is an example of performing magnetic flux weakening control corresponding to field weakening control (constant output control) in a DC machine. The configuration is basically the same as that in FIG. 2, but the circuit configuration leading to the deviation amplifiers 10 and 14 is partially changed. That is, a magnetic flux command circuit 22 generates a command signal of secondary magnetic flux φ ₂ for the output of the speed detector 6, a divider 23 divides the output of the amplifier 7 by the output of the magnetic flux command circuit 22, and a reactive component of the primary current. This is a circuit configuration consisting of a first-order lag circuit 24 for. The output of the magnetic flux command circuit 22 is a constant value depending on the rotation speed of the electric motor, for example, from the stop of rotation to the base speed, and changes in inverse proportion to the rotation speed from the base speed to the top speed. The outputs of the magnetic flux command circuit 22 and the first-order delay circuit 24 become input signals to the deviation amplifier 14, and the output of the divider 23 becomes an input signal to the deviation amplifier 10.

The embodiment of FIG. 7 consists in continuously changing the secondary magnetic flux φ ₂ above the base velocity. Torque τ
changes in proportion to the secondary magnetic flux φ ₂ as shown in equation (3), so it is necessary to change the torque τe in proportion to the torque command. For this purpose, the torque command τe is set to 2
The command signal of i ₁ 〓 is obtained by dividing by the command signal of the next magnetic flux φ ₂ . The first-order delay circuit 24 has a function of calculating a detection signal of φ ₂ from i ₁ 〓 according to equation (1), and the amplifier 14 performs matching amplification of this output and the output of the magnetic flux command circuit 22 . According to this PWM
The output voltage of inverter 3 will be controlled.

In general, φ ₂ is a quadratic time constant T ₂ for changes in I ₁ 〓
However, in this device, I ₁ 〓 is controlled to compensate for the delay in change of φ ₂ , so φ ₂ quickly follows the command signal. In this way, rapid magnetic flux weakening control is possible.

In the embodiment shown in FIG. 6, an example was shown in which control is performed using only the effective component of the primary current, but conversely, control may be performed using only the inactive component. In this way, the output voltage of the inverter is not directly detected as in the conventional method, so it is possible to control the output voltage without waveform distortion.Also, for the same reason, the rotational speed of the motor is low, making it difficult to detect the voltage. Even in such cases, φ ₂ can be controlled to a predetermined value. The configuration in this case is based on the speed command circuit 5 according to the embodiment shown in FIG.
The circuit from to adder 11 is removed, and adder 11
It is only necessary to give a frequency control command as an alternative to the output of .

Although the operations of regenerative braking and reverse operation were not mentioned in this embodiment, in the case where a reversible converter or a DC power supply capable of power regeneration is connected instead of the diode rectifier 1, the above-mentioned control circuit particularly controls the circuit. Regenerative braking and reverse operation can be performed without the need to add.

〔Effect of the invention〕

As explained above, the present invention detects the torque current component of the primary current, and adjusts the primary frequency of the induction motor according to the deviation between the torque current command and the torque current component obtained by comparing the speed command value and the detected speed value. It's in control. Therefore, the secondary current, which has a proportional relationship with the slip frequency, is controlled in accordance with the torque current command, making it possible to perform high-response speed control without being affected by changes in the secondary resistance.

[Brief explanation of the drawing]

FIG. 1 is a block diagram of a conventional device, FIG. 2 is a block diagram of a first embodiment of the present invention, FIGS. 3, 4, and 5 are explanatory diagrams explaining the operation of the present invention, and FIG. 6 7 is a block diagram of the second embodiment of the present invention.
The figure is a block diagram of a third embodiment of the present invention. 1... Rectifier, 3... PWM inverter, 4...
...Induction motor, 5...Speed command circuit, 6...Speed detector, 7...Speed deviation amplifier, 8...Current detector, 9...Current component detector, 10, 14...Difference amplifier, 11... ... Adder, 12 ... Two-phase oscillator,
13... Excitation current command circuit, 15, 16, 17...
... Multiplier, 18 ... Phase number converter, 19 ... Oscillator, 20 ... Comparator, 21 ... Gate amplifier, 2
2... Magnetic flux command circuit, 23... Divider, 24...
First-order delay circuit.

Claims (1)

[Claims] 1. In a speed control device for an induction motor in which a primary frequency is controlled by a frequency converter according to a speed command value, a phase command of the primary current supplied from the frequency converter to the induction motor and the output voltage of the frequency converter. The torque current component of the primary current is detected by a signal, and the deviation between the torque current command obtained by comparing the speed detection value of the rotational speed of the induction motor and the speed command value and the torque current component is used. A speed control device for an induction motor, characterized by controlling the output frequency of a frequency converter.