JPS6330236Y2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to an improvement in a control device for an induction motor having a high-speed response control system for controlling the magnitude and frequency of the motor's primary current.

In general, an induction motor control device that uses a voltage source inverter to perform high-speed response slip frequency control not only detects the primary current of the motor to be driven and controls this magnitude to a desired value. The frequency also needs to be controlled so that the detected value becomes a desired value. However, in an inverter that converts alternating current to variable frequency through thyristor commutation, it is difficult to detect the frequency of the fundamental current component of the current to be controlled without delay because the primary current has a distorted current waveform with many ripples. , a system that compensates for changes in each current phase during transient periods has become practically adopted.

However, in this type of transient compensation method, in addition to the steady slip frequency ω _s1 , there is also a transient slip frequency ω _s2 that performs phase compensation due to changes in the torque current or magnetizing current component, and a change in the power factor of the current. Since the transient slip frequency ω _s3 to be compensated is added and the entire slip frequency is calculated, the calculation configuration becomes complicated. Further, there was a difficulty in converting the in-phase component current and the reactive component current to the power factor angle or the voltage related thereto.

The present invention has been developed with the above-mentioned points in mind, and it is an object of the present invention to provide a control device for an induction motor in which the calculation structure of the slip frequency is simplified without deteriorating the quick response.

To explain the present invention in detail, first, the relationship between phase compensation and slip frequency will be explained with reference to the vector diagram shown in FIG. In other words, Figure 1a shows the transition in which the secondary current component that generates torque perpendicular to the magnetizing current component _I0 , which provides a rotating magnetic field, changes from _I21 to _I22 , with the dotted line indicating the state before the change. The rear is shown by a solid line. Primary current like this
I ₁₁ and I ₁₂ are given as their combined vectors, and the phase difference angle between the primary currents I ₁₁ and I ₁₂ and the reference magnetizing current component I ₀ also changes from φ _q1 to φ _q2 , that is, by Δφ _q . Furthermore, FIG. 1b shows the vector relationship after the change shown in _FIG . _1a more _clearly . is the excitation current at the gap shown in the equivalent circuit of an induction machine.
I _0G and the primary converted secondary current I ₂₂ ′ are corrected by the internal phase difference angle φ〓 ₂₂ of the secondary circuit. The secondary circuit that generates torque is represented by the secondary resistor r ₂ , and the induced voltages E ₂₁ and E ₂₂ and the secondary current components I ₂₁ and I ₂₂ before and after the change in the secondary current are the same.

In addition, as shown in Figure 1a, the induced voltage in the primary circuit
In the voltage relationship between E ₁₁ and E ₁₂ and the terminal voltages V ₁₁ and V ₁₂ , they have internal phase difference angles φ〓 ₁ and φ〓 ₂ , respectively, and have a phase difference angle difference of Δφ〓. In other words, between the steady state before and after the change in the secondary current, both the phase difference angle and the internal phase difference angle of the current are out of phase by Δφ _q and Δφ〓, so as the current command changes, these phases also change according to the change. By being able to adjust the phase, the current can be controlled while satisfying the phase relationship in a steady state, and rapid control is possible without generating a transient state and without interaction between the magnetizing current component and the secondary current component. As is well known, the temporal changes in these phase angles are given as angular velocities, and conventionally the phase compensation when the current command changes _is calculated as follows _.
It was supposed to be controlled using equations (1) and (2).

ω _s2 = d/d _t φ _q = d/d _t tan ^-1 I ₂ /I ₀ …(1) ω _s3 = d/d _t φ ₁ = d/d _t tan ^-1 I _sio /I _cps …( 2) However, φ ₁ is the power factor angle on the primary side of the motor,
As shown in FIG. 1b, I _cps and I _sio are the in-phase component and the reactive component current orthogonal to the terminal voltage V ₁₁ of the primary circuit, respectively. Note that in equations (1) and (2) above, the suffix symbols representing before and after the change are omitted because the instantaneous is taken into account. Also this
In equations (1) and (2), the state where the phase difference angle φ _q and the power factor angle φ ₁ do not change, that is, I ₀ , I _{2 ,} of each component of the primary current,
In a steady state where I _cps and I _sio do not change, there are no temporal changes in these, so ω _s1 and ω _s2 become zero and act only during transient times.

Therefore, during the transient period of such slip frequency, After all, the command signal for the inverter frequency can be obtained by adding the more steady slip frequency ω _s1 and the rotational frequency ω _o . Next, we will introduce a conventional induction motor control system with a high-speed response control system using slip frequency compensation.
As shown in the figure.

In FIG. 2, a voltage source inverter 1 receives a gate signal from a gate controller 2 to which a voltage command V〓 and a frequency command〓 are given, and converts the DC power supplied from a DC power supply 3 into variable voltage/variable frequency AC power. to drive the induction motor 4. The rotational speed of this induction motor 4 is the setting value of the setting device 5.
n〓 and the actual rotational speed n sent from the speed detector 6
are compared by a comparator 7, and controlled by a regulator 8 to achieve a predetermined speed. The output signal of the regulator 8 becomes a command for the secondary current I ₂ proportional to the torque, and the command for the magnetizing current component I ₀ set by the setting device 5' is given to the arithmetic units 9, 10, 11, 12. The calculation output will be sent out.

Here, the magnitude of the primary current I ₁ , the steady slip frequency ω _s1 , the slip frequency ω _s2 for phase compensation against changes in the secondary current and magnetizing current, and the slip frequency ω _s3 against changes in the power factor angle are expressed as ( 4) Equation,
The signal is generated from equations (5), (1), and (2). Note that 13 is a coordinate converter that converts the above-mentioned I ₂ and I ₀ into terminal voltage, synchronous component, and reactive component signals.

I ₁ = √ ( ₀ ) ² + ( ₂ ) ² ... (4) ω _s1 = r ₂ /L ₂ · I ₂ / I ₀ ... (5) Now, ω _s2 and ω _s3 of the outputs of computing units 11 and 12 are added. The output of the adder 14 and the output of the calculator 10, ω _s1 , are added together in the adder 14', and the adder 14'' adds the rotational frequency _ωo , and finally the frequency command 〓 is given. On the other hand, a detection signal indicating the magnitude of the primary current is obtained from the current detector 15 via the rectifier 16, and the output of the rectifier 16 is compared with the primary current command value I ₁ 〓 obtained by the calculator 9. This I ₁ 〓 is adjusted to the desired current value by the regulator 8', and the voltage command is
A signal is generated to the gate controller 2 as V〓.

In the vector relationship in such a control system, as shown in Figure 1b, φ ₁₂ = (π/2) + φ〓 ₂ −φ _q2 ...(6), and in the general equation considering each instant, (6' )
It is given by Eq.

φ ₁ =(π/2)+φ〓−φ _q (6′) Therefore, equation (6)′ can be expressed in a simple form as follows.

Furthermore, the equation (7) as described above does not require conversion of the in-phase component or reactive component or calculation of the non-linearly changing power factor angle, and the transient compensation term for the slip frequency can be obtained by simple calculation. be.
That is, the internal phase difference angle φ of the motor is relatively small compared to the power factor angle and the above-mentioned current phase difference angle, and this is also a quantity proportional to the internal impedance of the motor, that is, proportional to the secondary current or slip frequency. Therefore, the rated secondary current component of the primary conversion is I ₂₀ ,
If the internal phase difference angle at that time is φ〓 ₀ , then the internal phase difference angle φδ at any secondary current component I ₂ is given as φ〓=(φ〓 ₀ /I ₂₀ )・I ₂ …(8) Become something.

Further, this technical idea can be supplemented from FIG. 1a. That is, even if the magnetizing current component I ₀ is taken as a reference in FIG. 1a and the secondary current command is changed from I ₂₁ to I ₂₂ while keeping this magnetizing current component I ₀ constant,
It can be seen that the vector position of the secondary current and therefore the secondary induced voltage does not change, but the terminal voltage component changes by the internal phase difference angle φ. A specific example of a system that satisfies the requirements for such a calculation configuration is shown in FIG.

FIG. 3 shows a control system of an embodiment of the present invention, in which 17 is a slip frequency transient term calculation section consisting of an internal phase difference angle calculation unit 17a and a differentiator 17b. In the figure, the same reference numerals as in FIG. 2 indicate parts having the same functions. Here, the slip frequency transient term calculation section 17 operates by calculating the internal phase difference angle calculation unit 17a and the output of the internal phase difference angle calculation unit 17a, which calculates the equation (8) as illustrated, without calculating the above-mentioned transient compensation terms individually. The structure is such that an integrated calculation output command is generated from the differentiator 17b that performs differentiation, thereby realizing a simple calculation in which the power factor angle term is removed. The detailed explanation of the control system will be limited to the signal generation portion of the frequency command 〓, but the output of the slip frequency transient term calculation section 17 and the slip frequency ω _s1 are added in an adder 14', and By adding the rotational frequency ω _o at 14″, the frequency command 〓 can be obtained.

As explained above, according to the present invention, it is possible to provide a device that has a simple calculation configuration that integrates transient compensation of slip frequency and can perform high-speed response control.

[Brief explanation of drawings]

Fig. 1 is a vector diagram showing the before and after states of secondary current change, Fig. 2 is a control system diagram showing a conventional induction motor control device, and Fig. 3 is a control system diagram showing an embodiment of the present invention. be. 1... Voltage type inverter, 4... Induction motor,
5, 5'... Setting device, 9, 10, 11, 12...
Arithmetic unit, 17...Slip frequency transient term computing unit,
V〓……Voltage command, 〓……Frequency command, ω _s1 ,
ω _s2 , ω _s3 ...Slip frequency.

Claims (1)

[Scope of utility model registration request] A command signal corresponding to the motor secondary current, a command signal corresponding to the magnetizing current, and a slip frequency signal obtained by adding the slip frequency in the steady state and the transient term of the slip frequency corresponding to the current phase change during the transient from the signals are given. 1. A control device for an induction motor, which controls the slip frequency of an induction motor, characterized in that the slip frequency during a transient period is given as a temporal rate of change in an internal phase difference angle of the motor.