JPH08224000A - Controller for induction motor - Google Patents

Abstract

PURPOSE: To achieve vector control of a three-phase induction motor without requiring a voltage detector or a speed detector. CONSTITUTION: Input currents i1u , i1v of an induction motor 1 are detected and subjected to rotational coordinate transformation to produce a flux axis component current I1d and a torque axis component current I1q . A phase angle θbeing employed in the rotational coordinate transformation is determined by operating a primary angular speed ω and then integrating the ω. The primary angular speed ω is determined from the flux axis component current I1d , a torque axis component voltage command value V1q *, a flux axis component voltage reference command value V1d ', and the constants of an induction motor 1.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for a three-phase induction motor which does not use a speed detector.

[0002]

2. Description of the Related Art In order to reduce the cost and size of a speed control device, a method for detecting the rotation speed of a three-phase induction motor without using a speed detector has already been proposed. In this type of method, the rotation speed is estimated based on the input current, voltage resistance, inductance, etc. of the induction motor.

[0003]

By the way, when estimating the primary angular velocity or the rotational speed by calculation, it is costly without requiring a drastic change in the device for actually controlling the speed by using the speed detector. It is important in controlling the rise.

Therefore, the present invention is to provide a control device for a three-phase induction motor which can omit the speed detector without drastically changing the conventional control device.

[0005]

The present invention for achieving the above object will be described with reference to the reference numerals of the drawings showing an embodiment.
Phase PWM control inverter and one of the three phases of the induction motor
Current detectors 5u, 5v for detecting primary currents of at least two phases of the secondary currents, and 1 detected by the current detectors
First conversion means 13 for converting the next current into a torque axis component current I _1q and a magnetic flux axis component current I _1d orthogonal thereto by rotational coordinate conversion synchronized with the rotation phase of the induction motor;
First command means 11a for giving a torque axis component current command value I _1q ^* for rotating the induction motor to a desired state
And second command means 1 for giving a magnetic flux axis component current command value I _1d ^* for rotating the induction motor to the desired state.
9 and the torque axis component current command value I _1q ^* and the difference between the torque axis component current I _1q, and the torque axis component forming the torque axis component voltage command value V _1q ^* for making the difference zero. Voltage command value forming means and the magnetic flux axis component current command value I _1d
^The magnetic flux axis component voltage command value forming means for forming a magnetic flux axis component voltage command value V _1d ^* for obtaining the difference between ^* and the magnetic flux axis component current I _1d and making the difference zero, and the torque axis component voltage. The command value V _1q ^* and the magnetic flux axis component voltage command value V _1d ^* are set to 3
Second conversion means 21 for converting the phase voltage command values V _1u ^* , V _1v ^* , V _1w ^* , and the three-phase voltage command values V _1u ^* , V _1v ^* ,
Inverter control circuit for controlling the inverter by forming a pulse width modulation (PWM) control signal corresponding to V _1w ^*
#, The torque axis component voltage command value V _1q ^* , the torque axis component current I _1q , the magnetic flux axis component current command value I _1d ^* , the primary winding resistance R 1 of the induction motor, and the leakage inductance Lσ.
Ω ₀ = (1 / L 1) [{V _1q ^* -(PLσ + R 1) I _1q } / I _1d ^* ] based on the first-order self-inductance L 1 and the differential operator P or an equivalent first-order angular velocity a primary angular velocity estimating means for obtaining an omega _0, and a phase angle signal forming means 20 providing the said form a signal indicative of the rotational phase based on the estimated primary angular velocity omega ₀ first converting means (13) The present invention relates to an induction motor control device characterized by being provided. In addition, it is desirable that the control device for the induction motor of claim 1 is configured as shown in claims 2 to 8.

[0006]

According to the invention of each claim, the primary angular velocity is estimated by changing only a part of the control device of the conventional three-phase induction motor using the velocity detector. Can be used for control. Therefore, not only is the cost and size reduced by not using the speed detector,
It becomes possible to easily achieve the estimation of the primary angular velocity.
More specifically, the estimation of the primary angular velocity can be performed without detecting the input voltage of the induction motor, so that the estimation is easily achieved. That is, the input voltage of the induction motor is PWM
Since it is the output of the inverter, it cannot be used as the voltage information used in the control device again, and the provision of the smoothing means causes an increase in cost. On the other hand, in the present invention, since the induced voltage is estimated, the actual voltage detecting means becomes unnecessary. According to claims 2 and 3, it is possible to improve the accuracy of the estimation of the primary angular velocity. According to claim 7, the rotational angular velocity ω _rm ′ can be easily estimated. According to claim 8, control can be easily achieved by digital processing.

[0007]

2. Description of the Related Art In order to facilitate understanding of a speed control device according to an embodiment of the present invention, a conventional speed control device using a rotation detector will be described first.

The conventional speed control device for the three-phase induction motor 1 shown in FIG. 1 has a three-phase bridge type inverter 3 connected between a DC power supply 2 and the motor 1. This inverter 3 is well known, and as shown in FIG. 2, switches Q1 to Q6 composed of six transistors are connected in a three-phase bridge, and a three-phase PWM control is performed on these switches to generate a three-phase AC voltage. Convert to voltages V _1u , V _1v , and V _1w ,
The speed control and the torque control are performed. Incidentally. Feedback diodes D1 to D in anti-parallel to each switch Q1 to Q6
6 is connected.

The speed control device shown in FIG.
Configured in accordance with the paper entitled "Non-interference control method for induction motors driven by VVVF power supply and its characteristics" described in Vol. 104, No. 11, pp. 79-86 of "Journal of the Institute of Electrical Engineers of Japan", published in November 2013. Has been done. In order to apply the vector control for controlling the slip frequency, the speed control device of FIG. 1 according to this control principle is provided with the speed detector 4 of the electric motor 1 and the current detectors 5u and 5v composed of CT. It is connected to the line between the inverter 3 and the electric motor 1. The speed detector 4 is composed of, for example, a pulse encoder or a tacho generator, and detects the actual speed ω _rm .
The current detectors 5u and 5v detect the u-phase and v-phase currents, that is, the primary currents I _1u and I _1v of the motor. A / D converters 6, 7, and 8 are provided for the speed detector 4 and the current detectors 5u and 5v.
Is connected. In the present application, for ease of explanation,
The same symbols are given to analog signals and digital signals. Also, three-phase control voltages V _1u ^* , V _1v for PWM control of the inverter 3 based on the output data of the speed detection A / D converter 6 and the output data of the current detection A / D converters 7 and 8 Creation of ^* , V _1w ^* data is executed by a digital signal processing device or a microcomputer, and this data creation device is shown in an analog (equivalent) manner in FIGS. 1 and 5 to 8.

In order to execute the speed control of the electric motor 1, the speed detecting A / D converter 6 is connected to a subtracter (error amplifier or comparator) 9 as a difference signal forming means. The subtractor 9 outputs a signal indicating the difference between the digital speed command (desired rotation speed) ω _rm * obtained from the line 10 as the speed command generation means and the detected rotation angular velocity (rotation speed) ω _rm . A speed controller (ASR) 11 is connected to the subtractor 9, and here, a torque axis component current command value i _1q ^* for controlling the torque T of the electric motor 1 so that the output of the subtractor 9 becomes zero ^. Is formed, and this is output to the line 11a as the first command means. As is well known, the speed controller 11 is composed of, for example, a PI (proportional integral) compensating circuit, and is configured to generate an output y according to the following equation. y = {Kp + (Ki / s)} x = Kp {1+ (1 / Tis)} x where x is an input, Kp is a proportional gain or constant, Ki is an integral gain, Ti is an integral constant, S Is the Laplace operator.

The velocity detection A / D converter 6 is also connected to the primary angular velocity calculation circuit 12. In order to easily obtain the primary angular velocity ω, the input currents i _1u and i _1v of the induction motor 1 are orthogonal to each other on the dq coordinate axes that rotate in synchronization with the rotation phase of the induction motor 1, that is, the phase θ of the primary voltage. It is necessary to convert the two currents I _1d and I _1q to Where I _1d
Is the torque axis component current, and I _1q is the magnetic flux axis component current. As the first conversion means for performing the above conversion, 2
A first coordinate conversion circuit 13 is connected to the two current detection A / D converters 7 and 8. The coordinate conversion circuit 13 is configured to obtain the currents I _1d and I _1q according to the following equation (1). Induction motor 1 u, v, w three-phase current i _1u ,
It is also possible to detect i _1v and i _1w and convert them into two I _1d and I _1q . In this case, I _1d and I _1q are obtained according to the equation (1 ′).

[0012]

[Equation 1]

The primary angular velocity calculation circuit 12 comprises a slip angular velocity calculation circuit 14, a pole pair number multiplier 15 and an adder 16. The slip angular velocity calculation circuit 14 is the first coordinate conversion circuit 1
3 is connected to the output line of the magnetic flux axis component current I _{1d and} the output line of the torque axis component current I _1q . The slip angular velocity calculation circuit 14 is configured to obtain the slip angular velocity ω _se by the calculation of the following equation (2). ω _se = R2 (M / L2) ( _I1q / _φ2d ^* ) (2) where R2 is the secondary winding resistance of the induction motor 1, M is the mutual inductance, and L2 is the secondary self-inductance. , Φ
2d ^* is a desired secondary interlinkage magnetic flux represented by the d-axis (magnetic flux axis) component of the dq coordinate. The constants of the induction motor 1 other than the above used in the following description are shown below. R1
Is the primary winding resistance, L1 is the primary self-inductance, Lσ
Is the leakage inductance, and Lσ = (L1 L2-M
² ) / L2, p is the number of pole pairs.

The reason why the slip angular velocity ω _se is obtained by the equation (2) will be described below. Based on the well-known vector control principle, the voltage equation of the induction motor 1 on the dq coordinates rotating at the angular frequency ω of the primary voltage of the three-phase induction motor 1 is V _{1d of the} primary voltage of the magnetic flux axis component. , The primary voltage of the torque axis component orthogonal to this is V _1q , and the primary current of the magnetic flux axis component is I
_1d , the primary current of the torque axis component is I _1q , the secondary magnetic flux of the magnetic flux axis component is φ _2d , and the secondary magnetic flux of the torque axis component is φ _2q ,
It is expressed by the following expression (3), and the torque T generated by the induction motor 1 is expressed by the following expression (4). It should be noted that P in Expression (3) represents a differential operator, and the secondary voltage becomes 0 because the terminal is short-circuited due to the structure of the squirrel cage induction motor.

[0015]

[Equation 2]

In vector control for controlling the slip frequency, the q-axis of the secondary magnetic flux is calculated by giving the primary angular velocity ω by the equation (5) while the d-axis component of the secondary magnetic flux is constant. The component becomes zero, and it becomes possible to generate the torque T proportional to the rotor magnetic flux φ _2d and the current I _1q which is orthogonal to the rotor magnetic flux φ _2d as shown in the equation (6), similarly to the principle of torque generation of the DC motor. ω = pω _rm + R2 (M / L2) (I _1q / φ _2d ) ... (5) T = p (M / L 2) I _1q φ _2d (6) Here, the d-axis direction is the magnetic flux direction. Since q is selected, the q-axis becomes the torque component axis. The second term on the right side of the equation means the slip angular velocity ω _se .

The secondary magnetic flux φ _2d in equation (5) is
Secondary magnetic flux command value φ _2d supplied by line 17 in FIG.
It can be given as ^* . The d-axis secondary magnetic flux command value φ _2d ^* of this line 17 is the 1 / M multiplier, that is, the M divider 1
By dividing by 8 by M, the d-axis component current command value I _1d ^* is obtained on this output line 19. In addition, line 1
7, the divider 18 and the line 19 are the magnetic flux axis component current command value I
_It functions as the second command means for giving _1d ^* .

The primary angular velocity calculation circuit 12 will be described again.
In this circuit 12, the slip angular velocity ω _se obtained according to the equation (2) is input to the adder 16. The other input of the adder 16 is that the rotational angular velocity ω _rm obtained from the velocity detection A / D converter 6 is multiplied by the pole pair number p in the multiplier 15 to convert a change in mechanical angle into a change in electrical angle. It was done. Therefore, the adder 16 calculates ω = pω _rm + ω _se to obtain the primary angular velocity ω. Here, ω is 1 from the viewpoint of rotational motion.
It is called the secondary angular velocity, which is the primary angular frequency from the electrical point of view based on the input voltage of the induction motor 1,
Further, ω _se is a slip angular frequency obtained by multiplying the slip frequency by 2π. The primary angular velocity (primary angular frequency) ω is integrated by the integrating circuit 20 indicated by 1 / s, converted into a phase angle θ, and sent to the first and second coordinate conversion circuits 13 and 21. Phase angle θ
Indicates the phase of the sine wave input voltage and is used to determine the values of sin θ and cos θ used for coordinate conversion.

In the control system shown in FIG. 1, the torque axis component current I _1q is controlled so as to match the torque axis component current command value I _1q ^* output from the speed controller 11, and the magnetic flux axis component current command value I _1d. ^The magnetic flux axis component current I _1d is controlled so as to coincide with ^* . In order to execute this control, the subtracters 22 and 23 as the first and second difference signal forming means, the first and second subtractors
Current controller (ACR) 24, 25 as compensation means for
, Torque axis component voltage compensation value calculation means 26, magnetic flux axis component voltage compensation value calculation means 27, adders 28 and 29 as first and second addition means, and second coordinate conversion circuit 21.
, D / A converters 29, 30, 31 and a PWM control signal forming circuit 32 are provided.

The first subtractor 22 forms a signal (error signal) indicating the difference between the torque axis component current command value I _1q ^* and the torque axis component current I _1q , which is the first current controller 24.
The torque axis component voltage command reference value V _1q ′ is compensated by. The second subtractor 23 forms a signal (error signal) indicating the difference between the magnetic flux axis component current command value I _1q ^* and the magnetic flux axis component current I _1d , which is compensated by the second current controller 25. Thus, the magnetic flux axis component voltage command reference value V _1d ′ is obtained. Like the speed controller 11, the first and second current controllers 24 and 25 are, for example, well-known proportional-plus-integral compensation circuits.

By the way, in principle, the inverter 3 can be controlled based on the outputs of the first and second current controllers 24 and 25 to obtain a target rotation state. But,
In FIG. 1, means for canceling the influence of disturbance is added. This disturbance is interference components (interference terms) ΔV _1d and ΔV _1q on the d axis (magnetic flux axis) and the q axis (torque axis). The torque axis component voltage compensation value calculation means 26 is connected to the primary angular velocity calculation circuit 12, and although not shown,
It is connected to the I _1d line 13b and the φ _2d ^* line 17, and the torque axis component voltage compensation value ΔV is calculated by the following equation (7).
Form _1q '. ΔV _1q ′ = −ΔV _1q = ω {LσI _1d + (M / L 2) φ _2d ^* } (7) The first adder 28 calculates the torque axis component voltage command by the following equation (8). Outputs the value V1q ^* . V _1q ^* = V _1q ′ + ΔV _1q ′ (8)

The magnetic flux axis component voltage compensation value calculation means 27 has a value of 1
The magnetic flux axis component voltage compensation value ΔV _1d ′ is formed by the calculation of the following equation (9), which is connected to the next angular velocity calculation circuit 12 and also connected to the I _1q line 13a (not shown). ΔV _1d ′ = −ΔV _1d = −LσωI _1q (9) The second adder 28 outputs the magnetic flux axis component voltage command value V _1d ^* by the calculation of the following equation (10). _{^{V1d * = V 1d '+ ΔV}} 1d' ··· (10)

The second coordinate conversion circuit 21 uses the following equation (1) based on the voltage command values V _1q ^* , V _1d ^{* of} each axis component and the phase angle θ.
The coordinate conversion from the two-axis orthogonal coordinates to the three-phase AC static coordinates is performed by the calculation according to 1), and the three-phase primary voltage command value V _1u ^* ,
Outputs V _1v ^* and V _1w ^* .

[0024]

(Equation 3)

The w-phase voltage command value V _1w ^* is expressed by the equation (11).
Based on the u and v phase voltage command values V _1u ^* and V _1v ^{* obtained} in step _S1 , the following equation (12) is used. _V1w ^* = _-V1u ^* _-V1v ^* ... (12)

The three-phase voltage command values V _1u ^* , V _1v ^* , and V _1w ^* obtained from the second coordinate conversion circuit 21 are the D / A converter 3
The signals are converted into analog signals at 0, 31, and 32 and sent to the PWM control signal forming circuit 33. The PWM control signal forming circuit 33 includes a well-known triangular wave generating circuit 34a and a comparator 34b shown in FIG. 3, and has voltage command values V _1u ^* , V _1v ^* , V
Fig. 4 (A) shows _1w ^* and a triangular wave voltage Vt of 10kHz, for example.
The PWM control pulse shown in FIG. 4B is generated by comparison as shown in FIG. Although only one phase is shown in FIGS. 3 and 4, a PWM control signal is similarly formed in each phase. The PWM control signal is supplied to the switches Q1 to Q6 of the inverter 3 shown in FIG. The triangular wave voltage V
The frequencies of t and the PWM pulse are set sufficiently higher than the frequencies of the target output voltages V _1u , V _1v , and V _1w .

The controller of FIG. 1 obtains the rotational angular velocity ω _rm and the primary angular velocity ω based on the output of the velocity detector 4. Therefore, it has been difficult to reduce the cost and size of the device.

[0028]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS The induction motor controller of FIG. 5 is obtained by omitting the rotation detector 4 from the controller of FIG. In FIG. 5, those common to FIG. 1, that is, reference numerals 1 to 3 and 7-1
1, 11a, 13, 14, 15 and 32 are shown in FIG.
Since they are substantially the same as those denoted by the same reference symbols in FIG.

The control apparatus of FIG. 5 is provided with a primary angular velocity estimation calculation circuit 50 in place of the primary angular velocity calculation circuit 12 in FIG. 1, and also in place of the rotation detector 4 and A / D converter 6 of FIG. The configuration is the same as that of FIG. 1 except that a rotational angular velocity estimation calculation circuit 51 is provided. The first-order angular velocity estimation calculation circuit 50 according to the present invention includes an I _1q line 13a, an I _1d line 13b, and a second line.
V _1d ′ output line of the current controller 25 and the first adder 2
It is connected to the V _1q ^* output line of 8 and outputs the primary angular velocity (primary angular frequency) ω. Primary angular velocity calculation circuit 1 of FIG.
The output ω of 2 is an actual measurement value, and the output of the primary angular velocity estimation calculation circuit 50 of FIG. 5 is an estimated value. Although both are not exactly the same, the estimated primary angular velocity is also shown in FIG. 5 for convenience of explanation. Denote by ω.

The primary angular velocity estimation calculation circuit 50 of FIG. 5 is shown in FIG.
As shown in detail in FIG.
And obtain the temporary primary angular velocity ω ₀ . This provisional first-order angular velocity estimation calculation circuit 60 uses PLσ + R1 multiplier 61, adder 62, M / (L1 φ) to calculate the following equation (13).
_2d ^* ) having a multiplier 63. _{ω 0 = (M / L1)} {V 1q * - (PLσ + R1) I 1q} / φ 2d * · (13)

Next, the reason why the provisional primary angular velocity ω ₀ is obtained by the equation (13) will be described. In the voltage equation of the equation (3), when the primary voltages V _1d and V _1q of the d-axis and q-axis components are extracted, the following equations (14) and (15) are obtained. _{V 1d = (PLσ + R1)} I 1d -LσωI 1q + (M / L2) Pφ 2d - (M / L2) ωφ 2q ··· (14) V 1q = LσωI 1d + (PLσ + R1) I 1q + (M / L2) ωφ _2d + (M / L 2) Pφ _2q (15) Here, in the primary voltage equation for the q axis of the equation (15), φ _2q = Pφ _2d = which is the condition for establishing the vector control theory.
By substituting 0 and obtaining ω, the following equation (16) is obtained. ω = (M / L1) { V 1q - (PLσ + R1) I 1q} / φ 2d ··· (16) the omega of the formula (16) ω _0, the φ _2d φ _2d ^*, the V _1q V _1q
^{If *} , then the above-mentioned equation (13) is obtained.

The equations (13) and (16) are arithmetic expressions derived on the assumption that vector control is established. However, in reality, the voltage V _1q and the current I
It is desirable to perform correction processing in consideration of the fact that vector control is not established due to factors such as detection errors such as _1q and calculation errors. In FIG. 6, a d-axis induced voltage estimation arithmetic circuit 64 is provided to execute this correction. This arithmetic circuit 64 uses PLσ + R1 in order to calculate the following equation (17).
It has a multiplier 65 and an adder 66. e _2d ′ = V _1d ′ − (PLσ + R ₁ ) I _1d (17) The multiplier 65 multiplies I _{1d on the} I _1d line 13 b by PL σ + R ₁ . The adder 66 outputs the output V of the second current controller 25.
_1d ′ and the output of the multiplier 65 are added.

Next, it will be explained that the equation (17) shows the correction amount. In the primary voltage equation indicating V1d shown in the equation (14), the third and fourth terms on the right side are the secondary induced voltage e2d of the d-axis component, which can be arranged as shown in the following equation (18). Note that LσωI _{1q in} the second term on the right-hand side of Expression (14) is the d-axis interference term ΔV _1d as shown in Expression (9). e _2d = (M / L 2) Pφ _2d − (M / L 2) ωφ _2q = V _1d + LσωI _1q − (PLσ + R ₁ ) I _1d (18) This induced voltage e _2d is under the condition that vector control is established. Paying attention to the fact that it is zero, the induced voltage is estimated by calculation, and the first-order angular velocity ω ₀ tentatively obtained so as to be zero is corrected. As shown in the following equation (19), the estimated calculation of the induced voltage is the voltage command value V _1d ^* of the d-axis component, the interference compensation value ΔV _1d ′, the d-axis component current I _1d , the leakage inductance Lσ, and the primary winding. This can be done by using the resistor R1. e _2d ′ = V _1d ^* + Δ _1d ′ − (PLσ + R ₁ ) I _1d (19) To further simplify the calculation, V _1d ′ = V _1d ^* + ΔV _1d ′ can be obtained by the formula ( ₁₎ ^. Substituting into (19), the above equation (17) is obtained.

In this embodiment, the temporary primary angular velocity ω _{0 is set} so that the induced voltage e _2d ′ estimated according to the equation (17) becomes _zero.
To obtain an estimated first-order angular velocity ω. A primary angular velocity error correction circuit 67 is provided to execute the correction by the d-axis induced voltage e _2d ′. In this correction circuit 67, a signal of the difference between the induced voltage command value e _2d ^* set to zero on the line 68 and the input d-axis induced voltage ed 'is formed by the subtractor 69, and this signal is proportional to integral (PI) compensation. Circuit (PI
It is sent to the adder 71 via the controller 70. The adder 71 of FIG. 6 adds the correction amount given from the primary angular velocity estimation error correction circuit 67 to the provisional primary angular velocity ω ₀ , and outputs the estimated primary angular velocity (primary angular frequency) ω. The correction of FIG. 6 can be expressed by the following equation (20). ω = ω ₀ + (e _2d ^* -e _2d ′) (Kp + Ki / s) ··· (20)

By the way, it is desirable to adopt a method according to the following equation (21) in order to perform correction without delay. _{ω = ω 0 + {sgn (} ω 0)} [{M / (L1 φ 2d)} e 2d '] ·· (21) [ in the formula (21) {M / (L1φ 2d)} e 2d ’]
Is a value obtained by converting the estimated induced voltage into an angular velocity amount (angular frequency amount), and sgn (ω ₀ ) is a temporary primary angular velocity 63
Is the polarity value of. FIG. 7 shows a circuit for executing the correction of the equation (21). The correction circuit 67 shown in FIG.
The configuration is the same as that of FIG. In FIG. 7, the multiplier 72 sets the d-axis induced voltage e _2d ′ to M /
(L1 φ _2d ^* ) is multiplied to obtain the angular velocity amount, which is input to the next multiplier 74. In addition to the angular velocity amount, the provisional primary angular velocity polarity sgn (ω ₀ ) obtained from the provisional primary angular velocity polarity circuit 73 is input to the multiplier 74, and the product of both is sent to the adder 71 as a correction value. .

[0036]

[Rotation Angular Velocity Estimation Calculation Circuit] The rotation angular velocity estimation calculation circuit 51 of FIG. 5 is formed to calculate the estimated rotation angular velocity ω _rm ′ by calculating the following equation (22). ω _rm ′ = (ω−ω _se ) / p (22) That is, in the subtractor 52, the primary angular velocity estimation calculation circuit 50
The slip angular velocity (slip angular frequency) ω _se obtained by the slip angular velocity calculation circuit 14 is subtracted from the estimated first-order angular velocity (angular frequency) ω obtained in step 1 to form a signal indicating the difference between them, and 1 /
By dividing ω-ω _se by the number of pole pairs p in the p multiplier 53, the electrical angle is converted into a mechanical angle to obtain the rotational angular velocity ω _rm ′.
This rotational angular velocity ω _rm ′ is used by the subtractor 9 similarly to the rotational angular velocity ω _rm obtained by the velocity detector 4 of FIG.

As described above, according to the control device of FIG. 5, the primary angular velocity (angular frequency) ω is obtained without using the velocity detector, and the coordinate conversion circuits 13 and 21 and the voltage compensation values of the q-axis and the d-axis are obtained. It can be used in the computing means 26 and 27, and vector control becomes easy. Further, the rotation angular velocity estimation calculation circuit 51
The speed control can be easily performed by _{obtaining the} estimated rotational angular speed ω _rm ′ and performing feedback control with.

[0038]

MODIFICATION The present invention is not limited to the above-mentioned embodiment, but can be variously modified according to well-known techniques. For example,
The torque axis component current command value I _1q ^* in FIG. 5 may be obtained by giving the torque command value T ^* from the line 80 as shown in FIG. In FIG. 8, a torque axis component current command value I _1q ^* is obtained by multiplying the torque command value T ^* on the line 80 by L 2 / (pMφ _2d ^* ) in the multiplier 81.
8, the same parts as those in FIG. 5 are designated by the same reference numerals and the description thereof will be omitted. The PWM control circuit 33 of the inverter 3 can be various known circuits. Further, although the digital control portion is performed by the microcomputer in the embodiment, it may be constituted by an individual circuit. Further, the line 17 and the 1 / M multiplier 18 can be omitted, and the d-axis component current command value I _1d ^* can be directly input from the line 19. Further, in the operation expression using the phi _2d ^*, may be used MI _1d ^* instead of phi _2d ^*. Moreover, each constant of the induction motor 1 may be a measured value, a design value, or a calculated value.

[Brief description of drawings]

FIG. 1 is a block diagram showing a conventional control device for an induction motor.

FIG. 2 is a circuit diagram showing the inverter of FIG.

FIG. 3 is a circuit diagram showing a part of the PWM control signal forming circuit of FIG.

FIG. 4 is a waveform diagram of each part of FIG.

FIG. 5 is a block diagram showing a control device for an induction motor according to an embodiment of the present invention.

6 is a block diagram showing a primary angular velocity estimation calculation circuit of FIG.

7 is a block diagram showing another example of the primary angular velocity estimation calculation circuit of FIG.

FIG. 8 is a block diagram showing a control device for an induction motor of a modified example.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Induction motor 3 Inverter 13 and 21 Coordinate conversion circuit 50 Primary angular velocity estimation arithmetic circuit

Claims (8)

[Claims] 1. A three-phase PWM control inverter connected between a DC power supply and a three-phase induction motor; and a primary current for at least two phases among three-phase primary currents of the induction motor. A current detector (5u, 5v) and a torque axis component current I _1q and a magnetic flux axis component orthogonal to the primary current detected by the current detector by rotational coordinate conversion synchronized with the rotation phase of the induction motor. First conversion means (13) for converting to a current I _1d , and first command means (11a ⁾ for giving a torque axis component current command value I _1q ^* for rotating the induction motor to a desired state.
) And second command means (1 ⁾ for giving a magnetic flux axis component current command value I _1d ^* for rotating the induction motor to the desired state.
9) and a torque shaft component voltage command value V _1q ^* for _obtaining a difference between the torque shaft component current command value I _1q ^* and the torque shaft component current I _1q, and making the difference zero. The component voltage command value forming means determines the difference between the magnetic flux axis component current command value I _1d ^* and the magnetic flux axis component current I _1d, and _sets the magnetic flux axis component voltage command value V _{1d *} for making this difference zero. The magnetic flux axis component voltage command value forming means for forming, the torque axis component voltage command value V _1q ^*, and the magnetic flux axis component voltage command value V _1d ^* are three-phase voltage command values V _1u ^* , V _1v ^* , V
Second conversion means for converting into _1w ^* and (21), forming a pulse width modulation (PWM) control signal corresponding to the three-phase voltage command values V _1u ^* , V _1v ^* , V _1w ^* An inverter control circuit (33) for controlling, the torque axis component voltage command value V _1q ^* , the torque axis component current I _1q , the magnetic flux axis component current command value I _1d ^*, and the primary winding resistance R 1 of the induction motor. Based on the leakage inductance Lσ, the primary self-inductance L1 and the differential operator P ω ₀ = (1 / L1) [{V _1q ^* -(PLσ + R1)
I _1q} / I _1d ^*] or a primary angular velocity estimating means for obtaining an estimated primary angular velocity omega ₀ This by equivalent calculation, to form a signal indicative of the rotational phase based on the estimated primary angular velocity omega ₀ A control device for an induction motor, comprising: a phase angle signal forming means (20) applied to the first converting means (13). 2. The torque shaft component voltage command value forming means forms a first difference signal corresponding to a difference between the torque shaft component current command value I _1q ^* and the torque shaft component current I _1q .
Difference signal forming means, first compensating means for compensating the first difference signal to form a torque axis component voltage command reference value V _1q ′, and primary angular velocity estimated by the primary angular velocity estimating means. ω ₀ or the estimated primary angular velocity ω _a composed of this corrected primary angular velocity ω, the leakage inductance Lσ, the magnetic flux axis component current I _1d , the mutual inductance M, and the secondary self-inductance L.
2 and ΔV _1q ′ = ω _a {LσI _1d + (M ² / I _1d ^* ) / L 2} based on the magnetic flux axis component current command value I _1d ^* or torque axis component voltage compensation by performing an equivalent calculation. A torque axis component voltage compensation value calculation means for obtaining a value ΔV _1q ′, and a torque axis component voltage command value V by adding the torque axis component voltage compensation value ΔV _1q ′ to the torque axis component voltage command reference value V _1q ′. _A first adding means for obtaining _1q ^* , wherein the magnetic flux axis component voltage command value forming means corresponds to a difference between the magnetic flux axis component current command value I _1d ^* and the magnetic flux axis component current I _1d . Difference signal forming means for forming a difference signal of the magnetic flux axis component, second compensating means for forming a magnetic flux axis component voltage command reference value V _1d ′ by compensating the second difference signal, and the primary angular velocity omega ₀ or the leakage inductor and the estimated primary angular velocity omega _a consisting of the corrected primary angular velocity omega Chest Lσ and the torque axis component current based on the _{I 1q ΔV 1d '= -Lσω a} I 1q or its equivalent operation to be performed flux axis component voltage compensation value Δ
'Flux axis component voltage compensation value calculating means and the magnetic flux axis component voltage command reference value V _1d to obtain a' flux axis component voltage command value V _1d plus the flux axis component voltage compensation value [Delta] V _1q 'to V _1d
2. The control device for an induction motor according to claim 1, further comprising a second adding means for obtaining ^* . 3. The primary angular velocity estimating means further obtains a magnetic flux axis component induced voltage estimated value e _2d ′ by e _2d ′ = V _1d ′ − (PLσ + R ₁ ) I _1d or a calculation equivalent thereto, and the magnetic flux Estimated axial component induced voltage e _2d ′
3. The control device for an induction motor according to claim 2, further comprising a correction unit that forms a primary angular velocity estimation error correction signal based on the above and adds it to the primary angular velocity ω ₀ . 4. The second command means supplies a secondary magnetic flux command value φ _2d ^* of the induction motor, and the secondary magnetic flux command value φ _2d ^* of the mutual inductance M.
4. The control device for the induction motor according to claim 1, 2 or 3, further comprising means for obtaining the magnetic flux axis component current command value I _1d ^* by dividing by. 5. The calculation of the estimated first-order angular velocity ω ₀ is performed by ω ₀ = (M / L1) [{V _1q ^* − (PLσ + R 1) I _1q } / φ _2d ^* ]. Induction motor control device described. 6. The torque axis component voltage compensation value ΔV1q ′
3. The control device for the induction motor according to claim 2, wherein ΔV _1q ′ = ω _a {LσI _1d + (M / L 2) φ _2d ^* } is calculated. 7. The first command means is R2 (M / L2) ( _I1q / _φ2d ^* ) or R2 (1 / L2) ( _I1q / _I1d ^* ) (where R2 is a secondary winding). line resistance, L2 is a means for obtaining a slip angular velocity omega _se by the secondary self-inductance), and means for subtracting the slip angular velocity omega _se from the estimated primary angular velocity omega ₀ or a correction value omega, the value obtained by the subtraction Is divided by the number p of pole pairs of the induction motor to obtain an estimated rotational angular velocity ω _rm ′, a means for supplying a rotational angular velocity command value ω _rm ^* for achieving a desired rotational angular velocity of the induction motor, and the rotational angular velocity Means for forming an error signal between the command value ω _rm ^* and the estimated angular velocity ω _rm ′, and the torque axis component current command value I corresponding to the error signal
7. The control device for an induction motor according to claim 1, 2 or 3 or 4 or 5 or 6, further comprising means for forming _1q ^* . 8. The first and second converting means, the torque axis component voltage command value forming means, and the magnetic flux axis component voltage command value forming means are formed by a microprocessor or a digital signal processing device. Claim 1 characterized by
A control device for an induction motor as described in the above.