JPS62264709A - Detection circuit for automatic gain control - Google Patents

Abstract

PURPOSE:To eliminate the temperature fluctuation and to attain a large amplitude input by adding outputs of two sets of differential amplifiers receiving signals of opposite phase to each other and making a supply current to the amplifiers and the input signal both proportional to temperature. CONSTITUTION:Transistors Q21-Q24 are constituted in such a way that two sets of the differential amplifiers share output resistors. The phase of a signal Vin1 inputted to one set and the phase of a signal Vin2 inputted to the other set are opposite to each other. Since an output voltage appearing at terminals 12A, 12B is proportional to a full wave rectifier voltage of the input Vin, the voltage is used as an AGC output signal through a low-pass filter. In making the input signal Vin and a supply current Iee proportional to the absolute temperature, the temperature fluctuation of the mutual conductance of the differential amplifiers is compensated. Further, the amplitude of the input signal is increased more than that using a conventional double balanced type multiplier.

Description

[Detailed description of the invention] [Purpose of the invention] (Industrial application field) The present invention relates to a detection circuit for automatic gain control.

(Prior Art) Conventionally, a circuit as shown in FIG. 7, for example, is known as an automatic gain control detection circuit (hereinafter referred to as AGC rough circuit). This conventional AGC detection circuit is composed of a well-known double balanced multiplier.

This double balanced multiplier includes first, second and third differential amplifiers D1. I)2. It consists of D3.

Second and third differential amplifier D2. D3 are connected in series to two current paths of the -th differential amplifier D1, respectively. Transistors Q1 and Q2 forming the first differential amplifier D1
The base is Toshi Jinriki 11A.

11B. The collectors of the transistors Q3 and Q4 forming the second differential amplifier D2 are connected to the output links 12 and 123, respectively. Furthermore, the collectors of transistors Q5 and C6 constituting the third differential amplifier D3 are also connected to the output terminals 12A and 12B, respectively.

A first reference voltage Vl is applied from a first reference voltage source El to the bases of transistors Q4 and C5 via a resistor R3. The second reference voltage Vl is connected to the resistor R4 from the second reference voltage source E2.
to the bases of transistors Q3 and C6. Further, the drive voltage V3 is supplied from the drive voltage source E3 to the resistor R1.
is applied to the collectors of transistors Q3 and C5 via resistor R2, and is also applied to the collectors of transistors Q4 and C6 via resistor R2. The base of transistor Q1 is connected to the bases of transistors Q3 and C6 via capacitor C1, while the base of transistor Q2 is connected to the bases of transistors Q4 and C5 via capacitor C2.

AGC detection output vout is transmitted by transistors Q3 and C5.
and the connection point between the collectors of transistors Q4 and C5, that is, between the output terminals 12A and 123. A connection point between the emitters of transistors Q1 and C2 is connected to a current source 13.

In the above circuit, the collector currents IO1 to Cce of each transistor Q1 to Qe have input signal voltages van,
Letting the current of the current source 13 be Ice, it is generated as follows. (Vt is the thermal voltage of the transistor) From equations (1) to (6), the AGC output voltage Vout is the output terminal 12A, 12
Assuming that the load resistance of the load circuit 12 connected between B and B is R1, the following is required.

(Left below) vout-R1・1out-81・((Icafu”C5
)−(+.e+ 'c4)] San R1・ ([Ic3−
IC et al.)-(IC4+1c5))-R1・I・(t
anh(Vin/2vt)) ・---
---mle Here, if Vin < Tt, Vout-ru ・(Vin/2 Vl2
......(12) e.

Here, the input signal voltage Vin is expressed as its instantaneous value vo sin
If it is replaced by ω↑ (blank space below), then if this AGC output voltage vout is taken out via a low-pass filter (not shown), the following will be obtained.

vout-”-RI 1 (=12dt27r
ee 2Vt As mentioned above, the conventional AGC detector has a DC AGC output voltage vou that is proportional to the input signal system width value ■0 squared and inversely proportional to the square of the transistor thermal voltage Vt.
It can be understood that t can be obtained.

(Problems to be Solved by the Invention) As mentioned above, the conventional AGC detector has 1/V t2
Obtain the AGC output voltage vout proportional to . The thermal voltage Vt of the transistor is Vt = kT/Q (k
: Holtzmann constant, q: electronic charge, T: absolute temperature). Therefore, the AGC output voltage VOLI t is
It is inversely proportional to the square of the absolute temperature T, and is subject to significant fluctuations due to temperature fluctuations. Moreover, the AGC output voltage is V.

The DC potential of ut becomes high, and the output system width easily saturates at its maximum value.

As a means to solve such problems, for example, current source 1
An attempt has been made to make the current Ice of 3 proportional to the absolute temperature to compensate for the variation in the AGC output voltage vout.

An example of a current source having an output current proportional to the absolute temperature is the circuit shown in FIG. 8, for example.

In FIG. 8, transistor Q11. Q13 are transistors of the same size, each emitter connected to a resistor R
11, R13, and transistor Q12 is grounded via transistors Q11.11 and R13. This transistor has an emitter size N times larger than Q13. The emitter of transistor Q12 is connected to the emitter of transistor Q11 via resistor R12. Transistor Q14.
Q15. Qle and resistors R14°R15 constitute a current mirror circuit, and transistors Q11. An equal current Io is applied to the collectors of Q12. Also, transistor Q17. Q18. Qle and resistance R16 to R20
constitutes a starter circuit.

Here, Vbe11, Vbe12 and Vbe13 are connected to transistors Q11. The voltage between the bases and the terminals of Q12 and Q13 is taken as the voltage between the bases of Q12 and Q13, and I2 is the voltage between the transistors Q11. Assuming the reverse saturation current value of Q13, Vbcll, Vbe1
2 and Vbe13 are respectively 'bei1- Vbe12"" 12 ""
””・(15)v −V −Vt−un (I
o/Is) −・−−−・−(16)beH
be+3 V −Vt−, Qn (Do/(N Is))]−
・----IA71e12 Hail.

(15) to (17) can be obtained from equations (15) to (17). From equation (16), since the upper drops of R11 and R13 are equal and β>>Q, the output current Ice(t) flowing through transistor Q13 is given by the following equation.

R13・Ie, (El k Itll・(2・Io)
By substituting ① and =kT/q into the attack, we get .

Therefore, with the above circuit, an output current Ice(t) proportional to the absolute temperature T can be obtained at the collector of the transistor Q13.

Using the above current source, the AGC detector can reduce the effects of temperature fluctuations to a downwardly proportional dimension. However, fluctuations due to temperature fluctuations will still remain. In other words, this is because the above current source cannot obtain a current output proportional to T2.

One possible method for further reducing the influence of temperature is to switch the differential circuits D2 and D3 shown in FIG. 7. In this case, equation (14) appears as follows.

In this way, the influence of temperature fluctuation can be reduced to the AGC output voltage V
It can be reduced to the point where out is proportional to the bottom.

However, the differential circuit D2. To completely switch D3, it is necessary to create a large-amplitude switching signal that is completely synchronized with the input signal @l/in. For this purpose, for example, a phase-locked loop (PLL) circuit is required, which complicates the overall circuit. As an example of a means for relatively easily obtaining the switching signal, it is possible to add a cut-off circuit to amplify the input signal. However, it is easy to cause a phase shift between the switching signal and the input signal N Vin, and therefore the AGC output voltage vout
Errors occur. In this case, the error increases particularly in the high frequency range. Further, in order to maintain appropriate DC levels of the input signal Vin and the switching signal, many level shift circuits are required, but these level shift circuits are instead affected by temperature fluctuations.

In addition, as a means to reduce the influence of temperature fluctuations, a method is also proposed in which a resistor of a size such that the thermal voltage Vt can be sufficiently ignored is connected to each emitter of the transistors 01 to Q6.

Letting their resistance values be Re, when Re >> 2Vt/Ice, the AGC output voltage Vout is as follows. In order to satisfy this formula, the above condition R6>>
2 Vt/lee is required. For example, Ice=
1 mA, Vt = 26 mV (at T = 300'), it is assumed that 2t/Iee≦0.01 Re is a condition where Vt can be ignored. In this case, Re≧5.2
requires a condition of Ω. However, the input signal Vin
= 0.5V, in order to obtain VOUT = 0.1V, R1 = 4.32 kΩ is sufficient in the circuit of Fig. 7, whereas when Ω is added to Re = 5.2,
R1=86, 5 kΩ is required. Also, in this case, considering the value of Ice-1mA, in the circuit of Figure 7, the voltage drop across each resistor R1 and R2 is only 2.16V, but if Ω is added to Re = 5.2. So,
It reaches as much as 47V. Also, taking into consideration the saturation conditions of the transistors etc., the required power supply voltage is only about 8V at most in the circuit of Fig. 7, whereas Re = 5.
．． If 2kG is added, a power supply voltage of 55V is required. Therefore, the means of connecting a resistor to the emitter of a transistor is not practical as a means of reducing the influence of temperature fluctuations.

As explained above, in the conventional AGC detector,
Sufficient measures to avoid the adverse effects of temperature fluctuations have not been obtained, and products with high precision and a wide range of use have not been obtained. In addition, although it is possible to partially avoid the negative effects of temperature fluctuations, this only increases the complexity of the circuit, increases power consumption, and
Problems such as an increase in the required power supply voltage arise, and the range of use is rather narrowed.

This invention was made in view of the above circumstances, and has few restrictions on the amplitude (direction) of input/output signals, can operate satisfactorily even at high input signal frequencies, and has high accuracy, making it easy to eliminate fluctuations in output signals due to temperature fluctuations. It is an object of the present invention to provide a detection circuit for automatic gain control, which has a simple configuration suitable for IC implementation, has low power consumption, is capable of low voltage operation, and has low power consumption.

[Structure of the invention]

(Means for Solving the Problems) The present invention, for example, as shown in FIG.
21. Q22 group and 023. The transistors Q21 . A DC voltage V2 is commonly supplied to the bases of transistors Q23 .
Each base of Q24 is supplied with an input signal having an amplitude proportional to the absolute temperature in opposite phases to each other by an input signal processing circuit as shown in FIG. There is a configuration in which a current proportional to temperature is supplied to the emitter of each of the transistors 021 to Q24, and the output is obtained from the collector.

(Function) In the above configuration, the input signal Vin and the current Iee of the current source 13 are both proportional to the absolute temperature and cancel each other out, so that the mutual conductances gm1 to gm4 of the transistors 021 to Q24 no longer have a temperature coefficient. It is possible to obtain an AGC output voltage that is not affected by temperature fluctuations.

(Embodiments of the Invention) Embodiments of the invention will be described below with reference to drawings 1 to 6. In order to simplify the explanation, the same or equivalent circuit elements will be referred to throughout the drawings. It uses signs and symbols.

FIG. 1 shows an embodiment of the present invention, in which a driving power source E3 is connected to the collectors of transistors Q21, Q22 through a resistor R21, and to the collectors of transistors Q23.Q22 through a resistor R22. It is connected to the collector of Q24 and applies a driving voltage 3 to each transistor Q21 to Q24. Transistor Q21. A reference voltage source E2 is connected to the base of Q22, and a voltage V2 is applied thereto. The bases of transistors Q23 and Q24 are connected to input terminals 11A and 11B, respectively, which are connected to input signal sources. Also, transistors Q 21 and Q 22 .

Q23. The respective emitters of Q24 are connected in common and are connected to a ground potential source G via a constant current source 13. That is, the collector-emitter paths of transistors Q21 and Q22 are connected in parallel with each other, and the collector-emitter paths of transistors Q23 and Q24 are connected in parallel with each other. Further, the bases of transistors Q21 and Q22 are both connected to base Q voltage source E2, and transistor Q
The bases of 23 and Q24 are connected to both ends of the input signal source 11. The AGC output voltage vout is obtained between the connection point between the collectors of transistors Q21 and Q22 and the connection point between the collectors of transistors Q23 and Q24.

In the actual circuit design, transistors Q21 and Q22 are created in the form of a single-circuit multibase transistor Qm1, and transistors Q23 and Q24 are also created in the form of another single-circuit multibase transistor Q-2. .

Both bases of one multi-base transistor Q+111 are connected to the reference voltage source E2, and both bases of the other multi-base transistor Q-a are connected to the input terminal 11, respectively.
A, connected to IIB.

Reference voltage V2 is supplied from reference voltage source E2 to transistor Q21.
and the bases of Q22, and the input signal Vin
is applied from input signal source 11 between the bases of transistors Q23 and Q24. The input signal sources 11 each have an input signal V having a phase opposite to each other and having a similar width proportional to the absolute temperature T.
Generates inl and Vin2. Note that for convenience of explanation, Vin will be used hereinafter to represent both Vinl and Vin2. Furthermore, a DC bias Vd is applied between the input terminals 11A and 11B superimposed on the input signal Vin. This DC bias Vd is made to have a value equivalent to the reference voltage V2 in order to harmonize the bias states of each transistor Q21 to Q24. Therefore, the input signal source 11 is equivalent to the input signal Vin
Two signal sources giving l and Vin2 and DC voltage Vd
It is represented as having a DC bias source Ed that gives . Therefore, the DC bias Vd is superimposed on the input signal Vin at each input terminal 11A and 113. That is, Vin-a -Vd+Vin-Vd-Nlo S
inωt ・−・−・−(23) However, Vi
na is the potential of the input terminals 11A and IIB, and VO is the amplitude of the input signal Vin as described above.

Input signal Vin and AGC output voltage vo in this circuit
The direct current (OC) transfer characteristic between u is as follows.

Here, the current of the constant current source 13 is ICe, and the transistor 0
The emitter potential of 21 to Q24 is set to VC1 for each transistor Q.
Let the collector currents of 211 to Q24 be IC21 to IC24.

Further, it is assumed that each of the transistors 021 to Q24 has the same characteristics. Then, each transistor 021 to Q24
The reverse saturation currents l321 to l524 and the current amplification factor α
21 to α24 are given as being equal to each other.

That is, 's21- 's22- 's23- 's24 ””
IS ・・・・・・・・・(24) 2+ 2
2 24 ・・・・・・・・・(25)
Open it. Next, the relationship between the collector currents of each transistor 021 to Q24 is required as follows.

Vd Ve , , , , , (261
1c21-1s-eXD -VE Vd-1/e (27
)1.22-Is-exp-V [Vd-ve+v.

Ic23-1s-ey, o 1Jt-・-・(2B
11 -Is・QXp “Go traverse” ・・・・・・
...(2) c24 Vt' ” ” 'c21 '''c22'''c2
3 ””c24 )8e α ”’c21”’c22 ”’c23 ”’c24
°゛-゛-(30)(26), 'C21''C22...""""1) is obtained from equations (26) and (27). Therefore, equation (30) can be converted to (2B), (29) and (31 ) is expressed as follows.

Vd -Ve 1- (2・Is -eXH-+ Is # exp
±yovote■ as vt
vtVd-Ve-VO + Is' GXD yt)Vd-Ve-Is+eXI) □ (2+
eXD-! '9+ eXp-"-]vt
vt vtVo
-V.

-'C21・(2+ QXD -VF-+ GXD v
t)■ 'C21'''C22''' 2+ eXI) (VO/
Vt) +eXll(-VO/Vt) “””(32
) Vd −Ve Vo
v.

'c23-IS"xpvl" expVt-I
c21 ・c×' t/l-I -exo(Vin/
Vt1 2yaeXD (VO/Vt) + 8XD (-VO/Vj
) ”””(33)■ ・exp(-Vin
/Vt) 'C24'' 2- eXI) (VO/Vtl + 6
XI) (-VO/Vt) """"'Here, + [Vin-0)-'c22 (Vin-01-I
c23(Vin-01C2+ Satin 1c, 4fVin-01-1ee/4 -=・
135) 1 (Vin-=oo) -1c24(V
in-2oa) -0-------(36)1(vln
a-Nimachi-Ic24 (Vinage-oa)-■ee...
...(37)1 (Vin −-O=)II −1
c24 (Vin-”oo) -0---(3F31 mutual conductance gI1 of each transistor 021 to Q24
11~Om4, QIN-gH2=”C21+JVin Ia.

■ = qx '-Vin/Vt-ex/Vin/song-,
-a , , , , , , . , C2-, 0XD (VQ/V
t) -, eXD(-'10/Vt))
V("°' To simplify the formula, Vin -Vin exp ■--A , ext) -V-E--8, 0m3 is gml 4Vin-01 from equations (39) to (41) -am2 (Vin-0)-0
・・・・・・(42) am3 (Vin-0)-io/
(4Vt) -------(4
31QmA (vin-ol-io/+4 vtl
・For...(4al where io is the DC component of the AGC output current.

From equations (32) to (44), the DC transfer characteristics are as shown in FIG. Then, the current Ir21. passing through the resistors R21 and R22. Ir22 is 'r21''c21''c22'
・・・・・・・・・(45)'r22"'c23"
'C24°-°-(4G), so the input signal Vi
n, and Ir21. The relationship of Ir22 is as shown in FIG.

Here, Ir21. shown in FIG. The relationship between Ir22 and Vin can be sufficiently approximated as shown below.

...(48) According to formulas C47) and C48), the relationship between the input signal yen of the circuit of the present invention and the AGC output voltage vou is as follows.
The result is as shown in FIG. Then, the following equation is obtained.

vout -R I, 22 1, 21) ・
-...(491 However, R1 is the resistance value of resistors v'cR11 and R12.

Therefore, the GC output voltage vout is obtained as a rectified signal as shown in FIG. AGC output voltage Vou finally obtained through a reduced pass filter
t is expressed as follows.

Here, we proceed with the examination of equation (50). input signal V
amplitude of in (in the conventional AGC detection circuit, the pack signal V
(in is limited in its amplitude to a relatively low level), the AGC detection circuit of the embodiment shown in FIG.
By applying some of the circuit design techniques described in connection with the conventional AGC detection circuit shown in the figure, an input with a relatively large width can be set to RV! It is possible to handle n. That is,
As in the case of a normal office amplifier, the current I of the current source 73
By taking measures such as changing ee, changing the load resistance R1, and connecting a resistor Re between the emitter of each transistor 021 to Q24 and the current source 13, the gain of the circuit can be increased.
This can be achieved by adjusting the dynamic range. Since the AGC detection circuit according to the embodiment is in the form of a differential amplifier circuit, the output thread width can be set large and it can operate at a relatively high input frequency. Furthermore, the AGC detection circuit according to this embodiment has a simple configuration, has a small number of elements, consumes low power, and can operate at low voltage, making it suitable for IC implementation.

Furthermore, the AGC output voltage v of the AGC detection circuit according to this embodiment
As can be seen from equation (50), out is proportional to the absolute temperature, so by applying a known band-capped current source as shown in FIG. 8 as the current source 13 to this circuit, The effects of temperature fluctuations are easily compensated for.

Regarding the characteristic fluctuation of the AGC output voltage vout due to temperature fluctuation, each transistor 021 to Q in the circuit of FIG.
24 mutual conductance C]m1~Qm4 is (39)
It is determined by the formula (41). As can be seen from equations (39) to (41), the current Iee of the current source 13 in the circuit of FIG.
When the input signal Vin and the input signal Vin are made proportional to the absolute temperature T, the characteristic fluctuations of the input signal Vin and the current ree of the current source 13 cancel each other out. Therefore, the mutual conductance of each transistor 021-Q24 is not affected by temperature fluctuations. Therefore, AGC output voltage vou
t is also no longer affected by temperature fluctuations.

An example of a current source having the characteristic that the current Iee is proportional to the absolute temperature [T] is shown in FIG. 8, as described above.

An example of the signal input circuit 11 that changes the input signal Vin in proportion to the absolute temperature will be described with reference to FIG. The circuit shown in FIG. 6 basically consists of a differential amplifier D4 and a current source 14.

In the differential amplifier D4, the emitters of the transistors Q31 and R32 are commonly connected to the current source 14. In addition, each transistor Q31. The collector of Q32 is resistor R3
1 and R32 with a voltage V4.
are connected to each other. Furthermore, their emitters are commonly connected to the current source 14. The input signal Vin is supplied to each transistor Q31. Q32, and the collectors of each transistor Q31 and R32 are connected to supply the aforementioned input signals Vin1(t) and Vin2(t) to the bases of transistors Q23 and R24 in the circuit of FIG. Ru.

Hey, these input signals Vin1(t) and 1n2(t)
have opposite phases and the same amplitude. For convenience of explanation, Vin(t) will be referred to as Vinl(t) in the following explanation.
and Vin2(t).

This 1n(t) is proportional to the absolute temperature T as described below.

Current source 14 is composed of transistors 033-038. Transistor 033 has its base and collector driven with voltage v5 f! 7I is connected to the power source E5, and its emitter is connected to the ground potential terminal G via the @current source 333 of the constant current I33. Transistor Q34 has its collector connected to drive power source E5, its base connected to the emitter of transistor 033, and its emitter connected to constant current I34.
are connected to the ground potential terminal G via current sources 334, respectively. The transistor Q35 has its collector connected to the driving power source E4, and its emitter connected to the ground potential terminal G via a current source S35 of a constant current I34. Transistor Q36 has its collector and base connected to drive power source E5, and its emitter connected to transistor Q3.
It is connected to the base of 5. The emitters of the transistors Q37 and 038 are connected in common and are connected to the ground potential terminal G via a current source Ea5 of a constant current Id5. That is, transistors Q37 and R38 constitute a differential amplifier D5. The collector of transistor Q35 is connected to the base of transistor Q35, and the base is connected to the emitter of transistor 034.

On the other hand, the collector of transistor Q38 is connected to differential amplifier D4.
It is connected to the emitters of transistors Q31 and R32 therein, and its base is connected to the emitter of transistor Q35.

Next, the operation of the signal input circuit shown in FIG. 6 will be explained. The output of the differential amplifier D4, ie, the input signal V 1n(t) to the AGC circuit shown in FIG. 1, is given by the following equation in terms of its amplitude.

Vinft)-Vin-gm-R1-1/in'(1,
4/Vt)・R31・−・−(511 where, 11
4 is a current applied from the current 14 to the differential amplifier D4.

Assuming that the transistors 033 to 038 have the same size, the reverse saturation currents 1s33 to 153B of these transistors 033 to Q38 are given by the following equations.

1 -1 -1 -1 Voice engineer Peng ■
・・・・・・(52) s33 s34
S35 s36 s37 s38
In this case, the emitter potentials \/e37 and \/e38 of transistors Q37 and 038 are the same, and the current source Sd
It can be expressed as a terminal potential Va5 of 5. The base-emitter voltage of each transistor 033 to Q38 is V
When be33 to be3B, the following equation holds true.

V5- Vd5-Vbe33 =-Vbe34 +Vb
e37-Vbe36 +Vbc35 +Vbe38 Here, Vd4 represents the emitter potentials Vbe5x and Vbe3z, which have the same value as each transistor Q31 and R32 in the office amplifier D4.

Each base-emitter voltage Vbe3a to Vbe3a is applied to each current source 333. Constant current I33°E34 of Ssi, S35 and Sa5. When expressed using I35 and L5, the following equation is obtained. From equation (52), the following equation is obtained.

'33°'34°"d5 '14""+4°'35"
'd5 '14''14"(I34・'33"'3
5 ......(53) Expression (53) is transformed into (5
1) Substituting into the equation yields the following equation.

If the current sources 333 and 834 are configured using bandgap constant current sources as shown in FIG. 8, the following equation is obtained.

Therefore, if a current source that is not affected by temperature coefficient factors is used as the constant current 335, the output of the signal input circuit shown in FIG. 6, that is, the input signal Vin(t ) becomes proportional to the thermal voltage Vt.

As described above, by connecting the signal Vin(t) output from the signal input circuit shown in FIG. 6, the input signal Vin is set to the absolute temperature l3
By making the current Ice of the current source 13 proportional to the absolute temperature T, the mutual conductance gm1 of the transistors 021 to Q24 of the AGC detection circuit is
~gm4 no longer has a temperature coefficient, allowing stable operation unaffected by temperature fluctuations.

According to an embodiment of the present invention, the AGC detection circuit shown in FIG. 1 is composed of only four transistors.

In order to satisfy the characteristics necessary for stability against temperature fluctuations, it is sufficient that the collector currents 1021 to 024 flowing through each of the transistors 021 to Q24 are the same when the input signal is zero.

In other words, holds true. Here, 0 represents each collector current IC21 to IO24 having the same value. Therefore, the following formula holds true.

'sl+'s2''``'s3-'' ■s4 As can be seen from the above equation, the reverse saturation currents l323 and l524 of the two transistors Q23 and Q24 in the differential amplifier connected to the signal input circuit 11 are the same. , and two transistors Q21 and Q in the differential amplifier or multibase transistor 90m1 connected to the reference voltage source E2.
The sum of the reverse saturation currents l521 and l522 of 22 is the reverse saturation current l of the transistor Q23 or Q24.
523 or twice l524 is sufficient. The reverse saturation current Is of a transistor depends on the emitter area of the transistor. Therefore, other characteristics of the transistor are not particularly limited as long as the above relationship is satisfied between the reverse saturation currents Is21 to I524.

Roughly speaking, the AGC output voltage Vo of the AGC detection circuit in FIG.
ut is shown as being taken out to output terminals 12A and 12B as two outputs having a differential relationship with each other,
Single output may be used. Furthermore, although the AGC detection output is shown to be taken out as a voltage output, it can also be taken out as a current output, as the characteristics of the circuit shown in FIG. 1 have been analyzed from the current relationship.

The invention is not limited to the embodiments described above. In the circuit shown in Figure 1, by changing the voltage of the constant voltage source or the thread width setting of the input signal, different types of AG
C output can be detected.

To simplify the explanation, we will now explain the circuit connection, resistance R11.
, R12, the current ■ee of the constant current source 13, the voltage V3 of the constant voltage source E3, and the input signal Vin of the signal input source 11 are assumed to be unchanged.

The potential Vin-a of the signal input source 11 is expressed as Vin-a −Vd+Vin −Vd+Vo −sin as shown in equation (23).
ωt ・−・−(23) On the other hand, it is assumed that the basic potential V2 of the reference voltage source E2 is determined as follows.

V2- Vd + Vx -=-(5
6) That is, the bias voltages for the multi-base transistors Qm1 and Q are made to differ from each other by the voltage Vx. The voltage Vx is white as the base bias shift voltage.

To simplify the explanation, Vx=4Vt (Vt:
thermal voltage), then I of equations (47) and (48)
) From the C transfer characteristics, the input/output relationship in this embodiment is as shown in FIG. However, the horizontal axis in the figure indicates the base potential difference between each multi-base transistor QIT+1 and 01712.

As shown in FIG. 5, the AGC output voltage vout is obtained by half-wave rectifying the input signal Vin.

The AGC detection output "vout" finally obtained by passing this half-wave rectified output through a low-pass filter is as follows.

-R1・Ice・(1-ai-vt)・
(57) When looking at the change in the output with respect to the change in the input signal from equation (57), the term R1·Ice on the right side is constant and can be ignored. In this state, input Vin-
If it is b, it will be as follows.

Comparing Equation (58) and Equation (50), they are completely consistent except that 2π in the term on the right side of Equation (50) is replaced with 4π in Equation (58). Therefore, also in this example,
The same characteristics and effects as the AGC detector described above can be obtained.

In addition, in the above explanation, the case where the base bias shift voltage is Vx = 4Vt was explained, but in case of other base bias shift voltage Vx, the basic There is no change in operation.

;Effects of the Invention] As explained above, the present invention has a wider output range and signal input conditions than conventional ones, has high accuracy, and can easily compensate for output fluctuations due to temperature fluctuations. It is possible to provide a detection circuit for automatic gain control that is simple in circuit configuration, consumes little power, is capable of low voltage operation, and is suitable for IC implementation.

[Brief explanation of drawings]

Figure 1 is a circuit diagram showing an embodiment of the present invention, Figure 2 is an input voltage vs. collector current characteristic diagram of the circuit in Figure 1, and Figure 3 is an input voltage vs. output current characteristic diagram of the circuit in Figure 1. , @4 is a DC transfer characteristic diagram of the circuit in FIG. 1, FIG. 5 is a DC transfer characteristic diagram in another embodiment of the present invention, FIG. 6 is a circuit showing an example of a current source circuit, and FIG. FIG. 8 is a circuit diagram showing a conventional detection circuit for automatic gain control, and FIG. 8 is a circuit diagram showing a differential amplifier circuit for temperature compensation. Q21-Q24...Transistor, R11, R
12...Resistor, 11... Constant current source. Agent Patent Attorney Ken Chika Hiroshi Uji Figure 3 Figure 4

Claims (1)

[Claims] first to fourth transistors having substantially the same characteristics and the same polarity, whose emitters are commonly connected to a current source; a common connection point between the collectors of the first and second transistors; and a drive power source. a first collector load device connected between the third and fourth transistors, and a second collector load device connected between the drive power source and a common connection point between the collectors of the third and fourth transistors.
a collector load device; a reference voltage source having the bases of the first and second transistors connected in common; and an input signal processing circuit having the bases of the third and fourth transistors connected to both ends thereof. , the input signal processing circuit transmits an input signal proportional to absolute temperature to the third and fourth
The current source applies a current proportional to the absolute temperature to each emitter of the first to fourth transistors, and output terminals are respectively drawn out from a common connection point of the respective collectors. A detection circuit for automatic gain control characterized by the following.