JPS6042709B2 - induction motor control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to an induction motor control device, and more particularly to an induction motor control device suitable for performing voltage-based torque control of an induction motor.

Conventionally, when controlling an induction motor, the value of the secondary flux linkage, which is the sum of the effective magnetic flux and the secondary leakage flux, is given as a reference value, and the reference value of the rotational speed or torque is also given, and the operation is performed based on this. The amount of data that should be calculated was calculated using vectors, and the amount of data was controlled based on this result.

According to such a conventional method, since control is performed to match the secondary magnetic flux linkage to the command value, the effective magnetic flux is not constant but changes.

However, if the effective magnetic flux is not set, it is not possible to make the torque proportional to the effective current, and it is difficult to perform constant output control. Furthermore, in order to perform constant output control, it is necessary to perform field weakening control, but this is achieved by manipulating the effective magnetism and flux, so there is no point in manipulating the secondary flux linkage. do not have. As mentioned above, when controlling the induction motor by commanding the secondary flux linkage, more current than necessary may flow to satisfy the torque command. There was a strong demand for a device that could control induction motors with limited current.

SUMMARY OF THE INVENTION Accordingly, an object of the present invention is to provide an induction motor control device that enables effective torque control of an induction motor based on an effective magnetic flux command. Another object of the present invention is to provide an inductor configured to calculate the non-sinusoidal current rate and adjust the voltage reference in order to reduce the generated torque ripple even when a converter with a non-sinusoidal voltage output is used. To provide electric motor control equipment.

Hereinafter, the induction motor control device of the present invention will be explained in more detail with reference to the drawings.

FIG. 1 is a system configuration diagram of an induction motor control device according to an embodiment of the present invention, particularly illustrating a case where a DC square wave voltage source inverter is the main component.

In the figure, 62 is an AC-to-DC converter, 63 is a DC-to-AC inverter, 64 is a three-phase induction motor, 65 is a tachometer generator pivoted to the three-phase induction motor 64, and 66 is the AC-to-DC converter. 62 is a voltage detection circuit for the output of the primary voltage, 67 is a converter phase control circuit that controls the AC-DC converter 62, 68 is a primary voltage control amplifier, and 69 is an inverter phase that controls the DC-AC inverter 63. Control circuit, 70 is an adjustment resistor for setting effective magnetic flux, 71 is an adjustment resistor for setting rotation speed, 72 is an amplifier for speed control, 73 is effective magnetic flux, torque/secondary flux linkage,
74 is an arithmetic circuit for suberi frequency, secondary magnetic flux linkage, suberi frequency/primary voltage, motor primary frequency arithmetic circuit, 75
teeth. An amplifier for controlling the primary frequency of the motor, 76 a voltage/frequency conversion circuit, 77 a 2-phase/3-phase conversion circuit, 78 a frequency detection circuit, 79 a speed detection circuit, 81 the output of the DC-AC inverter 63, i.e. A voltage detection circuit for the motor primary voltage, 82 is a voltage non-sinusoidal wave rate. Each shows an arithmetic circuit. In addition, FIG. 2 shows the arithmetic circuit 73 shown in FIG.
This is a circuit configuration diagram showing a detailed configuration of , and shows a configuration for calculating the secondary flux linkage value φ2 and the sub-frequency SWe based on the effective magnetic flux value φ and the torque command value T.

In the figure, 8 is a bias for φ2 selection, 9 and 10 are diodes for φ2 selection, 11 is a square root function generator, 12, 14,
19 is a square function generator, 13 is a 1+T2P calculation circuit, 1
Reference numeral 5 designates a T2 calculation circuit, 16 and 20 multipliers, 17 an inverse proportional function generator, 18 a torque reference input, 21 an R2 calculation circuit, and 22 a polarity conversion circuit.
FIG. 3 is a circuit configuration diagram showing the detailed configuration of the arithmetic circuit 74 shown in FIG. This shows a configuration for calculating frequency We.

In the same figure, 31 is the L1/M calculation circuit, 32 is 1 + loan/T4
・P calculation circuit, 33 and 36 are P calculation circuits, 34 is T4
Arithmetic circuit, 35, 37, 38, 43 are multipliers, 39 is 1
/M calculation circuit, 40 is 1+T3P calculation circuit, 41, 44
is r1 arithmetic circuit, 42 is T3/M arithmetic circuit, 45, 46
4 represents a square function generator, and 47 represents a square root function generator. FIG. 4 is a circuit configuration diagram showing the detailed configuration of the voltage non-sinusoidal wave factor calculation circuit 82 shown in FIG. This constitutes a circuit that calculates the difference between .

In the same figure, 101, 102, 105, 106, 131, 1
11, 112, 115, 116, 121, 122, 12
5, 126 is a resistor, 103, 132, 113, 123
is an operational amplifier, 104, 108, 114, 118, 12
4, 128 are multipliers, 107, 117, 127 are filters, and 109, 119, 129 are average output zero control amplifiers.

The operation of the configuration as described above will be explained in detail below.

Note that the correspondence relationship with the induction motor 64 for various quantities used in the calculation formula used in the explanation is shown in the explanatory diagram of FIG. 5. In Fig. 5, δ1 is the primary voltage vector,
is the secondary voltage vector, i1 is the primary current vector, I2 is the secondary current vector, L1 is the primary inductance, M is the mutual inductance, ! is the secondary inductance, 11 is 1
Leakage inductance: 1. is the secondary leakage inductance, r is the primary resistance, R2 is the secondary resistance, φ1 is the primary flux linkage vector, φ is the effective flux vector, φ2 is the secondary flux linkage vector, We is the motor primary frequency, S has a lot, ω
r is the rotational angular velocity of the rotor, P is a differential operator, θe is the primary rotation angle of the motor, and 0r is the rotation angle. Now, the basic equation of an induction motor is expressed as follows: the first order is expressed in fixed coordinates, and the second order is expressed in rotating coordinates that rotate at the same speed as the rotational speed of the rotor.

In order to unify the coordinates to coordinates that rotate at the same speed as the rotating magnetic flux due to the primary voltage, a transformation spinner (Spi
When both sides are multiplied by nOr) and O is attached to the variable represented by the rotating magnetic flux coordinate, when this is decomposed into linear and quadratic, the following is obtained.

Here, if becomes.

If the three-phase induction motor 64 is a squirrel cage motor, e1=r1
・i1+pφ1+jωeφ1 ・(13) According to the definition of φ2.

Here, if the coordinates are fixed to the φ2 axis and decomposed into the D and q axes, then the following can be calculated.

On the other hand, the torque is.

Also, the effective magnetic flux is as follows.

To control with torque and effective flux instructions, other variables must be determined by T and φ.

Here, from equation (22), it becomes.

Now, in equation (23), if we use an amplifier with a response speed that does not cause problems as a system, as shown in Figure 2, then T, φ
(=lφ1), the smaller φ2 can be determined.

To explain the above according to the vector diagram in Figure 6,
The figure shows the effective magnetic flux φ of the induction motor 64 and the secondary interlinkage magnetic flux φ2.
This shows that two types of values of the interlinkage magnetic flux φ2 can be taken for a certain torque setting value depending on the phase relationship.

That is, the effective magnetic flux vector φ and the secondary interlinkage magnetic flux vector φ2
The difference vector (L2-M) I2 must be parallel to the secondary current vector φ2, and the secondary current vector φ
2 lags the effective magnetic flux vector φ by more than 90 volts, so
The smaller value of the secondary flux linkage vector φ2 is the value to be sought.

Therefore, the circuit configuration shown in FIG. 2 is such that when determining the secondary magnetic flux linkage φ2, the smaller value is determined.

Further, from equation (21), it becomes as follows. Therefore, SO)e can be determined based on the instructions of T and φ with a normal arithmetic circuit configuration as shown in FIG.

By the way, Iell (=
e), ωe, it is necessary to find these from φ2, s0)e.

Using the motor speed ωr, WV view W ~ Ren W &
It becomes \yυノ.

If calculated based on φ2 axis, (a), (16) to (19)
From the formula.

However, T4? It is. Also, from formulas (16) and (17), Mr2 becomes, and from formula (13), ゛-''
becomes.

This value can be determined through the circuit configuration shown in FIG. Note that in order to satisfy the instructions of T and φ, it is necessary and sufficient to control El and ωe, which can be proven as described below.

First, Sωe is determined from ωe according to equation (25).

Also, using equations (26) and (28), ωE, SωE,
φ and φ are determined depending on el. Using equations (18) and (19), i based on the φ2 axis is determined based on SO)e and φ2. Further, using equation (21), T is determined depending on SO)e and φ2. Then, using equation (22), S
Based on ωe and φ2, φ based on the φ2 axis is determined. Second
The configurations shown in the drawings and FIG. 3 constitute a circuit for executing each of the above operations, and the operation thereof will be explained in detail below.

First, the configuration shown in FIG. 2 calculates φ2 and SO) based on the effective magnetic flux φ given by the adjustment resistor 70 and the torque T which is the torque reference input 18 based on equations (23) and (24).
This is to find e.

Here, the operational amplifier 4 constitutes a φ2 control circuit, and includes a square root function generator 11, square function generators 12, 14,
The 1+T2P calculation circuit 13, the T2 calculation circuit 15, the multiplier 16, and the inverse proportional function generator 17 execute the calculation of equation (23), calculate the size φ from φ2 and T, and further convert the polarity conversion circuit. 22 performs polarity conversion and feeds back to the operational amplifier 4 as a feedback signal with respect to the reference φ.

The above configuration constitutes a φ control loop, but since two values are obtained for φ2, which is the output of the operational amplifier 4, in order to selectively output only the smaller value. , adjustment resistor 8, and diodes 9 and 10 provide bias.

Therefore, in the end, the required φ2 output can be obtained. On the other hand, the inverse proportional function generator 17, the square function generator 19, the multiplier 20, and the R2 arithmetic circuit 21 are (24)
Since the calculation of the formula is executed, it constitutes a system that calculates SO)e from φ2,T.

Also, the configuration in Figure 3 is (25), (28), (29)
This is to calculate El and ωe from the inputs of φ2, s6)E and ωr by executing the calculations of the equations.

Here, L1/M calculation circuit 31, 1+? Y4P calculation circuit 32 and P calculation circuit 33 calculate the third term of the real part of equation (28), and T4 calculation circuit 3 multipliers 35 and 38 calculate (28)
Calculates the second term of the real part of the equation, and also a 1/M calculation circuit 39,
1+T3P calculation circuit 40 and r1 calculation circuit 41 are (28)
The first term of the real part of the equation is calculated. The sum of the terms calculated by each of the circuit groups described above is input to the square function generator 46. On the other hand, the T4 arithmetic circuit 3 multiplier 35 and the P arithmetic circuit 36 calculate the third term of the imaginary part of the equation (28), and the T4 arithmetic circuit 34, the multiplier 35, and the multiplier 37 calculate the imaginary part of the equation (28). The second term is calculated, and the T3/M calculation circuit 42, multiplier 43, and r1 calculation circuit 44 calculate the first term of the imaginary part of equation (28). The sum of each term calculated in each of the L circuit groups described above is input to the square function generator 45. The sum of the outputs of the square function generators 45 and 46 is inputted to a square root function generator 47, and the output of the function generator 47 has a magnitude e1 of e1.

On the other hand, each input of Sωr and ωr is summed according to equation (25), and is output as ωe. Next, the configuration of FIG. 4, which is the main circuit of the present invention, will be explained.

As mentioned above, FIG. 4 is a circuit that calculates the difference in waveform between a sine wave voltage reference and an actual voltage when an induction motor is driven with a voltage that is not a sine wave. In the configuration shown in FIG. 4, ElR input to the operational amplifier 103 through the resistor 101 is a primary R-phase sine wave voltage reference.

On the other hand, the actual waveform signal e''1R of the primary R phase is input to the negative side of the operational amplifier 103 through the resistor 106, and as a result, the difference signal between ElR and e''1R is amplified by the resistor 102 and the operational amplifier 103. It is amplified by the circuit. In this case, average output zero control is performed in order to obtain only the difference as the output of the operational amplifier 103. Specifically, the output of the operational amplifier 103 is passed through the filter 107 to the amplifier 10.
Enter 9. Then, by multiplying the output signal of the amplifier 109 and the e''1R signal in a multiplier 108 and applying this to the negative side of the operational amplifier 103 via the resistor 106, the output of the operational amplifier 103 is :d.e″
Only the difference is 1R. The difference signal obtained as described above is multiplied by the ElR signal in the multiplier 104, a weighted average is taken, and the resultant signal is applied to the operational amplifier 132 through the resistor 105. On the other hand, regarding the S phase, 11
Exactly the same calculations are performed through the circuit groups indicated by 1 to 119, and the results are applied to the operational amplifier 132.

Further, the T-phase is also subjected to exactly the same calculation through the circuit group shown by 121 to 129, and is provided to the operational amplifier 132.

In the operational amplifier 132, the weighted average value of the differences between the R, S, and T phases obtained as described above is summed, and the amplification factor determined by the resistors 105, 115, 125, and 131 is calculated. This is amplified according to , and an output signal e″1 is obtained.

The output signal e''1 is used as a signal for voltage reference adjustment. The induction motor control device shown in FIG. 1 was used as the frequency calculation circuit 73, secondary flux linkage, slip frequency/primary voltage, motor primary frequency, calculation circuit 74, and voltage non-sinusoidal wave rate calculation circuit 82. As mentioned above, this is an example of the case where a DC type square wave voltage type inverter is used.

In the configuration of FIG. 1, an adjustment resistor 70 provides an effective magnetic flux reference φ, and an adjustment resistor 71 provides a rotational speed reference N.

The speed control amplifier 72 calculates the difference between the rotation speed reference N and the speed signal detected by the speed detection circuit 79 based on the output signal of the tachometer generator 65, and outputs this as a torque signal T. . The effective magnetic flux reference φ from the adjustment resistor 70 and the output T of the speed control amplifier 72 are input to the arithmetic circuit 73, but here, as mentioned earlier, φ
Also, φ2 and Sωe are calculated from T and output. The outputs φ2 and Sωe of the arithmetic circuit 73 are input to the arithmetic circuit 74.

In the arithmetic circuit 74, based on the outputs φ2, s4)e of the arithmetic circuit 73 and the output ωr of the speed detection circuit 79, the arithmetic operations as described above are performed to calculate e1 and ωe. On the other hand, the input voltage of the three-phase induction motor 64 is detected by the voltage detection circuit 81, and the voltage non-sinusoidal wave factor calculation circuit 82
In this case, the difference e″1 from the sine wave voltage reference is calculated by performing the calculation as described above.The calculation circuit 7
The output e1 of the AC-DC converter 62 is corrected by the output e''1 signal of the voltage non-sine wave factor calculation circuit 82, and this signal is used as the output voltage reference of the AC-DC converter 62. The output DC circuit voltage of
The voltage detection circuit 66 detects this and uses it as a voltage feedback signal. The primary voltage control amplifier 68 compares the voltage feedback signal with the output voltage reference, and inputs the difference signal to the converter phase control circuit 67. The phase control circuit 67 controls the firing pulse of each component of the AC-DC converter 62 based on the difference signal, and controls the output of the AC-DC converter 62 to a required voltage. On the other hand, the difference between the output signal ωe of the arithmetic circuit 74 and the frequency signal output from the frequency detection circuit 78 is input to the frequency control amplifier 75 and amplified.

The output of the frequency control amplifier 75 is given to a voltage/frequency conversion circuit 76 and converted into an oscillation signal of two-phase outputs D and q with a phase difference of 900. The two-phase/three-phase converter 77
The phase outputs D and q are converted into three-phase oscillation signals, which are applied to the phase control circuit 69 of the DC-AC inverter 63. As a result, the firing pulse signals of each component of the DC-AC inverter 63 are controlled, and the three-phase induction motor 64 is driven by the output thereof. Through the control circuit group as described above, the commercial power source is converted into DC with the required voltage by the AC-DC converter 62, smoothed by the capacitor C, and converted into AC with the required frequency by the DC-DC inverter 63. A three-phase induction motor 64 is provided.

As a result, driving torque is generated in the three-phase induction motor, and rotational force is generated, but the rotational speed at this time is
It will be detected and returned at 5. With the above-described configuration, when controlling the torque of the three-phase induction motor 64, it is possible to perform efficient control with the minimum necessary current and with less torque ripple.

In the above embodiment, an induction motor control circuit using a DC type wave voltage type inverter is illustrated, but the implementation of the present invention is not limited to this, and a cycloconverter or a PWM converter may be used as the converter. Of course, a (pulse width modulation) type converter may also be used.

Furthermore, in the case of square wave current control, the fundamental wave current acts as an effective component, but it is also possible to increase the ratio of effective current by using a converter with a sine wave output.

Further, in the above embodiment, the closed loop includes a speed control loop, a primary voltage control loop, and a primary motor frequency control loop, but it is possible to increase or decrease the number of closed loops as necessary.

As described above, according to the present invention, when performing vector control of an induction motor, since conventionally secondary flux linkage control was used, the effective magnetic flux is not necessarily constant, and the torque is not necessarily equal to the effective current. In contrast, with effective magnetic flux control, the torque is proportional to the effective current, so it is now possible to control the current with the minimum necessary current, and it is possible to control the current accurately. It is possible to obtain a new induction motor control device that can perform output control, and it meets the requirement of controlling a large-capacity system using small-capacity equipment, which is a necessary condition for automatically controlling rotating machines. It is possible to satisfy the following.

[Brief explanation of drawings]

FIG. 1 is a system configuration diagram of an induction motor control device according to an embodiment of the present invention, and FIGS. 2, 3, and 4 are partial circuit configuration diagrams showing the detailed configuration of the system in FIG. FIG. 5 is an explanatory diagram showing the correspondence between calculation formula quantities and induction motors, and FIG. 6 is a vector diagram explaining the operation of the circuit shown in FIG. 2. 62...AC-DC converter, 63...DC-DC inverter, 64...3-phase induction motor, 66...
Voltage detection circuit, 67...Converter phase control circuit, 6
8... Primary voltage control amplifier, 69... Inverter phase control circuit, 70, 71... Adjustment resistor, 72...
・Speed control amplifier, 73, 74... Arithmetic circuit, 75
...Amplifier for motor primary frequency control, 76...Voltage/
Frequency conversion circuit, 82... Voltage non-sinusoidal wave rate calculation circuit.

Claims (1)

[Claims] 1. A converter for supplying alternating current power of variable voltage and variable frequency to an induction motor, a setting means for respectively setting effective magnetic flux and rotational speed standards of the induction motor, and setting means for setting the primary voltage, rotational speed and rotational angular velocity of the induction motor, respectively. a detection means for detecting; an effective magnetic flux vector is a product of a vector sum of a primary current vector and a secondary current vector; and a primary and secondary mutual inductance; Since the primary current and secondary current can be separated into directional components, the primary current and secondary current are separated into a component in the direction of the secondary interlinkage flux and a component in a direction perpendicular to it, and the set effective magnetic flux standard and the set rotational speed standard and detection are a first calculation circuit that calculates a secondary magnetic flux linkage from a torque reference given as a difference in rotational speeds, and calculates a slip frequency based on the secondary magnetic flux linkage and the torque reference; Calculate the primary magnetic flux linkage from the magnetic flux linkage and the slip frequency, and calculate the primary voltage from the sum of the primary resistance drop, a component proportional to the rate of change in the primary magnetic flux linkage, and a component proportional to the primary linkage reference. and a second calculation circuit that calculates a primary frequency from the sum of the calculated slip frequency and the detected rotational angular velocity, and a sine wave oscillation circuit that converts the calculated primary frequency into a voltage/frequency to create a sine wave voltage reference. , comparing the detected primary voltage of the induction motor and the sinusoidal voltage reference;
and a third calculation circuit that calculates a non-sinusoidal wave factor by adding the average value of the difference of each phase, and the primary voltage calculated by the second calculation circuit is applied to the third calculation circuit. An induction motor control device characterized in that the magnitude of the voltage of the converter is controlled by a signal corrected by a non-sinusoidal wave factor of an arithmetic circuit, and the frequency of the converter is controlled by the sine wave voltage reference.