JPH09215398A - Inverter controller - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce the influence of a harmonic wave in an overmodulation drive region at the time of the feedback current control. SOLUTION: In the controller of a current control inverter 2 by which the primary current of an induction motor is divided into an excitation current and a torque current and controlled, if the output of the inverter 2 is saturated, an excitation current iγs' and a torque current iδs' which are obtained by removing harmonic components corresponding to a primary frequency ω from an excitation current iγs and a torque current iδs which are in accordance with the output of a current detection unit 25 are calculated by a filter arithmetic unit 22. In accordance with the calculated excitation current iγs* and torque current iδs' and an excitation current instruction value iγs* and a torque current instruction value iδs* which are calculated by a vector control instruction value arithmetic unit 11, an excitation voltage instruction value Vγs* and a torque voltage instruction value Vδs* are calculated by a current control unit 21 and the inverter 2 is controlled in accordance with the excitation voltage instruction value Vγs* and the torque voltage instruction value Vδs*.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an inverter control device, and more particularly to a current control type inverter control device.

[0002]

2. Description of the Related Art In recent years, many AC motors such as induction motors and synchronous motors have been used instead of DC motors. An inverter is mainly used to drive an AC motor, but the output voltage of the inverter is limited by the applied input DC voltage. In order to output a larger voltage, it is necessary to increase the input DC voltage, and accordingly the capacity of the inverter must be increased. Further, for example, in an electric vehicle, the power source of the inverter that drives the AC electric motor is a battery, and the output voltage of the inverter decreases as the battery capacity decreases. Therefore, the effective use of the output voltage is the performance of the vehicle, especially one charging range. Will be heavily involved in.

As a conventional inverter control device for effectively utilizing the output voltage of the inverter, for example, a device having a configuration shown in the block diagram of FIG. 14 or JP-A-60-12197 is used.
There is a device described in Publication No. 9. In FIG.
Here, the secondary magnetic flux direction of the induction motor 1 to which the primary voltage is supplied from the inverter 2 is the γ axis, and the direction orthogonal to it is δ axis.
Consider a coordinate system that rotates in synchronism with an axial primary voltage. The subscript of each symbol represents the γ-axis or δ-axis component. This conventional inverter control device includes a current control unit 12 that performs feedforward current control on the excitation current command value iγs ^* and the torque current command value iδs ^* calculated by the vector control command calculation unit 11, Excitation voltage command value vγs ^* and torque voltage command value vδs which are the outputs
^{From *} and, the modulation degree calculator 15 calculates the modulation degree μ. The feedforward current control unit 12, the excitation current command value Aiganmaesu ^*, the excitation voltage command value so as to realize the torque current command value iδs ^* vγs ^*, a torque voltage command value Buiderutaesu ^* intended to calculation using the motor constants is there. The correction coefficient calculator 16 calculates a correction coefficient h corresponding to the modulation degree μ. The correction coefficient h is a coefficient for correcting the output decrease due to the saturation of the output voltage of the inverter 2 so that the fundamental wave amplitude of the output voltage of the inverter 2 becomes equal, and h = 1 in the range of the modulation degree μ ≦ 1. , H> 1 in the range of μ> 1. According to this control device, as shown in the figure, when controlling a drive type inverter using a PWM generator 14 with a high voltage utilization ratio, the output voltage of the inverter is about 10% greater than when it is not controlled. The fundamental wave amplitude can be obtained. For example, when controlling the inverter of the drive system in which each phase voltage to the induction motor is a sine wave, the output voltage has a large fundamental wave amplitude of about 27%.

[0004]

In the above conventional control device, current control is performed by feedforward. The feedforward current control calculates a voltage command value from a current command value using various constants of the motor when driving the motor with an inverter. However, it is quite difficult to identify those motor constants accurately, and since the motor constants change with temperature, etc., the motor constants used in feedforward current control include some error with respect to the actual values. Conceivable.

For example, when the feedforward current control is performed by using the primary resistance, the secondary resistance, the primary inductance, the secondary inductance and the mutual inductance of the induction motor which are 10% larger than the actual values, The exciting current command value iγs ^* and the actually flowing exciting current iγs are shown in FIG. 15A, and the torque current command value i
δs ^* and the actually flowing torque current iδs are shown in FIG. In each figure, the exciting current iγs
Also, the torque current iδs vibrates in a transient state and deviates from the command value in a steady state.

Feedback current control may be performed in order to correct the influence of such an error in the motor constant. For example, in the feedforward current control, the exciting current command value iγs ^* and the actual exciting current iγ when the proportional-plus-integral control is performed as the feedback current control
FIG. 15B shows s, and FIG. 16B shows the torque current command value iδs ^* and the actual torque current iδs. As can be seen from the figure, by performing the feedback current control, the transient and steady-state current control can be performed well.

However, when the output voltage of the inverter is saturated when the feedback current control is performed, the fundamental wave of the inverter output voltage matches the command voltage due to the correction coefficient h, but harmonics increase. As a result, the harmonics are also superposed on the current flowing through the electric motor, and the feedback current control does not function sufficiently. FIG. 17 shows an example of voltage and current waveforms when the output voltage is saturated. In FIG. 17, the input DC voltage Vdc of the inverter decreases,
We are considering the case of voltage saturation. For example, in an electric vehicle that uses a battery as a DC voltage source, discharge causes a decrease in battery voltage as shown in FIG. As the battery voltage drops, FIG.
As shown in (h), the modulation degree μ exceeds 1 and enters the overmodulation drive region (voltage saturation region). In the overmodulation drive area,
7 (b) to indicate the current command value iγs ^*, with respect to the torque current command value Aiderutaesu ^* shown in FIG. 17 (e), the actual excitation current Aiganmaesu shown in FIG. 17 (c), the actual torque shown in FIG. 17 (f) The harmonic current increases in the current iδs. Moreover, in order to feed back the current in which these harmonics are superposed,
It can be seen that the harmonic components are also superposed on the excitation voltage command value vγs ^* shown in (d) and the torque voltage command value vδs ^* shown in FIG. 17 (g), and it is apparent that the current control is disturbed.

As described above, the conventional feedforward current control alone has a problem that the current control in the transient and steady states does not sufficiently function due to an error in the motor constant. Further, when the feedback current control is performed, in the region where the output voltage of the inverter is not saturated, it is possible to perform the current control without the influence of the error of the motor constant, but
In the region where the output voltage of the inverter is saturated, current control is disturbed because the current with harmonics superimposed is fed back.
As a result, the output torque of the electric motor is disturbed and the efficiency is deteriorated.

The present invention has been made in view of the above problems, and an object of the present invention is to provide a control device for a current control type inverter in which the influence of harmonics in the overmodulation drive region during feedback current control is reduced. .

[0010]

Therefore, according to the first aspect of the present invention, the control device for the current control type inverter for controlling the primary current of the AC motor by dividing the primary current into the exciting current and the torque current. In, a vector control command value calculation for calculating an exciting current command value, a torque current command value, and a primary frequency for realizing a preset drive target value of the AC motor according to the rotation state of the AC motor and the drive target value Means, a motor current detection means for detecting a primary current of the AC motor and calculating the exciting current and the torque current based on the primary current, and a frequency component corresponding to the primary frequency from the exciting current and the torque current And a magnetizing voltage based on the calculation output of the filter calculating means, the exciting current command value, and the torque current command value. Current control means for calculating a command value and a torque voltage command value, and two-phase / three-phase conversion means for calculating a three-phase voltage command value for controlling the output of the inverter from the excitation voltage command value and the torque voltage command value. And when the output of the inverter is saturated, the filter calculation means executes the calculation.

According to this structure, the excitation voltage command value and the torque current command value calculated by the vector control command value calculation means are such that the current control means realizes these command values. And the torque voltage command value are calculated, the exciting current and the torque current actually flowing in the AC motor detected by the motor current detection means are input to the current control means via the filter calculation means and feedback control is performed. . When the output of the inverter is saturated, the filter calculation means removes the frequency component corresponding to the primary frequency from the exciting current and the torque current. The excitation voltage command value and the torque voltage command value output from the current control means are
The 2-phase / 3-phase conversion means converts the voltage into a 3-phase voltage, and the inverter is controlled according to the 3-phase voltage.

According to a second aspect of the present invention, in the first aspect of the invention, the filter calculation means uses the excitation current and the torque current to obtain the primary frequency of 6
It is characterized in that it is configured to remove frequency components that are natural multiples of each. According to such a configuration, in the filter calculating means, when the output of the inverter is saturated, the frequency components of natural number times 6 of the primary frequency are removed from the exciting current and the torque current.

According to a third aspect of the present invention, in the first aspect of the invention, the filter calculation means uses the excitation current and the torque current to obtain the primary frequency of 6
It is characterized in that only double frequency components are removed. According to such a configuration, in the filter calculation means, when the output of the inverter is saturated, only the frequency component of 6 times the primary frequency is removed from the exciting current and the torque current.

According to a fourth aspect of the present invention, in the first aspect of the invention, the filter calculation means uses the excitation current and the torque current to obtain the primary frequency of 6
The feature is that all frequency components more than double are removed. According to such a configuration, in the filter calculation means, when the output of the inverter is saturated, all the frequency components of 6 times or more the primary frequency are removed from the exciting current and the torque current.

According to a fifth aspect of the invention, in the invention according to any one of the first to fourth aspects, the filter computing means executes the computation in synchronization with the rotation of the AC electric motor. It is characterized by According to such a configuration, the calculation of the filter calculation means is performed in synchronization with the rotation of the AC electric motor, and the control is performed according to the change of the rotation of the AC electric motor.

In a sixth aspect of the present invention, in a controller for a current control type inverter that controls a primary current of an AC motor by dividing it into an exciting current and a torque current, a preset drive target of the AC motor is set. An exciting current command value, a torque current command value, and a primary frequency that realize the value are calculated according to the rotation state of the AC motor and the drive target value,
A frequency component corresponding to the primary frequency is removed from the excitation current and torque current detected when the output of the inverter is saturated, and the excitation current and torque current and the excitation current command value and the frequency component corresponding to the primary frequency are removed. An exciting voltage command value and a torque voltage command value are calculated based on the torque current command value, and the inverter is controlled according to the exciting voltage command value and the torque voltage command value.

[0017]

As a result, according to the invention described in claim 1 or 2 of the present invention, the frequency component corresponding to the primary frequency from the exciting current and the torque current fed back in the filter calculating means, the invention according to claim 2 is provided. In the described invention, by removing the frequency component which is a natural multiple of 6 of the primary frequency, it is possible to perform feedback current control with a small influence of the harmonic component, so that the output voltage of the inverter is stable and the output of the AC motor is stable. It is possible to stabilize the torque and improve the efficiency.

In addition to the above effects, the invention according to claim 3 has a simple structure by using a filter calculating means as a filter for removing frequency components of 6 times the primary frequency. Therefore, the calculation load of the filter calculation can be reduced. In addition to the effect of the invention described in claim 1 or 2, the invention described in claim 4
By using a filter that removes frequencies that are 6 times or more of the primary frequency as the filter calculation means, the filter calculation means can have a simpler configuration and the calculation load of the filter calculation can be further reduced.

The invention described in claim 5 is the same as claim 1.
In addition to the effect of the invention described in any one of 1 to 4, by performing the calculation of the filter calculation means in synchronization with the rotation of the AC motor, regardless of the change in the rotation speed of the AC motor,
The same filter operation can always be performed.

[0020]

BEST MODE FOR CARRYING OUT THE INVENTION Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a block diagram showing the configuration of the first embodiment. However, the same parts as those in the configuration of the conventional device described above are designated by the same reference numerals. In FIG.
The control device for the inverter according to the first embodiment is based on the output of the motor rotation speed detection unit 4 that detects the rotation speed of the induction motor 1 as an AC motor, the motor rotation command value N ^*, and the magnetic flux command value Φγr ^*. Excitation current command value iγs ^* , torque current command value iδs ^* , primary frequency ω and power supply phase angle θ
Using the vector control command value calculation unit 11 as the vector control command value calculation unit and the power supply phase angle θ calculated by the vector control command value calculation unit 11, the current detection unit 25 as the motor current detection unit. The detected input three-phase current to the induction motor 1 is used as the exciting current iγs and the torque current i.
Excitation current command value iγ calculated by the three-phase / two-phase conversion unit 23 for converting into δs and the vector control command value calculation unit 11
s ^* , the torque current command value iδs ^* , the excitation current iγs ′ after the filter calculation and the torque current iδs ′ output from the filter calculation unit 22 as the filter calculation means described later.
To the excitation voltage command value vγs ^* and the torque voltage command value vδs ^* as the current control unit, and the output of the current control unit 21 and the voltage of the DC power supply 3 detected by the voltage detector 5. Modulation degree calculation unit 1 as a modulation degree calculation means for obtaining the modulation degree μ by inputting the value Vdc
5, the correction coefficient calculator 16 that calculates the correction coefficient h corresponding to the modulation degree μ obtained by the modulation degree calculator 15, and the primary frequencies ω and 3 calculated by the vector control command value calculator 11.
The excitation current iγs and the torque current iδs output from the two-phase / two-phase conversion unit 23 and the modulation degree μ calculated by the modulation degree calculation unit 15 are used as inputs to perform a filter calculation and the excitation current iγs ′.
And the excitation voltage command value v generated by the current control unit 21 and the filter calculation unit 22 that outputs the torque current iδs ′.
A two-phase / three-phase conversion unit 13 as a two-phase / three-phase conversion unit that converts γs ^* and the torque voltage command value vδs ^* into a three-phase voltage using the power supply phase angle θ, and the two-phase / 3-phase conversion unit 13. PWM using the voltage utilization rate improvement calculation unit 24 that calculates the corrected three-phase voltage command values Vuo, Vvo, Vwo from the output of
And a PWM generator 14 that outputs a signal to the inverter 2.

The vector control command value calculation unit 11 inputs a motor rotation command value N ^* and a magnetic flux command value Φγr ^* given from the outside as preset drive target values of the induction motor 1, and realizes these command values. Excitation current command value iγ
The s ^* , the torque current command value iδs ^* , the primary frequency ω, and the power supply phase angle θ are calculated and output using the input and the motor rotation speed detection value N that is the output of the motor rotation speed detection unit 4, respectively. The calculation of the vector control command value performed here is a widely used technique, and the torque current command value i
δs ^* is calculated using the following equation (1).

I δs ^* (s) = {k ₁ + (k ₂ / s)} {N ^* (s) −N (s)} (1) where k ₁ is the proportional gain of the motor speed detection unit 4. , K ₂ are integral gains of the motor speed detection unit 4, and N
Is a motor rotation speed detection value, and s is a Laplace operator. The exciting current command value iγs ^* is calculated using the following equation (2).

Iγs ^* (s) = (Lr / M / Rr) {s + (Rr / Lr)} Φγr ^* (s) (2) where Lr is the secondary inductance of the induction motor 1 and M is the induction motor 1 Rut is the mutual inductance of
Is the secondary resistance of the induction motor 1. Also, the primary frequency ω
Is calculated using the following equations (3) and (4).

Ωr = pN (3) ω = ωr + (MRr / Lr) (iδs ^* / iγs ^* ) (4) where ωr is the rotation speed (electrical angle) of the induction motor 1,
Let p be the number of pole pairs of the induction motor 1. In addition, the power supply phase angle θ
Is calculated using the following equation (5).

Θ (s) = ω / s (5) The current control unit 21 controls the excitation current command value iγs ^* and the torque current command value iδs ^* calculated by the vector control command value calculation unit 11 and the filter calculation unit 22. The exciting current iγs 'and the torque current iδs', which are calculation outputs, are matched by proportional-plus-integral control, and the exciting voltage command value vγs ^* and the torque voltage command value vδs ^* are calculated using the constants of the induction motor 1 ^.
Is calculated.

The modulation degree calculator 15 calculates the excitation voltage value vγ.
From s ^* , the torque voltage command value vδs ^*, and the voltage value Vdc of the DC power supply 3 detected by the voltage detector 5, the following equation 1
The modulation degree μ is calculated according to the equation (6) shown in FIG.

[0027]

(Equation 1)

The filter calculation unit 22 includes a modulation degree calculation unit 15
According to the modulation degree μ which is the output of the above, the filter operation as shown in the following equation (7) is executed and the exciting current iγs ′ and the torque current iδs ′ are output. When μ ≦ 1, iγs ′ = iγs iδs ′ = iδs μ> 1, iγs ′ (s) = G (s) iγs iδs ′ (s) = G (s) iδs (7) However, the exciting current iγs And the torque current iδs are the three-phase current iu of the induction motor 1 detected by the current detector 25.
It is obtained by coordinate conversion of s, ivs, and iws by the three-phase / two-phase conversion unit 23. Further, G (s) is a transfer function having a filter characteristic for removing a frequency component of 6m times the primary frequency ω (m is a natural number) as described later.

The two-phase / three-phase conversion unit 13 sets the excitation voltage command value vγs ^* and the torque voltage command value vδs ^* , which are the outputs of the current control unit 21, to 3 by the following equation (8). The phase voltage command values Vuo ^* , Vvo ^* , and Vwo ^* are converted.

[0030]

(Equation 2)

The voltage utilization rate improving operation unit 24 calculates the corrected three-phase voltage command values Vuo, Vvo, V according to the following equation (9).
Calculate wo. Vuo = Vuo ^* -(1/2) {max (Vuo ^* , Vvo ^* , Vwo ^* ) + min (Vuo ^* , Vvo ^* , Vwo ^* )} Vvo = Vvo ^* -(1/2) {max (Vuo ^* , Vvo ^* , Vwo ^* ) + min (Vuo ^* , Vvo ^* , Vwo ^* )} Vwo = Vwo ^* -(1/2) {max (Vuo ^* , Vvo ^* , Vwo ^* ) + min (Vuo ^* , Vvo ^* , Vwo ^*) )} (9) The inverter 2 uses the DC generator 3 based on the corrected three-phase voltage command values Vuo, Vvo, and Vwo via the PWM generator 14.
Of the DC voltage Vdc of the three-phase AC voltage Vuo, Vvo ',
It is converted into Vwo ′ and applied to the induction motor 1. However,
The amplitude of the three-phase AC voltage is limited to Vdc / 2.

Next, the operation of the first embodiment will be described.
First, a region where the output voltage of the inverter 2 is not saturated (μ ≦
In 1), with respect to the excitation current command value iγs ^* and the torque current command value iδs ^* calculated by the vector control command value calculation unit 11, the current control unit 21 generates an excitation voltage that realizes these command values. When the command value vγs ^* and the torque voltage command value vδs ^* are calculated using the motor constant,
The input 3-phase current to the induction motor 1 detected by the current detection unit 25 is input to the current control unit 21 via the 3-phase / 2-phase conversion unit 23 and the filter calculation unit 22, and feedback current control is performed. Then, the modulation degree calculator 15 calculates the modulation degree μ based on the excitation voltage command value vγs ^* and the torque voltage command value vδs ^* , which are the outputs of the current controller 21, and the correction coefficient calculator 16 determines the modulation degree μ. A corresponding correction coefficient h is calculated. However, since the output voltage of the inverter 2 is not saturated here, h = 1. The excitation voltage command value vγs ^* and the torque voltage command value vδs ^* are converted into a three-phase voltage by the two-phase / three-phase conversion unit 13 and are processed by the voltage utilization rate improvement calculation unit 24, and then multiplied by the correction coefficient h. Being P
It is input to the inverter 2 via the WM generator 14 and the inverter 2 is controlled.

Next, consider the overmodulation drive region (μ> 1) where the output voltage of the inverter 2 is saturated. FIG. 2 shows the corrected three-phase voltage command value Vuo and the three-phase AC voltage V in the overmodulation drive region.
uo 'and the voltage waveforms of the fundamental wave components of the three-phase AC voltage Vuo' are shown, and FIG. 3 shows each voltage waveform when the modulation degree μ in FIG. 2 is further increased.

In the figure, 1 / th of the three-phase AC voltage Vuo '
If the saturation section in the four cycles is β, the saturation section β is
In FIG. 2, it is in the range of 0 ≦ β ≦ π / 3, and in FIG. 3, it is π / 3.
It is in the range of ≦ β ≦ π / 2. Saturation interval β is 0 ≦ β ≦ π /
In the case of 3, the three-phase AC voltage Vuo ′ can be expressed by the following equation (10).

[0035]

[Equation 3]

When the saturation section β is π / 3 ≦ β ≦ π / 2, the three-phase AC voltage Vuo ′ is expressed by the following equation (1)
It can be represented by 1).

[0037]

[Equation 4]

Next, the excitation voltage vγs and the torque voltage vδs actually applied to the induction motor 1 in the overmodulation drive region.
Is expressed by the following equation (12) using the three-phase AC voltages Vuo ′, Vvo ′, and Vwo ′.

[0039]

(Equation 5)

However, α is the phase of the γ-δ axis coordinates.
From the equations (10) and (11), the corrected three-phase voltage command value Vu is obtained.
When o, Vvo, and Vwo are expressed by the following equation (13),

[0041]

[Equation 6]

From equations (12) and (13), the excitation voltage vγ
s and the torque voltage vδs can be expressed by the following equation (14).

[0043]

(Equation 7)

That is, the excitation voltage vγs and the torque voltage vδ
s is composed of a direct current component corresponding to the fundamental wave component and a 6mth order harmonic component (m is a natural number). Since this voltage is applied to the induction motor 1, the exciting current iγs and the torque current iδs flowing through the induction motor 1 also include the 6m-th harmonic component. Therefore, by removing the frequency component 6 m times the primary frequency ω by the filter calculation unit 22, the excitation current iγs ′ and the torque current iδs ′ after the filter calculation are
Even in the over-modulation drive region, the influence of the harmonic component is reduced, so that the feedback current control can be performed well.

For example, when the primary frequency ω is 200 Hz and the transfer function G (s) of the filter calculation unit 22 is given by the following expression (15), the frequency characteristic at this time is as shown in FIG. G (s) = 0.893137 (s ² + 40.9172s + 5.69303e ⁷ ) (s ² + 229.414s + 2.25579e ⁸ ) / (s ² + 943.876s + 5.43664e ⁷ ) / (s ² + 1386.72s + 2.10974e ⁸ ) (15) For example, when the primary frequency ω is 300 Hz and the transfer function G (s) shown in the following equation (16) is used, the frequency characteristic at this time is as shown in FIG.

G (s) = 0.885555 (s ² + 75.0339s + 1.26885e ⁸ ) (s ² + 619.982s + 5.11693e ⁸ ) / (s ² + 754.698s + 1.22273e ⁸ ) / (s ² + 1768.44s + 4.70223e ⁸ ) (16) That is, the filter calculation unit 22 in this case removes frequency components 6 times and 12 times the primary frequency ω.
FIG. 6 is a waveform diagram showing respective voltages and currents when the DC voltage Vdc decreases in the present embodiment. Compared with the waveform of the conventional control device shown in FIG. 17, vibration of the excitation voltage command value vγs ^* , the torque voltage command value vδs ^{*, the} excitation current iγs, and the torque current iδs in the overmodulation drive region (μ> 1) It can be seen that is decreasing.

As described above, according to the first embodiment, in the overmodulation drive region, the filter operation unit 22 removes the harmonic current of 6 m times the primary frequency ω, so that the influence of the harmonic component is small. Current control can be performed. Therefore, the output voltage of the inverter 2 becomes stable, and the output torque of the induction motor 1 can be stabilized and the efficiency thereof can be improved.

Next, the second embodiment will be described.
In the second embodiment, a case will be described in which the characteristics of the filter calculation unit 22 of the first embodiment are changed to those that remove only the sixth harmonic component of the primary frequency ω. The other configurations are similar to those of the first embodiment, and therefore the description is omitted here. For example, the filter calculation unit 22 determines that the primary frequency ω is 2
When the frequency is 00 Hz, the transfer function G (s) is calculated by the following equation (17).
The frequency characteristics are as shown in FIG.

G (s) = 0.954965 (s ² + 40.9172s + 5.69303e ⁷ ) / (s ² + 943.876s + 5.43664e ⁷ ) (17) For example, when the primary frequency ω is 300 Hz, the transfer function is, for example, When G (s) is represented by the following equation (18), the frequency characteristic is as shown in FIG. G (s) = 0.963654 (s ² + 75.0338s + 1.26884e ⁸ ) / (s ² + 754.698s + 1.22273e ⁸ ) (18) FIG. 9 is a graph of the second embodiment when the DC voltage Vdc is lowered. It is a wave form diagram which shows the voltage and current of. Among the harmonic components of the exciting current iγs and the torque current iδs, the dominant sixth-order harmonic component is. This is the formula (1
As can be seen from 0), (11), (13), and (14), the harmonic components of the excitation voltage vγs and the torque voltage vδs dominate the sixth harmonic component, and the induction motor 1
This is because it is difficult for high-order harmonic currents to flow due to the filter characteristics of. Therefore, even if only the 6th harmonic component is removed by the filter calculation unit 22, as shown in FIG.
7, the excitation voltage command value vγ in the overmodulation drive region
s ^* , torque voltage command value vδs ^* and exciting current iγs,
It can be seen that the vibration of the torque current iδs is decreasing.

As described above, according to the second embodiment, just by removing the harmonic current of 6 times the primary frequency ω in the filter operation section 22 in the overmodulation drive region, the same as in the first embodiment. It is possible to perform feedback current control that is less affected by harmonic components. In addition, the filter calculation unit 2
In 2, the required filter order for one band to be removed is at least second order. Therefore, in the first embodiment that removes the 6m-th order harmonic, at least the fourth-order filter is required, whereas in the second embodiment that removes only the sixth-order harmonic, the filter operation is performed by the second-order filter. Part 2
2 can be configured, the configuration of the filter calculation unit 22 can be simplified and the calculation load can be reduced.

Next, a third embodiment will be described.
In the third embodiment, the characteristic of the filter calculation unit 22 of the first embodiment is that the harmonic components of the sixth order or higher of the primary frequency ω are all removed, that is, the cutoff frequency is 6ω.
The following low pass characteristics were used. Other configurations are the same as the configurations of the first embodiment, and thus the description thereof will be omitted.

For example, when the transfer function G (s) of the filter is set as shown in equation (19) and the frequency characteristic is set as shown in FIG. 10, even when the primary frequency ω is 200 Hz, 30
Even at 0 Hz, the frequency component of 6 times or more of ω can be sufficiently removed. G (s) = 0.11216 (s + 22647.6) / (s + 2540.16) (19) FIG. 11 is a waveform diagram showing the voltage and current when the DC voltage Vdc is reduced in the third embodiment. 11, the excitation voltage command value vγs ^* and the torque voltage command value vδs ^{* in the} overmodulation drive region (μ> 1) are different from those in FIG. 17 described above.
It can be seen that the vibrations of the exciting current iγs and the torque current iδs are reduced.

As described above, according to the third embodiment, in the overmodulation drive region, the filter operation unit 22 removes all the harmonic currents of 6 times or more of the primary frequency ω, thereby realizing the first and second embodiments. It is possible to perform the feedback current control in which the influence of the harmonic component is small as in the case of the form. Further, since the filter order required to realize the low-pass filter characteristic is at least the first order, the configuration of the filter is simpler and the calculation load can be reduced as compared with the first and second embodiments. Further, it is possible to reduce the change in the calculation content of the filter calculation unit 22 due to the change of the primary frequency ω.

Next, a fourth embodiment will be described.
In the fourth embodiment, the calculation in the filter calculation unit 22 is performed in synchronization with the rotation of the induction motor 1. First to third
In the above embodiment, the filter calculation unit 22 having the filter characteristic according to the primary frequency ω removes the harmonic component in the overmodulation drive region. Therefore, if the primary frequency ω changes, the calculation content of the filter calculation unit 22 is necessarily changed, and the calculation amount increases. Therefore, if the filter calculation unit 22 is driven by a trigger signal that is synchronized with the rotation of the induction motor 1, the filter characteristics corresponding to the primary frequency ω can always be realized without changing the calculation content of the filter calculation unit 22.

FIG. 12 is a diagram showing the configuration of the fourth embodiment. In FIG. 12, the motor rotation speed detection unit 4 ′ is
The rotation speed N of the induction motor 1 is detected, and a trigger signal is output 100 times, for example, every time the induction motor 1 makes one rotation. Then, the filter calculation unit 22 performs a predetermined calculation each time this trigger signal is output. The calculation is performed, for example, according to the following equation (20).

G (Z) = 0.940809 (Z ² -1.46082Z + 1) / (Z ²
-1.37435Z + 0.881619) (20) However, let Z be a shift operator. When the number p of pole pairs of the induction motor 1 is 2, even if the rotation speed N of the induction motor 1 changes, the frequency characteristic of the filter calculation unit 22 is always as shown in FIG. 13, and the frequency component of 6ωr is removed. . In the case of the induction motor 1, since the primary frequency of the inverter is ω≈ωr, almost the same effect as in the second embodiment can be obtained.

As described above, according to the fourth embodiment, by synchronizing the operation of the filter calculation unit 22 with the rotation of the induction motor 1, the calculation content of the filter calculation unit 22 is changed to the rotation speed N of the induction motor 1. It can always be the same regardless of changes.
In the first to fourth embodiments, the case of driving the induction motor 1 has been described, but the present invention is not limited to this, and can be applied to the case of driving a synchronous motor or the like, for example.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a configuration of a first embodiment of the present invention.

FIG. 2 is a diagram showing a voltage waveform in the case of 0 ≦ β ≦ π / 3 in the overmodulation region.

FIG. 3 is a waveform diagram showing a voltage in the case of π / 3 ≦ β ≦ π / 2 in the overmodulation region.

FIG. 4 is a diagram showing frequency characteristics of a filter calculation unit when ω = 200 Hz in the first embodiment of the present invention.

FIG. 5 is a diagram showing frequency characteristics of a filter calculation unit when ω = 300 Hz in the first embodiment.

FIG. 6 is a waveform diagram showing voltage and current in the first embodiment of the above.

FIG. 7 is a diagram showing frequency characteristics of a filter calculation unit when ω = 200 Hz in the second embodiment of the present invention.

FIG. 8 is a diagram showing frequency characteristics of a filter calculation unit when ω = 300 Hz in the second embodiment.

FIG. 9 is a waveform diagram showing voltage and current in the second embodiment of the above.

FIG. 10 is a diagram showing frequency characteristics of a filter calculation unit according to the third embodiment of the present invention.

FIG. 11 is a waveform chart showing voltage and current in the third embodiment of the above.

FIG. 12 is a block diagram showing a configuration of a fourth exemplary embodiment of the present invention.

FIG. 13 is a diagram showing frequency characteristics of a filter calculation unit in the fourth embodiment.

FIG. 14 is a block diagram showing a configuration of a conventional inverter control device.

FIG. 15 is a waveform diagram showing the exciting current control performance of the conventional inverter control device.

FIG. 16 is a waveform diagram showing torque current control performance of a conventional inverter control device.

FIG. 17 is a waveform diagram showing voltage and current in a conventional inverter control device.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Induction motor 2 Inverter 3 DC power supply 4, 4'Motor speed detection unit 11 Vector control command value calculation unit 13 Two-phase / 3-phase conversion unit 15 Modulation degree calculation unit 16 Correction coefficient calculation unit 21 Current control unit 22 Filter calculation unit 23 3 Phase / 2 Phase Converter 25 Motor Current Detector

─────────────────────────────────────────────────── ─── Continuation of the front page (51) Int.Cl. ⁶ Identification number Office reference number FI technical display location H02P 7/63 302 H02P 7/63 302R

Claims (6)

[Claims] 1. A controller of a current control type inverter for controlling a primary current of an alternating current motor by dividing it into an exciting current and a torque current, wherein an exciting current command value for realizing a preset drive target value of the alternating current motor, Vector control command value computing means for computing a torque current command value and a primary frequency according to the rotation state of the AC motor and the drive target value, and the exciting current based on the primary current of the AC motor detected by the primary current. And a motor current detection unit that calculates the torque current, a filter calculation unit that performs a calculation to remove a frequency component corresponding to the primary frequency from the excitation current and the torque current, a calculation output of the filter calculation unit, and the excitation Current control means for calculating an excitation voltage command value and a torque voltage command value based on the current command value and the torque current command value; A two-phase / three-phase conversion unit that calculates a three-phase voltage command value that controls the output of the inverter from the command value and the torque voltage command value; and when the output of the inverter is saturated, the filter calculation unit performs the calculation. A control device for an inverter, which is configured to execute. 2. The inverter according to claim 1, wherein the filter calculation means is configured to remove a frequency component of a natural number multiple of 6 of the primary frequency from the exciting current and the torque current. Control device. 3. The filter calculating means is configured to remove only a frequency component of 6 times the primary frequency from the exciting current and the torque current.
Inverter control device according to. 4. The control of an inverter according to claim 1, wherein the filter calculation means is configured to remove all frequency components of 6 times or more of the primary frequency from the exciting current and the torque current. apparatus. 5. The inverter control device according to claim 1, wherein the filter calculation means is configured to execute the calculation in synchronization with the rotation of the AC motor. 6. A controller of a current control type inverter for controlling a primary current of an alternating current motor by dividing it into an exciting current and a torque current, and an exciting current command value for realizing a preset drive target value of the alternating current motor, The torque current command value and the primary frequency are calculated according to the rotation state of the AC motor and the drive target value, and the frequency component corresponding to the primary frequency is removed from the exciting current and the torque current detected when the output of the inverter is saturated. Then, the excitation voltage command value and the torque voltage command value are calculated based on the excitation current and the torque current from which the frequency component corresponding to the primary frequency is removed, and the excitation current command value and the torque current command value, and the excitation voltage An inverter control device for controlling the inverter according to a command value and a torque voltage command value.