JP2019083673A - Inverter and motor drive control method - Google Patents

Abstract

To suppress an unstable phenomenon of current control when controlling drive of a motor.SOLUTION: During drive control of a motor 3, a PWM inverter 2 performs dead beat control to calculate a d, q axis voltage command value from d, q axis current command values i, iof the motor 3, d, q axis current detection values i, icoordinate-converted to a synchronous coordinate system, and a frequency command value ω, calculates a second d, q axis voltage command value to suppress harmonics by calculation on the basis of a discretized model of motor 3 from the d, q axis current command values i, iof the motor 3, the d, q axis current detection values i, icoordinate-converted to the synchronous coordinate system, and the frequency command value ω, and adds the second d, q-axis voltage command values to the first d, q-axis voltage command values.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a technology for suppressing current distortion synchronized with the frequency of a power source generated when a motor (an induction motor, a permanent magnet motor) is driven.

  When a motor (induction motor, permanent magnet motor) is driven by a PWM inverter, distortion occurs in the inverter current (motor current) due to a dead time provided to a gate signal (on / off command signal) of a switching element in the inverter. This distortion needs to be suppressed because it has adverse effects such as torque pulsation and deterioration of control performance.

  Distortion appears as current harmonics synchronized to the fundamental frequency of the power supply. The current harmonics are mainly -5, 7, 11, and 13 times the frequency of the fundamental wave, and when viewed in coordinates synchronized with the fundamental wave, they have a frequency of 6 and 12 times.

  The harmonic distortion using PR control (proportional resonance control) of Non Patent Literatures 1 and 2 as a method of suppressing distortion by feedback of current utilizing the fact that harmonic distortion of the above frequency has a frequency synchronized with the fundamental wave. There is a wave suppression method.

  The PR control is easy to implement because the calculation load and memory used are small, but high gain can be provided for a specific frequency, and distortion components can be efficiently suppressed.

M. Liserre, R. Teodorescu and F. Blaabjerg. “Multiple harmonics control for three-phase grid converter systems with the use of PI-RES current controller in a rotating frame.” IEEE Transactions on Power Electronics, vol. 21, no. 3, pp. 836-841, May 2006. C. Liu, F. Blaabjerg, W. Chen and D. Xu. "Stator Current Harmonic Control with Resonant Controller for Doubly Fed Induction Generator." IEEE Transactions on Power Electronics, vol. 27, no. 7, no. 7, pp. 3207-3220 , July 2012. O. Kukrer. “Discrete-time current control of voltage-fed three-phase pwm inverters.” IEEE Transactions on Power Electronics, 11: 260-269, 1996. Yasuhiro Yamamoto, Koji Kodama, Tetsuo Yamada, Tadahiro Ichioka, Minoru Niwa, "Digital Current Control Method of Induction Motor by PWM Synchronous Current Sample", Journal of the Institute of Electrical Engineers of Japan, Vol. 112 (1992), No. .7, P. 613-622

JP, 2016-10311, A

  As described in Non-Patent Document 2, when current control and harmonic suppression by PR control are used in combination, it is known that the frequency to be suppressed by PR control becomes unstable when it exceeds the current control band. .

  The upper limit of the PI (proportional integration) current control band in digital control is determined by the current control cycle (generally, it is about 1/10 of the sampling frequency of current control). In the PWM inverter, the current control is performed in synchronization with the carrier signal (a triangular wave signal for generating the on / off signal of the switching element as compared to the voltage command), so the current control period is the carrier frequency (that is, switching of the switching element It depends on the frequency). Since the switching loss of the switching element increases when the switching frequency is increased, the switching frequency also has an upper limit.

  Therefore, when driving a high-speed motor whose fundamental frequency is high and the frequency of harmonics is high, or a large-capacity motor whose switching frequency can not be increased from the viewpoint of efficiency and thermal design, PR control stably suppresses harmonics. Can not secure the current control band of Therefore, there is a problem that the combination of the conventional PI current control and the PR control harmonic suppression becomes unstable when the high speed motor or the large capacity motor is rotated at the rated rotational speed.

  An object of the present invention is to suppress the unstable phenomenon of current control at the time of drive control of a motor, in view of the above-mentioned circumstances.

  Therefore, one aspect of the present invention is an inverter that drives and controls a motor, and the d and q axis current detection values of the motor at the current time, the previous d and q axis voltage command values, and the previous d and q axis disturbance The d and q axis current predicted values at the next time are calculated from the voltage estimated value and the discretized model of the motor, and the d and q axis current predicted values at the calculated next time and the stator of the motor are calculated. The d and q axis disturbance voltage estimated values of the next time are calculated based on the resistance and the magnetic flux and the frequency command value, and the d and q axis disturbance voltage estimated values of the calculated next time and the d and q of the motor are calculated. A dead beat control unit that calculates a first d, q axis voltage command value at the next time based on the axis current command value and the d, q axis current predicted value of the next time; d, q of the motor The motor is separated from the axis current command value and the d, q axis current detection values that are coordinate-converted to the synchronous coordinate system and the frequency command value. A proportional resonance control unit for calculating a second d, q-axis voltage command value for suppressing harmonics by calculation based on a simulation model, and the second d, q axis as the first d, q-axis voltage command value And an adding unit for calculating a d, q-axis voltage command value of the motor by adding the voltage command value.

  In one embodiment of the present invention, in the inverter, the proportional resonance control unit provides a frequency command value that is six times the frequency command value to an operation based on the discretization model to suppress harmonics of the six times the frequency. The second d and q axis voltage command values to be calculated are calculated.

  One aspect of the present invention is the inverter according to claim 1 or 2 which drives and controls a motor of an induction motor specification, and a detection unit that detects a mechanical angular velocity of the motor based on the rotation speed of the motor; A speed control unit that outputs a torque command value of the motor based on the speed command value of the motor and the mechanical angular velocity, an angular velocity conversion unit that converts the mechanical angular velocity into an electrical angular velocity, the torque command value, and the electrical angular velocity of the motor A torque control unit that outputs the d, q axis current command value and the frequency command value based on the following equation, and an integration unit that integrates a time series signal of the frequency command value from the torque control unit to calculate an electrical angle; The multiplication unit that calculates the product of the frequency command value from the torque control unit and the 1.5-fold value of the cycle of current control, the three-phase current detection value on the output side of the inverter is the electrical angle from the integration unit Group A first coordinate conversion unit for converting the d and q axis current detection values; and the electric angle from the integration unit and the product from the multiplication unit; and the d and q axis voltage command values from the addition unit A second coordinate conversion unit for converting into a three-phase voltage command value based on the sum of the values and a three-phase generated by performing pulse width modulation based on the three-phase voltage command value supplied from the second coordinate conversion unit And a pulse width modulation unit for outputting an alternating voltage to the motor.

  One aspect of the present invention is an inverter according to claim 1 or 2 which drives and controls a motor of a permanent magnet synchronous motor specification, and detects an angular velocity and an electrical angle of the motor based on the rotation speed of the motor. A speed control unit that outputs a torque command value of the motor based on an external speed command value and the mechanical angular velocity, an angular velocity conversion unit that converts the mechanical angular velocity into a frequency command value, and the torque command value A torque control unit that outputs the d, q axis current command value based on the multiplication unit that calculates a product of the frequency command value from the angular velocity conversion unit and a value 1.5 times the period of current control; A first coordinate transformation unit for converting the three-phase current detection value on the output side of the sensor into the d, q axis current detection value based on the electrical angle from the detection unit; and the d, q axis voltage command from the addition unit Values from the detector A second coordinate conversion unit for converting into a three-phase voltage command value based on the sum of an electrical angle and the value of the product from the multiplication unit; and a three-phase voltage command value provided from the second coordinate conversion unit And a pulse width modulation unit for outputting a generated three-phase AC voltage to the motor by performing pulse width modulation on the basis of the pulse width modulation.

  One aspect of the present invention is a method of driving control of a motor by an inverter, which includes: d and q axis current detection values of a motor at a current time, previous d and q axis voltage command values, and previous d and q axis disturbance voltages The d and q axis current predicted values of the next time are calculated from the estimated value and the discretized model of the motor, and the d and q axis current predicted values of the calculated next time and the stator resistance of the motor are calculated. The d and q axis disturbance voltage estimated value of the next time is calculated based on the magnetic flux and the frequency command value, and the d and q axis disturbance voltage estimated value of the calculated next time and the d and q axes of the motor are calculated. Calculating a first d, q axis voltage command value at the next time based on the current command value and the d, q axis current predicted value at the next time; and d, q axis current command values of the motor Based on the discretized model of the motor concerned from the d, q axis current detection values and the frequency Calculating a second d, q axis voltage command value for suppressing harmonics by calculation; adding the second d, q axis voltage command value to the first d, q axis voltage command value Calculating a d, q axis voltage command value of the motor.

  According to the present invention as described above, it is possible to suppress the unstable phenomenon of current control at the time of drive control of the motor.

BRIEF DESCRIPTION OF THE DRAWINGS The block block diagram of the system which has an inverter which drive-controls the motor of the induction motor specification which is 1 aspect of this invention. BRIEF DESCRIPTION OF THE DRAWINGS The block block diagram of the system which has an inverter which drive-controls the motor corresponding to the permanent magnet synchronous motor specification which is 1 aspect of this invention. The block block diagram of the current control part comprised to the inverter of FIG.1, 2. FIG. FIG. 4 is a block diagram of a PR control unit provided in the current control unit of FIG. 3; FIG. 4 is a block diagram of a dead beat control unit provided in the current control unit of FIG. 3; Block diagram of PI control. (A) is a characteristic diagram comparing gains ([dB]) of PI control (PI) and dead beat control (DB), (b) is a phase of PI control (PI) and dead beat control (DB) ([deg. The Bode diagram which compared]. Fifth and seventh harmonic components when the permanent magnet synchronous motor is driven at 100 Hz (frequency command value ω _e = 2π × 100) with an inverter with a carrier frequency of 5 kHz, current control cycle T _s = 100 μs, and dead time 6 μs Table comparing the amplitudes of. The wave form diagram of the three-phase current at the time of performing harmonic suppression by PR control using PI control for current control. The wave form diagram of the three-phase current at the time of trying harmonic suppression by PR control using dead beat control for current control.

  Embodiments of the present invention will be described below with reference to the drawings.

  In the motor drive system 1 illustrated in FIGS. 1 and 2, the PWM inverter 2 and the motor 3 are connected by the three-phase wire 5, and the motor 3 is coupled to the load 4.

(Aspect 1 of the PWM inverter 2)
The motor drive system 1 illustrated in FIG. 1 includes a PWM inverter 2 that drives and controls a motor 3 of an induction motor specification. Each functional unit of the PWM inverter 2 will be described below.

  The detection unit 20 detects the mechanical angular velocity of the motor 3 based on the rotational speed of the motor 3.

  The speed control unit 21 outputs a torque command value of the motor 3 based on the external speed command value and the mechanical angular velocity.

The angular velocity converter 22 converts the mechanical angular velocity into an electrical angular velocity ω _r .

The torque control unit 23 outputs d, q axis current command values i ^* _d , i ^* _q and a frequency command value ω _e based on the torque command value and the electric angular velocity ω _r .

The integration unit 24 integrates the time-series signal of the frequency command value ω _e from the torque control unit 23 to calculate an electrical angle θ _e .

The multiplying unit 25 calculates the product of the frequency command value ω _e from the torque control unit 23 and the 1.5-fold value of the cycle Ts of the current control.

The coordinate conversion unit 26 (corresponding to the first coordinate conversion unit) converts the three-phase current detection values i _u , i _v , i _w detected at the output side of the PWM inverter 2 into the electrical angle θ _e from the integration unit 24. Based on this, conversion is made to d, q axis current detection values _id , _iq .

Three-phase current detection values i _u , i _v , i _w are detected by a current sensor 7 attached to the three-phase electric wire 5. The current sensor 7 detects three-phase current detection values i _u , i _v and i _w by the current detection method of Non-Patent Document 4.

The current control unit 27, d from the torque control unit 23, q-axis current command value i ^* _d, i ^* _q and the frequency command value omega _e and d from the coordinate conversion unit 26, q-axis current detection value i _d, i _The d, q axis voltage command values v ^* _d and v ^* _q are calculated based on _q .

The coordinate conversion unit 28 (corresponding to a second coordinate conversion unit) calculates the d, q axis voltage command values v ^* _d and v ^* _q from the current control unit 27 and the electric angle θ _e from the integration unit 24 and the multiplication unit 25. _Are converted into three-phase voltage command values v ^* _u , v ^* _v , and v ^* _w based on the sum of the product and the value of the product.

The pulse width modulation unit 29 (hereinafter, the PWM unit 29) performs pulse width modulation (PWM) based on the three-phase voltage command values v ^* _u , v ^* _v and v ^* _w supplied from the coordinate conversion unit 28 and generates them. A three-phase alternating voltage is output to the motor 3.

(Example 2 of PWM inverter 2)
The motor drive system 1 illustrated in FIG. 2 includes a PWM inverter 2 that drives and controls a motor 3 of a permanent magnet synchronous motor specification. Hereinafter, each functional unit of the PWM inverter 2 will be described.

Detector 20 detects the mechanical angular velocity and the electrical angle theta _e of the motor 3 based on the rotation speed of the motor 3.

  The speed control unit 21 outputs the torque command value of the motor 3 based on the external speed command and the mechanical angular velocity, as in the first embodiment.

The angular velocity conversion unit 22 converts the mechanical angular velocity into a frequency command value ω _e .

The torque control unit 23 outputs d and q axis current command values i ^* _d and i ^* _q based on the torque command value.

The multiplication unit 25 calculates the product of the frequency command value ω _e from the angular velocity conversion unit 22 and the 1.5-fold value of the current control period Ts.

The coordinate conversion unit 26 sets the three-phase current detection values i _u , i _v , i _w detected by the current sensor 7 on the output side of the PWM inverter 2 to d, q based on the electrical angle θ _e from the detection unit 20. The axis current detection values i _d and i _q are converted.

Similar to Embodiment 1, the current control unit 27 controls the d, q-axis current command values i ^* _d and i ^* _q from the torque control unit 23 and the frequency command value ω _e and the d and q axes from the coordinate conversion unit 26. based current detection value i _d, to the i _q, calculates d, q-axis voltage command value v ^* _d, the v ^* _q.

The coordinate conversion unit 28 sets the d, q axis voltage command values v ^* _d and v ^* _q from the current control unit 27 to the sum of the electrical angle θ _e from the detection unit 20 and the product value from the multiplication unit 25. It converts into three-phase voltage command value v ^* _u , v ^* _v , v ^* _{w based} on.

Similar to Embodiment 1, the PWM unit 29 generates and generates three-phase pulse width modulation (PWM) based on the three-phase voltage command values v ^* _u , v ^* _v and v ^* _w supplied from the coordinate conversion unit 28. The AC voltage is output to the motor 3.

(Aspect Example of Current Control Unit 27)
The current control unit 27 mounts the dead beat control unit 11 and the proportional resonance control unit 12 (hereinafter, the PR control unit 12) in parallel as shown in FIG.

In both the dead beat control unit 11 and the PR control unit 12, the d, q-axis current command values i ^* _d , i ^* _q and the d, q-axis current detection values i _d , i _q coordinate-converted to the synchronous coordinate system And the frequency command value ω _e are input.

  Each functional unit of the current control unit 27 will be described below.

Deadbeat control unit 11, d of the motor 3, q-axis current command value i ^* _d, i ^* _q and the coordinate transformation to the synchronous coordinate system the d, q-axis current detection value i _d, i _q and the frequency command value ω The first d, q-axis voltage command values v ^* _{d, DB} , v ^* _{q, DB} are calculated from _e .

The PR controller 12 detects the d, q-axis current command values i ^* _d , i ^* _q of the motor 3 and the d, q-axis current detection values _id , _iq and the frequency command value ω _e which are coordinate-converted to the synchronous coordinate system. The second d, q-axis voltage command values v ^* _{d, PR} , v ^* _{q, PR} for suppressing predetermined harmonics are calculated by calculation based on the discretization model of the motor 3 from

The adding unit 13 adds the first d, q-axis voltage command values v ^* _{d, DB} , v ^* _{q, DB} from the dead beat control unit 11 to the second d, q-axis voltage from the PR control unit 12. The command values v ^* _{d, PR} , v ^* _{q, PR} are added to calculate the d, q-axis voltage command values v ^* _d , v ^* _q of the motor 3.

(Example of PR control unit 12)
The PR control unit 12 has a block configuration of control illustrated in FIG. 4 and suppresses predetermined harmonics (harmonics having a frequency six times the frequency command value in the present embodiment) in the synchronous coordinate system.

The PR controller 12 detects the d, q-axis current command values i ^* _d , i ^* _q of the motor 3 and the d, q-axis current detection values _id , _iq and the frequency command value ω _e which are coordinate-converted to the synchronous coordinate system. And a voltage command value v ^* _PR for suppressing predetermined harmonics by calculation based on a discretized model of the motor 3 (corresponding to the second d, q axis voltage command values v ^* _{d, PR} , v ^* _{q, PR)} Calculate).

  In the above calculation, the following equation (1) is provided by discretization at a period Ts of digital current control by bilinear conversion. In equation (1), s is a differential operator, and KPR is a gain of current proportional control.

Note that although the PR control unit 12 is a secondary IIR filter, the frequency command value ω _e fluctuates according to the rotation speed of the motor in the case of a permanent magnet motor, and the rotation speed and load of the motor 3 in the induction motor. , the coefficient is adapted to be changed in accordance with the frequency command value omega _e.

(Example of the dead beat control unit 11)
The dead beat control unit 11 mounts the current prediction unit 111, the disturbance voltage estimation unit 112, and the voltage command calculation unit 113 as illustrated in FIG. Illustrated current, voltage, Yes denoted using vector, for convenience of explanation, a current _{i s = [i d, i} q] T, the voltage _{v s = [v d, v} q] T ( Equation (2 The same applies to the above 4). The subscript k represents the sampling time of digital current control.

  The current prediction unit 111 generates a discretized model of the motor from the d and q axis current detection values measured at the current time, the previous d and q axis voltage command values, and the previous d and q axis disturbance voltage estimated values. Based on the above calculation, the d, q axis current predicted value at the next time k + 1 is calculated.

That is, the current prediction unit 111 detects the current time of d, q axis current detection value is _{, k} and the previous d, q axis voltage command value v ^* _{s, k} and the previous d, q axis disturbance voltage estimated value ^ From v _{l, k} , the current predicted value ^ i _{s, k + 1} after one sample (next time k + 1) is calculated by the following equation (2).

In equation (2), L ⁻¹ is “diagonal matrix having a reciprocal of leakage inductance L _σ at diagonal when motor 3 is an induction motor” “d-axis inductance L _d when motor 3 is a permanent magnet synchronous motor Is a diagonal matrix having a first row and a first column and a q-axis inductance L _q and a second row and a second column, respectively. Ts is the “period of digital d, q axis current control”.

The disturbance voltage estimation unit 112 calculates the d, q axis current predicted value ^ i _{s, k + 1} at the next time k + 1 supplied from the current prediction unit 111, the stator resistance R _{s of the} motor 3, the magnetic flux _{r r} and the frequency Based on the command value ω _e , the d, q-axis disturbance voltage estimated value ^ v _{l, k + 1} at the next time _{k + 1} is calculated.

That is, the disturbance voltage estimation unit 112 calculates the following equation (3) from the d, q axis current predicted value ^ i _{s, k + 1} and the frequency command value ω _e supplied from the angular velocity conversion unit 22: The d, q axis disturbance voltage estimated value ^ v _{l, k + 1} at time _{k + 1} is calculated.

In equation (3), Rs is “stator resistance of motor 3”, L is “diagonal matrix having Lσ diagonally when motor 3 is induction motor specification”, and “when motor 3 is permanent magnet synchronous motor specification” A diagonal matrix having Ld in the first row and first column and Lq in the second row and second column. When the motor 3 is an induction motor, φ _r is a secondary magnetic flux, and either an estimated value or a command value may be used. When the motor 3 is a permanent magnet synchronous motor specification, φ _r is represented by a permanent magnet magnetic flux λ m, and φ _r = [λ m, 0] ^T.

The voltage command calculation unit 113 calculates the d and q axis disturbance voltage estimated values ^ v _{l, k + 1 of} the next calculated time, the d and q axis current command values i ^* _{s, k of the} motor 3 _, and the next Based on the time d, q axis current predicted value ^ i _{s, k + 1} , the d, q axis voltage command value v ^* _{sDB, k + 1} at the next time is calculated by the following equation (4) Do.

(Description of the operation and operation of the first and second embodiments)
The deadbeat current control unit of the motor is applied to the controller of Non-Patent Document 3 or the like if the motor is a permanent magnet synchronous motor, or to the controller of Non-Patent Document 4 or Patent Document 1 if it is an induction motor.

  In particular, the dead beat control unit 11 of the present embodiment predicts the current at the next sample time k + 1 at the sample time k of digital current control, and calculates the d and q axis voltage command values calculated based on the predicted current. By outputting the current to the PWM unit 29 of FIG. 1 from time k + 1 to k + 2, the current at time k + 2 is made to coincide with the current command value. Therefore, the gain is constant up to the high frequency region.

FIG. 7 shows Bode diagrams of the dead beat control (DB) of FIG. 5 and the PI control (PI) of FIG. Bode plots are those seen the frequency response of the input (d of FIG. 3, q-axis current command value i ^* _d, i ^* _q) output for (d in FIG. 3, q-axis current detection value i _d, i _q) It is.

In particular, FIG. 7A compares the gains (gain [dB]) of PI control (PI) and dead beat control (DB). While the gain of PI control (PI) attenuates as the frequency rises, the gain of dead beat control (DB) becomes almost constant up to the high frequency region. The bandwidth of the PI control shown in FIG. 7B is approximately 700 Hz, since the bandwidth of the PI controller is a frequency whose gain is about 3 dB lower than the flat value near DC. Therefore, when “PI control” is applied instead of “dead beat control” in the current control of FIG. 3, PR of 6th harmonic suppression is performed when the harmonic that is 6 times the frequency command value ω _e reaches 700 Hz. Control becomes unstable. On the other hand, when “dead beat control” is applied to current control, the gain does not decrease even if the frequency exceeds 700 Hz, and thus PR control becomes stable.

The table shown in FIG. 8 is 5 when the permanent magnet synchronous motor is driven at 100 Hz (frequency command value ω _e = 2π × 100) by an inverter with a carrier frequency of 5 kHz, a digital current control cycle Ts = 100 μs, and a dead time of 6 μs. The amplitudes of the next and seventh harmonic components are compared. It can be seen that the harmonics are suppressed by the PR control when performing the dead beat control. When PI control is combined with PR control, it becomes unstable.

  FIG. 9 is a waveform diagram of a three-phase current when harmonic suppression is attempted by PR control using PI control for current control under the same conditions as FIG. PR control is started at 0.17 seconds. It can be seen that the current becomes unstable and diverges with the start of PR control.

  FIG. 10 is a waveform diagram of a three-phase current when harmonic suppression is attempted by PR control using dead beat control for current control under the same conditions as FIG. PR control is started at 0.17 seconds. The distortion of the current is suppressed with the start of the PR control, and there is no instability.

(Effect of this embodiment)
As described above, the PWM inverter 2 and the drive control method of the motor 3 by this control the deadbeat current control and PR whose gain is constant up to the high frequency region when driving and controlling the motor of the induction motor specification or the permanent magnet same motor specification. Current harmonic suppression by control is performed. Therefore, it is possible to suppress the unstable phenomenon of current control at the time of drive control of the motor. Thus, the region where stable operation can be performed while suppressing harmonics can be expanded to a high rotational speed. In addition, the high speed motor can be operated stably. Furthermore, stable operation can be performed even when driving a large capacity motor whose switching frequency can not be increased.

  The present invention is not limited to the embodiments described above, and can be implemented in various aspects within the scope of the claims of the present invention.

DESCRIPTION OF SYMBOLS 1 motor drive system 2 PWM inverter 3 motor 4 load 5 three-phase electric wire 20 detection unit 21 speed control unit 22 angular velocity conversion unit 23 torque control unit 24 integration unit 25 Multiplication unit 26: Coordinate conversion unit (first coordinate conversion unit) 27: Current control unit 28: Coordinate conversion unit (second coordinate conversion unit) 29: PWM unit (pulse width modulation unit)
11 ... dead beat control unit, 12 ... PR control unit (proportional resonance control unit), 13 ... addition unit 111 ... current prediction unit, 112 ... disturbance voltage estimation unit, 113 ... voltage command calculation unit

Claims (5)

An inverter for driving and controlling a motor,
Based on the motor's d, q-axis current detection value, the previous d, q-axis voltage command value, and the previous d, q-axis disturbance voltage estimated value, the next time The d and q axis current predicted values are calculated, and the d and q axes of the next time are calculated based on the d and q axis current predicted values of the calculated next time, the stator resistance and the magnetic flux of the motor, and the frequency command value. The disturbance voltage estimated value is calculated, and the calculated next time d, q axis disturbance voltage estimated value, the d, q axis current command value of the motor, and the next time d, q axis current predicted value A dead beat control unit that calculates a first d, q axis voltage command value at the next time based on
A second method of suppressing harmonics by calculation based on a discretization model of the motor from the d, q-axis current command values of the motor and the d, q-axis current detection values coordinate-converted to the synchronous coordinate system and the frequency command value A proportional resonance control unit that calculates a d, q axis voltage command value of
An adder configured to calculate the d, q-axis voltage command value of the motor by adding the second d, q-axis voltage command value to the first d, q-axis voltage command value;   The second proportional resonance control unit supplies a frequency command value that is six times the frequency command value to a calculation based on the discretized model to suppress harmonics of the frequency that is six times the second d and q axis voltage commands The inverter according to claim 1, wherein the value is calculated. The inverter according to claim 1 or 2, which drives and controls a motor of an induction motor specification,
A detection unit that detects a mechanical angular velocity of the motor based on the rotation speed of the motor;
A speed control unit that outputs a torque command value of the motor based on an external speed command value and the mechanical angular velocity;
An angular velocity conversion unit that converts the mechanical angular velocity into an electrical angular velocity;
A torque control unit that outputs the d, q-axis current command value and the frequency command value based on the torque command value and the electric angular velocity of the motor;
An integrating unit that integrates the time-series signal of the frequency command value from the torque control unit to calculate an electrical angle;
A multiplication unit that calculates a product of the frequency command value from the torque control unit and a value 1.5 times the cycle of current control;
A first coordinate conversion unit that converts a three-phase current detection value on the output side of the inverter into the d, q-axis current detection value based on the electrical angle from the integration unit;
A second coordinate transformation for converting the d, q axis voltage command value from the adding unit into a three-phase voltage command value based on the sum of the electrical angle from the integrating unit and the product value from the multiplying unit Department,
An inverter characterized by further comprising: a pulse width modulation unit for performing pulse width modulation on the basis of the three-phase voltage command value supplied from the second coordinate conversion unit and outputting a three-phase AC voltage generated to the motor. . The inverter according to claim 1 or 2, wherein a motor of a permanent magnet synchronous motor specification is driven and controlled,
A detection unit that detects a mechanical angular velocity and an electrical angle of the motor based on the rotational speed of the motor;
A speed control unit that outputs a torque command value of the motor based on an external speed command value and the mechanical angular velocity;
An angular velocity conversion unit that converts the mechanical angular velocity into a frequency command value;
A torque control unit that outputs the d, q axis current command value based on the torque command value;
A multiplication unit that calculates a product of the frequency command value from the angular velocity conversion unit and a 1.5-fold value of the period of current control;
A first coordinate conversion unit that converts a three-phase current detection value on the output side of the inverter into the d, q axis current detection value based on the electrical angle from the detection unit;
A second coordinate conversion unit for converting the d, q axis voltage command value from the addition unit into a three-phase voltage command value based on the sum of the electrical angle from the detection unit and the product value from the multiplication unit When,
An inverter characterized by further comprising: a pulse width modulation unit for performing pulse width modulation on the basis of the three-phase voltage command value supplied from the second coordinate conversion unit and outputting a three-phase AC voltage generated to the motor. . It is a drive control method of a motor by an inverter, and
Based on the motor's d, q-axis current detection value, the previous d, q-axis voltage command value, and the previous d, q-axis disturbance voltage estimated value, the next time The d and q axis current predicted values are calculated, and the d and q axes of the next time are calculated based on the d and q axis current predicted values of the calculated next time, the stator resistance and the magnetic flux of the motor, and the frequency command value. The disturbance voltage estimated value is calculated, and the calculated next time d, q axis disturbance voltage estimated value, the d, q axis current command value of the motor, and the next time d, q axis current predicted value Calculating a first d, q axis voltage command value at the next time based on
A second method of suppressing harmonics by calculation based on a discretization model of the motor from the d, q-axis current command values of the motor and the d, q-axis current detection values coordinate-converted to the synchronous coordinate system and the frequency command value Calculating the d and q axis voltage command values of
Calculating the d, q-axis voltage command value of the motor by adding the second d, q-axis voltage command value to the first d, q-axis voltage command value; Drive control method.

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