JP2019083672A - Inverter, and drive control method for motor - Google Patents

Abstract

To suppress increase in capacity of a lookup table for compensation current control and to reduce a size and a control cost of a control device, in drive control of a motor.SOLUTION: An inverter 2 that drives and controls a motor 3 comprises: a first lookup table from which a d-axis current command value depending on a torque command value and a rotational angle of the motor 3 is drawn; and a second lookup table from which a q-axis current command value depending on the torque command value and the rotational angle is drawn. The inverter 2 performs deadbeat current control on the basis of a d-axis current detection value and the d-axis current command value to calculate a d-axis voltage command value. The inverter 2 performs deadbeat current control on the basis of a q-axis current detection value and the q-axis current command value to calculate a q-axis voltage command value. On the basis of the d-axis voltage command value and the q-axis voltage command value, a voltage to be supplied to the motor is generated.SELECTED DRAWING: Figure 1

Description

  The present invention relates to control of a compensation current that suppresses space harmonic ripple of a motor (synchronous motor) driven by an inverter.

  The technology for generating the compensation current command to suppress the torque ripple derived from the space harmonics of the motor is roughly classified into the following two methods. That is, there is provided a shaft torque detection means to learn a compensation current command at the time of motor operation (Patent Document 1), and a method to feed compensation current command in a feedforward manner based on a ripple compensation look-up table prepared in advance. It is divided roughly into 2). In particular, while the latter method detects the axial torque only when creating the table, it does not require the detection of the axial torque when the motor is used.

O. Kukrer. “Discrete-time current control of voltage-fed three-phase pwm inverters.” IEEE Transactions on Power Electronics, 11: 260-269, 1996. Lothar Springob and Jachim Holth. “High-bandwidth current control for torque-ripple compensation in pm synchronous machines.” IEEE Transactions on Industrial Electronics, 45: 713-721, 1998. Hyung-Tae Moon, Myung-Joong Youn and Hyun-Soo Kim. “A discrete-time predictive current control for pmsm.” IEEE Transactions on Power Electronics, 18: 464-472, 2003. Yasuhiro Yamamoto, Koji Kodama, Tetsuo Yamada, Tadahiro Ichioka, Minoru Niwa, "Digital Current Control Method of Induction Motor by PWM Synchronous Current Sample", Journal of the Institute of Electrical Engineers of Japan, Vol. 112 (1992), No. .7, P. 613-622

JP, 2013-198221, A JP, 2011-50118, A

  Since ripples derived from space harmonics of the motor are periodically generated according to the rotation angle (position) of the motor, the look-up table for compensation also generates the compensation current command with the rotation angle of the motor as an input. Therefore, when compensating for high-order torque pulsations and torque pulsations generated during high-speed rotation, the compensation current command also has a high frequency, and the current control band of the inverter becomes a problem.

  In general, as the current control becomes higher frequency, the amplitude (gain) in the Bode diagram is attenuated and the phase delay becomes larger. For this reason, when the compensation current command becomes high frequency, the deviation between the compensation current command waveform and the actual output current waveform is enlarged, and the ripple can not be accurately compensated.

  On the other hand, in the method of Patent Document 2, information on amplitude attenuation and phase delay according to the rotational angular velocity (rotational frequency) is also learned at the time of creation of the lookup table, and compensation is performed by including in the lookup table. . This method has an advantage that accurate compensation can be performed because the lookup table is learned including not only the mechanical structure of the motor but also the delay characteristics of the inverter.

  However, compensation can only be made with an inverter having the same gain for current control and dead time as the inverter used for learning the look-up table.

  In addition, since the delay amount changing according to the rotational angular velocity is also compensated by the look-up table, there is a problem that the size (capacity) of the look-up table becomes large.

  Furthermore, in order to create a look-up table corresponding to high-speed rotation, it is necessary to have a broadband axial torque detection means capable of detecting high frequency torque pulsations.

  In view of the above circumstances, the present invention has an object to suppress increase in capacity of a look-up table for compensating current control and to achieve miniaturization of a control device and reduction of control cost in drive control of a motor. .

  Therefore, one aspect of the present invention is an inverter that drives and controls a motor, and a first look-up table from which a d-axis current command value corresponding to a torque command value and a rotation angle of the motor is derived; A second look-up table from which a q-axis current command value corresponding to the value and the rotational angle is derived, a d-axis current detection value measured at the current time, a previous d-axis voltage command value, and a previous d-axis The d-axis current predicted value at the next time is calculated from the disturbance voltage estimated value by calculation based on the discretization model of the motor, and the d-axis current predicted value at the next calculated time and the winding resistance of the motor; The d-axis disturbance voltage estimated value at the next time is calculated based on the rotational angular velocity and the permanent magnet magnetic flux, and the d-axis disturbance voltage estimated value at the calculated next time and the d-axis current at the calculated next time Predicted value and said extracted before The first control unit that calculates the d-axis voltage command value at the next time based on the d-axis current command value, the q-axis current detection value measured at the current time, the previous q-axis voltage command value, and the previous time The q-axis current predicted value at the next time is calculated from the q-axis disturbance voltage estimated value of the above by the discretization model of the motor, and the calculated q-axis current predicted value at the next time and the winding of the motor The q-axis disturbance voltage estimated value at the next time is calculated based on the line resistance, the rotational angular velocity, and the permanent magnet flux, and the calculated q-axis disturbance voltage estimated value at the next time and the calculated next time and d) calculating the q-axis voltage command value at the next time based on the q-axis current predicted value and the extracted q-axis current command value. A voltage to be supplied to the motor is generated based on the q-axis voltage command value.

  In one embodiment of the present invention, in the inverter, a product of the rotation angle provided to the first look-up table and the second look-up table, the product of the rotation angular velocity of the motor and a double value of the current control period. It further comprises an addition unit that adds

  One aspect of the present invention is a first look-up table from which a d-axis current command value corresponding to a torque command value and a rotational angle of a motor is derived, and a q-axis current corresponding to the torque command value and the rotational angle A drive control method of a motor by an inverter having a second look-up table from which a command value is derived, which is a d-axis current detection value measured at a current time, a previous d-axis voltage command value, and a previous d-axis The d-axis current predicted value at the next time is calculated from the disturbance voltage estimated value by calculation based on the discretization model of the motor, and the d-axis current predicted value at the next calculated time and the winding resistance of the motor; The d-axis disturbance voltage estimated value at the next time is estimated based on the rotational angular velocity and the permanent magnet magnetic flux, and the d-axis disturbance voltage estimated value at the calculated next time and the d-axis current at the calculated next time Predicted value and said derived Calculating the d-axis voltage command value at the next time based on the d-axis current command value, the q-axis current detection value measured at the current time, the previous q-axis voltage command value, and the previous q-axis The q-axis current predicted value at the next time is calculated from the disturbance voltage estimated value by calculation based on the discretized model of the motor, and the calculated q-axis current predicted value at the next time and the winding resistance of the motor; The q-axis disturbance voltage estimated value at the next time is calculated based on the rotational angular velocity and the permanent magnet magnetic flux, and the calculated q-axis disturbance voltage estimated value at the next time and the q-axis current at the calculated next time Calculating the q-axis voltage command value at the next time based on the predicted value and the extracted q-axis current command value; and the motor based on the d-axis voltage command value and the q-axis voltage command value And generating a voltage to be provided.

  According to the present invention described above, in drive control of a motor, it is possible to suppress an increase in the capacity of a look-up table for compensating current control and to achieve downsizing of a control device and reduction of control cost.

BRIEF DESCRIPTION OF THE DRAWINGS The block block diagram of the motor drive system provided with the inverter which is 1 aspect of this invention. The block block diagram of the compensation current control apparatus in the inverter of FIG. FIG. 3 is a block diagram of a deadbeat current control unit in the compensation current control device of FIG. 2; (A) is a Bode diagram comparing the gains (gain [dB]) of the deadbeat current controller (DB) and PI (proportional integration) current controller (PI); (b) is a deadbeat current controller (DB And Bode diagram comparing the phase delay (phase [degree]) of the PI current controller (PI). (A) is a d-axis current waveform diagram when phase delay compensation is not applied, (b) is a q-axis current waveform diagram in that case, and (c) is a torque waveform diagram in that case. (A) is a d-axis current waveform diagram when phase delay compensation is applied, (b) is a q-axis current waveform diagram in that case, and (c) is a torque waveform diagram in that case.

  Embodiments of the present invention will be described below with reference to the drawings.

  The motor drive system 1 illustrated in FIG. 1 is an inverter system to which the inverter 2 according to one aspect of the present invention is applied, and the inverter 2 and the motor 3 are connected by a three-phase electric wire 5. Coupled to the

The inverter 2 controls the drive of the motor 3 by controlling the current flowing through the three-phase wire 5 so that a torque command value T _e ^* given from the outside is generated in the shaft of the motor 3. Each functional unit of the inverter 2 will be described below.

  The rotation angle / rotation angular velocity detection unit 20 outputs the rotation angle θ of the motor 3 provided from the rotation angle sensor 6 provided in the motor 3 and the rotation angular velocity ω based on the rotation angle θ to the torque / current control unit 21. Do.

Torque / current control unit 21 receives torque command value T _e ^* from the outside, rotation angle θ and rotation angular velocity ω from rotation angle / rotation angular velocity detection unit 20, and d-axis current detection value i from coordinate conversion unit 22. _{A d-} axis current command value _id ^* and a q-axis current command value _iq ^* are output based on the _d and q-axis current detection values _iq .

The coordinate conversion unit 22 is based on the rotation angle θ from the rotation angle / rotation angular velocity detection unit 20 on the three-phase current detection values i _u , i _v , i _w provided from the current sensor 7 attached to the three-phase electric wire 5. The coordinate conversion is performed to convert the d-axis current detection value _id and the q-axis current detection value _iq .

The coordinate conversion unit 23 coordinates the d-axis voltage command value v _d ^* and the q-axis voltage command value v _q ^* supplied from the torque and current control unit 21 based on the rotation angle θ from the rotation angle and rotation angular velocity detection unit 20. It converts into three-phase voltage command value v _u ^* , v _v ^* , v _w ^* by conversion.

The PWM unit 24 generates on the basis of the three-phase voltage command values v _u ^* , v _v ^* , v _w ^* from the coordinate conversion unit 23 by turning on and off the switching elements of each phase in the PWM unit 24. Output to the motor 3 through the three-phase electric wire 5.

  Further, the torque / current control unit 21 mounts the compensation current control device 10 shown in FIG. 2 for suppressing torque ripple derived from motor space harmonics.

(Example embodiment of the compensation current control device 10)
As shown in FIG. 2, the compensation current control device 10 implements the adder 11, two-dimensional lookup tables 12d and 12q, and dead beat current controllers 13d and 13q.

  When the addition unit 11 receives the rotation angle θ and the rotation angular velocity ω from the rotation angle and rotation angular velocity detection unit 20, the product of the rotation angular velocity ω and the double value of the current control period Ts is converted to the rotation angle θ as phase delay compensation. Add to The rotation angle θ to which this product has been added is used for current control in the deadbeat current controllers 13d and 13q using the two-dimensional lookup tables 12d and 12q.

The two-dimensional lookup tables 12 d and 12 q are current command values (d (d) derived according to torque command values T _e ^* for the d axis and q axis of the synchronous coordinate system from the outside and the rotation angle θ from the adder 11. The axis current command values _id ^* and q-axis current command values _iq ^* ) are respectively learned and stored in advance. The two-dimensional lookup table 12d corresponds to the first lookup table of the present invention, and the two-dimensional lookup table 12q corresponds to the second lookup table of the present invention.

The dead beat current controllers 13d and 13q are the two-dimensional lookup tables 12d, “d-axis current detection value _id ” and “q-axis current detection value _iq ” from the current sensor 7 via the coordinate conversion unit 22; "d-axis current command value i _d ^*" from 12q, based on the "q-axis current command value i _q ^*" and "d axis voltage value v _d ^*" and ^"* q-axis voltage command value v _q" Output. The dead beat current controller 13 d corresponds to a first control unit of the present invention, and the dead beat current controller 13 q corresponds to a second control unit of the present invention.

The d-axis voltage command value v _d ^* and the q-axis voltage command value v _q ^* are supplied to the PWM unit 24 via the coordinate conversion unit 23 of FIG. At the output side of the PWM unit 24, three-phase current detection values i _u , i _v , i _w are outputted from the current sensor 7 attached to the three-phase electric wire 5. The current sensor 7 calculates three-phase current detection values i _u , i _v and i _{w according} to the current detection method of Non-Patent Document 4. Then, the three-phase current detection values i _u , i _v , i _w are converted into “d-axis current detection value _id ” and “q-axis current detection value i _q ” by the coordinate conversion unit 22, and then torque · Feedback to the current control unit 21 controls the d and q axis currents.

  The dead beat current controllers 13d and 13q predict the current value at the next sample time k + 1 at the sample time k of digital control. Then, the voltage command value calculated based on this current value is output from the time k + 1 to k + 2 to the PWM unit 24 through the coordinate conversion unit 23, so that the current at time k + 2 matches the current command value. (For example, Non-Patent Documents 1-4).

  The dead beat current controllers 13 d and 13 q implement the current prediction unit 131, the disturbance voltage estimation unit 132, and the voltage command calculation unit 133 as illustrated in FIG. 3.

In the illustrated embodiment, a voltage command value v ^* _{dq, k + 1} to be supplied to the PWM unit 24 via the coordinate conversion unit 23 is calculated from time k + 1 to time k + 2 starting at time k.

  The current prediction unit 131 is based on the “current value measured at current time k”, “previous voltage command value”, and “previous disturbance voltage estimated value” based on the discretization model of the synchronous motor (motor 3): The “current predicted value at the next time k + 1” is calculated by the calculation of equation (1).

  The disturbance voltage estimation unit 132 calculates the following equation (2) from the calculated “current predicted value at the next time k + 1” and “the winding resistance of the motor 3 and the rotational angular velocity and the permanent magnet flux”. “A disturbance voltage estimated value at the next time k + 1” is calculated.

The voltage command calculation unit 133 is extracted from the calculated “disturbed voltage estimated value at time k + 1”, the calculated “current predicted value at next time k + 1”, and the two-dimensional lookup tables 12 d and 12 q. From the “current command value at time k” corresponding to the torque command value T _e ^* and the rotation angle θ, “voltage command value at time k + 1” is calculated by the following equation (3).

  The “voltage command value at time k + 1” is output to the PWM unit 24 through the coordinate conversion unit 23 from time k + 1 to k + 2.

  As described above, “the current command value set at the current time k” is realized at time k + 2, and the current is controlled to follow the command value with a two sample cycle delay.

(Operation example of the present embodiment)
The dead beat current controller 13 of this embodiment utilizes the feature that the gain is constant up to the high frequency region, and the phase delay is constant at twice the current control period Ts at any frequency, to obtain torque ripple. Output the current command waveform to be compensated without deviation.

  FIG. 4 is a Bode diagram of the dead beat current controller (DB) and PI (proportional integration) current controller (PI) of this embodiment.

  FIG. 4A compares the gains (gain [dB]) of the dead beat current controller (DB) and the PI current controller (PI). While the gain of the PI current controller (PI) attenuates as the frequency rises, the gain of the deadbeat current controller (DB) becomes constant up to the high frequency region. Therefore, the deadbeat current controller (DB) can output a current waveform with the amplitude according to the command value even in the high frequency region.

FIG. 4B compares the phase delay (phase [degree]) of the dead beat current controller (DB) and the PI current controller (PI). The phase delay of the PI current controller changes according to the setting of the proportional / integral gain of the controller, while the phase delay of the deadbeat current controller is constant at 2 × Ts. In FIG. 4, the period Ts is set to 100 μs, and for example, the delay at a value of 1400 Hz on the horizontal axis is 1400 × 360 × 2 × 100 × 10 ⁻⁶ = 100.8 °. When the motor is rotating at the rotational angular velocity ω, if current control is performed without compensating for phase delay, the current command value generated by the lookup table is actually output from the rotation angle θ after 2 × Ts The rotational angle at this time is not θ but θ + 2 × ω × Ts. Therefore, as shown in FIG. 2, if the phase obtained by multiplying the rotational angular velocity ω by 2 × Ts is compensated, the current can be output in agreement with the respective rotational angles of the two-dimensional lookup tables 12d and 12q.

  FIG. 5 shows d-axis current waveform (a), q-axis current waveform (b) and torque waveform (c) when phase delay compensation is not applied. Further, FIG. 6 shows a d-axis current waveform (a), a q-axis current waveform (b), and a torque waveform (c) when phase delay compensation is applied. If compensation is not performed, the relationship between the current and the rotation angle deviates 2 × ω × Ts from that specified in the look-up table, so space harmonics can not be accurately compensated, and a large torque ripple is generated. When compensation is added, the relationship between the rotation angle and the current as specified in the look-up table is maintained, so that torque ripple derived from space harmonics is suppressed.

(Effect of this embodiment)
In the conventional control method (for example, Patent Document 2), the same gain as current control and dead time are applied to learning of a look-up table (torque ripple compensation current table), which depends on the characteristics of the inverter.

  On the other hand, the two-dimensional look-up tables 12d and 12q provided to the inverter 2 of the present embodiment and the drive control method of the motor 3 by this depend on only the rotation angle of the motor 3 and depend on the characteristics of the inverter 2. do not do. Therefore, if the two-dimensional look-up tables 12d and 12q are created for a motor of a new design, various inverters can be supported without being limited to a specific inverter.

  Further, in the conventional control method, the size of the look-up table is increased because the delay amount that changes according to the rotational angular velocity is also compensated by the look-up table.

  On the other hand, in the inverter 2 of this embodiment and the drive control method of the motor 3 based on this, the delay amount changing according to the rotational angular velocity is compensated not by the look-up table but by calculation of ω × 2 × Ts. Because the size of the look-up table is small. As a result, the control device of this system can be miniaturized and cost reduced.

  Furthermore, since the conventional control method learns the look-up table including the delay characteristics of the inverter according to the frequency, the table depends on the dead time of the inverter and the gain of the current control.

  On the other hand, the inverter 2 of this embodiment and the drive control method of the motor 3 based on this use the feature of dead beat current control that the phase delay is constant in two cycles of current control. Therefore, it is not necessary to adjust the phase delay compensation according to the frequency or the delay compensation according to the gain of the inverter or the current controller. Therefore, the control configuration is simplified, and the cost of the control device of the present system can be reduced. In addition, the number of adjustment steps of the controller can be reduced.

  Further, in the conventional control method, in order to create a look-up table corresponding to high speed rotation, a wide band axial torque detection means capable of detecting high frequency torque pulsation is required.

  On the other hand, the inverter 2 of this embodiment and the drive control method of the motor 3 based thereon do not require the look-up table of the rotational angular velocity. Therefore, although the rotational angular velocity of the motor at the time of look-up table generation may be low speed within the range of the shaft torque meter, high frequency torque in the high speed range which can not be detected by the shaft torque meter can be similarly compensated It is. Furthermore, since it is not necessary to create a lookup table for each rotational angular velocity, it is possible to reduce the number of test steps when creating the lookup table.

  The present invention is not limited to the embodiments described above, and can be implemented in various aspects within the scope of the claims of the present invention.

DESCRIPTION OF SYMBOLS 1 ... Motor drive system 2 ... Inverter 3 ... Motor 4 ... Load 5 ... Three-phase electric wire 10 ... Compensation current control apparatus 11 ... Addition part 12 d ... Two-dimensional look-up table (1st look-up table), 12 q ... Two-dimensional look Up table (second lookup table)
13 ... dead beat current controller, 13 d ... dead beat current controller (first control unit), 13 q ... dead beat current controller (second control unit)
131 ... current prediction unit, 132 ... disturbance voltage estimation unit, 133 ... voltage command calculation unit

Claims (3)

An inverter for driving and controlling a motor,
A first lookup table from which a d-axis current command value corresponding to a torque command value and a rotation angle of the motor is derived;
A second lookup table from which a q-axis current command value corresponding to the torque command value and the rotation angle is derived;
The d-axis current predicted value at the next time by calculation based on the discretization model of the motor from the d-axis current detection value measured at the current time, the previous d-axis voltage command value, and the previous d-axis disturbance voltage estimated value Calculating the d-axis disturbance voltage estimated value at the next time based on the calculated d-axis current predicted value at the next time, the winding resistance of the motor, the rotational angular velocity, and the permanent magnet flux, The d-axis voltage command at the next time is calculated based on the d-axis disturbance voltage estimated value at the next time calculated, the d-axis current predicted value at the calculated next time, and the d-axis current command value extracted. A first control unit that calculates a value;
From the q-axis current detection value measured at the current time, the previous q-axis voltage command value, and the previous q-axis disturbance voltage estimated value, the q-axis current prediction value at the next time by calculation based on the discretization model of the motor Calculating the q-axis disturbance voltage estimated value at the next time based on the calculated q-axis current predicted value at the next time, the winding resistance of the motor, the rotational angular velocity, and the permanent magnet magnetic flux; The q-axis voltage command at the next time based on the calculated q-axis disturbance voltage estimated value at the next time, the q-axis current predicted value at the calculated next time, and the extracted q-axis current command value A second control unit that calculates a value;
An inverter characterized in that a voltage to be supplied to the motor is generated based on the d-axis voltage command value and the q-axis voltage command value.   It further comprises an addition unit for adding the product of the rotational angular velocity of the motor and a double value of the period of current control to the rotational angle provided to the first look-up table and the second look-up table. The inverter according to claim 1, characterized in that: A first lookup table from which a d-axis current command value corresponding to a torque command value and a rotation angle of a motor is derived; and a second look-up table from which a q-axis current command value corresponding to the torque command value and the rotation angle is derived Drive control method of a motor by an inverter having a look-up table of
The d-axis current predicted value at the next time by calculation based on the discretization model of the motor from the d-axis current detection value measured at the current time, the previous d-axis voltage command value, and the previous d-axis disturbance voltage estimated value Calculating the d-axis disturbance voltage estimated value at the next time based on the calculated d-axis current predicted value at the next time, the winding resistance of the motor, the rotational angular velocity, and the permanent magnet flux, The d-axis voltage command at the next time is calculated based on the d-axis disturbance voltage estimated value at the next time calculated, the d-axis current predicted value at the calculated next time, and the d-axis current command value extracted. Process of calculating the value,
From the q-axis current detection value measured at the current time, the previous q-axis voltage command value, and the previous q-axis disturbance voltage estimated value, the q-axis current prediction value at the next time by calculation based on the discretization model of the motor Calculating the q-axis disturbance voltage estimated value at the next time based on the calculated q-axis current predicted value at the next time, the winding resistance of the motor, the rotational angular velocity, and the permanent magnet magnetic flux; The q-axis voltage command at the next time based on the calculated q-axis disturbance voltage estimated value at the next time, the q-axis current predicted value at the calculated next time, and the extracted q-axis current command value Process of calculating the value,
Generating a voltage to be supplied to the motor based on the d-axis voltage command value and the q-axis voltage command value.

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