JPH11206171A - Brushless dc motor and its drive controlling method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a brushless DC motor with a small quantity of its vibration and noise and with its excellent power efficiency, by performing smoothly the phase changeover of the windings of its stator in the state of suppressing to low levels the power losses of its driving transistors. SOLUTION: A second driving transistor group 5b is so controlled that an operational voltage of a first transistor group 5a is made equal to a reference voltage E1 of a first reference-voltage generating means 9. Also, the DC output voltage of a voltage converting means 4 is so controlled that the operational voltage of the second driving transistor group 5b is made equal to a reference voltage E2 of a second reference-voltage generating means 10. Further, the operational voltages of the driving transistor groups are varied responding to the positional signals of the plural phases of a positional-signal synthesizing means 40 and to the shaping signals of buffer amplifiers 31, 32, 33 of Hall elements 1, 2, 3.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a brushless DC motor, and more particularly to a brushless DC motor that efficiently uses electric power supplied from a power supply.

[0002]

2. Description of the Related Art A brushless DC motor has no mechanical contacts and therefore has a longer life and less electrical noise than a DC motor with a brush. Due to this advantage, brushless DC motors are widely used in industrial equipment and video / audio equipment that require high reliability in recent years.

In a conventional brushless DC motor,
A voltage supplied from a DC power supply having a constant output voltage is variably controlled using a voltage control transistor or the like, and a voltage corresponding to, for example, a rotation speed is supplied to a motor. Therefore,
The effective voltage used for driving the motor is always lower than the voltage of the DC power supply, and the difference between the DC power supply voltage and the voltage actually supplied to the motor is almost equal to the collector loss (heat loss) of the voltage control transistor. Power efficiency was reduced.

In order to improve the power efficiency of the brushless DC motor, several methods have been proposed for reducing the collector loss of the voltage control transistor by switching the voltage control transistor.

In one example, a first current path forming one end of a DC power supply and a current feed terminal of a stator winding is formed.
Is controlled in accordance with the current command and the position signal. Then, the second drive transistor group forming the current path between the other end of the DC power supply and the current supply terminal is adjusted so that the minimum operating voltage of the first drive transistor group becomes equal to a predetermined reference voltage. Control. Further, the on / off control of the switching transistor for voltage control is performed so that the operating voltage of the second drive transistor group becomes equal to a predetermined reference voltage. In this way, the voltage supplied to the motor is controlled. By performing such control, the power efficiency of the motor is greatly improved (see, for example, Japanese Patent Application Laid-Open No. 58-198189).

In the above-described configuration, since the position switching is performed by applying the position signal to the base input of the differential transistor having the emitter commonly connected, the driving current of the stator winding can be switched stably. However, the drive current flowing through the stator winding has a rectangular waveform with a conduction width of approximately 120 degrees in electrical angle, and is sharply turned on and off. For this reason, vibration and noise are easily generated.

In order to smoothly switch the phase of the stator winding, when performing current switching from a certain phase to the next phase, there is a period in which current is supplied to two phases at the same time, that is, a method of performing overlap driving. (For example, see JP-A-62-22).
1894).

[0008]

In the above-mentioned prior art, the first and second drive transistors are overlap-driven to improve the power efficiency (to reduce power loss). If the operating voltage of the transistor is reduced by reducing the remaining voltage between the emitter and the collector of the driving transistor group, the current switching is not performed smoothly, and the driving current waveform is distorted. Vibration and noise are generated (the details of this problem will be described later with reference to FIG. 7 in an embodiment of the invention). In other words, it is difficult to achieve both smooth drive current phase switching and reduction of power loss.

SUMMARY OF THE INVENTION The present invention has been made in view of the above-mentioned conventional problems, and has as its object to provide a brushless DC motor having less vibration and noise and having excellent power efficiency.

[0010]

According to a first aspect of the present invention, there is provided a brushless DC motor for detecting a rotor having a plurality of magnetic poles, a plurality of phase stator windings, and a rotational position of the rotor. A plurality of Hall elements, a position signal synthesizing means for generating a plurality of phase position signals from outputs of the plurality of Hall elements, a voltage converting means for obtaining a variable output DC voltage from a DC power supply, and an output of the voltage converting means. A first drive transistor group including a plurality of transistors forming a current path between one of the terminal pairs and a power supply terminal of each phase of the stator winding, and instructs a current supply to the stator winding. First distribution control means for distributing and controlling the current flowing through the first drive transistor group in accordance with the command signal and the output of the position signal synthesizing means; the other of the output terminal pair of the voltage conversion means and the stator winding; Of each phase of the line A second driving transistor group including a plurality of transistors forming a current path between the second driving transistor group and the first driving transistor group according to an output of the position signal synthesizing means; Second distribution control means for distributing and controlling the current flowing through the second drive transistor group so as to match the voltage, and so that the minimum operating voltage of the second drive transistor group matches the second reference voltage. Voltage control means for controlling an output voltage of the voltage conversion means, wherein the first and second reference voltages are:
In the first and second drive transistor groups, current switching is performed from one phase of the stator winding to the next phase, and current flows only in one phase in a phase switching energized state in which current flows simultaneously in two phases. It is configured so as to be larger than the one-phase energized state and to change its size in accordance with the rotation speed of the rotor.

With such a configuration, the plurality of driving transistors constituting the first driving transistor group and the second driving transistor group correspond to each other during the phase switching period in which the driving current is switched. The operating voltage between the emitter and the collector of the driving transistor is switched higher according to the multi-phase position signal output from the position signal synthesizing means and the rotation speed of the rotor to prevent saturation of the driving transistor, while the driving current is reduced. In the one-phase energizing period in which the phase switching is completed, the operating voltage between the emitter and the collector of the driving transistor that is supplying the driving current can be set as low as possible.

Therefore, power loss in the drive transistor can be reduced during the one-phase energizing period. On the other hand, in the phase switching period, a sufficient operating voltage is applied to the drive transistor, so that phase switching of the drive current can be smoothly performed. Done,
Motor vibration and noise are significantly reduced. In this way, it is possible to realize a brushless DC motor with low vibration and noise and excellent power efficiency.

According to a second aspect of the present invention, in the brushless DC motor according to the first aspect, the first and second reference voltages are each equal to one of the stator windings in the first and second drive transistor groups. The current is switched from one phase to the next phase, and the phase switching energized state in which two phases flow simultaneously is larger than the one-phase energized state in which current flows only in one phase, and the magnitude thereof is substantially proportional to the rotation speed of the rotor. Then, it was configured to increase continuously.

[0014] This makes it possible to reduce the amount of ripple caused by the high-speed rotation of the motor and to perform smoother phase switching.

According to a third aspect of the present invention, in the brushless DC motor according to the first aspect, the first and second reference voltages are continuously changed, and the current flowing in the two phases is substantially reduced in the phase switching energized state. The triangular waveform was changed so that the point at which it became equal was the vertex.

In this configuration, the first and second reference voltages, which are switched higher in the phase switching energized state, are changed in a symmetrical triangular waveform, so that the effect of the back electromotive force induced in the stator winding is reduced. As a result, the phase switching operation of the drive current can be performed more smoothly, so that the vibration and noise at the time of driving the brushless DC motor can be further reduced.

According to a fourth aspect of the present invention, in the brushless DC motor according to the first aspect, the first and second reference voltages change in response to a command signal for commanding a current supply to the stator winding. The configuration was adopted.

Since the first and second reference voltages are changed in accordance with the magnitude of the current supply to the stator winding, the current is always smoothly switched regardless of the change in the drive current, and the waveform of the drive current is changed. The motor can be driven in a state where no distortion occurs. As a result, the effect of being excellent in power efficiency and reducing vibration and noise is further enhanced.

According to a fifth aspect of the present invention, in the brushless DC motor according to the first aspect, the position signal synthesizing means includes a plurality of buffer amplifiers for amplifying respective outputs of the plurality of Hall elements, and the plurality of buffer amplifiers. And a plurality of subtraction circuits for generating a difference between the outputs of
The second reference voltage is generated from a multi-phase position signal output from the position signal synthesizing unit and a shaped signal obtained by shaping the output of the buffer amplifier.

Thus, the first and second reference voltages can be generated with a simple circuit configuration.

According to a sixth aspect of the present invention, there is provided a brushless DC motor comprising: a rotor having a plurality of magnetic poles; a plurality of stator windings; a plurality of Hall elements for detecting a rotational position of the rotor; Position signal synthesizing means for generating a plurality of phase position signals from outputs of the plurality of Hall elements, voltage converting means for obtaining a variable output DC voltage from a DC power supply, and one of the output terminal pairs of the voltage converting means and the fixed A first drive transistor group including a plurality of transistors forming a current path between a power supply terminal of each phase of the stator winding, a command signal for commanding current supply to the stator winding, and the position signal synthesis First distribution control means for distributing and controlling the current flowing through the first drive transistor group in accordance with the output of the means, and the other of the output terminal pair of the voltage conversion means and the power supply terminal of each phase of the stator winding Between A second driving transistor group including a plurality of transistors forming a path, and a minimum operating voltage of the first driving transistor group according to an output of the position signal synthesizing means so as to match a first reference voltage. Second distribution control means for distributing and controlling the current flowing through the second drive transistor group; and an output voltage of the voltage conversion means such that a minimum operating voltage of the second drive transistor group matches a second reference voltage. Voltage control means for controlling the
The first and second reference voltages are the first and second reference voltages.
Of the stator windings in the drive transistor group of
When the current is switched from one phase to the next phase and the phase switching energized state in which two phases flow simultaneously is larger than the one-phase energized state in which current flows in only one phase, and at least the rotor is rotated at a high speed, The magnitudes of the first reference voltage and the second reference voltage are changed so that the waveform of the output voltage of the voltage conversion means becomes substantially flat.

Thus, even at the time of high-speed rotation, the phase switching operation of the drive current is performed smoothly without waveform distortion, so that the motor drive with very little vibration and noise is realized.

According to a seventh aspect of the present invention, in the brushless DC motor according to the sixth aspect, the first and second reference voltages at the time of high-speed rotation are set to be higher than the level at the time of low-speed rotation. did.

With this configuration, even during high-speed rotation,
The ripple of the output voltage of the voltage conversion means (the operating voltage of the transistor group) is extremely suppressed, and a flat waveform is obtained.

According to an eighth aspect of the present invention, in the brushless DC motor according to the sixth aspect, the first and second reference voltages continuously change, and the current flowing in the two phases in the phase switching energized state is reduced. It is configured to change into a triangular wave shape with the time point at which they are substantially equal to each other.

By optimizing the level of the triangular reference voltage, the effect of the back electromotive force generated in the stator winding can be reduced, and a smooth phase switching can be performed.

According to a ninth aspect of the present invention, in the invention of the brushless DC motor according to the sixth aspect, the first and second brushless DC motors are provided.
Is changed in accordance with a command signal for commanding a current supply to the stator winding.

Thus, the first and second reference voltages are changed according to the magnitude of the current supply to the stator winding.
Irrespective of the change in the drive current, the current is always smoothly switched, and the motor can be driven in a state where the waveform of the drive current is not distorted.

According to a tenth aspect of the present invention, in the brushless DC motor according to the sixth aspect, the position signal synthesizing means includes a plurality of buffer amplifiers for amplifying respective outputs of the plurality of Hall elements, and a plurality of the buffer amplifiers. And a plurality of subtraction circuits for generating a difference between the outputs of the buffer amplifiers. The first and second reference voltages are used to calculate a multi-phase position signal output by the position signal synthesizing means and an output of the buffer amplifier. It is configured to be generated from a waveform-shaped shaped signal.

Thus, the first circuit can be constructed with a simple circuit configuration.
And a second reference voltage can be generated.

[0031] In the drive control method for a brushless DC motor according to the eleventh aspect, a plurality of phase position signals are generated based on a signal indicating the rotational position of the rotor output from the plurality of Hall elements, and the position signal is generated. To generate first and second reference voltages whose voltage values increase in a phase switching state in which current flows simultaneously in the two phases of the stator winding as compared with the other states. 2 reference voltages
And the non-inverting terminal of the second operational amplifier,
And controlling the voltage of the power supply terminal based on the first and second reference voltages by including a power supply terminal for supplying power to the stator winding in a negative feedback loop of the second operational amplifier, whereby: A desired operating voltage in the first and second drive transistor groups that supplies current to the stator winding via the power supply terminal is ensured.

As a result, a necessary (desired) operating voltage of the drive transistor group is secured at the phase switching timing of the stator winding, smooth phase switching can be performed, and power loss can be reduced.

According to a twelfth aspect of the present invention, in the drive control method for a brushless DC motor according to the eleventh aspect, when the first and second reference voltages are generated, the voltage value is changed according to the rotation speed of the motor. I tried to make it.

Thus, it is possible to prevent a decrease in the current amplification factor of the transistor even during high-speed rotation.

[0035]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, a specific embodiment of a brushless DC motor according to the present invention will be described with reference to the drawings.

FIG. 1 shows a circuit configuration of a brushless DC motor according to an embodiment of the present invention.

In FIG. 1, reference numeral 27 denotes a permanent magnet rotor having a plurality of magnetic poles, and 11, 12, and 13 denote stator windings provided with a predetermined gap between the rotor and the rotor.
a is a first drive transistor group, 5b is a second drive transistor group, 20 is a DC power supply, and 4 is a switching control type voltage conversion circuit for obtaining a variable output DC voltage from the DC power supply 20.

The first driving transistor group 5a includes three N
PN-type drive transistors 21, 22, 23,
Each of the drive transistors 21, 22, 23 is interposed in a current path between the current supply terminals A, B, C of the stator windings 11, 12, 13 and the negative terminal (GND) of the voltage conversion circuit 4, respectively. ing. The second drive transistor group 5b has 3
PNP-type drive transistors 24, 25, and 26. Each of the drive transistors 24, 25, and 26 has a positive terminal of the voltage conversion circuit 4 and a stator winding 11, 12,
13 are provided in a current path between the current supply terminals A, B, and C.

1, 3, and 3 are three Hall elements disposed so as to form a predetermined gap between the rotor 27 and the rotor 1, 2, and 3. Reference numeral 30 denotes three Hall elements 1
DC power supplies 2 and 3 and buffer amplifiers 31, 32 and 33 output outputs H1, H2 and H3 proportional to the differential outputs of the Hall elements 1, 2 and 3, respectively. 41, 42 and 43 are subtraction circuits.

The subtraction circuit 41 receives the output H1 of the buffer amplifier 31 and the output H3 of the buffer amplifier 33,
A current Ip1 proportional to the difference (H1-H3) is output. The subtraction circuit 42 outputs the output H2 of the buffer amplifier 32.
And the output H1 of the buffer amplifier 31, and outputs a current Ip2 proportional to the difference (H2−H1). The subtraction circuit 43 receives the output H3 of the buffer amplifier 33 and the output H2 of the buffer amplifier 32, and calculates the difference (H3
-H2) is output as a current Ip3. Each output current Ip1, Ip2, Ip3 is a three-phase position signal. Buffer amplifiers 31, 32, 33 and subtraction circuits 41, 42, 4
3 together form a position signal synthesizing circuit 40.

Each of the subtraction circuits 41, 42 and 43 has 3
It is configured to obtain two output currents. That is, the current signals Ip1, Ip2, Ip3, the current signals Ip1 ', Ip2', Ip3 'having the same polarity as these current signals,
And current signals Ip1 ", Ip2", Ip with inverted polarities
3 "output currents are obtained.

The current signals Ip1, Ip2, Ip
3 corresponds to a three-phase position signal, and diodes 54, 55,
56, only the negative side is taken out and currents g, i, e
And is input to the first distribution circuit 6a. The branched output currents Ip1, Ip2, Ip3 are
Only the positive polarity side is taken out by 1, 52 and 53 to become currents d, f and h, which are input to the second distribution circuit 6b.

The bases of the driving transistors 21, 22, 23 are supplied with the three-phase current signals g ', i', e 'generated by the first distribution circuit 6a, respectively. 23 is controlled. Similarly, three-phase current signals d ', f', and h 'generated by the second distribution circuit 6b are supplied to the bases of the driving transistors 24, 25, and 26, respectively. 26 is controlled. A current in a flowing direction is applied to the bases of the NPN transistors 21, 22, 23, and the PNP transistors 24, 25,
A current in a drawing direction is applied to the base 26.

In FIG. 1, reference numeral 57 denotes a current detection resistor, which converts the current flowing through the three-phase stator windings 11, 12, 13 into a voltage. 36 is a first comparison control circuit, which is a command terminal 50
Is compared with the voltage obtained at the current detection resistor 57. The resulting control signal CL is the first
And the input position signals g,
The current signals g ′, i ′, and e ′ are generated by controlling the magnitudes of i and e. These current signals g ′, i ′, and e ′ control the currents of the drive transistors 21, 22, and 23, so that the magnitudes of the currents supplied to the three-phase stator windings 11, 12, and 13 are controlled. Is done.

In FIG. 1, reference numeral 7 denotes driving transistors 21, 22, 23 constituting the first driving transistor group 5a.
A first operating voltage detection circuit for detecting the minimum operating voltage L of
9 is a first circuit for supplying a reference voltage to the second comparison control circuit 34;
Reference voltage generating circuit 37, buffer amplifiers 31, 3
This is a shaping circuit that shapes the waveforms of the outputs H1, H2, and H3 of the second and the third and outputs the shaped signals D1, D2, and D3.

The first reference voltage generation circuit 9 includes a subtraction circuit 4
Output currents Ip1 ", Ip2", Ip3 "of one of each of the three output currents 1, 42, 43
And a shaping signal D1, D2, D3 of the shaping circuit 37 and a switching signal input to the command terminal 60 to form a reference voltage E1.

The second comparison control circuit 34 compares the first reference voltage E1 generated by the first reference voltage generation circuit 9 with the minimum operation voltage L output by the first operation voltage detection circuit 7. As a result, the obtained control signal CU is supplied to the second distribution circuit 6b, and the current signals d ', f', h 'are generated by controlling the magnitudes of the input position signals d, f, h. . These current signals d ', f', h 'are
By controlling the currents of 4, 25, and 26, the magnitude of the current supplied to the three-phase stator windings 11, 12, and 13 is controlled.

In FIG. 1, reference numeral 8 denotes a second driving transistor group 5
b is a second operating voltage detection circuit that detects the minimum operating voltage U of the driving transistors 24, 25, and 26 that constitutes b.
10 is the second voltage control circuit 35 for the voltage conversion circuit 4
Is a second reference voltage generating circuit that supplies the reference voltage E2 of FIG. The second reference voltage generation circuit 10 includes subtraction circuits 41 and 4
2 of each of three output currents of 43
A reference voltage E2 is generated from each of the output currents Ip1 ', Ip2', Ip3 ', the shaping signals D1, D2, D3 of the shaping circuit 37 and the switching signal input to the command terminal 60.

The voltage control circuit 35 compares the reference voltage E2 generated by the second reference voltage generation circuit 10 with the minimum operation voltage U obtained by the second operation voltage detection circuit 8, and outputs a control signal C
S is output to the voltage conversion circuit 4. The voltage conversion circuit 4 is connected between the positive terminal of the DC power supply 20 and the stator windings 11, 12, 13
And is configured to control the output voltage VM of the voltage conversion circuit 4 in accordance with the control signal CS of the voltage control circuit 35.

The voltage conversion circuit 4 includes a power supply control switching transistor 10 inserted in series in a power supply path from the positive terminal of the DC power supply 20 to the stator windings 11, 12, and 13.
1, a switching control circuit 100 for controlling the on / off of the switching transistor 101 based on a control signal CS from the voltage control circuit 35, and a freewheeling diode 105.
, The inductance coil 106 and the smoothing capacitor 1
07.

The voltage control circuit 35 generates a control signal CS corresponding to the minimum operation voltage U obtained by the second operation voltage detection circuit 8. The switching control circuit 100 controls ON / OFF of the switching transistor 101 by a pulse signal corresponding to the control signal CS. As a result, the voltage conversion circuit 4 converts the voltage VS of the DC power supply 20 into an output voltage VM, outputs the converted voltage, and supplies the output voltage VM to the second drive transistor group 5b. The switching control circuit 100
May be configured using a known circuit such as a triangular wave generating circuit that generates a triangular wave voltage signal of 200 kHz and a comparator that compares the control signal CS of the voltage control circuit 35 with the triangular wave voltage signal.

According to such a configuration, the first driving transistor group 5a and the second driving transistor group 5b are respectively distributed and controlled by the position signal output from the position signal synthesizing circuit 40, and the stator windings 11 to 11 are controlled. 13 are sequentially switched, and the rotor 27 is rotationally driven.

At this time, the comparison control circuit (operational amplifier) 3
4, the first reference voltage E1 generated by the first reference voltage generation circuit 9 is input to the non-inverting terminal of the comparison control circuit 3.
4, a negative feedback control loop via the second distribution circuit 6b, the second drive transistor group 5b, and the first operating voltage detection circuit 7 is formed, and the non-inverting terminal and the inverting terminal of the comparison control circuit 34 Is virtually grounded, the operating voltage between the emitter and the collector of the first driving transistor group 5a (that is, the collector voltage of each of the transistors 21 to 23) becomes the first.
The current flowing through the second driving transistor group is controlled so as to be equal to the reference voltage E1 output from the reference voltage generation circuit 9 of FIG.

Similarly, a voltage control circuit (operational amplifier) 35
The second reference voltage E2 is supplied to the non-inverting terminal of the voltage control circuit 35, the output terminal of the voltage control circuit 35, the voltage conversion circuit 4,
Drive transistor group 5b, second operating voltage detection circuit 8
And a non-inverting terminal and an inverting terminal of the voltage control circuit 35 are virtually grounded. Therefore, the operating voltage between the emitter and the collector of the second driving transistor group 5b (that is, the driving transistor The DC output voltage of the voltage conversion circuit 4 is controlled so that the collector voltages (24 to 26) become equal to the reference voltage E2 output from the second reference voltage generation circuit 10.

The operating voltage between the emitter and the collector of the first driving transistor group 5a and the second driving transistor group 5b is converted into the multi-phase position signal output from the position signal synthesizing circuit 40 and the rotation speed of the rotor 27. Vary accordingly.
That is, in each of the plurality of drive transistors constituting the first drive transistor group 5a and the second drive transistor group 5b, the emitter of the corresponding drive transistor is provided during the "phase switching period" in which the drive current is switched. The operating voltage between the collectors is switched to a higher value in accordance with the multi-phase position signals output from the position signal synthesizing circuit 40 and the rotation speed of the rotor, so that the operating voltage shortage of the driving transistor groups 5a and 5b due to the influence of the back electromotive force is reduced. Thus, a smooth drive current phase switching is realized.

On the other hand, in the “one-phase energizing period” in which the phase switching of the drive current is completed, the operating voltage between the emitter and the collector of the drive transistor through which the drive current is flowing is set as low as possible, and the power of the drive transistor is reduced. Keep losses low. In this way, the configuration of FIG. 1 realizes a brushless DC motor that is excellent in power efficiency and has less vibration and noise by performing smooth phase switching while suppressing power loss in overlap driving.

Next, a specific circuit example of the first operating voltage detection circuit 7 in FIG. 1 will be described with reference to FIG.

In FIG. 3, reference numerals 81, 82, 83, and 84 denote diodes, whose anode terminals are connected in common, and the cathode terminals of the diodes 81, 82, and 83 have three-phase stator windings 11, 12, respectively. Thirteen current supply terminals A, B, and C. A resistor 85 has one end grounded and the other end connected to the cathode terminal of the diode 84. A constant current source 86 supplies a constant current to the anode common terminals of the diodes 81, 82, 83, 84. 87 is an output terminal of the first operating voltage detection circuit 7. When the diode having the lowest cathode potential among the three diodes 81, 82, and 83 connected to the current supply terminals A, B, and C is turned on, the potential of the anode common terminal of the diode is changed to the forward direction of the diode in the on state. It becomes higher than the cathode potential by the voltage. The output current of the constant current source 86 is also supplied to the resistor 85 via the diode 84. Therefore, a voltage lower than the potential of the anode common terminal by the forward voltage of the diode is generated in the resistor 85. Therefore, the three-phase stator windings 11,
The minimum operating voltage L, which is the minimum voltage among the current supply terminals A, B, and C of the power supply terminals 12 and 13, is output from the output terminal 87 of the first operating voltage detection circuit 7.

Next, FIG. 4 shows a specific circuit example of the second operating voltage detection circuit 8 in FIG. In FIG.
Diodes 91, 92 and 93 have their cathode terminals connected in common. Diodes 91 and 9
The anode terminals 2 and 93 each have a three-phase stator winding 1
1, 12, and 13 are connected to current supply terminals A, B, and C, respectively. Reference numeral 94 denotes a PNP type transistor. The emitter is connected to the output voltage VM of the switching control type voltage conversion circuit 4 via a resistor 96, and the collector is grounded via a resistor 95.

The base of the transistor 94 is a diode 9
1, 92 and 93 are connected to the commonly connected cathode terminals. 97 is a constant current source, and diodes 91 and 9
A constant current is drawn from the cathode common terminals 2 and 93 and the base of the transistor 94. Reference numeral 98 denotes an output terminal of the second operating voltage detection circuit 8.

When only the diode having the highest anode potential among the three diodes 91, 92, 93 connected to the current supply terminals A, B, C is turned on, the cathode common terminals of the diodes 91, 92, 93 are turned on. Is lower than the anode potential by the forward voltage of the diode in the ON state. The base of the transistor 94 is a diode 91,
Since the voltage between the emitter and the base of the transistor 94 is substantially equal to the diode forward voltage in the ON state, the emitter potential VE of the transistor 94 is equal to that of the current supply terminals A, B and C. It is almost equal to the highest potential.

Since one end of the resistor 96 is connected to the output potential VM of the voltage conversion circuit 4, the potential difference (V
M-VE) flows, and a current substantially equal to this current flows through the collector of the transistor 94. Therefore, if the resistance values of the resistor 95 and the resistor 96 are selected to be equal, the resistor 9
5 generate the same (VM-VE) potential difference as the both ends of the resistor 96. Therefore, the difference between the output voltage VM of the voltage conversion circuit 4 and the highest potential VE among the current supply terminals A, B, and C of the three-phase stator windings 11, 12, and 13 (V
M-VE), that is, the minimum operating voltage U having the smallest difference from the output voltage VM of the voltage conversion circuit 4 among the current supply terminals A, B, and C is output from the output terminal 98 of the second operating voltage detection circuit 8. You.

In FIG. 1, the first distribution circuit 6a is composed of one type of multiplier, and inputs the position signals g,
It outputs current signals g ′, i ′, and e ′ in which i and e are changed in magnitude according to the control signal CL of the first comparison control circuit 36. These three-phase current signals g ′, i ′, and e ′ are applied to the bases of the drive transistors 21, 22, and 23 constituting the first drive transistor group 5 a, and the currents of the drive transistors 21, 22, and 23 are provided. Control. The first comparison control circuit 36 compares the command signal input to the command terminal 50 with the voltage obtained by the current detection resistor 57, and supplies the obtained control signal CL to the first distribution circuit 6a. Thus, the magnitude of the current supplied to the three-phase stator windings 11, 12, 13 is controlled in accordance with the command signal input to the command terminal 50.

In FIG. 1, the second distribution circuit 6b is also composed of one type of multiplier. The position signals d, f, and h input to the second distribution circuit 6b are
The magnitude is changed according to the control signal CU which is the output of
The current signals d ', f', h 'are output. The three-phase current signals d ′, f ′, h ′ are supplied to the second drive transistor group 5 b
Are provided to the respective bases of the driving transistors 24, 25, and 26 to control the currents of the driving transistors 24, 25, and 26.

The second comparison control circuit 34 compares the reference voltage E1 generated by the first reference voltage generation circuit 9 with the minimum operation voltage L obtained by the first operation voltage detection circuit 7, and based on the result, A certain control signal CU is output to the second distribution circuit 6b. The current signal d 'output from the second distribution circuit 6b,
f 'and h' are input to the bases of the driving transistors 24, 25 and 26, and control the output current of the second driving transistor group 5b. As a result, the three-phase stator windings 11, 1
Control is performed so that the minimum voltages of the current supply terminals A, B, and C of the second and the thirteenth become equal to the reference voltage E1.

Similarly, the voltage control circuit 35 compares the reference voltage E2 generated by the second reference voltage generation circuit 10 with the minimum operation voltage U obtained by the second operation voltage detection circuit 8, and The resulting control signal CS is output to the switching control circuit 100 of the voltage conversion circuit 4. The switching control circuit 100 controls the switching transistor 101 to adjust the output voltage VS of the DC power supply 20,
M is output to the second drive transistor group 5b. Therefore, the output voltage VM and the three-phase stator windings 11, 12,.
The control is performed so that the voltage difference between the thirteen current supply terminals A, B, and C becomes equal to the second reference voltage E2. That is, the second
Drive transistor 2 forming transistor group 5b
Control is performed so that the minimum operating voltage between the emitters and the collectors of 4, 25 and 26 becomes equal to the reference voltage E2.

FIG. 2 shows signal waveforms of various parts relating to the operation of the position signal synthesizing circuit 40 when the above-described brushless DC motor is in a steady rotation state. Fig. 2 (1)
2 shows the back electromotive force waveforms a, b, and c induced in the stator windings 11, 12, and 13. FIG.
The output waveforms H1, H2, and H3 after the differential outputs of a few are amplified by the buffer amplifiers 31, 32, and 33 are shown. The output waveforms H1, H2, and H3 are out of phase by 120 degrees. The output waveforms H1, H2, and H3 are advanced by 30 degrees with respect to the respective back electromotive force waveforms a, b, and c.

FIG. 2C shows subtraction circuits 41, 42, and 43.
Output current waveforms Ip1, Ip2, and Ip3, and are three-phase position signals. FIG. 2D shows the subtraction circuits 41 and 42,
43 show current waveforms d, f, and h in which the negative sides of the output currents Ip1 ', Ip2', and Ip3 'are cut by diodes 51, 52, and 53, and FIG.
current signals Ip1 ″, Ip in which the polarities of p2 and Ip3 are inverted.
The positive side of 2 ″, Ip3 ″ is connected to diodes 54, 55, 5
6 shows current waveforms g, i, and e cut at 6. FIG.
(6), (7) and (8) are buffer amplifiers 31 and 3
2 shows shaping signals D1, D2, and D3 obtained by shaping the waveforms of outputs H1, H2, and H3 of H2 and H3, respectively. Shaping signals D1, D
2, the rising edges of D3 are buffer amplifiers 31, 3
2, 33 correspond to the rising zero-cross points of the outputs H1, H2, H3, and the falling edges of the shaping signals D1, D2, D3 correspond to the falling zero-cross points of the outputs H1, H2, H3. ing.

FIG. 5 shows signal waveforms at various parts when the brushless DC motor of the present invention is driven by the signals obtained by the above signal processing. For comparison, FIGS. 6 and 7 show similar signal waveforms in a conventional brushless DC motor.

FIG. 6 is used to explain the following problem in the conventional brushless DC motor. When the reference voltages E1 and E2 are set sufficiently high, the stator winding 1
The drive currents Ia, Ib, and Ic flowing through 1, 12, and 13 are trapezoidal currents that change smoothly during the phase switching period without distortion. However, on the other hand, the driving transistor 2
1, 22, 23 and drive transistors 24, 25, 26
Therefore, the operating voltage between the emitter and the collector is set to be large, so that the power loss in each driving transistor increases.

FIG. 7 is used to explain the following problem in the conventional brushless DC motor. When the reference voltages E1 and E2 are set to sufficiently small values E1 'and E2' to suppress the power loss, the driving transistor operates in the saturation region, and the operating voltage between the emitter and the collector becomes insufficient. Stator winding 11, 1
The drive currents Ia, Ib, Ic flowing through the second and the third 13 cause distortion during the phase switching period.

FIG. 6 shows the signal waveforms of the respective components when the reference voltage E1 generated by the first reference voltage generation circuit 9 and the reference voltage E2 generated by the second reference voltage generation circuit 10 are set to sufficiently large constant voltages. Is shown.

In FIG. 6, a, b, and c are stator windings 1
V, V, V
B and VC are voltage waveforms at the current supply terminals A, B, and C of the three-phase stator windings 11, 12, and 13, respectively. Also, g ',
i ′ and e ′ are driving transistors 21, 22, 22 constituting the first driving transistor group 5 a from the first distribution circuit 6 a.
23 is a trapezoidal wave-shaped three-phase current signal flowing into the base of 23, and d ′, f ′, h ′ are the second distribution signals from the bases of the driving transistors 24, 25, 26 constituting the second driving transistor group 5b. This is a trapezoidal waveform three-phase current signal drawn to the circuit 6b. Ia, Ib, and Ic are trapezoidal drive currents that are supplied to the three-phase stator windings 11, 12, and 13, respectively.

Since the driving transistors 21, 22, 23 and the driving transistors 24, 25, 26 use transistors having uniform electric characteristics, when the magnitudes of the reference voltages E1 and E2 are set sufficiently large, the driving transistors 21 , 22, 23 and drive transistors 24, 2
Since the operating voltage between the emitter and the collector of each of the transistors 5 and 26 is sufficiently large, the current amplification factor (hf
e) is sufficiently large with little variation. Therefore, 3
Drive currents Ia, Ia for phase stator windings 11, 12, 13
b and Ic are trapezoidal three-phase current signals g ′, i ′, input to the bases of the driving transistors 21, 22, 23, respectively.
e ′ and trapezoidal three-phase current signals d ′, f ′, h ′ input to the bases of the driving transistors 24, 25, 26
Can be equally amplified. As a result, accurate trapezoidal drive currents Ia, Ib, and Ic without distortion can be supplied to the three-phase stator windings 11, 12, and 13.

In the period (1) shown in FIG. 6, the driving transistor 24 is turned on because the current signal d 'drawn from the base is large. On the other hand, since the current signals f 'and h' drawn from the base are zero, the drive transistors 25 and 26 are turned off. Further, since the current signal g ′ flowing into the base is zero, the drive transistor 21 is turned off. In this way, simultaneous ON of the drive transistors 24 and 21 connected in cascade is prevented.

The ON state of the drive transistor 22 in which the current signal i ′ flowing into the base gradually decreases gradually decreases, and the ON state of the drive transistor 23 in which the current signal e ′ flows into the base gradually increases gradually. . And a pair of driving transistors 22 and 23
Is kept constant. Therefore, the current I flowing from the drive transistor 24 into the stator winding 11
The current “a” is divided into the stator windings 12 and 13 at the neutral point o, and the current Ib flows through the stator winding 12 and the current Ic flows through the stator winding 13. Drive currents Ib and Ic flowing through the stator windings 12 and 13 are extracted via drive transistors 22 and 23.

In the period (1) of FIG. 6, the three drive transistors 2 forming the second drive transistor group 5b
Of the drive transistors 4, 25, and 26, only the drive transistor 24 that flows the drive current Ia is turned on, and the drive transistor 24 is in the one-phase conduction period T1. In addition, of the three drive transistors 21, 22, 23 constituting the first drive transistor group 5a, two of the drive transistors 22, 23 are on, and the phase switching of the drive currents Ib, Ic is performed. Therefore, these drive transistors 22 and 23 are in the phase switching period T2.

In the period (2) of FIG. 6, the current signal d 'drawn from the base gradually decreases, so that the ON state of the drive transistor 24 gradually decreases, and the drive current Ia flowing into the stator winding 11 flows. Gradually decreases. On the other hand, since the current signal f ′ drawn from the base gradually increases, the on state of the drive transistor 25 gradually increases, and the drive current Ib flowing into the stator winding 12 increases.
Gradually increases. Then, the sum of the two drive currents (Ia
+ Ib) is constant. Drive current I flowing into stator windings 11 and 12 via drive transistors 24 and 25
a and Ib merge at the neutral point o, and the driving current Ic (= Ia + Ib) flows through the stator winding 13. This drive current Ic
Is drawn out via the drive transistor 23.

In the period (2), the three drive transistors 21 and 2 constituting the first drive transistor group 5a
Of the drive transistors 23 and 23, only the drive transistor 23 that flows the drive current Ic is turned on, and the drive transistor 23 is in the one-phase conduction period T1.
Further, of the three drive transistors 24, 25, 26 constituting the second drive transistor group 5b, two of the drive transistors 24, 25 are on, and the phase of the drive currents Ia, Ib is switched. Therefore, these drive transistors 24 and 25 are in the phase switching period T2.

The same operation is performed in periods other than the periods (1) and (2).

As described above, the first reference voltage E1 and the second
When the reference voltage E2 is set sufficiently high, the operating voltage between the emitter and the collector of each drive transistor is also sufficiently large and the respective current amplification factors (hfe) are sufficiently large, so that the three-phase stator windings 11, 12 , 13 can be supplied with drive currents Ia, Ib, Ic in a trapezoidal waveform in which switching can be smoothly performed without distortion even in the phase switching period T2.

However, the reference voltage E1 and the reference voltage E
2 is set to be sufficiently large, and the driving transistor 2
1, 22, 23 and drive transistors 24, 25, 26
When the operating voltage between the emitter and the collector is set high, there is a problem that the power loss in the driving transistor increases.

Therefore, in order to improve the power efficiency of the motor drive, as will be described below, the drive transistors 21 and
It is necessary to set the operating voltages between the emitters and collectors of the driving transistors 22, 23 and the driving transistors 24, 25, 26 as low as possible.

FIG. 7 shows that the magnitudes of the reference voltage E1 generated by the first reference voltage generation circuit 9 and the reference voltage E2 generated by the second reference voltage generation circuit 10 are sufficiently small.
It shows the signal waveform of each part when it is set to 2 '.

In FIG. 7, a, b, and c are stator windings 1
This is the back electromotive force induced at 1, 12, and 13 and has the same waveform as FIG. VA, VB, and VC indicate voltage waveforms at the current supply terminals A, B, and C of the three-phase stator windings 11, 12, and 13, respectively, and the voltage values are lower than those in FIG. Ia, Ib, Ic correspond to the first reference voltage generation circuit 9
And the second reference voltage generating circuit 10
Is a substantially trapezoidal drive current supplied to the three-phase stator windings 11, 12, and 13 when the reference voltage E2 at which the occurs is set to a small voltage.

In period (1) shown in FIG. 7, current I flowing from drive transistor 24 to stator winding 11 is
a is a constant current as in FIG. On the other hand, the current Ia flowing into the stator winding 11 is divided into the stator windings 12 and 13 at the neutral point o, and from the stator winding 12 (current Ib) to the stator winding 13 (current Ic). When the phase switching of the current is performed, the driving transistors 22 and 23 for controlling the current flowing through the stator windings 12 and 13 operate in the saturation region because the magnitude of the reference voltage E1 'is set sufficiently small. ing. Therefore, the current amplification factor (hfe) of the driving transistors 22 and 23 is equal to the operating voltage between the emitter and the collector of the transistors (substantially equal to VA, VB and VC in FIG. 7).
Depends on. As a result, the drive currents Ib and Ic flowing through the stator windings 12 and 13 cause waveform distortion due to the current switching not being performed smoothly. That is, no distortion occurs in the drive current Ia during the one-phase conduction period T1 for the drive transistor 24, but the drive transistors 22 and 2
3 during the phase switching period T2.
Distortion occurs in b and Ic.

Similarly, in period (2) shown in FIG. 7, the sum (Ia + Ib) of drive currents Ia and Ib flowing from drive transistors 24 and 25 to stator windings 11 and 12, respectively, is constant. Current Ia, Ib
Merge at the neutral point o, and the current Ic (= I
a + Ib) flows. This current Ic is drawn from the stator winding 13 via the drive transistor 23.

In period (2) of FIG. 7, current Ic drawn from stator winding 13 via drive transistor 23
Is a constant current waveform as in FIG. On the other hand, when current phase switching is performed from stator winding 11 (drive current Ia) to stator winding 12 (drive current Ib), drive transistor 2 that controls the current flowing through stator windings 11 and 12
Since the operating voltages between the emitters and the collectors of the transistors 4 and 25 set the reference voltage E2 'to a sufficiently small value, the driving transistors 24 and 25 operate in the saturation region of the transistors in the period (2). I have. Therefore, the current amplification factor (hfe) of the driving transistors 24 and 25 depends on the operating voltage between the emitter and the collector of the transistors (substantially equal to VM-VA and VM-VB in FIG. 7). As a result, the drive currents Ia and Ib flowing through the stator windings 11 and 12 are not smoothly switched, and the drive currents Ia and Ib generate waveform distortion. That is, in the one-phase conduction period T1 of the driving transistor 23, no distortion occurs in the driving current Ic.
In the phase switching period T2 for 4 and 25, distortion occurs in the drive currents Ia and Ib.

The same operation is performed in periods other than the periods (1) and (2). That is, when the reference voltage E1 and the reference voltage E2 are set sufficiently small to suppress the power loss, the driving transistor operates in the saturation region, and the operating voltage between the emitter and the collector becomes insufficient. Drive current I flowing through lines 11, 12, and 13
a, Ib, and Ic cause distortion in the phase switching period T2,
When the motor is driven in such a state, vibration and noise are generated.

In order to solve such a problem, in the brushless DC motor of the present invention, control is performed so that signal waveforms of various parts as shown in FIG. 5 are obtained. That is, the magnitudes of the reference voltage E1 generated by the first reference voltage generation circuit 9 and the reference voltage E2 generated by the second reference voltage generation circuit 10 are expressed by:
Back electromotive force a induced in the stator windings 11, 12, 13
It is configured to change continuously in synchronization with the timings b and c.

As shown in FIG. 5, the reference voltage E1 generated by the first reference voltage generation circuit 9 gradually increases from the timing when each signal of the back electromotive force a, b, c changes from positive to negative. , A point at which two of the back electromotive forces a, b, and c intersect with each other as a vertex, and thereafter, gradually decrease. On the other hand, the second reference voltage E2 generated by the second reference voltage generation circuit 10 gradually increases from the timing at which the signals of the back electromotive forces a, b, and c change from positive to negative. Power a,
A point at which two of b and c intersect is set as a vertex, and thereafter, it gradually decreases.

Next, the signal waveform of each part when the brushless DC motor of the present invention is rotated at low speed and high speed is shown in FIG.
Shown in For comparison, FIG. 8 is a signal waveform diagram of each part when the brushless DC motor of the present invention was rotated at a low speed.
This is shown in (1). This is the same as the waveform diagram shown in FIG.
A, VB, and VC indicate voltage waveforms at the current supply terminals A, B, and C, and VM indicates an output voltage waveform of the voltage conversion circuit 4. In FIG. 8A, the ripple of the output voltage VM of the voltage conversion circuit 4 has an extremely flat waveform. Fig. 8 (2)
Is a reference voltage E1 generated by the first reference voltage generation circuit 9.
And the reference voltage E2 generated by the second reference voltage generation circuit 10.
Shows the signal waveforms of the respective parts when the brushless DC motor of the present invention is rotated at a high speed, with the same setting as that of the low-speed rotation described above. The magnitude of the back electromotive force a, b, c induced in each of the stator windings 11, 12, 13 is determined by the rotor 2
7, the voltage waveforms VA, VB, VC of the current supply terminals A, B, C also increase.

As a result, output voltage VM of voltage conversion circuit 4
Also increases in size. However, since the magnitudes of the reference voltage E1 generated by the first reference voltage generation circuit 9 and the reference voltage E2 generated by the second reference voltage generation circuit 10 are set to be the same at the time of high-speed rotation as at the time of low-speed rotation, The ripple of the output voltage VM to be output from the voltage conversion circuit 4 shown in FIG. 8 (2) is larger than the ripple of the output voltage VM shown in FIG. 8 (1). In FIG. 8, the horizontal axis is represented by the electrical angle ωt, so that the waveform period is the same between the low-speed rotation and the high-speed rotation. However, the waveform period is actually inversely proportional to the rotation speed. Be shorter.

In order to smooth the output voltage VM as much as possible, the voltage conversion circuit 4 generally includes an inductance coil 106 having a large inductance with respect to the operating frequency;
Since it is configured to include the smoothing capacitor 107 having a large electrostatic capacity, the control band of the output voltage of the voltage conversion circuit 4 is reduced. Therefore, the generation of the output voltage ripple as shown by VM in FIG. Since the output voltage of the voltage conversion circuit 4 cannot be sufficiently followed, a phase delay occurs, and in the worst case, the peaks of the envelope waveforms of the voltage waveforms VA, VB, and VC (6 times per cycle) are output. The valley portion of the waveform of the voltage VM overlaps.

When the peaks of the envelope waveforms of the voltage waveforms VA, VB, and VC overlap with the valleys of the waveform of the output voltage VM, the driving transistors 21, 22, 23 constituting the first driving transistor group 5a. In addition, since the voltage between the emitter and the collector of the driving transistors 24, 25, and 26 constituting the second driving transistor group 5b is not sufficiently secured, the current amplification operation of the transistors is not performed, and the driving current Ia,
Waveform distortion occurs in Ib and IC. When the motor is driven in such a state, vibration and noise are generated. In the related art, the peaks of the envelope waveforms of the voltage waveforms VA, VB, and VC do not overlap with the valleys of the waveform of the output voltage VM. First
The operating voltage between the emitter and collector of the driving transistor group 5a and the second driving transistor group 5b is set to be higher with a sufficient margin. As a result, it has not been possible to realize a brushless DC motor with less vibration and noise in the range before low-speed and high-speed rotation and excellent in power efficiency.

In order to solve such a problem, the brushless DC motor according to the present embodiment has a structure in which the first and second drive transistor groups 5a and 5b operate at high speed.
The current is switched from one phase of the stator winding to the next phase, and the phase switching energized state in which two phases flow simultaneously is made larger than the one phase energized state in which current flows only in one phase, and the size is high speed. At the time of rotation, control is made to be larger than at the time of low-speed rotation.

That is, according to the rotation speed of the motor, the first
The magnitudes of the reference voltage E1 generated by the reference voltage generation circuit 9 and the reference voltage E2 generated by the second reference voltage generation circuit 10 are set higher during high-speed rotation than during low-speed rotation.

When the motor is rotating at high speed, the reference voltage E
FIG. 8 (3) shows signal waveforms of the respective units when 1 and the reference voltage E2 are changed from the settings at the time of low-speed rotation. FIG. 8C shows that the ripple of the output voltage VM has an extremely flat waveform. By changing the reference voltage E1 and the reference voltage E2 according to the low-speed and high-speed rotations in this manner, the vibration and noise are low even when the voltage conversion circuit 4 having a low control band is used and the low-speed and high-speed rotations are performed. A highly efficient brushless DC motor can be realized.

Hereinafter, a specific circuit for generating such reference voltages E1 and E2 and its operation will be described in detail.

FIG. 9 shows an example of the first reference voltage generating circuit 9 in the brushless DC motor of the present invention.
Shows signal waveforms of various parts in a steady rotation state. In FIG. 9, 111, 112 and 113 are diodes,
The respective cathode terminals are connected in common, and the switching circuit 120
Is connected to the input terminal of The switching circuit 120 has two output terminals. One output terminal X is connected to a resistor 114a, and the other output terminal Y is connected to a resistor 114b. The other ends of the resistor 114a and the resistor 114b are commonly connected, and are grounded via a reference voltage E1 'set lower than the reference voltage source 115. The switching circuit 120 is connected to the output terminal X when the switching signal input to the command terminal 60 is at the "L" level, and the switching signal is "L".
When the signal is at H level, the changeover switch is connected to the output terminal Y. The anode terminals of the diodes 111, 112, and 113 are connected to one ends of the switch circuits 116, 117, and 118, respectively. The other end is grounded.

In FIG. 9, 121, 122, 123
Is a two-input AND circuit, and one input terminal has
Inverted signals obtained by inverting the shaped signals D1, D2, and D3 obtained by the shaping circuit 37 by respective inverter circuits 124, 125, and 126 are input. 2-input AND circuit 1
Shaping signals D3, D1, and D2 are input to the other input terminals of 21, 122, and 123, respectively.

The switch circuits 116, 117, 118
The circuit is configured to close when the outputs of the AND circuits 121, 122 and 123 are at “H” level, and to open when the outputs are at “L” level. And diodes 111, 112, 11
3 are connected to subtraction circuits 31 and 3 respectively.
2, 33 output current signals Ip1, Ip2, Ip3 with inverted polarity.
Is entered. Reference numeral 119 denotes an output terminal of the first reference voltage generation circuit 9, which outputs a reference voltage E1.

Next, the operation of the first reference voltage generation circuit 9 shown in FIG. 9 will be described with reference to the waveform diagram of FIG. FIG.
(1) shows current signal waveforms Ip1 ″, Ip input from the subtraction circuits 31, 32, 33 to the first reference voltage generation circuit 9.
2 ", Ip3", and FIG. 10 (2), (3), (4)
Are the shaping signal waveforms D1, D2,
D3 is shown. FIGS. 10 (5), (6) and (7)
10 shows output waveforms (logical product of two inputs) of the AND circuits 121, 122, and 123 of FIG. FIG. 10 (8) shows a reference voltage signal waveform E1 output from the output terminal 119 of the first reference voltage generation circuit 9.

The switch circuits 116, 117, 118
The current signals Ip1 ", Ip2", Ip3 "input to the first reference voltage generating circuit 9 are opened and closed by the signals shown in FIGS. , 117, and 118 only flow to the switching circuit 120 via the diodes 111, 112, and 113.

When the switching signal input to the command terminal 60 of the switching circuit 120 is at "L" level, the current input to the switching circuit 120 is applied to the resistor 114a because the switching switch is connected to the output terminal X. Flows. As a result, the output terminal 119 of the first reference voltage generation circuit 9 is connected to the reference voltage source 1
15 reference voltages E1 ′, diodes 111, 112,
A voltage generated by adding a voltage drop generated by the resistor 114a is generated by a current supplied through the resistor 113, and FIG.
(8) A mountain-shaped reference voltage signal waveform E1 as shown by a solid line.
Is output.

When the switching signal input to command terminal 60 of switching circuit 120 is at "H" level, the switching switch is connected to output terminal Y, so that the current input to switching circuit 120 flows to resistor 114b. . If the size of the resistor 114b is selected to be larger than that of the resistor 114a, the output terminal 119 is connected to the reference voltage E1 'of the reference voltage source 115 by a current flowing through the diodes 111, 112 and 113. A voltage is generated by adding the voltage drop generated at 114b, and a mountain-shaped reference voltage signal waveform E1 as indicated by a dotted line in FIG. 10 (8) is output.

Next, an example of the second reference voltage generating circuit 10 in the brushless DC motor of the present invention is shown in FIG. FIG. 1 shows the signal waveform of each part in the steady rotation state.
It is shown in FIG.

In FIG. 11, 131, 132, 133
Are diodes, each having a cathode terminal commonly connected and connected to an input terminal of the switching circuit 140. The switching circuit 140 has two output terminals, a resistor 134a is connected to one output terminal X, and a resistor 13a is connected to the other output terminal Y.
4b is connected. The other ends of the resistors 134a and 134b are connected in common, and are grounded via a reference voltage source 135, which is a lower reference voltage E2 '.
The switching circuit 140 is connected to the output terminal X when the switching signal input to the command terminal 60 is at "L" level, and is connected to the output terminal Y when the switching signal is at "H" level. You. Diode 131, 132, 1
33 are connected to the switch circuits 136, 13
7, 138, and the switch circuits 136, 137,
It is grounded via 138. 141, 142, 14
Reference numeral 3 denotes a two-input AND circuit.
The shaping signals D1, D2, and D3 obtained by the shaping circuit 37 are input, and the shaping signals D3, D1, and D are input to the other input terminals.
2 is input by an inverter circuit 144, 145, 146.

The switch circuits 136, 137, and 138
The circuit is configured to be opened and closed by the outputs of the AND circuits 141, 142, 143, closed when the output is at “H” level, and opened when the output is at “L” level. The current signals Ip output from the subtraction circuits 31, 32, and 33 are connected to the anode terminals of the diodes 136, 137, and 138, respectively.
1 ', Ip2', Ip3 'are input. 139 is the second
The output terminal of the reference voltage generation circuit 10
2 is output.

Next, the operation of the second reference voltage generation circuit 10 shown in FIG. 11 will be described with reference to the waveform diagram of FIG. FIG.
(1) shows the current signal waveforms Ip1 ′, Ip1 input from the subtraction circuits 31, 32, 33 to the second reference voltage generation circuit 10.
12 (2), (3), and p2 'and Ip3'.
(4) is a shaping signal waveform D1, output by the shaping circuit 37.
D2 and D3 are shown. FIGS. 12 (5), (6) and (7)
12 illustrates output waveforms (logical product of two inputs) of the AND circuits 141, 142, and 143 of FIG. FIG. 12 (8) shows a reference voltage signal waveform E2 output from the output terminal 139 of the second reference voltage generation circuit 10. Switch circuits 136, 13
7 and 138 are opened and closed by the signals shown in FIGS. 11 (5), (6) and (7), so that the current signals Ip1 ′, Ip2 ′ and Ip3 ′ inputted to the second reference voltage generating circuit 10 are Only when the respective switch circuits 136, 137, 138 are open, the current flows to the switching circuit 140 via the diodes 131, 132, 133. When the switching signal input to the command terminal 60 of the switching circuit 140 is at "L" level, the current input to the switching circuit 140 flows through the resistor 134a because the switching switch is connected to the output terminal X. As a result, at the output terminal 139 of the second reference voltage generating circuit 10, the voltage drop generated at the resistor 134a by the current supplied to the reference voltage E2 'of the reference voltage source 135 through the diodes 131, 132, 133 is output. A voltage resulting from the addition of the voltage is generated, and a mountain-shaped reference voltage signal waveform E2 as shown by a solid line in FIG.

When the switching signal input to the command terminal 60 of the switching circuit 140 is at "H" level, the switching switch is connected to the output terminal Y, so that the current input to the switching circuit 140 flows to the resistor 134b. . If the size of the resistor 134b is selected to be larger than that of the resistor 134a, the output terminal 139 is connected to the reference voltage E2 'of the reference voltage source 135 by a current flowing through the diodes 131, 132, 133. A voltage is generated by adding the voltage drop generated at 134b, and a mountain-shaped reference voltage signal waveform E2 as indicated by a dotted line in FIG. 12 (8) is output.

As is apparent from FIGS. 10 (8) and 12 (8), the first reference voltage E1 output from the first reference voltage generation circuit 9 and the output from the second reference voltage generation circuit 10 are output. Is 180 degrees out of phase with the second reference voltage E2.

In the period (1) shown in FIG. 5, the current signal d 'drawn from the base is the same as described with reference to FIG.
Is large, the driving transistor 24 is turned on, while the current signals f 'and h' to be drawn are zero, and the driving transistors 25 and 26 are turned off. Further, since the current signal g 'flowing into the base is zero, the drive transistor 21 is turned off, thereby preventing the two cascade-connected drive transistors 24 and 21 from being simultaneously turned on. Then, the ON state of the driving transistor 22 in which the current signal i ′ flowing into the base gradually decreases, the ON state of the driving transistor 23 in which the current signal e ′ flowing gradually increases gradually increases, and the two driving transistors 22 and 23 is kept constant. Accordingly, a driving current Ia is supplied to the stator winding 11 from the driving transistor 24, and the driving current Ia is supplied to the stator winding 1 at the neutral point o.
It is diverted into 2 and 13. Then, a drive current Ib flows through the stator winding 12, and a drive current Ic flows through the stator winding 13. These drive currents Ib and Ic are drawn through drive transistors 22 and 23.

In the period (1), the three drive transistors 24, 2 forming the second drive transistor group 5b
Only one of the drive transistors 24, through which the drive current Ia flows, is turned on among the drive transistors 5 and 26. The drive transistor 24 is in the one-phase conduction period T1. In the one-phase conduction period T1 for the drive transistor 24, the drive current Ia flowing into the stator winding 11 has a constant value. This one-phase conduction period T1
In the above, the reference voltage output from the second reference voltage generating circuit 10 becomes the reference voltage E2 'set lower, and the operating voltage between the emitter and the collector of the driving transistor 24 is low, so that the power loss is small.

In the period (1), three drive transistors 2 forming the first drive transistor group 5a are used.
1, 2, 23, two driving transistors 22,
23 is turned on, and the phase switching of the drive currents Ib and Ic is performed.
23 has a phase switching period T2. In this phase switching period T2, the reference voltage output from the first reference voltage generation circuit 9 becomes a higher reference voltage E1 obtained by adding a chevron voltage signal to a lower reference voltage E1 ', and the driving transistor The operating voltage between the emitters and the collectors 22 and 23 is sufficiently large.

Therefore, as in the case of FIG. 7, the current amplification factors (hfe) of the driving transistors 22 and 23 are different from the operating voltage (VB, VB of FIG. 7) between the emitter and collector of the transistors.
(Approximately equal to VC). In addition, since the waveforms of the voltages VB and VC of the current supply terminals B and C are smooth, the drive currents Ib and Ic flowing through the stator windings 12 and 13 are smoothly switched, so that the phase is changed. No waveform distortion occurs even during the switching period T2.

Further, in the period (2) shown in FIG. 5, the current signal d 'drawn from the base gradually decreases, as described with reference to FIG. 6, so that the ON state of the drive transistor 24 gradually decreases and becomes fixed. The drive current Ia flowing into the slave winding 11 gradually decreases. On the other hand, since the current signal f 'drawn from the base gradually increases, the ON state of the drive transistor 25 gradually increases, and the stator winding 1
2 gradually increases. The sum of the two drive currents (Ia + Ib) is constant. Driving transistors 24 and 25 are connected to stator windings 11 and 12, respectively.
Drive currents Ia and Ib flowing through the winding are combined at the neutral point o, and the drive current Ic (= Ia +
Ib) flows, and the drive current Ic is applied to the drive transistor 2
Withdrawn through 3.

In the period (2), the three drive transistors 21 and 2 constituting the first drive transistor group 5a
Of the drive transistors 23 and 23, only the drive transistor 23 that flows the drive current Ic is turned on, and the drive transistor 23 is in the one-phase conduction period T1.
In the one-phase conduction period T1, the drive current Ic flowing through the stator winding 13 keeps a constant value. In the one-phase energizing period T1, the reference voltage output from the first reference voltage generation circuit 9 is the first reference voltage E set at a lower level.
1 ', the operating voltage between the emitter and the collector of the driving transistor 23 is low, so that the power loss is small.

In the period (2), the three drive transistors 2 forming the second drive transistor group 5b
4, 25, 26, two driving transistors 24,
25 is turned on and the drive currents Ia and Ib are being phase-switched.
25 is a phase switching period T2. In the phase switching period T2 for these drive transistors 24 and 25, the reference voltage output from the second reference voltage generation circuit 10 is obtained by adding a chevron voltage signal to the second reference voltage E2 'set lower. The second reference voltage E2 becomes higher and the operating voltage between the emitter and collector of the driving transistors 24 and 25 becomes sufficiently large. Therefore,
The fact that the current amplification factor (hfe) of the driving transistors 24 and 25 depends on the operating voltage between the emitter and collector of the transistors (substantially equal to VM-VA and VM-VB in FIG. 7) as in the case of FIG. In addition, the waveforms of the voltages VA and VB of the current supply terminals A and B also become smooth. As a result, the drive currents Ia and Ib applied to the stator windings 11 and 12 are smoothly switched, and the drive currents Ia and Ib are switched even during the phase switching period T2.
No waveform distortion occurs in b.

Similar operations are performed in periods other than periods (1) and (2).

With the above configuration, the stator windings 11, 12, and 13 are formed by the three drive transistors 21, 22, and 23 forming the first drive transistor group 5a.
In the phase switching period T2 during which the phase switching of the driving current is performed, the first reference voltage generation circuit 9 outputs a higher first reference voltage E1 to drive the driving transistors 21 and
The operating voltages between the emitters and the collectors of the drive transistors 21, 22, and 23 are set to a sufficiently high level to avoid operation in the saturation region. Operating voltage during the first reference voltage E, which was originally set lower
It is set low enough based on the output of 1 '.

Similarly, in the phase switching period T2 in which the driving currents of the three driving transistors 24, 25, and 26 constituting the second driving transistor group 5b are switched.
By outputting a higher second reference voltage E2 from second reference voltage generating circuit 10, drive transistors 24 and 2 are output.
The operating voltage between the emitter and the collector is set to be sufficiently high to avoid operation in the saturation region. On the other hand, in the one-phase conduction period T1 in which the phase switching is completed, the drive transistor 2
The operating voltage between the emitter and collector of 4, 25, 26 is
It is set sufficiently low based on the output of the second reference voltage E2 'which was originally set low.

When the brushless DC motor of the present invention is rotated at a high speed, an "H" level switching signal is input to the command terminal 60 shown in FIG. Since it is connected to the output terminal Y, the reference voltage E1 generated by the first reference voltage generation circuit 9 and the reference voltage E2 generated by the second reference voltage generation circuit 10 are respectively shown in FIGS.
As shown by the dotted line in (8), a voltage waveform having a mountain shape is formed. The size of the mountain shape portion is made higher at high speed rotation of the motor than at low speed rotation, so that the ripple of the output voltage VM to be output from the voltage conversion circuit 4 is obtained. The minute has a very flat waveform.

As is apparent from the above description, the first reference voltage generator 9 and the second reference voltage generator 10 basically set the lower reference voltage E1 'and the lower reference voltage E1.
2 ′ are output, so that the driving transistors 21, 22, 23 and the driving transistors 24, 25, 2
6, the operating voltage between the emitter and the collector can be set small, so that the power loss in the driving transistor can be suppressed small. Moreover, the higher reference voltage E1
And the higher reference voltage E2, the stator windings 11, 1
By outputting a signal changed in a mountain shape in synchronization with the timing of the back electromotive forces a, b, c induced in the stator windings 2, 13 and 13, the phase switching operation of the drive current of the stator windings 11, 12, and 13 is performed. Since the operation is performed smoothly without the occurrence of waveform distortion, it is possible to drive a brushless DC motor with very little vibration and noise.

When the brushless DC motor of the present invention is rotated at high speed, an "H" level switching signal is input to the command terminal 60, and the first reference voltage generating circuit 9 and the reference voltage generating circuit 10 Reference voltage E1 and reference voltage E
(2) The size of the chevron portion of each voltage waveform is made higher during high-speed rotation of the motor than at low-speed rotation, so that the ripple of the output voltage VM to be output from the voltage conversion circuit 4 has an extremely flat waveform. Since the phase switching operation of the drive currents of the stator windings 11, 12, and 13 can be smoothly performed without generating waveform distortion even at low speed and high speed rotation, the brushless DC motor with very little vibration and noise can be used. Driving becomes possible.

The output voltage VM to be output from the voltage conversion circuit 4 in the low-speed and high-speed rotations of the motor shown in FIGS. 8 (1) and 8 (3) is flat in both cases with extremely little ripple. The magnitudes of the peaks of the voltage waveforms of the reference voltage E1 and the reference voltage E2 are determined so that the output voltage VM has a flat waveform with a very small amount of ripple only during high-speed rotation of the motor. Is also good.
When the motor is rotating at low speed, the voltage waveforms VA, V
Since the ripple frequencies of the envelope waveforms of B and VC are low and sufficiently lower than the control band of the output voltage of the voltage conversion circuit 4,
The output voltage VM can sufficiently follow the ripples of the envelope waveforms of the voltage waveforms VA, VB, and VC. Therefore,
Only during high-speed rotation when the ripple frequency of the envelope waveforms of the voltage waveforms VA, VB, and VC is higher than the control band of the output voltage of the voltage conversion circuit 4, the output voltage VM is set to be a flat waveform with extremely little ripple. The size of the chevron of the voltage waveform of the voltage E1 and the reference voltage E2 may be determined.
In this case, the size of the peaks of the voltage waveforms of the reference voltage E1 and the reference voltage E2 when the motor rotates at a low speed can be set smaller than the waveforms of E1 and E2 shown in FIG. 23 and drive transistor 2
Since the operating voltage between the emitter and the collector of 4, 25, 26 can be set still smaller, the power loss in the driving transistor can be further reduced.

In the first reference voltage generation circuit 9 shown in FIG. 9 and the second reference voltage generation circuit 10 shown in FIG. 11, the changeover switches of the changeover circuits 120 and 140 operate at low and high speeds of the motor. Although the switching is performed only in two stages according to the rotation, the switching may be performed in any number of stages according to the rotation speed of the motor. Further, without providing the switching circuit 120 and the switching circuit 140, the resistance 114 and the resistance 1
The resistance value of the resistor 34 may be configured to increase continuously according to the rotation speed.

The resistance values of the resistors 114 and 134 are fixed, and the current signals Ip1 ", Ip2", Ip3 "input to the first reference voltage generation circuit 9 and the second reference voltage generation circuit 10 The input current signal waveform Ip1 ′,
The size of Ip2 ′ and Ip3 ′ may be configured to increase continuously according to the rotation speed.

In the first reference voltage generation circuit 9 shown in FIG. 9 and the second reference voltage generation circuit 10 shown in FIG. 11, the lower reference voltage E1 'and the lower reference voltage E1 are set.
Although the size of 2 ′ is fixed, the operating voltage between the emitter and collector of the driving transistors 21, 22, 23, 24, 25, and 26 constituting the first driving transistor group 5a and the second driving transistor group 5b. Is changed according to a command signal for commanding the current supply to the stator winding, and when the supply current increases, the reference voltage signal E
1 'and the reference voltage signal E2' may be configured to increase in magnitude.

In the above embodiment, the stator winding 1
The sum of the drive currents flowing out of the drive currents 1, 12, and 13 is detected by the current detection resistor 57, and the drive currents Ia, Ib, and Ic are controlled so that the sum of the drive currents becomes constant, so that the waveforms of the drive currents become trapezoidal. Although it is configured, the sum of the drive currents flowing out of the stator windings 11, 12, and 13 is modulated in addition to the trapezoidal drive current (for example, see Japanese Patent Application Laid-Open No. 61-150695).
In this case, the waveforms of the drive currents Ia, Ib, and Ic may be sinusoidal.

Further, the present invention is not limited to the three-phase motor as in the above embodiment, but can be applied to a four-phase motor or more.

[0132]

As described above, according to the brushless DC motor of the present invention, the operating voltage between the emitter and the collector of the first driving transistor group becomes equal to the reference voltage output from the first reference voltage generating circuit. Current control of the second driving transistor group, and voltage conversion so that the operating voltage between the emitter and the collector of the second driving transistor group becomes equal to the reference voltage output from the second reference voltage generating circuit. The DC output voltage of the circuit is controlled, and the operating voltage between the emitter and the collector of the first driving transistor group and the second driving transistor group is converted into a multi-phase position signal output from the position signal synthesis circuit and each output of the Hall element. The output of the buffer amplifier to be amplified is changed according to the shaped signal whose waveform has been shaped.
In the phase conduction period, the operating voltage between the emitter and the collector of the driving transistor through which the driving current is flowing is set to be as low as possible, so that the power loss in the driving transistor can be reduced.

Further, during the phase switching period in which the phase switching of the driving current is performed, the operating voltage between the emitter and the collector of the corresponding driving transistor is changed in accordance with the rotation speed of the motor, and is switched higher. The phase switching operation of the drive current of the stator winding is smoothly performed, and the brushless DC motor can be driven with very little vibration and noise.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of a brushless DC motor according to an embodiment of the present invention.

FIG. 2 is a signal waveform diagram of each part for explaining the operation of a position signal synthesizing circuit and a shaping circuit constituting the brushless DC motor of FIG.

FIG. 3 is a circuit diagram showing an example of a first operating voltage detection circuit in the brushless DC motor of FIG.

FIG. 4 is a circuit diagram showing an example of a second operating voltage detection circuit in the brushless DC motor of FIG.

FIG. 5 is a signal waveform diagram of each part for describing an operation when an operating voltage of a driving transistor is changed according to a position signal in the brushless DC motor of FIG. 1;

FIG. 6 is a signal waveform diagram of each section for explaining a problem that power loss increases in a conventional brushless DC motor.

FIG. 7 is a diagram for explaining a problem that waveform distortion occurs in a driving current of a stator winding during a phase switching period when a first reference voltage and a second reference voltage are set low in a conventional brushless DC motor. Signal waveform diagram of each part

FIGS. 8 (1), (2), and (3) each explain the operation of the brushless DC motor of FIG. 1 when the operating voltage of the drive transistor is changed according to the position signal and the rotation speed of the motor. Signal waveform diagram of each part for

9 is a circuit diagram showing an example of a first reference voltage generation circuit in the brushless DC motor of FIG.

FIG. 10 is a signal waveform diagram of each section for explaining the operation of the first reference voltage generation circuit shown in FIG. 8;

11 is a circuit diagram showing an example of a second reference voltage generation circuit in the brushless DC motor of FIG.

12 is a signal waveform diagram of each part for describing the operation of the second reference voltage generation circuit shown in FIG.

[Explanation of symbols]

 1, 2, 3 ... Hall element 4 ... voltage conversion circuit 5a ... first drive transistor group 5b ... second drive transistor group 6a ... first distribution control circuit 6b ... second distribution control Circuit 9 First reference voltage generating circuit 10 Second reference voltage generating circuit 11, 12, 13 Stator winding 27 Rotor 40 Position signal synthesizing circuit

Claims (12)

[Claims] 1. A rotor having a plurality of magnetic poles, a plurality of stator windings, a plurality of Hall elements for detecting a rotational position of the rotor, and a plurality of phases from outputs of the plurality of Hall elements. Position signal synthesizing means for generating a position signal, a voltage converting means for obtaining a variable output DC voltage from a DC power supply, one of an output terminal pair of the voltage converting means and a power supply terminal for each phase of the stator winding, A first driving transistor group including a plurality of transistors forming a current path between the first and second driving transistors, a first command signal for commanding a current supply to the stator winding, and the output of the position signal synthesizing means. First distribution control means for distributing and controlling the current flowing through the drive transistor group; and a plurality of current distribution means for forming a current path between the other of the output terminal pair of the voltage conversion means and the power supply terminal of each phase of the stator winding. The second including the transistor of Distribution control of a current flowing through the second driving transistor group so that a minimum operating voltage of the first driving transistor group coincides with a first reference voltage according to an output of the driving transistor group and the position signal synthesizing means. A second distribution control unit that controls the output voltage of the voltage conversion unit such that a minimum operating voltage of the second drive transistor group matches a second reference voltage;
Wherein the first and second reference voltages are the first
In the second driving transistor group, the current is switched from one phase of the stator winding to the next phase, and the phase switching energized state in which the two phases flow simultaneously is better than the one phase energized state in which the current flows in only one phase. A brushless DC motor characterized in that it is large and its size changes according to the rotation speed of the rotor. 2. A method according to claim 1, wherein the first and second reference voltages are switched simultaneously from one phase of a stator winding to the next phase in the first and second driving transistor groups, and are simultaneously switched to two phases. The flowing phase-switching energized state is larger than the one-phase energized state in which current flows only in one phase, and the magnitude thereof is continuously increased substantially in proportion to the rotation speed of the rotor. The brushless DC motor according to claim 1, wherein 3. The method according to claim 1, wherein the first and second reference voltages continuously change, and in the phase switching energized state, change in a triangular waveform having a peak at a time when currents flowing in the two phases become substantially equal. The brushless DC motor according to claim 1. 4. The brushless DC motor according to claim 1, wherein the first and second reference voltages change in accordance with a command signal for commanding a current supply to the stator winding. 5. A position signal synthesizing means, comprising: a plurality of buffer amplifiers for amplifying respective outputs of the plurality of Hall elements;
A plurality of subtraction circuits for generating a difference between the outputs of the plurality of buffer amplifiers, wherein the first and second reference voltages are a plurality of phase position signals output by the position signal synthesizing means and the buffer amplifier. 2. A brushless DC motor according to claim 1, wherein said output is generated from a shaping signal obtained by shaping the output of said brushless DC motor. 6. A rotor having a plurality of magnetic poles, a plurality of stator windings, a plurality of Hall elements for detecting a rotational position of the rotor, and a plurality of phases from outputs of the plurality of Hall elements. Position signal synthesizing means for generating a position signal, a voltage converting means for obtaining a variable output DC voltage from a DC power supply, one of an output terminal pair of the voltage converting means and a power supply terminal for each phase of the stator winding, A first driving transistor group including a plurality of transistors forming a current path between the first and second driving transistors, a first command signal for commanding a current supply to the stator winding, and the output of the position signal synthesizing means. First distribution control means for distributing and controlling the current flowing through the drive transistor group; and a plurality of current distribution means for forming a current path between the other of the output terminal pair of the voltage conversion means and the power supply terminal of each phase of the stator winding. The second including the transistor of Distribution control of a current flowing through the second driving transistor group so that a minimum operating voltage of the first driving transistor group coincides with a first reference voltage according to an output of the driving transistor group and the position signal synthesizing means. A second distribution control unit that controls the output voltage of the voltage conversion unit such that a minimum operating voltage of the second drive transistor group matches a second reference voltage;
Wherein the first and second reference voltages are the first
In the second driving transistor group, the current is switched from one phase of the stator winding to the next phase, and the phase switching energized state in which the two phases flow simultaneously is better than the one phase energized state in which the current flows in only one phase. When the rotor is large and at least the rotor is rotated at a high speed, the magnitudes of the first reference voltage and the second reference voltage are changed so that the waveform of the output voltage of the voltage conversion means becomes substantially flat. A brushless DC motor. 7. The brushless DC motor according to claim 6, wherein the levels of the first and second reference voltages during high-speed rotation are higher than the levels during low-speed rotation. 8. The method according to claim 1, wherein the first and second reference voltages are continuously changed, and in the phase switching energized state, the first and second reference voltages are changed into a triangular waveform having a peak at a time when currents flowing in the two phases become substantially equal. The brushless DC motor according to claim 6. 9. The brushless DC motor according to claim 6, wherein said first and second reference voltages change in accordance with a command signal for commanding current supply to said stator winding. 10. The position signal synthesizing means includes a plurality of buffer amplifiers for amplifying respective outputs of the plurality of Hall elements, and a plurality of subtraction circuits for generating a difference between respective outputs of the plurality of buffer amplifiers. 7. The method according to claim 6, wherein the first and second reference voltages are generated from a multi-phase position signal output by the position signal synthesizing unit and a shaped signal obtained by shaping the output of the buffer amplifier. The brushless DC motor as described. 11. A multi-phase position signal is generated based on a signal indicating a rotational position of a rotor output from a plurality of Hall elements, and a current is simultaneously supplied to two phases of a stator winding using the position signal. Generates a first and a second reference voltage whose voltage value increases in the phase switching state as compared with the other states, and uses the first and the second reference voltages as the first and second operational amplifiers, respectively. The voltage at the power supply terminal is input to the non-inverting terminal and the power supply terminal for supplying power to the stator winding is included in the negative feedback loops of the first and second operational amplifiers. Control based on voltage,
Thus, a drive control method for a brushless DC motor, wherein a desired operating voltage in the first and second drive transistor groups for supplying a current to the stator winding via the power supply terminal is secured. 12. The drive control method for a brushless DC motor according to claim 11, wherein, when generating the first and second reference voltages, a voltage value is changed according to a rotation speed of the motor.