JP3300637B2 - Brushless DC motor - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a brushless DC motor, and more particularly, to a brushless DC motor that efficiently uses power supplied from a power supply.

[0002]

2. Description of the Related Art A brushless DC motor has no mechanical contact and thus has a long life and has little electrical noise compared to a DC motor with a brush. Because of this advantage, brushless DC motors are widely used in industrial equipment and video / audio equipment that require high reliability in recent years.

In a conventional brushless DC motor,
A voltage supplied from a DC power supply having a constant output voltage is variably controlled using a voltage control transistor or the like, and for example, a voltage corresponding to a rotation speed is supplied to a motor. Therefore,
The effective voltage used for driving the motor is always lower than the voltage of the DC power supply, and the difference between the DC power supply voltage and the voltage actually supplied to the motor is almost equal to the collector loss (heat loss) of the voltage control transistor. Power efficiency was reduced.

In order to improve the power efficiency of a brushless DC motor, several methods have been proposed for reducing the collector loss of the voltage control transistor by controlling the switching of the voltage control transistor.

In one example, a first current path which forms a current path between one end of a DC power supply and a current supply terminal of a stator winding is provided.
Is controlled in accordance with the current command and the position signal. Then, the second drive transistor group forming the current path between the other end of the DC power supply and the current supply terminal is adjusted so that the minimum operating voltage of the first drive transistor group becomes equal to a predetermined reference voltage. Control. Further, the on / off control of the switching transistor for voltage control is performed so that the operating voltage of the second driving transistor group becomes equal to a predetermined reference voltage. In this way, the voltage supplied to the motor is controlled. By performing such control, the power efficiency of the motor is greatly improved (see, for example, Japanese Patent Application Laid-Open No. 58-198189).

[0006] In the above-described configuration, the position switching is performed by applying a position signal to the base input of the differential transistor having the emitter commonly connected, so that the stator winding driving current can be switched stably. However, the drive current flowing through the stator winding has a rectangular waveform having a conduction width of approximately 120 degrees in electrical angle, and is rapidly turned on and off. For this reason, vibration and noise are easily generated.

In order to smoothly perform phase switching of the stator winding, when performing current switching from one phase to the next phase, there is a period in which current is supplied to two phases simultaneously, that is, a method of performing overlap driving. (For example, see JP-A-62-22).
1894).

[0008]

In the above prior art, the first and second driving transistor groups are overlap-driven to improve the power efficiency, so that the emitter and collector of the first and second driving transistor groups are improved. If the operating voltage of the transistor is reduced by reducing the remaining voltage during this time, current switching is not performed smoothly, and the drive current waveform is distorted. When the motor is driven in this state, vibration and noise are generated. The details of this problem will be described later with reference to FIG. 7 in an embodiment of the invention.

In view of the above-mentioned conventional problems, the present invention uses a voltage control of a switching control method capable of outputting a variable output DC voltage, and performs smooth phase switching of the stator winding to thereby reduce vibration. It is an object of the present invention to provide a brushless DC motor with low noise and excellent power efficiency.

[0010]

SUMMARY OF THE INVENTION A brushless DC motor according to the present invention comprises a rotor having a plurality of magnetic poles, a plurality of stator windings, and a plurality of Hall elements for detecting the rotational position of the rotor. A position signal synthesizing unit that generates a plurality of phase position signals from outputs of the plurality of Hall elements, a voltage conversion unit that obtains a variable output DC voltage from a DC power supply, and one of output terminal pairs of the voltage conversion unit. A first drive transistor group including a plurality of drive transistors forming a current path between the power supply terminal of each phase of the stator winding and a command for supplying a current to the current supply terminal; First distribution control means for distributing and controlling a current flowing through the first drive transistor group in accordance with a signal and a position signal from the position signal synthesizing means; and the other of the output terminal pair of the voltage conversion means and the stator Each of the windings A second driving transistor group including a plurality of driving transistors forming a current path between the power supply terminals of driving a plurality of constituting the first driving transistor group
Detects the minimum voltage among the operating voltages of the
First operating voltage detecting means for outputting as a first operating voltage
And a plurality of drivers constituting the second drive transistor group.
Detects the minimum voltage among the operating voltages of the
Second operating voltage detecting means for outputting as a second operating voltage
And according to the position signal from the position signal synthesizing means,
So that the first operating voltage matches the first reference voltage.
Distribution control of current flowing through the second driving transistor group
Second distribution control means, wherein the second operating voltage is a second reference voltage;
The output voltage of the voltage conversion means is adjusted to match the reference voltage.
And voltage control means for controlling.

The first and second reference voltages are respectively equal to the first and second drive transistors.
In the group, each of the stator windings is separated into two phases.
Drive train that is in the phase switching energized state to allow the
The operating voltage of the transistor is in the 1-phase energized state after the phase switching is completed.
Higher than the operating voltage of the drive transistor
Is set variably.

According to the above configuration, the first drive transistor group and the second drive transistor group are respectively distributed and controlled by the position signal output from the position signal synthesizing means, and the current supplied to the stator winding is controlled. Are sequentially performed, and the rotor is rotationally driven. Controlling the current supplied to the second driving transistor group so that the operating voltage between the emitter and the collector of the first driving transistor group becomes equal to the reference voltage output from the first reference voltage generating means; The operating voltage between the emitter and collector of the group is
The DC output voltage of the switching control type voltage conversion means is controlled to be equal to the reference voltage signal output from the reference voltage generation means.

The first drive transistor group and the second drive transistor group
The operating voltage between the emitter and the collector of the driving transistor group is changed according to the position signal output from the position signal synthesizing means. That is, the first driving transistor group and the second driving transistor group
In each of the plurality of drive transistors constituting the drive transistor group, during the phase switching period in which the drive current is switched, the operating voltage between the emitter and collector of the corresponding drive transistor is switched higher. The phase switching operation of the drive current of the stator winding is performed smoothly. As a result, the brushless DC motor can be driven with very little vibration and noise.

On the other hand, when the phase switching of the drive current has been completed 1
In the phase conduction period, the operating voltage between the emitter and the collector of the driving transistor through which the driving current is flowing is set as low as possible, so that the power loss in the driving transistor can be reduced. Thus, according to the present invention, it is possible to realize a brushless DC motor with less vibration and noise and excellent in power efficiency.

In the above configuration, preferably, the first
And the second reference voltage, respectively,
The state continuously changes between the state and the one-phase energized state, and in the phase-switched energized state, changes in a triangular waveform having a peak at a point in time at which drive currents flowing through the two phases become substantially equal. Since the first and second reference voltages, which are switched higher in the phase switching energized state, are changed in a symmetrical triangular waveform, the phase switching operation of the drive current of the stator winding can be performed more smoothly, and the brushless DC motor Vibration and noise at the time of driving can be further reduced.

It is also preferable that the first and second reference voltages change according to a command signal for commanding a current supply to the stator winding. Since the first and second reference voltages are changed in accordance with the magnitude of the current supply to the stator winding, current switching is always performed smoothly irrespective of the change in drive current, and waveform distortion of the drive current is reduced. The motor can be driven in a state where no occurrence occurs. As a result, power efficiency is high,
The effect of reducing vibration and noise is further enhanced.

More preferably, the position signal synthesizing means includes a plurality of buffer amplifiers for amplifying respective outputs of the plurality of Hall elements, and a plurality of subtraction circuits for generating a difference between respective outputs of the plurality of buffer amplifiers. The said first
And the second reference voltage is generated from a multi-phase position signal output by the position signal synthesizing means and a shaped signal obtained by shaping the output of the buffer amplifier. Thus, the first and second reference voltages can be generated with a simple circuit configuration.

Another configuration of the brushless DC motor according to the present invention includes a rotor having a plurality of magnetic poles, a stator winding having a plurality of phases, a plurality of Hall elements for detecting a rotational position of the rotor, and Position signal synthesizing means for generating a plurality of phase position signals from the outputs of the plurality of Hall elements, voltage converting means for obtaining a variable output DC voltage from a DC power supply, and one of the output terminal pairs of the voltage converting means and the fixed A first drive transistor group including a plurality of drive transistors forming a current path between the power supply terminal of each phase of the slave winding and a command signal for commanding current supply to the current supply terminal; First distribution control means for distributing and controlling the current flowing through the first drive transistor group in accordance with the position signal from the position signal synthesizing means; and the other of the output terminal pair of the voltage conversion means and the stator winding Salary of each phase of A plurality of forming a current path between the terminal
A second group of driving transistors including a driving transistor;
A plurality of driving transistors constituting the first driving transistor group
Operates by detecting the minimum voltage among the operating voltages of the transistor
Operating voltage detection means for outputting a voltage, and the position signal
The operating voltage is set to the reference voltage according to the position signal from the combining means.
To the second drive transistor group so as to match the voltage.
And second distribution control means for controlling distribution of current to be supplied.
The reference voltage is applied to the first driving transistor group.
The two phases of the stator windings are in the phase switching state respectively.
Driving transistor in phase-switching energized state for driving current
Drive with the operating voltage of
Variable setting to be higher than transistor operating voltage
Is done.

In the above configuration, the plurality of drive transistors constituting the second drive transistor group are directly connected to the DC power supply. In this case, the voltage conversion means of the switching control method can be omitted, and the power loss of the plurality of drive transistors constituting the first drive transistor group can be reduced. A plurality of drive transistors constituting a second drive transistor group having a large power loss are constituted by external discrete transistors, and a plurality of drive transistors constituting a first drive transistor group having a small power loss and a small heat generation are provided. Can be easily included in an integrated circuit including a drive circuit.

In this configuration, preferably, the reference voltage
The pressure is the phase switching energized state and the one-phase energized state, respectively.
And continuously changes between the two states.
It changes in a triangular waveform with the point at which the drive currents flowing through the phases become substantially equal to each other. Since the reference voltage, which is switched higher in the phase switching energized state, is changed in a symmetrical triangular wave shape, the phase switching operation of the stator coil drive current can be performed more smoothly, and vibration and noise in driving the brushless DC motor. Suppression is promoted.

Preferably, the reference voltage changes according to a command signal for commanding a current supply to the stator winding. Since the reference voltage is changed according to the change in the current supply to the stator winding, regardless of the drive current,
The current can be smoothly switched, and the motor can be driven in a state where no waveform distortion of the drive current occurs. Therefore, it is possible to provide a brushless DC motor having excellent power efficiency and low vibration and noise.

Further, the position signal synthesizing means includes a plurality of buffer amplifiers for amplifying respective outputs of the plurality of Hall elements, and a plurality of subtraction circuits for generating a difference between respective outputs of the plurality of buffer amplifiers. Preferably, the reference voltage signal is generated from a multi-phase position signal output by the position signal synthesizing means and a shaped signal obtained by shaping the output of the buffer amplifier. Thereby, the reference voltage signal can be generated with a simple circuit configuration.

[0023]

DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, a specific embodiment of a brushless DC motor according to the present invention will be described with reference to the drawings.

FIG. 1 shows a circuit configuration of a brushless DC motor according to an embodiment of the present invention. In FIG. 1, reference numeral 27 denotes a permanent magnet rotor having a plurality of magnetic poles;
Is a stator winding provided so as to form a predetermined gap with the rotor 27;
b denotes a second drive transistor group, 20 denotes a DC power supply, and 4 denotes a switching control type voltage conversion circuit for obtaining a variable output DC voltage from the DC power supply 20.

The first driving transistor group 5a includes three N
PN-type drive transistors 21, 22, 23,
Each of the drive transistors 21, 22, 23 is interposed in a current path between the current supply terminals A, B, C of the stator windings 11, 12, 13 and the negative terminal (GND) of the voltage conversion circuit 4, respectively. ing. Also, the second driving transistor group 5b has 3
PNP-type drive transistors 24, 25, and 26. Each drive transistor 24, 25, and 26 includes a positive terminal of the voltage conversion circuit 4 and stator windings 11, 12, and
13 are provided in the current path between the current supply terminals A, B, and C.

1, 2 and 3 located at the left end of FIG.
7 are three Hall elements arranged so as to form a predetermined gap. Reference numeral 30 denotes three Hall elements 1, 2, 3,
, DC power supplies 31, 32 and 33 are buffer amplifiers, which output outputs H1, H2 and H3 proportional to the differential outputs of the Hall elements 1, 2 and 3, respectively. 41, 42 and 43 are subtraction circuits.

The subtraction circuit 41 receives the output H1 of the buffer amplifier 31 and the output H3 of the buffer amplifier 33, and outputs a current Ip1 proportional to the difference (H1-H3). The subtraction circuit 42 receives the output H2 of the buffer amplifier 32 and the output H1 of the buffer amplifier 31, and outputs a current Ip2 proportional to the difference (H2−H1). The subtraction circuit 43 receives the output H3 of the buffer amplifier 33 and the output H2 of the buffer amplifier 32, and calculates the difference (H3-
A current Ip3 proportional to H2) is output. Each output current I
p1, Ip2, and Ip3 are three-phase position signals. Buffer amplifiers 31, 32, 33 and subtraction circuits 41, 42, 43
These form the position signal synthesis circuit 40.

Each of the subtraction circuits 41, 42 and 43 has 3
It is configured to obtain two output currents. That is, the current signals Ip1, Ip2, Ip3, the current signals Ip1 ', Ip2', Ip3 'having the same polarity as these current signals,
And current signals Ip1 ″, Ip2 ″, I
Three types of output currents of p3 ″ are obtained.

The current signals Ip1, Ip2, Ip
3 corresponds to a three-phase position signal, and diodes 54, 55,
56, only the negative side is taken out and currents g, i, e
And input to the first distribution circuit 6a. The branched output currents Ip1, Ip2, Ip3 are
Only the positive polarity side is taken out by 1, 52 and 53 to become currents d, f and h, which are input to the second distribution circuit 6b.

The bases of the driving transistors 21, 22, 23 are supplied with three-phase current signals g ', i', e 'formed by the first distribution circuit 6a, respectively. 23 is controlled. Similarly, three-phase current signals d ', f', and h 'formed by the second distribution circuit 6b are supplied to the bases of the driving transistors 24, 25, and 26, respectively. 26 is controlled. A current in a flowing direction is applied to the bases of the NPN transistors 21, 22, 23, and the PNP transistors 24, 25,
An electric current in a drawing direction is applied to the base 26.

In FIG. 1, reference numeral 57 denotes a current detection resistor, which converts a current supplied to the three-phase stator windings 11, 12, 13 into a voltage. A first comparison control circuit 36 compares a command signal input to the command terminal 50 with a voltage obtained by the current detection resistor 57. The control signal CL obtained as a result is supplied to the first distribution circuit 6a, and controls the magnitudes of the input position signals g, i, e to control the current signals g ', i',
Generate e '. The current signals g ′, i ′, and e ′ control the operation of the drive transistors 21, 22, and 23 to control the magnitude of the current supplied to the three-phase stator windings 11, 12, and 13. Is done.

In FIG. 1, reference numeral 7 denotes drive transistors 21, 22, 23 constituting the first drive transistor group 5a.
A first operating voltage detection circuit for detecting the minimum operating voltage L of
9 is a first circuit for supplying a reference voltage to the second comparison control circuit 34;
The reference voltage generation circuit 37 has buffer amplifiers 31 and 3
This is a shaping circuit for shaping the waveforms of the outputs H1, H2, and H3 of the H.2, 33 and outputting shaping signals D1, D2, and D3.

The first reference voltage generation circuit 9 includes a subtraction circuit 4
1, 42, and 43, and the shaping signal D of the shaping circuit 37.
1, D2 and D3 to form a reference voltage signal E1. The second comparison control circuit 34 compares the first reference voltage signal E1 generated by the first reference voltage generation circuit 9 with the minimum operation voltage L output by the first operation voltage detection circuit 7. The control signal CU obtained as a result is supplied to the second distribution circuit 6b, and controls the magnitudes of the input position signals d, f, h to generate current signals d ', f', h '. These current signals d ', f', h 'are supplied to the drive transistors 24, 25, 2
6 to control the three-phase stator winding 1.
The magnitude of the current supplied to 1, 12, 13 is controlled.

In FIG. 1, reference numeral 8 denotes a second drive transistor group 5
b is a second operating voltage detection circuit that detects the minimum operating voltage U of the driving transistors 24, 25, and 26 that constitutes b.
Reference numeral 10 denotes a second voltage control circuit 35 for the voltage conversion circuit 4.
Is a second reference voltage generating circuit that supplies the reference voltage signal E2 of FIG. The second reference voltage generation circuit 10 includes a subtraction circuit 4
1, 42, and 43, and the shaping signal D of the shaping circuit 37.
1, D2 and D3 to form a reference voltage signal E2.

The voltage control circuit 35 compares the reference voltage signal E2 generated by the second reference voltage generation circuit 10 with the minimum operation voltage U obtained by the second operation voltage detection circuit 8, and outputs the control signal CS as a voltage. Output to the conversion circuit 4. Voltage conversion circuit 4
Are connected from the positive terminal of the DC power supply 20 to the stator windings 11, 1
It is inserted in series in the power supply paths leading to 2 and 13, and is configured to control the output voltage VM of the voltage conversion circuit 4 according to the control signal CS of the voltage control circuit 35.

The voltage conversion circuit 4 includes a power supply control switching transistor 10 inserted in series in a power supply path from the positive terminal of the DC power supply 20 to the stator windings 11, 12, and 13.
1, a switching control circuit 100 for controlling the switching transistor 101 to turn on and off based on a control signal CS from the voltage control circuit 35, and a free-wheeling diode 105.
, The inductance coil 106 and the smoothing capacitor 1
07.

The voltage control circuit 35 generates a control signal CS corresponding to the minimum operation voltage U obtained by the second operation voltage detection circuit 8. The switching control circuit 100 controls on / off of the switching transistor 101 by a pulse signal corresponding to the control signal CS. As a result, the voltage conversion circuit 4 converts the voltage VS of the DC power supply 20 into an output voltage VM and outputs it, and supplies the output voltage VM to the second drive transistor group 5b. The switching control circuit 100
May be configured using a known circuit such as a triangular wave generating circuit that generates a triangular wave voltage signal of 200 kHz and a comparator that compares the control signal CS of the voltage control circuit 35 with the triangular wave voltage signal.

Next, a specific circuit example of the first operating voltage detection circuit 7 in FIG. 1 is shown in FIG. In FIG.
Diodes 81 to 84 have their anode terminals connected in common, and the cathode terminals of the diodes 81, 82, and 83 have three-phase stator windings 11, 12, 1 respectively.
3 are connected to the three current feed terminals A, B, and C. 85 is a resistor, one end of which is grounded and the other end of which is a diode 8
4 are connected to the cathode terminal. A constant current source 86 supplies a constant current to the anode common terminals of the diodes 81 to 84. 87 is an output terminal of the first operating voltage detection circuit 7. When the diode having the lowest cathode potential among the three diodes 81, 82, and 83 connected to the current supply terminals A, B, and C is turned on, the potential of the anode common terminal of the diode is changed to the forward direction of the diode in the on state. It becomes higher than the cathode potential by the voltage. The output current of the constant current source circuit 86 is also supplied to the resistor 85 via the diode 84. Therefore, a voltage lower than the potential of the anode common terminal by the forward voltage of the diode is generated in the resistor 85. Therefore, the minimum operating voltage L, which is the minimum voltage among the current feeding terminals A, B, and C of the three-phase stator windings 11, 12, and 13, is output from the output terminal 87 of the first operating voltage detection circuit 7. You.

Next, FIG. 4 shows a specific circuit example of the second operating voltage detection circuit 8 in FIG. In FIG.
Diodes 91, 92 and 93 have their cathode terminals connected in common. Diodes 91 and 9
Anode terminals 2 and 93 each have a three-phase stator winding 1
1, 12, and 13 are connected to current supply terminals A, B, and C, respectively. Reference numeral 94 denotes a PNP transistor. The emitter is connected to the output voltage VM of the switching control type voltage conversion circuit 4 via a resistor 96, and the collector is grounded via a resistor 95.

The base of the transistor 94 is a diode 9
1, 92 and 93 are connected to the commonly connected cathode terminals. 97 is a constant current source circuit,
1, 92, 93 and a transistor 94
A constant current is drawn from the base. Reference numeral 98 denotes an output terminal of the second operating voltage detection circuit 8.

When only the diode having the highest anode potential among the three diodes 91, 92 and 93 connected to the current supply terminals A, B and C is turned on, the potential of the cathode common terminal of the diode is turned on. Becomes lower than the anode potential by the forward voltage of the diode. The base of the transistor 94 is connected to the common cathode terminal of the diodes 91, 92 and 93. Since the voltage between the emitter and the base of the transistor 94 is substantially equal to the diode forward voltage in the ON state, the emitter potential VE of the transistor 94 is obtained.
Becomes almost equal to the highest potential among the current supply terminals A, B, and C.

Since one end of the resistor 96 is connected to the output potential VM of the voltage conversion circuit 4, the potential difference (V
M-VE), and a current substantially equal to this current flows through the collector of the transistor 94. Therefore, if the resistance values of the resistor 95 and the resistor 96 are selected to be equal, the resistor 9
5 generate the same potential difference (VM−VE) as both ends of the resistor 96. Therefore, the difference between the output voltage VM of the voltage conversion circuit 4 and the highest potential VE among the current supply terminals A, B, and C of the three-phase stator windings 11, 12, 13 (V
M-VE), that is, the minimum operating voltage U having the smallest difference from the output voltage VM of the voltage conversion circuit 4 among the current supply terminals A, B, and C is output from the output terminal 98 of the second operating voltage detection circuit 8. You.

In FIG. 1, the first distribution circuit 6a is composed of one type of multiplier, and inputs the position signals g,
It outputs current signals g ′, i ′, and e ′ in which i and e are changed in magnitude according to the control signal CL of the first comparison control circuit 36. These three-phase current signals g ′, i ′, and e ′ are applied to the bases of the drive transistors 21, 22, and 23 that form the first drive transistor group 5 a, and are supplied to the drive transistors 21, 22, and 23. Control. The first comparison control circuit 36 compares the command signal input to the command terminal 50 with the voltage obtained at the current detection resistor 57, and supplies the obtained control signal CL to the first distribution circuit 6a. Thus, the magnitude of the current supplied to the three-phase stator windings 11, 12, 13 is controlled in accordance with the command signal input to the command terminal 50.

In FIG. 1, the second distribution circuit 6b is also formed of a kind of multiplier. The magnitudes of the position signals d, f, and h input to the second distribution circuit 6b are changed in accordance with the control signal CU output from the second comparison control circuit 34, and the current signals d ', f', h ′ is output. The three-phase current signals d ', f', h 'are applied to the bases of the driving transistors 24, 25, 26 constituting the second driving transistor group 5b, and control the energization of the driving transistors 24, 25, 26. I do.

The second comparison control circuit 34 compares the reference voltage signal E1 generated by the first reference voltage generation circuit 9 with the minimum operation voltage L obtained by the first operation voltage detection circuit 7, and as a result, Is transmitted to the second distribution circuit 6b.
Output to The current signals d ′, f ′, h ′ output from the second distribution circuit 6b are applied to the drive transistors 24, 25, 26
To control the output current of the second drive transistor group 5b. As a result, the three-phase stator winding 1
Control is performed so that the minimum voltages of the current supply terminals A, B, and C of the power supply terminals 1, 12, and 13 are equal to the reference voltage signal E1.

Similarly, the voltage control circuit 35 compares the reference voltage signal E2 generated by the second reference voltage generation circuit 10 with the minimum operation voltage U obtained by the second operation voltage detection circuit 8, The resulting control signal CS is supplied to the voltage conversion circuit 4
To the switching control circuit 100 of FIG. The switching control circuit 100 controls the switching transistor 101 to adjust the output voltage VS of the DC power supply 20, and outputs the output voltage VM to the second drive transistor group 5b. Therefore, the output voltage VM and the three-phase stator winding 1
The voltage difference between the current supply terminals A, B, C of 1, 12, 13 is the second
Is controlled to be equal to the reference voltage signal E2. That is, control is performed so that the minimum operating voltage between the emitter and the collector of the driving transistors 24, 25, and 26 constituting the second transistor group 5b becomes equal to the reference voltage signal E2.

FIG. 2 shows signal waveforms of various parts relating to the operation of the position signal synthesizing circuit 40 when the above-described brushless motor is in a steady rotation state. In FIG.
(1) shows the back electromotive force waveforms a, b, and c induced in the stator windings 11, 12, and 13, and (2) shows the Hall elements 1 and 2.
The output waveforms H1, H2, and H3 after the differential outputs of a few are amplified by the buffer amplifiers 31, 32, and 33 are shown. The output waveforms H1, H2, and H3 are out of phase by 120 degrees. The output waveforms H1, H2, and H3 are advanced by 30 degrees with respect to the respective back electromotive force waveforms a, b, and c.

In FIG. 2, (3) shows subtraction circuits 41 and 4
2, 43 output current waveforms Ip1, Ip2, Ip3, which are three-phase position signals. (4) is subtraction circuits 41 and 4
The current waveforms d, f, and h are obtained by cutting the negative polarity sides of the output currents Ip1 ', Ip2', and Ip3 'of the output currents Ip1, Ip2, and Ip3' with diodes 51, 52, and 53, respectively.
Current signals Ip1 ″, Ip obtained by inverting the polarities of Ip2 and Ip3.
The positive side of 2 ″, Ip3 ″ is connected to diodes 54, 55,
The current waveforms g, i, and e cut at 56 are shown. (6),
(7) and (8) are shaping signals D1 and D3 obtained by shaping the waveforms of the outputs H1, H2 and H3 of the buffer amplifiers 31, 32 and 33.
2 and D3 are shown, respectively. The rising edges of the shaping signals D1, D2, and D3 correspond to the zero-cross points on the rising sides of the outputs H1, H2, and H3 of the buffer amplifiers 31, 32, and 33, and the falling edges of the shaping signals D1, D2, and D3 correspond to the outputs. It corresponds to the zero cross point on the falling side of H1, H2, and H3.

FIG. 5 shows signal waveforms at various parts when the brushless DC motor of the present invention is driven by signals obtained by the above-described signal processing. For comparison, FIGS. 6 and 7 show similar signal waveforms in a conventional brushless DC motor.

FIG. 6 is used to explain the following problem in the conventional brushless DC motor. When the reference voltage signals E1 and E2 are set sufficiently large, the drive currents Ia, Ib, and Ic flowing through the stator windings 11, 12, and 13 become trapezoidal currents that change smoothly during the phase switching period without distortion. However, on the other hand, the driving transistor 2
Since the operating voltages between the emitters and the collectors of 1, 22, 23 and 24, 25, 26 are set to be large, the power loss in each drive transistor increases.

FIG. 7 is used to explain the following problem in the conventional brushless DC motor. When the reference voltage signals E1 and E2 are set to sufficiently small values E1 'and E2' in order to suppress power loss, the driving transistor operates in the saturation region, and the operating voltage between the emitter and the collector becomes insufficient. , Stator windings 11, 1
The drive currents Ia, Ib, Ic flowing through the second and the third 13 cause distortion during the phase switching period.

FIG. 6 is a circuit diagram of each part when the reference voltage signal E1 generated by the first reference voltage generation circuit 9 and the reference voltage signal E2 generated by the second reference voltage generation circuit 10 are set to a sufficiently large constant voltage. 3 shows a signal waveform.

In FIG. 6, a, b, and c are stator windings 1
V, V, V
B and VC are voltage waveforms at the current supply terminals A, B, and C of the three-phase stator windings 11, 12, and 13, respectively. Also, g ',
i ′ and e ′ are drive transistors 21, 22, 22 that form the first drive transistor group 5 a from the first distribution circuit 6 a.
23, a trapezoidal three-phase current signal flowing into the base of the second driving transistor group 5 d ′, f ′, h ′
The trapezoidal wave 3 drawn out from the bases of the driving transistors 24, 25 and 26 constituting the second distribution circuit 6b to the second distribution circuit 6b.
This is a phase current signal. Ia, Ib, and Ic are trapezoidal drive currents that are supplied to the three-phase stator windings 11, 12, and 13, respectively.

Driving transistors 21, 22, 23 and 2
4, 25 and 26 use transistors having the same electrical characteristics. Therefore, when the magnitudes of the reference voltage signals E1 and E2 are set to be sufficiently large, the driving transistor 2
Since the operating voltages between the emitters and collectors of 1, 22, 23 and 24, 25, 26 are also sufficiently large, the current amplification factors (hfe) of the respective transistors are sufficiently large and have little variation. Therefore, the three-phase stator windings 11, 12,
The drive currents Ia, Ib, and Ic of the drive transistor 21 correspond to three-phase trapezoidal wave current signals g ′, i ′, and e ′ input to the bases of the drive transistors 21, 22, and 23.
The trapezoidal three-phase current signals d ', f', and h 'input to the bases of 4, 25, and 26 can be equally amplified. As a result, accurate trapezoidal drive currents Ia, Ib, Ic without distortion are applied to the three-phase stator windings 11, 12, 1
3 can be supplied.

In the period (1) shown in FIG. 6, the drive transistor 24 is turned on because the current signal d 'drawn from the base is large. On the other hand, since the current signals f 'and h' drawn from the base are zero, the drive transistors 25 and 26 are turned off. Further, since the current signal g ′ flowing into the base is zero, the drive transistor 21 is turned off. In this way, simultaneous ON of the cascaded drive transistors 24 and 21 is prevented.

Then, the on-resistance of the driving transistor 22 whose current signal i 'flowing into the base gradually decreases gradually increases, and the on-resistance of the driving transistor 23 whose current signal e' flowing into the base gradually increases gradually decreases. The combined on-resistance of the pair of drive transistors 22 and 23 is kept constant. Therefore, the current Ia flowing from the drive transistor 24 into the stator winding 11 is divided into the stator windings 12 and 13 at the neutral point o, and the current Ib flows through the stator winding 12 and the stator winding 12 13 has a current I
c flows. Drive current I flowing through stator windings 12 and 13
b and Ic are extracted through the driving transistors 22 and 23.

In the period (1) of FIG. 6, the three drive transistors 2 forming the second drive transistor group 5b
Of the drive transistors 4, 25, and 26, only the drive transistor 24 that flows the drive current Ia is turned on, and the drive transistor 24 is in the one-phase conduction period T1. Further, out of the three drive transistors 21, 22, 23 constituting the first drive transistor group 5a, two drive transistors 22, 23 are turned on, and the phase switching of the drive currents Ib, Ic is performed. Therefore, these drive transistors 22 and 23 are in the phase switching period T
It is 2.

In the period (2) of FIG. 6, the current signal d 'drawn from the base gradually decreases, so that the on-resistance of the drive transistor 24 gradually increases, and the drive current Ia flowing into the stator winding 11 is increased. Gradually decreases. On the other hand, since the current signal f ′ drawn from the base gradually increases, the on-resistance of the drive transistor 25 gradually decreases, and the drive current Ib flowing into the stator winding 12 is reduced.
Gradually increases. Sum of two drive currents (Ia + I
b) is constant. Drive currents Ia, which flow into stator windings 11, 12 via drive transistors 24, 25,
Ib joins at the neutral point o, and the drive current Ic (= Ia + Ib) flows through the stator winding 13. This drive current Ic is drawn through the drive transistor 23.

In the period (2), the three drive transistors 21 and 2 constituting the first drive transistor group 5a
Of the drive transistors 23 and 23, only the drive transistor 23 for supplying the drive current Ic is turned on, and the drive transistor 23 is in the one-phase conduction period T1.
Further, of the three drive transistors 24, 25, and 26 constituting the second drive transistor group 5b, two of the drive transistors 24 and 25 are on, and the phase of the drive currents Ia and Ib is switched. Therefore, these drive transistors 24 and 25 are in the phase switching period T2.

The same operation is performed in periods other than the periods (1) and (2). As described above, when the first reference voltage signal E1 and the second reference voltage signal E2 are set to be sufficiently large, the operating voltages between the emitter and the collector of each driving transistor are also sufficiently large, and the respective current amplification factors (hf
e) is sufficiently large so that the three-phase stator windings 11, 1
The drive currents Ia, Ib, I of the trapezoidal waveforms with smooth and smooth switching even in the phase switching period T2 for the phase switching periods T2 and I3.
c can be supplied.

However, when the magnitudes of the reference voltage signal E1 and the reference voltage signal E2 are set sufficiently large, and the operating voltages between the emitters and collectors of the driving transistors 21, 22, 23 and 24, 25, 26 are set large. There is a problem that power loss in the driving transistor increases.

Therefore, in order to improve the power efficiency of the motor drive, as will be described below, the drive transistors 21 and
It is necessary to set the operating voltage between the emitters and the collectors of 22, 23 and 24, 25, 26 as small as possible.
FIG. 7 shows each unit when the reference voltage signal E1 generated by the first reference voltage generation circuit 9 and the reference voltage signal E2 generated by the second reference voltage generation circuit 10 are set to sufficiently small voltages E1 'and E2'. 3 shows a signal waveform of the first embodiment.

In FIG. 7, a, b, and c are stator windings 1
This is the back electromotive force induced in 1, 12, and 13 and has the same waveform as in FIG. VA, VB, and VC indicate voltage waveforms at the current supply terminals A, B, and C of the three-phase stator windings 11, 12, and 13, respectively, and the voltage values are lower than those in FIG. Ia, Ib, Ic correspond to the first reference voltage generation circuit 9
, And the reference voltage signal E2 generated by the second reference voltage generation circuit 10 are set to a small voltage, and are substantially trapezoidal waveforms supplied to the three-phase stator windings 11, 12, and 13. Is the driving current.

In the period (1) shown in FIG. 7, the current I flowing from the drive transistor 24 into the stator winding 11 is
a is a constant current similarly to FIG. On the other hand, the current Ia flowing into the stator winding 11 is divided into the stator windings 12 and 13 at the neutral point o, from the stator winding 12 (current Ib) to the stator winding 13 (current Ic). When current phase switching is performed, drive transistors 22 and 23 for controlling the current flowing through stator windings 12 and 13 operate in the saturation region because the magnitude of reference voltage signal E1 'is set sufficiently small. are doing. Therefore, the current amplification factor (hfe) of the driving transistors 22 and 23 depends on the operating voltage between the emitter and collector of the transistors (substantially equal to VA, VB and VC in FIG. 7). As a result, the drive currents Ib and Ic flowing through the stator windings 12 and 13 cause waveform distortion due to the current not being smoothly switched. That is, no distortion occurs in the drive current Ia during the one-phase conduction period T1 for the drive transistor 24, but the drive transistor 2
In the phase switching period T2 for the phases 2 and 23, the drive currents Ib and Ic are distorted.

Similarly, in the period (2) shown in FIG.
Drive transistors 2 are connected to stator windings 11 and 12, respectively.
The sum (Ia + Ib) of the drive currents Ia and Ib flowing from 4, 25 is constant, and these currents Ia and Ib are constant.
Merge at the neutral point o, and the current Ic (= I
a + Ib) flows. This current Ic is drawn from the stator winding 13 via the drive transistor 23.

In the period (2) of FIG. 7, the current Ic drawn from the stator winding 13 via the drive transistor 23
Is a constant current waveform as in FIG. On the other hand, when the current phase is switched from the stator winding 11 (drive current Ia) to the stator winding 12 (drive current Ib), the stator winding 1
Driving transistor 2 for controlling the current flowing through 1 and 12
The operating voltages between the emitter and the collector of 4, 25 are set sufficiently small in the magnitude of the reference voltage signal E2 '.
The drive transistors 24 and 25 operate in the saturation region of the transistors during the period (2). Therefore,
Current amplification factor (hfe) of drive transistors 24 and 25
Depends on the operating voltage between the emitter and the collector of the transistor (substantially equal to VM-VA and VM-VB in FIG. 7). As a result, the drive currents Ia and Ib flowing through the stator windings 11 and 12 are not smoothly switched, and the drive currents Ia and Ib generate waveform distortion. That is, the one-phase energizing period T1 for the drive transistor 23
However, no distortion occurs in the drive current Ic, but distortion occurs in the drive currents Ia and Ib during the phase switching period T2 for the drive transistors 24 and 25.

The same operation is performed in periods other than the periods (1) and (2). That is, when the reference voltage signal E1 and the reference voltage signal E2 are set sufficiently small in order to suppress the power loss, the driving transistor operates in the saturation region, and the operating voltage between the emitter and the collector becomes insufficient. The drive currents Ia, Ib, and Ic flowing through the slave windings 11, 12, and 13 cause distortion during the phase switching period T2, and when the motor is driven in such a state, vibration and noise are generated.

In order to solve such a problem, in the brushless DC motor of the present invention, control is performed so that signal waveforms of various parts as shown in FIG. 5 are obtained. That is, the magnitudes of the reference voltage signal E1 generated by the first reference voltage generation circuit 9 and the reference voltage signal E2 generated by the second reference voltage generation circuit 10 are induced in the stator windings 11, 12, and 13. It is configured to continuously change in synchronization with the timing of the back electromotive forces a, b, and c.

As shown in FIG. 5, the reference voltage signal E1 generated by the first reference voltage generation circuit 9 includes counter electromotive forces a, b,
The signal c gradually increases from the timing when it changes from positive to negative, and changes so as to gradually decrease from the point where two of the back electromotive forces a, b, and c intersect. On the other hand, the second reference voltage generation circuit 10
Conversely, the reference voltage signal E2 gradually increases from the timing when the signals of the back electromotive forces a, b, and c change from positive to negative,
The point at which two of the back electromotive forces a, b, and c intersect is a vertex, and thereafter, the voltage gradually decreases.

Hereinafter, such reference voltage signals E1 and E1
2 and its operation will be described in detail. FIG. 8 shows an example of the first reference voltage generating circuit 9 in the brushless DC motor according to the present invention, and FIG. 9 shows signal waveforms at various portions in a steady rotation state. In FIG. 8, 1
Reference numerals 11, 112, and 113 denote diodes, each having a cathode terminal commonly connected and connected to one end of a resistor 114. The other end of the resistor 114 is grounded via a lower reference voltage E1 'of the reference voltage source 115.
The anode terminals of the diodes 111, 112, and 113 are connected to one ends of switch circuits 116, 117, and 118, respectively, and the other ends of the switch circuits 116, 117, and 118 are grounded.

In FIG. 8, 121, 122, 123
Is a two-input AND circuit, and one input terminal has
Inverted signals obtained by inverting the shaped signals D1, D2, and D3 obtained by the shaping circuit 37 by respective inverter circuits 124, 125, and 126 are input. 2-input AND circuit 12
Shaping signals D3, D1 and D2 are input to the other input terminals of 1, 122 and 123, respectively.

The switch circuits 116, 117, 118
The outputs are closed when the outputs of the AND circuits 121, 122 and 123 are at H level, and are opened when the outputs are at L level. Then, current signals Ip1 ″, Ip2 ″, Ip3 ″ obtained by inverting the polarities of the current signals Ip1, Ip2, Ip3 output from the subtraction circuits 31, 32, 33 are input to the respective anode terminals of the diodes 111, 112, 113. An output terminal 119 of the first reference voltage generation circuit 9 outputs a reference voltage signal E1.

Next, the operation of the first reference voltage generating circuit 9 shown in FIG. 8 will be described with reference to the waveform diagram of FIG. 9 in the case where the rotor 27 is rotating normally. FIG.
(1) shows current signal waveforms Ip1 ″, Ip input from the subtraction circuits 31, 32, 33 to the first reference voltage generation circuit 9.
2 ", Ip3", and FIG. 9 (2), (3), (4)
Are the shaping signal waveforms D1, D2,
D3 is shown. 9 (5), (6) and (7) show output waveforms (2) of the AND circuits 121, 122 and 123 of FIG.
Input logical product). FIG. 9 (8) shows a reference voltage signal waveform E1 output from the output terminal 119 of the first reference voltage generation circuit 9.

The switch circuits 116, 117, 118
Since the switch is opened and closed by the signals shown in FIGS. 9 (5), (6) and (7), the current signal I input to the first reference voltage generating circuit 9 is
p1 ", Ip2", and Ip3 "flow through the resistor 114 via the diodes 111, 112, and 113 only when the switch circuits 116, 117, and 118 are open. As a result, the first reference voltage generation circuit 9 Output terminal 1
19, a voltage is generated by adding a voltage drop generated in the resistor 114 by the current supplied through the diodes 111, 112, and 113 to the reference voltage E1 'of the reference voltage source 115, and FIG. A reference voltage signal waveform E1 having a mountain shape as shown in FIG.

Next, an example of the second reference voltage generating circuit 10 in the brushless DC motor of the present invention is shown in FIG. FIG. 1 shows the signal waveform of each part in the steady rotation state.
1 is shown.

In FIG. 10, 131, 132, 133
Are diodes, each having a cathode terminal commonly connected and connected to one end of the resistor 134. Resistance 134
Is grounded via a reference voltage E2 ′ set lower than the reference voltage source 135. Diode 131, 1
32 and 133 are connected to the switch circuit 13
6, 137, 138 and the switch circuit 136,
137 and 138 are grounded. 141,14
Reference numerals 2 and 143 denote a two-input AND circuit. One of the input terminals is provided with the shaped signals D1, D2 and D2 obtained by the shaping circuit 37.
D3 is input, and the shaping signal D3 is input to the other input terminal.
Inverted signals obtained by inverting D1 and D2 by inverter circuits 144, 145 and 146 are input.

The switch circuits 136, 137, and 138
The circuits are opened and closed by the outputs of the AND circuits 141, 142, and 143, closed when the output is at the H level, and opened when the output is at the L level. And the diode 13
6, 137, and 138 have current signals Ip1 ',
Ip2 'and Ip3' are input. Reference numeral 139 denotes an output terminal of the second reference voltage generation circuit 10, which outputs a reference voltage signal E2.

Next, the operation of the second reference voltage generation circuit 10 shown in FIG. 10 will be described with reference to the waveform diagram of FIG. 11 in the case where the rotor 27 is rotating normally. FIG.
(1) shows the current signal waveforms Ip1 ′, Ip1 input from the subtraction circuits 31, 32, 33 to the second reference voltage generation circuit 10.
11 (2), (3), and p2 'and Ip3'.
(4) is a shaping signal waveform D1, output by the shaping circuit 37.
D2 and D3 are shown. FIGS. 11 (5), (6) and (7)
11 shows output waveforms (logical product of two inputs) of the AND circuits 141, 142, 143 of FIG. FIG. 11 (8) shows a reference voltage signal waveform E2 output from the output terminal 139 of the second reference voltage generation circuit 10. Switch circuits 136, 13
7, 138 are opened and closed by the signals shown in FIGS. 10 (5), (6), (7), so that the current signals Ip1 ′, Ip2 ′, Ip3 ′ inputted to the second reference voltage generating circuit 10 are Only when the respective switch circuits 136, 137 and 138 are open, the current flows to the resistor 134 via the diodes 131, 132 and 133. As a result, the output terminal 139 of the second reference voltage generating circuit 10 supplies the reference voltage E2 'of the reference voltage source 135 with the voltage drop generated by the resistor 134 due to the current flowing through the diodes 131, 132, 133. The added voltage is generated, and a mountain-shaped reference voltage signal waveform E2 as shown in FIG. 11 (8) is output.

As is clear from FIGS. 9 (8) and 11 (8), the first reference voltage generation circuit 9 outputs the first
The reference voltage signal E1 and the second reference voltage signal E2 output from the second reference voltage generation circuit 10 are 180 degrees out of phase.

In the period (1) shown in FIG. 5, the current signal d 'drawn from the base is the same as described with reference to FIG.
Is large, the driving transistor 24 is turned on. On the other hand, the current signals f ′ and h ′ to be extracted are zero, and the driving transistors 25 and 26 are turned off. Further, since the current signal g 'flowing into the base is zero, the drive transistor 21 is turned off, thereby preventing the two cascade-connected drive transistors 24 and 21 from being simultaneously turned on. Then, the on-resistance of the driving transistor 22 in which the current signal i 'flowing into the base gradually decreases gradually increases, and the on-resistance of the driving transistor 23 in which the current signal e' flowing gradually increases gradually decreases. 23 is kept constant. Accordingly, a driving current Ia is supplied to the stator winding 11 from the driving transistor 24, and the driving current Ia is supplied to the stator winding 1 at the neutral point o.
It is diverted into 2 and 13. The drive current Ib flows through the stator winding 12, and the drive current Ic flows through the stator winding 13. These drive currents Ib and Ic are drawn through drive transistors 22 and 23.

In the period (1), the three drive transistors 24, 2 forming the second drive transistor group 5b
Only one of the drive transistors 24, through which the drive current Ia flows, is turned on among the drive transistors 5 and 26. The drive transistor 24 is in the one-phase conduction period T1. In the one-phase conduction period T1 for the drive transistor 24, the drive current Ia flowing into the stator winding 11 has a constant value. This one-phase conduction period T1
, The reference voltage output from the second reference voltage generation circuit 10 becomes a reference voltage signal E2 ′ set lower,
Since the operating voltage between the emitter and the collector of the driving transistor 24 is low, the power loss is small.

In the period (1), the three drive transistors 2 forming the first drive transistor group 5a
1, 2, 23, two driving transistors 22,
23 are on and the phase switching of the drive currents Ib and Ic is being performed, so that these drive transistors 22 and
23 has a phase switching period T2. In the phase switching period T2, the reference voltage output from the first reference voltage generating circuit 9 is the lower reference voltage signal E
The reference voltage signal E1 is obtained by adding a chevron-shaped voltage signal to 1 ′, and becomes a higher reference voltage signal E1.
The operating voltage between the collectors is large enough.

Therefore, as in the case of FIG. 7, the current amplification factor (hfe) of the driving transistors 22 and 23 changes with the operating voltage (VB,
(Substantially equal to VC). In addition, since the waveforms of the voltages VB and VC of the current supply terminals B and C become smooth, the drive currents Ib and Ic flowing through the stator windings 12 and 13 are smoothly switched, and the phase is changed. No waveform distortion occurs even in the switching period T2.

In the period (2) shown in FIG. 5, the current signal d 'drawn from the base gradually decreases as described with reference to FIG. 6, so that the on-resistance of the drive transistor 24 gradually increases and becomes fixed. The drive current Ia flowing into the slave winding 11 gradually decreases. On the other hand, since the current signal f 'drawn from the base gradually increases, the on-resistance of the drive transistor 25 gradually decreases, and the stator winding 1
2 gradually increases. The sum of the two drive currents (Ia + Ib) is constant. Driving transistors 24 and 25 are connected to stator windings 11 and 12, respectively.
Drive currents Ia and Ib flowing through the stator winding 13 are combined at the neutral point o, and the drive current Ic (= Ia +
Ib) flows, and the drive current Ic is
Withdrawn through 3.

In the period (2), the three drive transistors 21 and 2 constituting the first drive transistor group 5a
Of the drive transistors 23 and 23, only the drive transistor 23 for supplying the drive current Ic is turned on, and the drive transistor 23 is in the one-phase conduction period T1.
In the one-phase conduction period T1, the drive current Ic flowing through the stator winding 13 keeps a constant value. In the one-phase energizing period T1, the reference voltage output from the first reference voltage generation circuit 9 becomes the first reference voltage signal E1 'set lower, and the emitter-
Since the operating voltage between the collectors is low, the power loss is small.

In the period (2), the three drive transistors 2 constituting the second drive transistor group 5b
4, 25, 26, two driving transistors 24,
25 is turned on and the phase switching of the drive currents Ia and Ib is being performed.
25 is a phase switching period T2. In the phase switching period T2 for these drive transistors 24 and 25, the reference voltage output by the second reference voltage generation circuit 10 is obtained by adding a chevron voltage signal to the second reference voltage signal E2 'set lower. The second reference voltage signal E2 becomes higher and is higher than the second reference voltage signal E2.
The operating voltage between the collectors is sufficiently high. Therefore, as in the case of FIG.
The current amplification factor (hfe) of the transistor 5
The operating voltage between the collectors (VM-VA, VM-VB in FIG. 7)
Is almost equal to), and
The waveforms of the voltages VA and VB at the current supply terminals A and B are also smoothed. As a result, the drive currents Ia and Ib applied to the stator windings 11 and 12 are smoothly switched, and the drive currents Ia and Ib have waveforms even during the phase switching period T2. No distortion occurs.

The same operation is performed in periods other than the periods (1) and (2). With the above-described configuration, the three drive transistors 21, 22, and 23 included in the first drive transistor group 5a can be used to form the stator winding 11,
Phase switching period T during which phase switching of the drive currents 12 and 13 is performed
2, the first reference voltage generating circuit 9 outputs a higher first reference voltage signal E1 to set the operating voltage between the emitter and the collector of the driving transistors 21, 22, 23 to a sufficiently high level so that the saturation region In the one-phase energizing period T1 in which the phase switching is completed, the operating voltage between the emitter and the collector of the driving transistors 21, 22, and 23 is lower than the first reference voltage signal. It is set sufficiently low based on the output of E1 '.

Similarly, during the phase switching period T2 in which the driving currents of the three driving transistors 24, 25, and 26 constituting the second driving transistor group 5b are switched.
By outputting a higher second reference voltage signal E2 from second reference voltage generation circuit 10, drive transistor 2
The operating voltage between the emitter and the collector of each of the transistors 4, 25 and 26 is set sufficiently high to avoid the operation in the saturation region. The operating voltage between the collectors is set sufficiently low based on the output of the second reference voltage signal E2 ', which was originally set lower.

As is apparent from the above description, the first and second reference voltage generating circuits 9 and 10 output the reference voltage signals E1 'and E2' which are basically set to be lower, respectively. The operating voltage between the emitters and the collectors of the transistors 21, 22, 23 and 24, 25, 26 can be set small, so that the power loss in the driving transistor can be reduced. In addition, as the higher reference voltage signals E1 and E2, the stator winding 1
A phase change of the drive current of the stator windings 11, 12, and 13 is performed by outputting a signal changed in a mountain shape in synchronization with the timing of the back electromotive forces a, b, and c induced in the stator windings 11, 12, and 13. Since the operation is performed smoothly without generating waveform distortion,
It becomes possible to drive a brushless DC motor with very little vibration and noise.

In the first reference voltage generator 9 shown in FIG. 8 and the second reference voltage generator 10 shown in FIG. 10, the lower reference voltage signal E1 'and the lower reference voltage signal E2' are set. Are constant, but the driving transistors 21, 22, 23, 24, 25, constituting the first driving transistor group 5 a and the second driving transistor group 5 b are
26, the operating voltage between the emitter and the collector is changed in accordance with a command signal for commanding the current supply to the stator winding. When the supply current increases, the magnitude of the reference voltage signals E1 'and E2' increases. It may be configured to increase the height.

In the above-described embodiment, the operating voltages between the emitters and the collectors of the three driving transistors 21, 22, 23 constituting the first driving transistor group 5a and the second driving transistor group 5b constitute the first driving transistor group 5b. The magnitude of the operating voltage between the emitters and collectors of the three drive transistors 24, 25, and 26 is obtained by changing the waveforms of the multi-phase position signals output from the position signal synthesizing circuit and the output of the buffer amplifier for amplifying each output of the Hall element. Although it is configured to change in response to the shaped signal that has been shaped, for example, the three drive transistors 21, 22 and 22 that constitute the first drive transistor group 5 a are configured.
The configuration may be such that only the operating voltage between the emitter and the collector 23 is changed, and the three driving transistors 24, 25 and 26 constituting the second driving transistor group 5b may be directly connected to the DC power supply 20. In this case, the switching control type voltage conversion circuit 4 can be omitted, and the three drive transistors 2 constituting the first drive transistor group 5a can be omitted.
Since the power loss of 1, 2, 23 can be reduced, the three drive transistors 24, 25, 26 constituting the second drive transistor group 5b having a large power loss are constituted by external discrete transistors, The three drive transistors 21, 22, and 23 constituting the first drive transistor group 5a with small power loss and small heat generation can be easily included in the drive integrated circuit.

In the above embodiment, the stator winding 1
The sum of the drive currents flowing out of 1, 12, and 13 is detected by a current detection resistor 57, and the drive currents Ia, Ib, and Ic are trapezoidal by controlling the sum of the drive currents to be constant. Although the configuration is not limited to the trapezoidal drive current, the drive currents Ia, Ib, and Ib are modulated by modulating the sum of the drive currents flowing out of the stator windings 11, 12, and 13 (see, for example, JP-A-61-150695). The waveform of Ic may be configured to be a sine wave.

Further, the present invention relates to the third embodiment as in the above embodiment.
The present invention can be applied not only to a phase motor but also to a single-phase motor or a motor having four or more phases.

[0094]

As described above, according to the brushless DC motor of the present invention, the operating voltage between the emitter and the collector of the first driving transistor group is made equal to the reference voltage signal output from the first reference voltage generating circuit. The current flowing through the second driving transistor group is controlled so that the operating voltage between the emitter and the collector of the second driving transistor group becomes equal to the reference voltage signal output from the second reference voltage generating circuit. The DC output voltage of the voltage conversion circuit is controlled, and the operating voltage between the emitter and the collector of the first drive transistor group and the second drive transistor group is changed to a multi-phase position signal output from the position signal synthesis circuit and each of the Hall elements. The output of the buffer amplifier for amplifying the output is changed according to the shaping signal obtained by shaping the waveform. Since configured to set as low as possible operating voltage between jitter collector, it is possible to reduce the power loss in the driving transistor. Furthermore, during the phase switching period in which the drive current is switched, the operating voltage between the emitter and the collector of the corresponding drive transistor is switched higher, so that the drive current phase switching operation of the stator winding is smoothly performed. Therefore, the brushless DC motor can be driven with very little vibration and noise.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of a brushless DC motor according to an embodiment of the present invention.

FIG. 2 is a signal waveform diagram of each part for explaining the operation of a position signal synthesizing circuit and a shaping circuit constituting the brushless DC motor of FIG.

FIG. 3 is a circuit diagram showing an example of a first operating voltage detection circuit in the brushless DC motor of FIG.

FIG. 4 is a circuit diagram showing an example of a second operating voltage detection circuit in the brushless DC motor of FIG.

FIG. 5 is a signal waveform diagram of each part for describing an operation when an operating voltage of a driving transistor is changed according to a position signal in the brushless DC motor of FIG. 1;

FIG. 6 is a signal waveform diagram of each section for explaining a problem that power loss increases in a conventional brushless DC motor.

FIG. 7 is a diagram for explaining the problem that waveform distortion occurs in the drive current of the stator winding during the phase switching period when the first and second reference voltage signals are set low in the conventional brushless DC motor. Signal waveform diagram

8 is a circuit diagram showing an example of a first reference voltage generation circuit in the brushless DC motor of FIG.

FIG. 9 is a signal waveform diagram of each section for explaining the operation of the first reference voltage generation circuit shown in FIG.

10 is a circuit diagram showing an example of a second reference voltage generation circuit in the brushless DC motor of FIG.

11 is a signal waveform diagram of each section for describing the operation of the second reference voltage generation circuit shown in FIG.

[Explanation of symbols]

 1, 2, 3 Hall element 4 Voltage conversion circuit 5a First drive transistor group 5b Second drive transistor group 6a First distribution control circuit 6b Second distribution control circuit 9 First reference voltage generation circuit 10 Second Reference voltage generation circuit 11, 12, 13 Stator winding 27 Rotor 40 Position signal synthesis circuit

──────────────────────────────────────────────────続 き Continued on front page (58) Field surveyed (Int.Cl. ⁷ , DB name) H02P 6/10 H02P 6/08

Claims (8)

(57) [Claims] 1. A rotor having a plurality of magnetic poles, a plurality of stator windings, a plurality of Hall elements for detecting a rotational position of the rotor, and a plurality of phases from outputs of the plurality of Hall elements. Position signal synthesizing means for generating a position signal, a voltage converting means for obtaining a variable output DC voltage from a DC power supply, one of an output terminal pair of the voltage converting means and a power supply terminal for each phase of the stator winding, A first driving transistor group including a plurality of driving transistors forming a current path between the first driving transistor group and a command signal for commanding current supply to a current supply terminal and a position signal from the position signal synthesizing means. First distribution control means for controlling the distribution of the current flowing through the first drive transistor group in response to the current, between the other of the output terminal pair of the voltage conversion means and the power supply terminal of each phase of the stator winding. a plurality of driving Trang forming a current path A second driving transistor group including a static, driving a plurality of constituting the first driving transistor group
Detects the minimum voltage among the operating voltages of the
First operating voltage detecting means for outputting as a first operating voltage
And a plurality of drivers constituting the second drive transistor group.
Detects the minimum voltage among the operating voltages of the
Second operating voltage detecting means for outputting as a second operating voltage
And according to the position signal from the position signal synthesizing means,
So that the first operating voltage matches the first reference voltage.
Distribution control of current flowing through the second drive transistor group
Second distribution control means, wherein the second operating voltage is a second reference voltage;
The output voltage of the voltage conversion means so as to match the reference voltage.
And voltage control means for controlling the first and second reference voltages, respectively.
In the first and second drive transistor groups, the fixed
Drive current in the phase switching state is applied to each of the two
Operating voltage of the drive transistor in the phase switching energized state
Is the one-phase energized drive
A brushless DC motor, wherein the brushless DC motor is variably set to be higher than an operating voltage of a star . 2. The method according to claim 1, wherein the first and second reference voltages are
Between the phase switching energized state and the one-phase energized state, respectively.
2. The brushless DC motor according to claim 1 , wherein the current changes continuously in a triangular waveform having a peak at a point in time when drive currents flowing in the two phases become substantially equal in the phase switching energized state. 3. The brushless DC motor according to claim 1, wherein the first and second reference voltages change in accordance with a command signal for commanding current supply to the stator winding. 4. A position signal synthesizing means, comprising: a plurality of buffer amplifiers for amplifying respective outputs of the plurality of Hall elements;
A plurality of subtraction circuits for generating a difference between outputs of the plurality of buffer amplifiers, wherein the first and second reference voltages are a plurality of phase position signals output by the position signal synthesizing means and the buffer amplifier. 2. The brushless DC motor according to claim 1, wherein the output is generated from a shaping signal obtained by shaping the output of the brushless DC motor. 5. A rotor having a plurality of magnetic poles, a plurality of stator windings, a plurality of Hall elements for detecting a rotational position of the rotor, and a plurality of phases from outputs of the plurality of Hall elements. Position signal synthesizing means for generating a position signal, a voltage converting means for obtaining a variable output DC voltage from a DC power supply, one of an output terminal pair of the voltage converting means and a power supply terminal for each phase of the stator winding, A first driving transistor group including a plurality of driving transistors forming a current path between the first driving transistor group and a command signal for commanding current supply to a current supply terminal and a position signal from the position signal synthesizing means. First distribution control means for controlling the distribution of the current flowing through the first drive transistor group in response to the current, between the other of the output terminal pair of the voltage conversion means and the power supply terminal of each phase of the stator winding. a plurality of driving Trang forming a current path A second driving transistor group including a static, driving a plurality of constituting the first driving transistor group
Detects the minimum voltage among the operating voltages of the
Operating voltage detecting means for outputting an operating voltage; and the position signal
The operating voltage is set to the reference voltage according to the position signal from the combining means.
To the second drive transistor group so as to match the voltage.
And a second distribution control means for performing distribution control of current to be supplied to the first driving transistor group.
The two phases of the stator windings are in the phase switching state respectively.
Driving transistor in phase-switching energized state for driving current
Operating voltage of the drive in the one-phase energized state after the phase switching is completed.
Variable to be higher than the operating voltage of the
Brushless DC motor, characterized in that it is a constant. 6. The phase switching circuit according to claim 1 , wherein
Continuously changing between an energized state and the one-phase energized state,
6. The brushless DC motor according to claim 5, wherein in the phase switching energized state, the brushless DC motor changes in a triangular waveform with a point at which the drive current flowing through the two phases becomes substantially equal. 7. The brushless DC motor according to claim 5, wherein the reference voltage changes according to a command signal for commanding current supply to the stator winding. 8. The position signal synthesizing means includes: a plurality of buffer amplifiers for amplifying respective outputs of the plurality of Hall elements;
A plurality of subtraction circuits for generating a difference between outputs of the plurality of buffer amplifiers, wherein the reference voltage signal is a waveform shaping of a plurality of phase position signals output by the position signal synthesizing means and an output of the buffer amplifier. The brushless DC motor according to claim 5, wherein the brushless DC motor is generated from the shaped signal.